MCP6N11 - 500 kHz, 800 uA Instrumentation Amplifier

MCP6N11500 kHz, 800 µA Instrumentation Amplifier

Features• Rail-to-Rail Input and Output• Gain Set by 2 External Resistors• Minimum Gain (GMIN) Options:

1, 2, 5, 10 or 100 V/V• Common Mode Rejection Ratio (CMRR): 115 dB

(typical, GMIN = 100)• Power Supply Rejection Ratio (PSRR): 112 dB

(typical, GMIN = 100)• Bandwidth: 500 kHz (typical, Gain = GMIN)• Supply Current: 800 μA/channel (typical)• Single Channel• Enable/VOS Calibration pin: (EN/CAL)• Power Supply: 1.8V to 5.5V• Extended Temperature Range: -40°C to +125°C

Typical Applications• High Side Current Sensor• Wheatstone Bridge Sensors• Difference Amplifier with Level Shifting• Power Control Loops

Design Aids• Microchip Advanced Part Selector (MAPS)• Demonstration Board• Application Notes

Block Diagram

DescriptionMicrochip Technology Inc. offers the single MCP6N11instrumentation amplifier (INA) with Enable/VOS Cali-bration pin (EN/CAL) and several minimum gainoptions. It is optimized for single-supply operation withrail-to-rail input (no common mode crossover distor-tion) and output performance.

Two external resistors set the gain, minimizing gainerror and drift-over temperature. The reference voltage(VREF) shifts the output voltage (VOUT).

The supply voltage range (1.8V to 5.5V) is low enoughto support many portable applications. All devices arefully specified from -40°C to +125°C.

These parts have five minimum gain options (1, 2, 5, 10and 100 V/V). This allows the user to optimize the inputoffset voltage and input noise for different applications.

Typical Application Circuit

Package Types

RF VFG

VOUT

Low Power

VSSVDD

EN/CAL

VOUT

VOS Calibration

VREF

RM4

GM2 ΣI2VREF

I4

GM3

I3

VTR

RG

VIP

VIM

GM1

I1VIP

VIM

POR

10 ΩVDD

IDD

VBAT+1.8V

to+5.5V

VOUT

VREF

VFG

RF

RG

200 kΩ

10 kΩ

U1MCP6N11

MCP6N11SOIC

VIP

VIM

VSS

VDD

VOUT

1

234

8

765 VREF

EN/CALVFG

MCP6N112×3 TDFN *

VIP

VIM

VSS

VDD

VOUT

1

2

34

8

7

65 VREF

EN/CALVFG

* Includes Exposed Thermal Pad (EP); see Table 3-1.

EP9

© 2011 Microchip Technology Inc. DS25073A-page 1

MCP6N11
Minimum Gain OptionsTable 1 shows key specifications that differentiatebetween the different minimum gain (GMIN) options.See Section 1.0 “Electrical Characteristics”,Section 6.0 “Packaging Information” and ProductIdentification System for further information on GMIN.
TABLE 1: KEY DIFFERENTIATING SPECIFICATIONS

Part No.GMIN(V/V)Nom.

VOS(±mV)Max.

∆VOS/∆TA(±µV/°C)

Typ.

CMRR (dB)Min.

VDD = 5.5V

PSRR(dB)Min.

VDMH(V)

Max.

GBWP(MHz)Nom.

Eni(µVP-P)Nom.

(f = 0.1 to 10 Hz)

eni(nV/√Hz)

Nom.(f = 10 kHz)

MCP6N11-001 1 3.0 90 70 62 2.70 0.50 570 950MCP6N11-002 2 2.0 45 78 68 1.35 1.0 285 475MCP6N11-005 5 0.85 18 80 75 0.54 2.5 114 190MCP6N11-010 10 0.50 9.0 81 81 0.27 5.0 57 95MCP6N11-100 100 0.35 2.7 88 86 0.027 35 18 35

DS25073A-page 2 © 2011 Microchip Technology Inc.

MCP6N11

1.0 ELECTRICAL CHARACTERISTICS

1.1 Absolute Maximum Ratings †VDD – VSS .......................................................................6.5VCurrent at Input Pins †† ...............................................±2 mAAnalog Inputs (VIP and VIM) †† ..... VSS – 1.0V to VDD + 1.0VAll Other Inputs and Outputs ......... VSS – 0.3V to VDD + 0.3VDifference Input Voltage....................................... |VDD – VSS|Output Short Circuit Current ................................ContinuousCurrent at Output and Supply Pins ............................±30 mAStorage Temperature ...................................-65°C to +150°CMax. Junction Temperature ........................................ +150°CESD protection on all pins (HBM, CDM, MM) .≥ 2 kV, 1.5 kV, 300V

† Notice: Stresses above those listed under “AbsoluteMaximum Ratings” may cause permanent damage tothe device. This is a stress rating only and functionaloperation of the device at those or any otherconditions above those indicated in the operationallistings of this specification is not implied. Exposure tomaximum rating conditions for extended periods mayaffect device reliability.†† See Section 4.2.1.2 “Input Voltage Limits” andSection 4.2.1.3 “Input Current Limits”.

1.2 Specifications

TABLE 1-1: DC ELECTRICAL SPECIFICATIONSElectrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = 1.8V to 5.5V, VSS = GND, EN/CAL = VDD, VCM = VDD/2, VDM = 0V, VREF = VDD/2, VL = VDD/2, RL = 10 kΩ to VL and GDM = GMIN; see Figure 1-6 and Figure 1-7.

Parameters Sym Min Typ Max Units GMIN Conditions

Input OffsetInput Offset Voltage, Calibrated

VOS -3.0 — +3.0 mV 1 (Note 2)-2.0 — +2.0 mV 2

-0.85 — +0.85 mV 5-0.50 — +0.50 mV 10-0.35 — +0.35 mV 100

Input Offset Voltage Trim Step

VOSTRM — 0.36 — mV 1— 0.21 — mV 2— 0.077 — mV 5— 0.045 — mV 10— 0.014 — mV 100

Input Offset Voltage Drift

ΔVOS/ΔTA — ±90/GMIN — µV/°C 1 to 10 TA= -40°C to +125°C (Note 3)— ±2.7 — µV/°C 100

Power Supply Rejection Ratio

PSRR 62 82 — dB 168 88 — dB 275 96 — dB 581 102 — dB 1086 112 — dB 100

Note 1: VCM = (VIP + VIM) / 2, VDM = (VIP – VIM) and GDM = 1 + RF/RG.2: The VOS spec limits include 1/f noise effects.3: This is the input offset drift without VOS re-calibration; toggle EN/CAL to minimize this effect.4: These specs apply to both the VIP, VIM input pair (use VCM) and to the VREF, VFG input pair (VREF takes VCM’s place).5: This spec applies to the VIP, VIM, VREF and VFG pins individually.6: Figure 2-11 and Figure 2-19 show the VIVR and VDMR variation over temperature.7: See Section 1.5 “Explanation of DC Error Specs”.


MCP6N11

Input Current and Impedance (Note 4)Input Bias Current IB — 10 — pA all

Across Temperature — 80 — pA TA= +85°CAcross Temperature 0 2 5 nA TA= +125°C

Input Offset Current IOS — ±1 — pAAcross Temperature — ±5 — pA TA= +85°CAcross Temperature -1 ±0.05 +1 nA TA= +125°C

Common Mode Input Impedance

ZCM — 1013||6 — Ω||pF

Differential Input Impedance

ZDIFF — 1013||3 — Ω||pF

Input Common Mode Voltage (VCM or VREF) (Note 4)Input Voltage Range VIVL — — VSS − 0.2 V all (Note 5, Note 6)

VIVH VDD + 0.15 — — VCommon Mode Rejection Ratio

CMRR 62 79 — dB 1 VCM = VIVL to VIVH, VDD = 1.8V69 87 — dB 2

75 101 — dB 579 107 — dB 1086 119 — dB 10070 94 — dB 1 VCM = VIVL to VIVH,

VDD = 5.5V78 100 — dB 280 108 — dB 581 114 — dB 1088 115 — dB 100

Common Mode Non-Linearity

INLCM -1000 ±115 +1000 ppm 1 VCM = VIVL to VIVH, VDM = 0V,VDD = 1.8V (Note 7)

-570 ±27 +570 ppm 2-230 ±11 +230 ppm 5-125 ±6 +125 ppm 10-50 ±2 +50 ppm 100

-400 ±42 +400 ppm 1 VCM = VIVL to VIVH, VDM = 0V,VDD = 5.5V (Note 7)

-220 ±10 +220 ppm 2-100 ±4 +100 ppm 5-50 ±2 +50 ppm 10-30 ±1 +30 ppm 100

TABLE 1-1: DC ELECTRICAL SPECIFICATIONS (CONTINUED)Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = 1.8V to 5.5V, VSS = GND, EN/CAL = VDD, VCM = VDD/2, VDM = 0V, VREF = VDD/2, VL = VDD/2, RL = 10 kΩ to VL and GDM = GMIN; see Figure 1-6 and Figure 1-7.




MCP6N11

Input Differential Mode Voltage (VDM) (Note 4)Differential Input Voltage Range

VDML -2.7/GMIN — — V all VREF = (VDD – GDMVDM)/2(Note 6)VDMH — — +2.7/GMIN V

Differential Gain Error gE -1 ±0.13 +1 % VDM = VDML to VDMH,Differential Gain Drift ΔgE/ΔTA — ±0.0006 — %/°C VREF = (VDD – GDMVDM)/2Differential Non-Linearity

INLDM -500 ±30 +500 ppm 1 (Note 7)-800 ±40 +800 ppm 2, 5

-2000 ±100 +2000 ppm 10, 100DC Open-Loop Gain AOL 61 84 — dB 1 VDD = 1.8V,

68 90 — dB 2 VOUT = 0.2V to 1.6V76 98 — dB 578 104 — dB 1086 116 — dB 10070 94 — dB 1 VDD = 5.5V,77 100 — dB 2 VOUT = 0.2V to 5.3V84 108 — dB 590 114 — dB 1097 125 — dB 100

OutputMinimum Output Voltage Swing

VOL — — VSS + 15 mV all VDM = -VDD/(2GDM),VDD = 1.8V,VREF = VDD/2 – 1V

— — VSS + 25 mV VDM = -VDD/(2GDM),VDD = 5.5V,VREF = VDD/2 – 1V

Maximum Output Voltage Swing

VOH VDD − 15 — — mV VDM = VDD/(2GDM),VDD = 1.8V,VREF = VDD/2 + 1V

VDD − 25 — — mV VDM = VDD/(2GDM),VDD = 5.5V,VREF = VDD/2 + 1V

Output Short Circuit Current

ISC — ±8 — mA VDD = 1.8V— ±30 — mA VDD = 5.5V

Power SupplySupply Voltage VDD 1.8 — 5.5 V allQuiescent Current per Amplifier

IQ 0.5 0.8 1.1 mA IO = 0

POR Trip Voltage VPRL 1.1 1.4 — VVPRH — 1.4 1.7 V

TABLE 1-1: DC ELECTRICAL SPECIFICATIONS (CONTINUED)Electrical Characteristics: Unless otherwise indicated, TA = +25°C, VDD = 1.8V to 5.5V, VSS = GND, EN/CAL = VDD, VCM = VDD/2, VDM = 0V, VREF = VDD/2, VL = VDD/2, RL = 10 kΩ to VL and GDM = GMIN; see Figure 1-6 and Figure 1-7.




MCP6N11

TABLE 1-2: AC ELECTRICAL SPECIFICATIONSElectrical Characteristics: Unless otherwise indicated, TA = 25°C, VDD = 1.8V to 5.5V, VSS = GND, EN/CAL = VDD, VCM = VDD/2, VDM = 0V, VREF = VDD/2, VL = VDD/2, RL = 10 kΩ to VL, CL = 60 pF and GDM = GMIN; see Figure 1-6 and Figure 1-7.


AC ResponseGain Bandwidth Product

GBWP — 0.50 GMIN — MHz 1 to 10— 35 — MHz 100

Phase Margin PM — 70 — ° allOpen-Loop Output Impedance

ROL — 0.9 — kΩ 1 to 10— 0.6 — kΩ 100

Power Supply Rejection Ratio

PSRR — 94 — dB all f < 10 kHz

Common Mode Rejection Ratio

CMRR — 104 — dB 1 to 10 f < 10 kHz— 94 — dB 100 f < 10 kHz

Step ResponseSlew Rate SR — 3 — V/µs 1 to 10 VDD = 1.8V

— 9 — V/µs VDD = 5.5V— 2 — V/µs 100 VDD = 1.8V— 6 — V/µs VDD = 5.5V

Overdrive Recovery,Input Common Mode

tIRC — 10 — µs all VCM = VSS – 1V (or VDD + 1V) to VDD/2,GDMVDM = ±0.1V, 90% of VOUT change

Overdrive Recovery,Input Differential Mode

tIRD — 5 — µs VDM = VDML – (0.5V)/GMIN(or VDMH + (0.5V)/GMIN) to 0V,

VREF = (VDD – GDMVDM)/2,90% of VOUT change

Overdrive Recovery, Output

tOR — 8 — µs GDM = 2GMIN, GDMVDM = 0.5VDD to 0V,VREF = 0.75VDD (or 0.25VDD),90% of VOUT change

NoiseInput Noise Voltage Eni — 570/GMIN — µVP-P 1 to 10 f = 0.1 Hz to 10 Hz

— 18 — µVP-P 100Input Noise Voltage Density

eni — 950/GMIN — nV/√Hz 1 to 10 f = 100 kHz— 35 — nV/√Hz 100

Input Current Noise Density

ini — 1 — fA/√Hz all f = 1 kHz


MCP6N11

TABLE 1-3: DIGITAL ELECTRICAL SPECIFICATIONSElectrical Characteristics: Unless otherwise indicated, TA = 25°C, VDD = 1.8V to 5.5V, VSS = GND, EN/CAL = VDD, VCM = VDD/2, VDM = 0V, VREF = VDD/2, VL = VDD/2, RL = 10 kΩ to VL, CL = 60 pF and GDM = GMIN; see Figure 1-6 and Figure 1-7.


EN/CAL Low SpecificationsEN/CAL Logic Threshold, Low

VIL VSS — 0.2 VDD V all

EN/CAL Input Current, Low

IENL — -0.1 — nA EN/CAL = 0V

GND Current ISS -7 -2.5 — µA EN/CAL = 0V, VDD = 5.5VAmplifier Output Leakage IO(LEAK) — 10 — nA EN/CAL = 0V

EN/CAL High SpecificationsEN/CAL Logic Threshold, High

VIH 0.8 VDD VDD V all

EN/CAL Input Current, High

IENH — -0.01 — nA EN/CAL = VDD

EN/CAL Dynamic SpecificationsEN/CAL Input Hysteresis VHYST — 0.2 — V all

EN/CAL Low to Amplifier Output High-Z Turn-off Time

tOFF — 3 10 µs EN/CAL = 0.2VDD to VOUT = 0.1(VDD/2), VDMGDM = 1 V, VL = 0V

EN/CAL High to Amplifier OutputOn Time

tON 12 20 28 ms EN/CAL = 0.8VDD to VOUT = 0.9(VDD/2), VDMGDM = 1 V, VL = 0V

EN/CAL Low to EN/CAL High low time

tENLH 100 — — µs Minimum time before externallyreleasing EN/CAL (Note 1)

Amplifier On to EN/CAL Low Setup Time

tENOL — 100 — µs

POR Dynamic SpecificationsVDD ↓ to Output Off tPHL — 10 — µs all VL = 0V, VDD = 1.8V to

VPRL – 0.1V step,90% of VOUT change

VDD ↑ to Output On tPLH 140 250 360 ms VL = 0V, VDD = 0V to VPRH + 0.1V step,90% of VOUT change

Note 1: For design guidance only; not tested.

TABLE 1-4: TEMPERATURE SPECIFICATIONSElectrical Characteristics: Unless otherwise indicated, all limits are specified for: VDD = 1.8V to 5.5V, VSS = GND.

Parameters Sym Min Typ Max Units ConditionsTemperature RangesSpecified Temperature Range TA -40 — +125 °C

Operating Temperature Range TA -40 — +125 °C (Note 1)Storage Temperature Range TA -65 — +150 °C

Thermal Package ResistancesThermal Resistance, 8L-SOIC θJA — 150 — °C/WThermal Resistance, 8L-TDFN (2×3) θJA — 53 — °C/WNote 1: Operation must not cause TJ to exceed the Absolute Maximum Junction Temperature specification (+150°C).


MCP6N11
1.3 Timing Diagrams
FIGURE 1-1: Common Mode Input Overdrive Recovery Timing Diagram.

FIGURE 1-2: Differential Mode Input Overdrive Recovery Timing Diagram.

FIGURE 1-3: Output Overdrive Recovery Timing Diagram.

FIGURE 1-4: POR Timing Diagram.

FIGURE 1-5: EN/CAL Timing Diagram.

VOUT

tIRC

VDM

VCM

±(1V)/GDM

VOUT

tIRD

VCM

VDM

VDD/2

VOUT

tOR

VCM

VDM

VDD/2

1.8V

VPRL – 0.1V

High-ZVOUT

VDD

tPHLtPLH

VPRH + 0.1V

0V

High-ZVOUT

EN/CAL

tOFFtON

tENLH

tENOL


MCP6N11
1.4 DC Test Circuits
1.4.1 INPUT OFFSET TEST CIRCUITFigure 1-6 is used for testing the INA’s input offseterrors and input voltage range (VE, VIVL and VIVH; seeSection 1.5.1 “Input Offset Related Errors” andSection 1.5.2 “Input Offset Common Mode Non-linearity”). U2 is part of a control loop that forces VOUTto equal VCNT; U1 can be set to any bias point.

FIGURE 1-6: Test Circuit for Common Mode (Input Offset).When MCP6N11 is in its normal range of operation, theDC output voltages are (where VE is the sum of inputoffset errors and gE is the gain error):

EQUATION 1-1:

Table 1-5 gives the recommended RF and RG valuesfor different GMIN options.

1.4.2 DIFFERENTIAL GAIN TEST CIRCUITFigure 1-7 is used for testing the INA’s differential gainerror, non-linearity and input voltage range (gE, INLDM,VDML and VDMH; see Section 1.5.3 “Differential GainError and Non-linearity”). RF and RG are 0.01% foraccurate gain error measurements.

FIGURE 1-7: Test Circuit for Differential Mode.The output voltages are (where VE is the sum of inputoffset errors and gE is the gain error):

EQUATION 1-2:

To keep VREF, VFG and VOUT within their ranges, set:

EQUATION 1-3:

Table 1-6 shows the recommended RF and RG. Theyproduce a 10 kΩ load; VL can usually be left open.

TABLE 1-5: SELECTING RF AND RGGMIN(V/V)Nom.

RF(Ω)

Nom.

RG(Ω)

Nom.

GDM(V/V)Nom.

GDMVOS(±V)Max.

BW(kHz)Nom.

1 100k 499 201.4 0.60 2.52 0.40 5.05 100k 100 1001 0.85 2.5

10 0.50 5.0100 0.35 35

RL

VCM 100 nF

VDD

2.2 µF

VREF

VL

12.7 kΩVM

100 nF CCNT

U1

MCP6N11

U2

MCP6H01

VCNT

63.4 kΩ

RG

RF

RCNT63.4 kΩ

VOUT

10 nF

1 kΩ1 kΩ

GDM 1 RF RG⁄+=VOUT VCNT=

VM VREF GDM 1 gE+( )VE+=

TABLE 1-6: SELECTING RF AND RGGMIN(V/V)Nom.

RF(Ω)

Nom.

RG(Ω)

Nom.

GDM(V/V)Nom.

1 0 Open 1.0002 4.99k 4.99k 2.0005 8.06k 2.00k 5.030

10 9.09k 1.00k 10.09100 10.0k 100 101.0

RL

6.34 kΩ

1 kΩ

1 kΩ

VCM + VDM/2

+100 nF

VOUT

RF

RG

VM–

100 nF

VDD

2.2 µF

6.34 kΩ

VREF

VFG

VL

VCM – VDM/2

0.01%

0.01%

U1

MCP6N11

GDM 1 RF RG⁄+=

VM VOUT VREF–=VOUT VREF GDM 1 gE+( ) VDM VE+( )+=

GDM 1 gE+( ) VDM VE+( )=

VREF VDD GDMVDM–( ) 2⁄=


MCP6N11
1.5 Explanation of DC Error Specs
1.5.1 INPUT OFFSET RELATED ERRORSThe input offset error (VE) is extracted from input offsetmeasurements (see Section 1.4.1 “Input Offset TestCircuit”), based on Equation 1-1:

EQUATION 1-4:

VE has several terms, which assume a linear responseto changes in VDD, VSS, VCM, VOUT and TA (all of whichare in their specified ranges):

EQUATION 1-5:

Equation 1-2 shows how VE affects VOUT.

1.5.2 INPUT OFFSET COMMON MODE NON-LINEARITY

The input offset error (VE) changes non-linearly withVCM. Figure 1-8 shows VE vs. VCM, as well as a linearfit line (VE_LIN) based on VOS and CMRR. The op ampis in standard conditions (ΔVOUT = 0, VDM = 0, etc.).VCM is swept from VIVL to VIVH. The test circuit is inSection 1.4.1 “Input Offset Test Circuit” and VE iscalculated using Equation 1-4.

FIGURE 1-8: Input Offset Error vs. Common Mode Input Voltage.

Based on the measured VE data, we obtain thefollowing linear fit:

EQUATION 1-6:

The remaining error (ΔVE) is described by the CommonMode Non-Linearity spec:

EQUATION 1-7:

The same common mode behavior applies to VE whenVREF is swept, instead of VCM, since both input stagesare designed the same:

EQUATION 1-8:

1.5.3 DIFFERENTIAL GAIN ERROR AND NON-LINEARITY

The differential errors are extracted from differentialgain measurements (see Section 1.4.2 “DifferentialGain Test Circuit”), based on Equation 1-2. Theseerrors are the differential gain error (gE) and the inputoffset error (VE, which changes non-linearly with VDM):

EQUATION 1-9:

These errors are adjusted for the expected output, thenreferred back to the input, giving the differential inputerror (VED) as a function of VDM:

EQUATION 1-10:

VEVM VREF–

GDM 1 gE+( )---------------------------------=

Where:

PSRR, CMRR and AOL are in units of V/V

ΔTA is in units of °C

VDM = 0

VE VOSΔVDD ΔVSS–

PSRR---------------------------------ΔVCMCMRR-----------------

ΔVREFCMRR-----------------+ + +=

ΔVOUT

AOL----------------- ΔTA

ΔVOSΔTA

-------------⋅+ +

V1

V3

VE, VE_LIN (V)

VCM (V)VIVL VIVHVDD/2

V2

VE_LIN

VE

ΔVE

Where:

VE_LIN VOSVCM VDD 2⁄–

CMRR-----------------------------------+=

VOS V2=

1CMRR-----------------

V3 V1–VIVH VIVL–------------------------------=

Where:

INLCMmax ΔVE

VIVH VIVL–------------------------------=

ΔVE VE VE_LIN–=

VE_LIN VOSVREF VDD 2⁄–

CMRR-------------------------------------+=

INLCMmax ΔVE

VIVH VIVL–------------------------------=

GDM 1 RF RG⁄+=VM GDM 1 gE+( ) VDM V+ E( )=

VEDVM

GDM------------ VDM–=


MCP6N11
Figure 1-9 shows VED vs. VDM, as well as a linear fitline (VED_LIN) based on VE and gE. The op amp is instandard conditions (ΔVOUT = 0, etc.). VDM is sweptfrom VDML to VDMH.
FIGURE 1-9: Differential Input Error vs. Differential Input Voltage.Based on the measured VED data, we obtain thefollowing linear fit:

EQUATION 1-11:

Note that the VE value measured here is not asaccurate as the one obtained in Section 1.5.1 “InputOffset Related Errors”.

The remaining error (ΔVED) is described by theDifferential Mode Non-Linearity spec:

EQUATION 1-12:

V1

V3

VED, VED_LIN (V)

VDM (V)VDML VDMH0

V2

VED_LIN

VED

ΔVED

Where:VED_LIN 1 gE+( )VE gEVDM+=

gEV3 V1–

VDMH VDML–----------------------------------- 1–=

VEV2

1 gE+----------------=

Where:

INLDMmax ΔVED

VDMH VDML–-----------------------------------=

ΔVED VED VED_LIN–=


MCP6N11
NOTES:

MCP6N11

2.0 TYPICAL PERFORMANCE CURVES

Note: Unless otherwise indicated, TA = +25°C, VDD = 1.8V to 5.5V, VSS = GND, EN/CAL = VDD, VCM = VDD/2, VDM = 0V, VREF = VDD/2, VL = VDD/2, RL = 10 kΩ to VL, CL = 60 pF and GDM = GMIN; see Figure 1-6 and Figure 1-7.

2.1 DC Voltages and Currents

FIGURE 2-1: Normalized Input Offset Voltage, with GMIN = 1 to 10.

FIGURE 2-2: Normalized Input Offset Voltage, with GMIN = 100.

FIGURE 2-3: Normalized Input Offset Voltage Drift, with GMIN = 1 to 10.

FIGURE 2-4: Normalized Input Offset Voltage Drift, with GMIN = 100.

Note: The graphs and tables provided following this note are a statistical summary based on a limited number ofsamples and are provided for informational purposes only. The performance characteristics listed hereinare not tested or guaranteed. In some graphs or tables, the data presented may be outside the specifiedoperating range (e.g., outside specified power supply range) and therefore outside the warranted range.

30%

35%

es

330 SamplesTA =

25%

30%

rren

ce

TA +25°CVDD = 1.8V and 5.5VRTO

20%

f Occ

u

GMIN = 1GMIN = 2 to 10

10%

15%

tage

of GMIN = 2 to 10

5%

10%

Perc

ent

0%

0 6 2 8 4 0 4 8 2 6 0

P

-2.0

-1.6

-1.2

-0.8

-0.4 0.0

0.4

0.8

1.2

1.6

2.0

Normalized Input Offset Voltage; GMINVOS (mV)

12%

14%

s

330 SamplesGMIN = 100

10%

12%

rren

ce

MINTA = +25°CVDD = 1.8V and 5.5VRTO

8%

f Occ

u

6%

tage

of

2%

4%

Perc

ent

0%

2%P

-18

-16

-14

-12

-10 -8 -6 -4 -2 0 2 4 6 8 10 12 14 16 18

Normalized Input Offset Voltage; GMINVOS (mV)

25%

ces No VOS Re-calibration

330 Samples20%

curr

enc p

GMIN = 1 to 10VDD = 5.5VRTO

15%

of O

cc

10%

enta

ge

5%

Perc

e

0%

600

500

400

300

200

100 0

100

200

300

400

500

600

- - -4 - -2 - 2 4

Normalized Input Offset Voltage Drift;GMIN(�VOS/�TA) (μV/°C)

16%18%

ces No VOS Re-calibration

330 Samples

12%14%

curr

enc p

GMIN = 100VDD = 5.5VRTO

8%10%

of O

cc

4%6%8%

enta

ge

0%2%4%

Perc

e

0%

1200

1000

-800

-600

-400

-200 0

200

400

600

800

1000

1200

-1 -1 1 1

Normalized Input Offset Voltage Drift;GMIN(�VOS/�TA) (μV/°C)


MCP6N11
FIGURE 2-5: Normalized Input Offset Voltage vs. Power Supply Voltage, with VCM = 0V and GMIN = 1 to 10.

FIGURE 2-6: Normalized Input Offset Voltage vs. Power Supply Voltage, with VCM = 0V and GMIN = 100.

FIGURE 2-7: Normalized Input Offset Voltage vs. Power Supply Voltage, with VCM = VDD and GMIN = 1 to 10.

FIGURE 2-8: Normalized Input Offset Voltage vs. Power Supply Voltage, with VCM = VDD and GMIN = 100.

FIGURE 2-9: Normalized Input Offset Voltage vs. Output Voltage, with GMIN = 1 to 10.

FIGURE 2-10: Normalized Input Offset Voltage vs. Output Voltage, with GMIN = 100.

-0.50.00.51.01.52.02.5

d In

put O

ffset

Vol

tage

;G

MIN

V OS

(mV)

-40°C

-2.5-2.0-1.5-1.0

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

6.5

Nor

mal

ized G

Power Supply Voltage

Representative PartVCM = VSSGMIN = 1 to 10RTO

+25°C+85°C

+125°C

2025

age;

1015

et V

olta

05

10

t Offs

eS

(mV)

-50

d In

put

GM

INV O

-15-10

mal

ized G

Representative PartVCM = VSS

-40°C25°C85°C

-25-20

Nor

m CM SSGMIN = 100RTO

85°C125°C

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

6.5


-0 4-0.20.00.20.40.60.81.01.2

d In

put O

ffset

Vol

tage

;G

MIN

V OS

(mV)

Representative PartVCM = VDDGMIN = 1 to 10RTO

-40°C+25°C+85°C+125°C

-1.2-1.0-0.8-0.6-0.4

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

6.5

Nor

mal

ize d G


-202468

10

d In

put O

ffset

Vol

tage

;G

MIN

V OS

(mV)

Representative PartVCM = VDDGMIN = 100RTO

-10-8-6-4

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

6.5

Nor

mal

ized G


-40°C+25°C+85°C+125°C

-0.5

0.0

0.5

1.0

1.5

2.0

d In

put O

ffset

Vol

tage

;G

MIN

V OS

(mV)

Representative PartGMIN = 1 to 10RTO

VDD = 5.5V

VDD = 1.8V

-2.0

-1.5

-1.0

0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5

Nor

mal

ized G

Output Voltage (V)

2-10123456

d In

put O

ffset

Vol

tage

;G

MIN

V OS

(mV)

Representative PartGMIN = 100RTO

VDD = 5.5V

VDD = 1.8V

-6-5-4-3-2

0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5

Nor

mal

ized G

Output Voltage (V)


MCP6N11
FIGURE 2-11: Input Common Mode Voltage Headroom vs. Ambient Temperature.

FIGURE 2-12: Normalized Input Offset Voltage vs. Common Mode Voltage, with VDD = 1.8V and GMIN = 1 to 10.

FIGURE 2-13: Normalized Input Offset Voltage vs. Common Mode Voltage, with VDD = 1.8V and GMIN = 100.

FIGURE 2-14: Normalized Input Offset Voltage vs. Common Mode Voltage, with VDD = 5.5V and GMIN = 1 to 10.

FIGURE 2-15: Normalized Input Offset Voltage vs. Common Mode Voltage, with VDD = 5.5V and GMIN = 100.

FIGURE 2-16: Normalized CMRR and PSRR vs. Ambient Temperature.

0.40.5

om

1 Wafer LotVIVH – VDD

0.20.3

Hea

droo

0 00.10.2

ange

HV) VDD = 1.8V

0 2-0.10.0

tage

Ra (V

DDVDD = 5.5V

-0.3-0.2

ut V

olt

-0.5-0.4In

p VIVL – VSS

-50 -25 0 25 50 75 100 125Ambient Temperature (°C)

-0.5

0.0

0.5

1.0

1.5

2.0

d In

put O

ffset

Vol

tage

;G

MIN

V OS

(mV)

VDD = 1.8VRepresentative PartGMIN = 1 to 10RTO

-2.0

-1.5

-1.0

-0.5 0.0 0.5 1.0 1.5 2.0 2.5

Nor

mal

ized G

Input Common Mode Voltage (V)

+125°C+85°C+25°C-40°C

-5

0

5

10

15

d In

put O

ffset

Vol

tage

;G

MIN

V OS

(mV)

VDD = 1.8VRepresentative PartGMIN = 100RTO

-15

-10

-5

-0.5 0.0 0.5 1.0 1.5 2.0 2.5

Nor

mal

ized G


+125°C+85°C+25°C-40°C

-0.5

0.0

0.5

1.0

1.5

2.0

d In

put O

ffset

Vol

tage

;G

MIN

V OS

(mV)

VDD = 5.5VRepresentative PartGMIN = 1 to 10RTO

+125°C+85

-2.0

-1.5

-1.0

-0.5 0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

Nor

mal

ized G


+85°C+25°C-40°C

-5

0

5

10

15

d In

put O

ffset

Vol

tage

;G

MIN

V OS

(mV)

VDD = 5.5VRepresentative PartGMIN = 100RTO

-15

-10

-5-0

.5 0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

Nor

mal

ized G


+125°C+85°C+25°C-40°C

105110

B) CMRR / GMIN, VDD = 1.8V:

GMIN = 1 100CMRR / GMIN, VDD = 5.5V:GMIN = 1 to 10

95100

PSR

R;

GM

IN(d

B GMIN 1, 100GMIN = 2 to 10

GMIN 1 to 10GMIN = 100

859095

MR

R,P

SRR

/ G

8085

ized

CM

GM

IN, P

S

7075

Nor

mal

RR

/ G

PSRR / GMIN:

6065

NC

M

PSRR / GMIN:GMIN = 1 to 10GMIN = 100

60-50 -25 0 25 50 75 100 125

Ambient Temperature (°C)


MCP6N11
FIGURE 2-17: Normalized DC Open-Loop Gain vs. Ambient Temperature.

FIGURE 2-18: The MCP6N11 Shows No Phase Reversal vs. Common Mode Voltage.

FIGURE 2-19: Normalized Differential Mode Voltage Range vs. Ambient Temperature.

FIGURE 2-20: Normalized Differential Input Error vs. Differential Voltage, with GMIN = 1.

FIGURE 2-21: Normalized Differential Input Error vs. Differential Voltage, with GMIN = 2 to 100.

FIGURE 2-22: The MCP6N11 Shows No Phase Reversal vs. Differential Voltage, with VDD = 5.5V.

105110

ain;

95100

oop

Ga

)

VDD = 5.5VVDD = 1.8V

859095

Ope

n-Lo

MIN

(dB

)

8085

d D

C O

AO

L/ G

M

7075

mal

ized A

GMIN = 1 to 10G = 100

6065

Nor

m GMIN = 100


5.56.0

Representative PartVDD = 5.5V

4 04.55.0

(V)

DD

GDM = 100

3 03.54.0

olta

ge

GDM 100GDM = 1

2.02.53.0

tput

Vo

VIM = -0.20V

1.01.52.0

Out VIM = VDD + 0.15V

0.00.5

-1.0

-0.5 0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

6.5

Non-inverting Input Voltage; VIP (V)

3.84.0

ut V)

1 Wafer LotGMINVDMH = -GMINVDML

3.43.6

ial I

npu

V DM

H(V RTO

3 03.23.4

ffere

nti

e; G

MIN

V

2.83.0

zed

Dif

Ran

ge

2.42.6

orm

aliz

olta

ge

2.02.2N

o Vo Note: For GMIN = 1,VDMH = minimum of plot value and VDD

-50 -25 0 25 50 75 100 125Axis Title

-2-1012345

ized

Diff

eren

tial I

nput

ror;

GM

INV E

D(m

V)

Representative PartVED = (VOUT – VREF)/GDM – VDMGMIN = 1RTO

VDD = 1.8V

VDD = 5.5V

-5-4-32

-5 -4 -3 -2 -1 0 1 2 3 4 5

Nor

mal Er

r

Normalized Differential Input Voltage;GMINVDM (V)

45

put Representative Part

VED = (VOUT – VREF)/GDM – VDM

23

tial I

npm

V)

VED (VOUT VREF)/GDM VDMGMIN = 2 to 100RTO

01

iffer

ent

MIN

V ED

(

-2-10

ized

Di

ror;

GM

-4-32

Nor

mal Er

r

-54

-5 -4 -3 -2 -1 0 1 2 3 4 5

N

-5 -4 -3 -2 -1 0 1 2 3 4 5Normalized Differential Input Voltage;

GMINVDM (V)

2.02.53.03.54.04.55.05.5

tput

Vol

tage

(V)

Representative PartVDD = 5.5VVREF = (VDD – GDMVDM)/2

0.00.51.01.5

-7 -6 -5 -4 -3 -2 -1 0 1 2 3 4 5 6 7

Out

Differential Input Voltage (V)

GMIN = 1GMIN = 2GMIN = 5GMIN = 10GMIN = 100


MCP6N11
FIGURE 2-23: Input Bias and Offset Currents vs. Ambient Temperature, with VDD = +5.5V.

FIGURE 2-24: Input Bias Current vs. Input Voltage (below VSS).

FIGURE 2-25: Input Bias and Offset Currents vs. Common Mode Input Voltage, with TA = +85°C.

FIGURE 2-26: Input Bias and Offset Currents vs. Common Mode Input Voltage, with TA = +125°C.

FIGURE 2-27: Output Voltage Headroom vs. Output Current.

FIGURE 2-28: Output Voltage Headroom vs. Ambient Temperature.

1.E-10

1.E-09

1.E-08

s, O

ffset

Cur

rent

s (A

) VDD = 5.5VVCM = VDD

IB100p

1n

10n

1.E-12

1.E-11

25 45 65 85 105 125

Inpu

t Bia

s


| IOS |

1p

10p

1 E 09

1.E-08

1.E-07

1.E-06

1.E-05

1.E-04

1.E-03

urre

nt M

agni

tude

(A)

+125°C+85°C+25°C-40°

1m

100μ

10μ

1μ

100n

10n

1n

1.E-12

1.E-11

1.E-10

1.E-09

-1.0 -0.9 -0.8 -0.7 -0.6 -0.5 -0.4 -0.3 -0.2 -0.1 0.0

Inpu

t Cu

Input Voltage (V)

-40 C1n

100p

10p

1p

-200

20406080

100

s, O

ffset

Cur

rent

s (p

A) Representative Part

TA = +85°CVDD = 5.5V

IB

IOS

-100-80-60-40

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

Inpu

t Bia

s

Common Mode Input Voltage (V)

1 0-0.50.00.51.01.52.02.5

s, O

ffset

Cur

rent

s (n

A) Representative Part

TA = +125°CVDD = 5.5V

IB

IOS

-2.5-2.0-1.5-1.0

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

Inpu

t Bia

s


1000

mV)

oom

(m

V = 1 8V

VDD = 5.5V

100Hea

dro VDD = 1.8V

100lta

geH

VDD – VOH

V V

put V

o VOL – VSS

10

Out

100.1 1 10

Output Current Magnitude (mA)

910

78

(mV) VDD – VOH

567

droo

m VDD = 5.5V

45

ut H

ead

23

Out

pu VDD = 1.8V

01

VOL – VSS0-50 -25 0 25 50 75 100 125



MCP6N11
FIGURE 2-29: Output Short Circuit Current vs. Power Supply Voltage.

FIGURE 2-30: Supply Current vs. Power Supply Voltage.

FIGURE 2-31: Supply Current vs. Common Mode Input Voltage.

-100

1020304050

ort C

ircui

t Cur

rent

(mA

)

+125°C+85°C+25°C-40°C

-50-40-30-20

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

6.5

Out

put S

ho

Power Supply Voltage (V)

400500600700800900

10001100

ply

Cur

rent

(μA

)

+125°C+85°C+25°C40

0100200300

0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0

6.5

Sup

Power Supply Voltage (V)

-40°C

10001100

800900

μA) VDD = 5.5V

600700

urre

nt (

VDD = 1.8V

400500

ply

Cu

200300Su

p

0100

-0.5 0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0



MCP6N11
2.2 Frequency Response

FIGURE 2-32: CMRR vs. Frequency.

FIGURE 2-33: PSRR vs. Frequency.

FIGURE 2-34: Normalized Open-Loop Gain vs. Frequency.

FIGURE 2-35: Normalized Gain Bandwidth Product and Phase Margin vs. Ambient Temperature.

FIGURE 2-36: Closed-Loop Output Impedance vs. Frequency.

FIGURE 2-37: Gain Peaking vs. Normalized Capacitive Load.

405060708090

100

CM

RR

(dB

)

GMIN = 1

VDD = 5.5V

0102030

1.E+03 1.E+04 1.E+05 1.E+06Frequency (Hz)

GMIN 1GMIN = 2GMIN = 5

GMIN = 10GMIN = 100

1k 10k 100k 1M

110120

VDD = 5.5V

8090

100

607080

R (d

B)

405060

PSR

R

203040

GMIN = 1GMIN = 2GMIN = 5

01020 MIN

GMIN = 10GMIN = 100


1k 10k 100k 1M

-90-60

100120

nn

-150-120

6080

op G

ai(°

)

op G

aiN

(dB

) �AOL/GMIN

210-180-150

204060

pen-

Loo

OL/G

MIN

pen-

Loo

AO

L/GM

IN

| AOL/GMIN |

-240-210

020

zed

Op

ase;

AO

zed

Op

tude

;A

-300-270

-40-20

orm

aliz

Pha

orm

aliz

Mag

nit

GMIN = 1GMIN = 2G = 5

-360-330

-80-60 N

o

No M GMIN = 5

GMIN = 10GMIN = 100


10k 100k 1M 10M

140150

0.450.50

h )

120130

0 350.40

°)ndw

ithN

(MH

z)

100110120

0 250.300.35

argi

n (

ain

Ban

WP/

GM

IN

GMIN = 1GMIN = 2 GBWP

90100

0.200.25

hase

Ma

ized

Ga

t; G

BW MIN

GMIN = 5GMIN = 10

GMIN = 100

GBWP

PM

7080

0.100.15 Ph

Nor

mal

iro

duct

5060

0.000.05N P


1.E+04

nce

10k

mpe

dan

GDM/GMIN = 10 GMIN = 1 to 101.E+03

tput

Im)

1kGDM/GMIN 10 MIN

1 E+02op O

ut (�

100

GMIN = 100

1.E+02

sed-

Loo 100

1 E+01

Clo

s

GDM/GMIN = 1101.E+011.E+03 1.E+04 1.E+05 1.E+06 1.E+07

Frequency (Hz)

101k 10k 100k 1M 10M

6

7

GMIN = GDM = 1 GMIN = 10

5

6

B)

GMIN GDM 1= 2= 5

= 10

GDM = 20= 50

4

5

king

(d = 100

3

n Pe

ak GMIN = 100GDM = 200

= 500

2Gai

0

1

01.E+1 1.E+2 1.E+3

Normalized Capacitive Load; CL(GMIN/GDM) (F)10p 100p 1n


MCP6N11
2.3 Noise

FIGURE 2-38: Normalized Input Noise Voltage Density vs. Frequency.

FIGURE 2-39: Normalized Input Noise Voltage Density vs. Input Common Mode Voltage, with f = 100 Hz.

FIGURE 2-40: Normalized Input Noise Voltage Density vs. Input Common Mode Voltage, with f = 10 kHz.

FIGURE 2-41: Normalized Input Noise Voltage vs. Time, with GMIN = 1 to 10.

FIGURE 2-42: Normalized Input Noise Voltage vs. Time, with GMIN = 100.

1000

age

1mRTO

100e Vo

lta/�

Hz)

100μ

10ut N

oise

Ne n

i(V/

10

GMIN = 100

10

ed In

puty

; GM

I 10μ

1

rmal

ize

Den

sit

1μ GMIN = 10GMIN = 5GMIN = 2

0.1

Nor

100n

GMIN 2GMIN = 1

1.E-1 1.E+0 1.E+1 1.E+2 1.E+3 1.E+4 1.E+5 1.E+6Frequency (Hz)

0.1 100 1k1 10 100k 1M10k

12

14

e

10

12

Volta

g�H

z)

G = 100

8

Noi

se

e ni(

μV/�


GMIN = 100GMIN = 10GMIN = 5GMIN = 2

6

Inpu

t ; G

MIN

e GMIN = 2GMIN = 1

2

4

mal

ized

en

sity

;

0

2

Nor

m De

f = 100 HzRTO

-0.5 0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0


3 5

4.0

e

3.0

3.5

Volta

g�H

z)

2 0

2.5

Noi

se

e ni(

μV/�


GMIN = 100GMIN = 10GMIN = 5

1.5

2.0

Inpu

t ; G

MIN

e MINGMIN = 2GMIN = 1

0 5

1.0

mal

ized

en

sity

;

0.0

0.5

Nor

m De

f = 10 kHzRTO

-0.5 0.0

0.5

1.0

1.5

2.0

2.5

3.0

3.5

4.0

4.5

5.0

5.5

6.0


0.40.5

Representative PartGMIN = 1 to 10

Analog NPBW = 0.1 HzSample Rate = 4 SPS

0 20.3

Noi

se;

)

MINRTO

p

0 00.10.2

nput

Nt)

(mV)

-0.10.0

aliz

ed I

GM

INe n

i(t

-0.3-0.2

Nor

ma G

-0.5-0.4

0 5 10 15 20 25 30 35Time (min)

1 5

2.0Representative PartGMIN = 100

Analog NPBW = 0.1 HzSample Rate = 4 SPS

1.0

1.5N

oise

;)

MINRTO

p

0 0

0.5

nput

Nt)

(mV)

-0.5

0.0

aliz

ed I

GM

INe n

i(t

-1.0

Nor

ma G

-2.0

-1.5

0 5 10 15 20 25 30 35Time (min)


MCP6N11
2.4 Time Response

FIGURE 2-43: Small Signal Step Response.

FIGURE 2-44: Large Signal Step Response.

FIGURE 2-45: Slew Rate vs. Ambient Temperature.

FIGURE 2-46: Maximum Output Voltage Swing vs. Frequency.

FIGURE 2-47: Common Mode Input Overdrive Recovery Time vs. Normalized Gain.

FIGURE 2-48: Differential Input Overdrive Recovery Time vs. Normalized Gain.

)

VDD = 5.5VGDM = GMIN

mV/

div) RF + RG = 10 k�

e (1

0 m

GMIN = 1 to 10

Volta

g GMIN = 1 to 10GMIN = 100

Out

put

O

0 2 4 6 8 10 12 14 16 18 20Time (μs)

5.05.5

)

VDD = 5.5VGDM = GMIN

4.04.5

mV/

div) DM MIN

RF + RG = 10 k�

3.03.5

e (1

0 m

2.02.5

Volta

ge GMIN = 1 to 10GMIN = 100

1.01.5

Out

put V

0.00.5

O

0.00.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0

Time (μs)

910

78

s)

567

te (V

/μs

G 1 t 10 V = 5 5V

45

ew R

at GMIN = 1 to 10GMIN = 100


23Sl

e

01


10

ing

ge S

w VDD = 5.5V

1t Vol

ta-P

) VDD = 1.8V1

Out

pu (VP

GMIN = 1 to 10GMIN = 100

xim

um

0

Max

01.E+4 1.E+5 1.E+6

Frequency (Hz)10k 100k 1M

1000

ge s)

GDMVDM = ±1V

100Vo

ltag

t IRC

(μs

VDD = 5.5V100

n M

ode

over

y;VDD = 1.8V

DD

10omm

onve

Rec

o

10

put C

ove

rdriv

GMIN = 100

1

Inp

Ov

GMIN = 1GMIN = 10

1 10 100Normalized Gain; GDM/GMIN

1000

age

s)

100e Vo

ltat IR

D(μ

s

VDD = 5.5V100

al M

ode

over

y;

VDD = 1.8V

VDD 5.5V

10eren

tiave

Rec

o

10

put D

iffe

verd

riv

GMIN = 100

1

Inp Ov

GMIN = 1GMIN = 10

1 10 100Normalized Gain; GDM/GMIN


MCP6N11
FIGURE 2-49: Output Overdrive Recovery Time vs. Normalized Gain.

FIGURE 2-50: The MCP6N11 Shows No Phase Reversal vs. Common Mode Input Overdrive, with VDD = 5.5V.

FIGURE 2-51: The MCP6N11 Shows No Phase Reversal vs. Differential Input Overdrive, with VDD = 5.5V.

10

100

1000

t Ove

rdriv

e R

ecov

ery;

t OR

(μs)

GMIN = 10

GDM = 2GMINVREF = 0.75VDD

GMIN = 1VDD = 1.8V

VDD = 5.5V

1

10

1 10 100

Out

pu

Normalized Gain; GDM/GMIN

GMIN = 100

5

6

ut

VDD = 5.5VGDMVDM = +0.1VVCM

4

5

, Out

pu

GDMVDM 0.1Vf = 10 kHz

3 Mod

e,es

(V)

2

mm

onVo

ltage

V G 1

0

1

put C

o V VOUT, GMIN = 1VOUT, GMIN = 100

-1

0

Inp

0 10 20 30 40 50 60 70 80 90 100Time (μs)

3

4VDD = 5.5V VIP

2

3

ges

(V)

0

1

Volta

g

VOUT, GMIN = 1VOUT, GMIN = 100

-1

0

Out

put

-2

nput

,O

-4

-3In

VIM

0 10 20 30 40 50 60 70 80 90 100Time (μs)


MCP6N11
2.5 Enable/Calibration and POR Responses

FIGURE 2-52: EN/CAL and Output Voltage vs. Time, with VDD = 1.8V.

FIGURE 2-53: EN/CAL and Output Voltage vs. Time, with VDD = 5.5V

FIGURE 2-54: EN/CAL Hysteresis vs. Ambient Temperature.

FIGURE 2-55: EN/CAL Turn On Time vs. Ambient Temperature.

FIGURE 2-56: Power Supply On and Off and Output Voltage vs. Time.

FIGURE 2-57: POR Trip Voltages and Hysteresis vs. Temperature.

0.60.81.01.21.41.61.82.0

, Out

put V

olta

ge (V

)

VDD = 1.8VVL = 0V

INAturns off

CalibrationStarts

INAturns on

-0.20.00.20.4

0 10 20 30 40 50 60 70 80 90 100

EN/C

AL

Time (ms)

EN/CAL VOUT

2.02.53.03.54.04.55.05.56.0

, Out

put V

olta

ge (V

)

VDD = 5.5VVL = 0V

INAturns off

CalibrationStarts

INAturns on

-0.50.00.51.01.5

0 10 20 30 40 50 60 70 80 90 100

EN/C

AL

Time (ms)

EN/CAL VOUT

0.550.60

0.450.50

s (V

)

VDD = 5.5V

0 300.350.40

ster

esi

0 200.250.30

AL

Hys

0.100.150.20

EN/C

A

VDD = 1.8V

0.000.050.10


30

ms)

25

; tO

N(m

VDD = 5.5V

15

20

n Ti

me;

VDD = 1.8V

10

15

urn

On

5

10

/CA

L T

0

5

EN/


1.61.8

e (V

) VL = 0V

1 21.4

Volta

ge

0 81.01.2

utpu

tV

On

0.60.8

ply,

Ou

VDD VOUT

On

0.20.4

er S

up

OffOff

-0.20.0

Pow

e

Calibrating OffOff

0.0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1.0Time (s)

0 06

0.08

0.10

0.12

0.14

0.16

0.18

1 1

1.2

1.3

1.4

1.5

1.6

1.7

R H

yste

resi

s (V

)

Trip

Vol

tage

s (V

)

VPRH – VPRL

VPRH

0.00

0.02

0.04

0.06

0.8

0.9

1.0

1.1

-50 -25 0 25 50 75 100 125

POR

POR


VPRL


MCP6N11
FIGURE 2-58: Quiescent Current in Shutdown vs. Power Supply Voltage.

FIGURE 2-59: Output Leakage Current vs. Output Voltage.

0.0

ent; EN/CAL = 0V

-0.5

y C

urre

-1.0

Supp

lyμA

)

-1.5

Pow

erS

I SS

(μ

-2.0ativ

eP

+125°C+85°C

-2.5

Neg

+85 C+25°C-40°C

0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0 6.5Power Supply Voltage (V)

1.E-07

A)

EN/CAL = 0VVDD = 5.5V

100n

1.E-08

rent

(A

+125°C

DD

10n

1.E-09

ge C

urr

+85°C1n

1.E-10

Leak

ag

100p

1.E-11utpu

tL

25°C10p

1.E-12

Ou +25°C

1p0.0 0.5 1.0 1.5 2.0 2.5 3.0 3.5 4.0 4.5 5.0 5.5 6.0

Output Voltage (V)

1p


MCP6N11

3.0 PIN DESCRIPTIONSDescriptions of the pins are listed in Table 3-1.

3.1 Analog Signal InputsThe non-inverting and inverting inputs (VIP, and VIM)are high-impedance CMOS inputs with low biascurrents.

3.2 Analog Feedback InputThe analog feedback input (VFG) is the inverting inputof the second input stage. The external feedbackcomponents (RF and RG) are connected to this pin. It isa high-impedance CMOS input with low bias current.

3.3 Analog Reference InputThe analog reference input (VREF) is the non-invertinginput of the second input stage; it shifts VOUT to itsdesired range. The external gain resistor (RG) isconnected to this pin. It is a high-impedance CMOSinput with low bias current.

3.4 Analog OutputThe analog output (VOUT) is a low-impedance voltageoutput. It represents the differential input voltage(VDM = VIP – VIM), with gain GDM and is shifted byVREF. The external feedback resistor (RF) is connectedto this pin.

3.5 Power Supply PinsThe positive power supply (VDD) is 1.8V to 5.5V higherthan the negative power supply (VSS). For normaloperation, the other pins are between VSS and VDD.

Typically, these parts are used in a single (positive)supply configuration. In this case, VSS is connected toground and VDD is connected to the supply; VDD willneed bypass capacitors.

3.6 Digital Enable and VOS Calibration Input

This input (EN/CAL) is a CMOS, Schmitt-triggeredinput that controls the active, low power and VOScalibration modes of operation. When this pin goes low,the part is placed into a low power mode and the outputis high-Z. When this pin goes high, the amplifier’s inputoffset voltage is corrected by the calibration circuitry,then the output is re-connected to the VOUT pin, whichbecomes low impedance, and the part resumes normaloperation.

3.7 Exposed Thermal Pad (EP)There is an internal connection between the ExposedThermal Pad (EP) and the VSS pin; they must beconnected to the same potential on the Printed CircuitBoard (PCB).

This pad can be connected to a PCB ground plane toprovide a larger heat sink. This improves the packagethermal resistance (θJA).

TABLE 3-1: PIN FUNCTION TABLEMCP6N11

Symbol DescriptionSOIC TDFN

1 1 VFG Feedback Input2 2 VIM Inverting Input3 3 VIP Non-inverting Input4 4 VSS Negative Power Supply5 5 VREF Reference Input6 6 VOUT Output7 7 VDD Positive Power Supply8 8 EN/CAL Enable/VOS Calibrate Digital Input— 9 EP Exposed Thermal Pad (EP); must be connected to VSS


MCP6N11
NOTES:

MCP6N11

4.0 APPLICATIONSThe MCP6N11 instrumentation amplifier (INA) ismanufactured using Microchip’s state of the art CMOSprocess. It is low cost, low power and high speed,making it ideal for battery-powered applications.

4.1 Basic Performance

4.1.1 STANDARD CIRCUITFigure 4-1 shows the standard circuit configuration forthese INAs. When the inputs and output are in theirspecified ranges, the output voltage is approximately:

EQUATION 4-1:

FIGURE 4-1: Standard Circuit.For normal operation, keep:

• VIP, VIM, VREF and VFG between VIVL and VIVH• VIP – VIM (i.e., VDM) between VDML and VDMH• VOUT between VOL and VOH

4.1.2 ARCHITECTUREFigure 4-2 shows the block diagram for these INAs.

FIGURE 4-2: MCP6N11 Block Diagram.

The input offset voltage (VOS) is corrected by thevoltage VTR. Each time a VOS Calibration event occurs,VTR is updated to the best value (at that moment).These events are triggered by either powering up(monitored by the POR) or by toggling the EN/CAL pinhigh. The current out of GM3 (I3) is constant and verysmall (assumed to be zero in the following discussion).

The input signal is applied to GM1. Equation 4-2 showsthe relationships between the input voltages (VIP andVIM) and the common mode and differential voltages(VCM and VDM).

EQUATION 4-2:

The negative feedback loop includes GM2, RM4, RF andRG. These blocks set the DC open-loop gain (AOL) andthe nominal differential gain (GDM):

EQUATION 4-3:

AOL is very high, so I4 is very small and I1 + I2 ≈ 0. Thismakes the differential inputs to GM1 and GM2 equal inmagnitude and opposite in polarity. Ideally, this gives:

EQUATION 4-4:

For an ideal part, changing VCM, VSS or VDD producesno change in VOUT. VREF shifts VOUT as needed.

The different GMIN options change GM1, GM2 and theinternal compensation capacitor. This results in theperformance trade-offs shown in Table 1.

VOUT ≈ VREF + GDMVDM

Where:

GDM = 1 + RF / RG

VOUT

VIP

VDD

VIM

VREF

VFGRF

RG

U1

MCP6N11

RF VFG

VOUT

Low Power

VSSVDD

EN/CAL

VOUT

VOS Calibration

VREF

RM4

GM2 ΣI2VREF

I4

GM3

I3

VTR

RG

VIP

VIM

GM1

I1VIP

VIM

POR

VIP VCM VDM 2⁄+=VIM VCM VDM 2⁄–=

VCM VIP VIM+( ) 2⁄=VDM VIP VIM–=

AOL GM2RM4=GDM 1 RF RG⁄+=

VFG VREF–( ) VDM=VOUT VDMGDM VREF+=


MCP6N11
4.1.3 DC ERRORSSection 1.5 “Explanation of DC Error Specs”defines some of the DC error specifications. Theseerrors are internal to the INA, and can be summarizedas follows:
EQUATION 4-5:

The non-linearity specs (INLCM and INLDM) describeerrors that are non-linear functions of VCM and VDM,respectively. They give the maximum excursion fromlinear response over the entire common mode anddifferential ranges.

The input bias current and offset current specs (IB andIOS), together with a circuit’s external input resistances,give an additional DC error. Figure 4-3 shows theresistors that set the DC bias point.

FIGURE 4-3: DC Bias Resistors.The resistors at the main input (RIP and RIM) and itsinput bias currents (IBP and IBM) give the followingchanges in the INA’s bias voltages:

EQUATION 4-6:

The best design results when RIP and RIM are equaland small:

EQUATION 4-7:

The resistors at the feedback input (RR, RF and RG)and its input bias currents (IBR and IBF) give thefollowing changes in the INA’s bias voltages:

EQUATION 4-8:

The best design results when GDMRR and RF are equaland small:

EQUATION 4-9:

Where:

VOUT VREF GDM 1 gE+( ) VDM ΔVED+( )+= GDM 1 gE+( ) VE ΔVE+( )+

Where:

PSRR, CMRR and AOL are in units of V/V

ΔTA is in units of °C

VE VOSΔVDD ΔVSS–

PSRR---------------------------------ΔVCMCMRR-----------------

ΔVREFCMRR-----------------+ + +=

ΔVOUT

AOL----------------- ΔTA

ΔVOSΔTA

-------------⋅+ +

ΔVED INLDM VDMH VDML–( )≤ΔVE INLCM VIVH VIVL–( )≤

VOUT

VIP

VDD

VIM

VREF

RF

RG

RIP

RIM

RR

IBP

IBM VFG

IBF

IBR

U1MCP6N11

Where:

CMRR is in units of V/V

ΔVIP IBPRIP– IB–IOS2--------–⎝ ⎠

⎛ ⎞RIP= =

ΔVIM IBMRIM– IB–IOS2--------+⎝ ⎠

⎛ ⎞RIM= =

ΔVCMΔVIP ΔVIM+

2---------------------------------=

I– BRIP RIM+

2-------------------------⎝ ⎠⎛ ⎞ I– OS

2-----------R– IP RIM+

2----------------------------⎝ ⎠⎛ ⎞+=

ΔVDM ΔVIP ΔVIM–=

IB R– IP RIM+( )IOS2-------- RIP RIM+( )–=

ΔVOUT GDM ΔVDMΔVCMCMRR-----------------+⎝ ⎠

⎛ ⎞=

Where:

RIP = RIM

εRTOL = tolerance of RIP and RIM

ΔVOUT GDMΔVDM≈ GDM 2IBεRTOL IOS–±( )RIP≈

Where:

IB2 meets the IB spec, but is not equal to IBIOS2 meets the IOS spec, but is not equal to IOS

ΔVREF IBRRR– IB2–IOS2

2----------–⎝ ⎠

⎛ ⎞RR= =

ΔVFG ΔVREF ,≈

ΔVOUT IB2 RF GDMRR–( )IOS2

2---------- RF GDMRR+( )+≈

due to high AOL

Where:

GDMRR = RF

εRTOL = tolerance of RR, RF and RG

ΔVOUT 2IB2εRTOL IOS2+( )±( )RF≈


MCP6N11
4.1.4 AC PERFORMANCEThe bandwidth of these amplifiers depends on GDMand GMIN:
EQUATION 4-10:

The bandwidth at the maximum output swing is calledthe Full Power Bandwidth (fFPBW). It is limited by theSlew Rate (SR) for many amplifiers, but is close to fBWfor these parts:

EQUATION 4-11:

CMRR is constant from DC to about 1 kHz.

4.1.5 NOISE PERFORMANCEAs shown in Figures 2-41 and 2-42, the 1/f noisecauses an apparent wander in the DC output voltage.Changing the measurement time or bandwidth has littleeffect on this noise.

We recommend re-calibrating VOS periodically, toreduce 1/f noise wander. For example, VOS could bere-calibrated at least once every 15 minutes; moreoften when temperature or VDD change significantly.

4.2 Functional Blocks

4.2.1 RAIL-TO-RAIL INPUTSEach input stage uses one PMOS differential pair at theinput. The output of each differential pair is processedusing current mode circuitry. The inputs show nocrossover distortion vs. common mode voltage.

With this topology, the inputs (VIP and VIM) operatenormally down to VSS – 0.2V and up to VDD + 0.15V atroom temperature (see Figure 2-11). The input offsetvoltage (VOS) is measured at VCM = VSS – 0.2V andVDD + 0.15V (at +25°C), to ensure proper operation.

4.2.1.1 Phase ReversalThe input devices are designed to not exhibit phaseinversion when the input pins exceed the supplyvoltages. Figures 2-18 and 2-50 show an input voltageexceeding both supplies with no phase inversion.

The input devices also do not exhibit phase inversionwhen the differential input voltage exceeds its limits;see Figures 2-22 and 2-51.

4.2.1.2 Input Voltage LimitsIn order to prevent damage and/or improper operationof these amplifiers, the circuit must limit the voltages atthe input pins (see Section 1.1 “Absolute MaximumRatings †”). This requirement is independent of thecurrent limits discussed later on.

The ESD protection on the inputs can be depicted asshown in Figure 4-4. This structure was chosen toprotect the input transistors against many (but not all)overvoltage conditions, and to minimize input biascurrent (IB).

FIGURE 4-4: Simplified Analog Input ESD Structures.

Where:

fBW = -3 dB bandwidth

fGBWP = Gain bandwidth product

fBWfGBWPGDM

---------------≈

0.50 MHz( ) GMIN GDM⁄( ),≈ 0.35 MHz( ) GMIN GDM⁄( ),≈

GMIN = 1, …, 10

GMIN = 100

Where:

VO = Maximum output voltage swing

≈ VOH – VOL

fFPBWSRπVO----------≈

fBW≈ , for these parts

BondPad

BondPad

BondPad

VDD

VIP

VSS

InputStage

BondPad

VIM

ofINA Input


MCP6N11
The input ESD diodes clamp the inputs when they tryto go more than one diode drop below VSS. They alsoclamp any voltages that go too far above VDD; theirbreakdown voltage is high enough to allow normaloperation, but not low enough to protect against slowovervoltage (beyond VDD) events. Very fast ESDevents (that meet the spec) are limited so that damagedoes not occur.
In some applications, it may be necessary to preventexcessive voltages from reaching the op amp inputs.Figure 4-5 shows one approach to protecting theseinputs. D1 and D2 may be small signal silicon diodes,Schottky diodes for lower clamping voltages or diode-connected FETs for low leakage.

FIGURE 4-5: Protecting the Analog Inputs Against High Voltages.

4.2.1.3 Input Current LimitsIn order to prevent damage and/or improper operationof these amplifiers, the circuit must limit the currentsinto the input pins (see Section 1.1 “Absolute Maxi-mum Ratings †”). This requirement is independent ofthe voltage limits previously discussed.

Figure 4-6 shows one approach to protecting theseinputs. The resistors R1 and R2 limit the possiblecurrent in or out of the input pins (and into D1 and D2).The diode currents will dump onto VDD.

FIGURE 4-6: Protecting the Analog Inputs Against High Currents.

It is also possible to connect the diodes to the left of theresistor R1 and R2. In this case, the currents throughthe diodes D1 and D2 need to be limited by some othermechanism. The resistors then serve as in-rush currentlimiters; the DC current into the input pins (VIP and VIM)should be very small.

A significant amount of current can flow out of theinputs (through the ESD diodes) when the commonmode voltage (VCM) is below ground (VSS); seeFigure 2-25.

4.2.1.4 Input Voltage RangesFigure 4-7 shows possible input voltage values(VSS = 0V). Lines with a slope of +1 have constant VDM(e.g., the VDM = 0 line). Lines with a slope of -1 haveconstant VCM (e.g., the VCM = VDD/2 line).

For normal operation, VIP and VIM must be kept withinthe region surrounded by the thick blue lines. Thehorizontal and vertical blue lines show the limits on theindividual inputs. The blue lines with a slope of +1 showthe limits on VDM; the larger GMIN is, the closer they areto the VDM = 0 line.

The input voltage range specs (VIVL and VIVH) changewith the supply voltages (VSS and VDD, respectively).The differential input range specs (VDML and VDMH)change with minimum gain (GMIN). Temperature alsoaffects these specs.

To take full advantage of VDML and VDMH, set VREF(see Figure 1-6 and Figure 1-7) so that the output(VOUT) is centered between the supplies (VSS andVDD).

FIGURE 4-7: Input Voltage Ranges.

VDD

V1

D1

V2

D2

U1MCP6N11

min(R1, R2) >VSS – min(V1, V2)

2 mA

VDD

V1R1

D1

V2R2

D2

U1MCP6N11

min(R1, R2) >max(V1, V2) – VDD

2 mA

VIP

VIM

V DM= 0

VIVH

VIVL0

VIV

H

VIV

L

0

V DM= V DMH

VCM = V

DD /2

V DM= V DMH

VDD

VD

D


MCP6N11
4.2.2 ENABLE/VOS CALIBRATION
(EN/CAL)These parts have a Normal mode, a Low Power modeand a VOS Calibration mode.

When the EN/CAL pin is high and the internal POR(with delay) indicates that power is good, the partoperates in its Normal mode.

When the EN/CAL pin is low, the part operates in itsLow Power mode. The quiescent current (at VSS) dropsto -2.5 µA (typical), the amplifier output is put into ahigh-impedance state. Signals at the input pins canfeed through to the output pin.

When the EN/CAL pin goes high and the internal POR(with delay) indicates that power is good, the amplifierinternally corrects its input offset voltage (VOS) with theinternal common mode voltage at mid-supply (VDD/2)and the output tri-stated (after tOFF). Once VOS Calibra-tion is completed, the amplifier is enabled and normaloperation resumes.

The EN/CAL pin does not operate normally when leftfloating. Either drive it with a logic output, or tie it highso that the part is always on.

4.2.3 POR WITH DELAYThe internal POR makes sure that the input offsetvoltage (VOS) is calibrated whenever the supplyvoltage goes from low voltage (< VPRL) to high voltage(> VPRH). This prevents corruption of the VOS trim reg-isters after a low-power event.

After the POR goes high, the internal circuitry adds afixed delay (tPLH), before telling the VOS Calibrationcircuitry (see Figure 4-2) to start. If the EN/CAL pin istoggled during this time, the fixed delay is restarted(takes an additional time tPLH).

4.2.4 PARITY DETECTORA parity error detector monitors the memory contentsfor any corruption. In the rare event that a parity error isdetected (e.g., corruption from an alpha particle), aPOR event is automatically triggered. This will causethe input offset voltage to be re-corrected, and the opamp will not return to normal operation for a period oftime (the POR turn on time, tPLH).

4.2.5 RAIL-TO-RAIL OUTPUTThe Minimum Output Voltage (VOL) and MaximumOutput Voltage (VOH) specs describe the widest outputswing that can be achieved under the specified loadconditions.

The output can also be limited when VIP or VIM exceedsVIVL or VIVH, or when VDM exceeds VDML or VDMH.

4.3 Applications Tips

4.3.1 MINIMUM STABLE GAINThere are different options for different Minimum StableGains (1, 2, 5, 10 and 100 V/V; see Table 1-1). Thedifferential gain (GDM) needs to be greater than orequal to GMIN in order to maintain stability.

Picking a part with higher GMIN has the advantages oflower Input Noise Voltage Density (eni), lower InputOffset Voltage (VOS) and increased Gain BandwidthProduct (GBWP); see Table 1. The Differential InputVoltage Range (VDMR) is lower for higher GMIN, but theoutput voltage range would limit VDMR anyway, whenGDM ≥ 2.

4.3.2 CAPACITIVE LOADSDriving large capacitive loads can cause stabilityproblems for amplifiers. As the load capacitanceincreases, the feedback loop’s phase margindecreases, and the closed-loop bandwidth is reduced.This produces gain peaking in the frequency response,with overshoot and ringing in the step response. Lowergains (GDM) exhibit greater sensitivity to capacitiveloads.

When driving large capacitive loads with theseinstrumentation amps (e.g., > 100 pF), a small seriesresistor at the output (RISO in Figure 4-8) improves thefeedback loop’s phase margin (stability) by making theoutput load resistive at higher frequencies. Thebandwidth will be generally lower than the bandwidthwith no capacitive load.

FIGURE 4-8: Output Resistor, RISO stabilizes large capacitive loads.Figure 4-9 gives recommended RISO values fordifferent capacitive loads and gains. The x-axis is thenormalized load capacitance (CL GMIN/GDM), whereGDM is the circuit’s differential gain (1 + RF / RG) andGMIN is the minimum stable gain.

RISOVOUT

CL

V1

VDD

V2

VREF

VFGRF

RG

U1MCP6N11


MCP6N11

FIGURE 4-9: Recommended RISO Values for Capacitive Loads.After selecting RISO for your circuit, double check theresulting frequency response peaking and stepresponse overshoot on the bench. Modify RISO’s valueuntil the response is reasonable.

4.3.3 GAIN RESISTORSFigure 4-10 shows a simple gain circuit with the INA’sinput capacitances at the feedback inputs (VREF andVFG). These capacitances interact with RG and RF tomodify the gain at high frequencies. The equivalentcapacitance acting in parallel to RG is CG = CDM + CCMplus any board capacitance in parallel to RG. CG willcause an increase in GDM at high frequencies, whichreduces the phase margin of the feedback loop (i.e.,reduce the feedback loop's stability).

FIGURE 4-10: Simple Gain Circuit with Parasitic Capacitances.

In this data sheet, RF + RG = 10 kΩ for most gains (0Ωfor GDM = 1); see Table 1-6. This choice gives goodPhase Margin. In general, RF (Figure 4-10) needs tomeet the following limits to maintain stability:

EQUATION 4-12:

4.3.4 SUPPLY BYPASSWith these INAs, the power supply pin (VDD for singlesupply) should have a local bypass capacitor (i.e.,0.01 µF to 0.1 µF) within 2 mm for good high frequencyperformance. Surface mount, multilayer ceramiccapacitors, or their equivalent, should be used.

These INAs require a bulk capacitor (i.e., 1.0 µF orlarger) within 100 mm, to provide large, slow currents.This bulk capacitor can be shared with other nearbyanalog parts as long as crosstalk through the suppliesdoes not prove to be a problem.

1.E+03

1.E+04om

men

ded

RIS

O(�

)10k

1k

1.E+021.E-10 1.E-09 1.E-08 1.E-07 1.E-06

Rec

o

Normalized Load Capacitance;CL GMIN/GDM (F)

100100p 1n 10n 100n 1μ

GMIN = 1 to 10GMIN = 100

VOUT

V1

VDD

V2

VREF

VFGRF

RGCDMCCMCCM

U1MCP6N11

Where:

α ≤ 0.25

GDM ≥ GMIN

fGBWP = Gain Bandwidth Product

CG = CDM + CCM + (PCB stray capacitance)

RF 0=

For GDM = 1:

RFαGDM

2

2πfGBWPCG------------------------------<

For GDM > 1:


MCP6N11
4.4 Typical Applications
4.4.1 HIGH INPUT IMPEDANCE DIFFERENCE AMPLIFIER

Figure 4-11 shows the MCP6N11 used as a differenceamplifier. The inputs are high impedance and give goodCMRR performance.

FIGURE 4-11: Difference Amplifier.

4.4.2 DIFFERENCE AMPLIFIER FOR VERY LARGE COMMON MODE SIGNALS

Figure 4-12 shows the MCP6N11 INA used as adifference amplifier for signals with a very largecommon mode component. The input resistor dividers(R1 and R2) ensure that the voltages at the INA’s inputsare within their range of normal operation. Thecapacitors C1, with the parasitic capacitances C2 (theresistors’ parasitic capacitance plus the INA’s inputcommon mode capacitance, CCM), set the samedivision ratio, so that high-frequency signals (e.g., astep in voltage) have the same gain. Select the INAgain to compensate for R1 and R2’s attenuation. SelectR1 and R2’s tolerances for good CMRR.

FIGURE 4-12: Difference Amplifier with Very Large Common Mode Component.

4.4.3 HIGH SIDE CURRENT DETECTORFigure 4-13 shows the MCP6N11 INA used as to detectand amplify the high side current in a battery powereddesign. The INA gain is set at 21 V/V, so VOUT changes210 mV for every 1 mA of IDD current. The best GMINoption to pick would be a gain of 10 (MCP6N11-010).

FIGURE 4-13: High Side Current Detector.

4.4.4 WHEATSTONE BRIDGEFigure 4-14 shows the MCP6N11 singleinstrumentation amp used to condition the signal froma Wheatstone bridge (e.g., strain gage). The overallINA gain is set at 201 V/V. The best GMIN option to pick,for this gain, is 100 V/V (MCP6N11-100).

FIGURE 4-14: Wheatstone Bridge Amplifier.

VOUT

VIP

VDD

VIM

VREF

VFGRF

RG

U1

MCP6N11

VOUT

VDD

VREF

VFGRF

RG

R2R1

V2

C1 C2

R2R1V1

C1 C2

U1MCP6N11

IDD =(VBAT – VDD)

(10 Ω)

=(VOUT – VREF)

(10 Ω) (21.0 V/V)

10 ΩVDD

IDD

VBAT+1.8V

to+5.5V

VOUT

VREF

VFG

RF

RG

200 kΩ

10 kΩ

U1MCP6N11

VOUT

VREF

VFG

RF

RG

200 kΩ

1 kΩ

VDD

RW1RW2

RW2RW1U1MCP6N11


MCP6N11
NOTES:

MCP6N11

5.0 DESIGN AIDSMicrochip provides the basic design aids needed forthe MCP6N11 instrumentation amplifiers.

5.1 Microchip Advanced Part Selector (MAPS)

MAPS is a software tool that helps efficiently identifyMicrochip devices that fit a particular designrequirement. Available at no cost from the Microchipwebsite at www.microchip.com/maps, the MAPS is anoverall selection tool for Microchip’s product portfoliothat includes Analog, Memory, MCUs and DSCs. Usingthis tool, a customer can define a filter to sort featuresfor a parametric search of devices and exportside-by-side technical comparison reports. Helpful linksare also provided for Data sheets, Purchase andSampling of Microchip parts.

5.2 Analog Demonstration BoardMicrochip offers a broad spectrum of AnalogDemonstration and Evaluation Boards that aredesigned to help customers achieve faster timeto market. For a complete listing of these boardsand their corresponding user’s guides and technicalinformation, visit the Microchip web site atwww.microchip.com/analog tools.

5.3 Application NotesThe following Microchip Application Notes areavailable on the Microchip web site at www.microchip.com/appnotes and are recommended as supplementalreference resources.

• AN884: “Driving Capacitive Loads With Op Amps”, DS00884

• AN990: “Analog Sensor Conditioning Circuits – An Overview”, DS00990

• AN1228: “Op Amp Precision Design: Random Noise”, DS01228

Some of these application notes, and others, are listedin the design guide:

• “Signal Chain Design Guide”, DS21825


www.microchip.com/maps

www.microchip.com/maps

www.micro chip.com/analog tools

www.micro chip.com/analog tools

www.microchip. com/appnotes

www.microchip. com/appnotes

MCP6N11
NOTES:

MCP6N11

6.0 PACKAGING INFORMATION

6.1 Package Marking Information

Legend: XX...X Customer-specific informationY Year code (last digit of calendar year)YY Year code (last 2 digits of calendar year)WW Week code (week of January 1 is week ‘01’)NNN Alphanumeric traceability code Pb-free JEDEC designator for Matte Tin (Sn)* This package is Pb-free. The Pb-free JEDEC designator ( )

can be found on the outer packaging for this package.

Note: In the event the full Microchip part number cannot be marked on one line, it willbe carried over to the next line, thus limiting the number of availablecharacters for customer-specific information.

3e

3e

8-Lead SOIC (150 mil) (MCP6N11)

8-Lead TDFN (2×3) (MCP6N11) Example

Device Code

MCP6N11-001 AAQMCP6N11-002 AARMCP6N11-005 AASMCP6N11-010 AATMCP6N11-100 AAU

Note: Applies to 8-Lead 2x3 TDFN

Note: The example is for aMCP6N11-001 part.

NNN

6N11001ESN^^ 1121

2563e

AAQ12125

Example


MCP6N11

Note: For the most current package drawings, please see the Microchip Packaging Specification located at http://www.microchip.com/packaging


MCP6N11



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MCP6N11
NOTES:

MCP6N11

APPENDIX A: REVISION HISTORY

Revision A (October 2011)• Original Release of this Document.


MCP6N11
NOTES:

MCP6N11

PRODUCT IDENTIFICATION SYSTEMTo order or obtain information, e.g., on pricing or delivery, refer to the factory or the listed sales office.

Device: MCP6N11 Single Instrumentation Amplifier

MCP6N11T Single Instrumentation Amplifier(Tape and Reel)

Gain Option: 001 = Minimum gain of 1 V/V002 = Minimum gain of 2 V/V005 = Minimum gain of 5 V/V010 = Minimum gain of 10 V/V100 = Minimum gain of 100 V/V

Temperature Range: E = -40°C to +125°C

Package: MNY = 2×3 TDFN, 8-lead *SN = Plastic SOIC (150mil Body), 8-lead

* Y = nickel palladium gold manufacturing designator. Onlyavailable on the TDFN package.

Examples:a) MCP6N11T-001E/MNY: Tape and Reel,

Minimum gain = 1,Extended temperature, 8LD 2×3 TDFN.

b) MCP6N11-002E/SN: Minimum gain = 2, Extended temperature,8LD SOIC.

PART NO. –X /XX

PackageGainOption

Device

X

TemperatureRange


MCP6N11
NOTES:

Note the following details of the code protection feature on Microchip devices:• Microchip products meet the specification contained in their particular Microchip Data Sheet.

• Microchip believes that its family of products is one of the most secure families of its kind on the market today, when used in the intended manner and under normal conditions.

• There are dishonest and possibly illegal methods used to breach the code protection feature. All of these methods, to our knowledge, require using the Microchip products in a manner outside the operating specifications contained in Microchip’s Data Sheets. Most likely, the person doing so is engaged in theft of intellectual property.

• Microchip is willing to work with the customer who is concerned about the integrity of their code.

• Neither Microchip nor any other semiconductor manufacturer can guarantee the security of their code. Code protection does not mean that we are guaranteeing the product as “unbreakable.”

Code protection is constantly evolving. We at Microchip are committed to continuously improving the code protection features of ourproducts. Attempts to break Microchip’s code protection feature may be a violation of the Digital Millennium Copyright Act. If such actsallow unauthorized access to your software or other copyrighted work, you may have a right to sue for relief under that Act.

Information contained in this publication regarding deviceapplications and the like is provided only for your convenienceand may be superseded by updates. It is your responsibility toensure that your application meets with your specifications.MICROCHIP MAKES NO REPRESENTATIONS ORWARRANTIES OF ANY KIND WHETHER EXPRESS ORIMPLIED, WRITTEN OR ORAL, STATUTORY OROTHERWISE, RELATED TO THE INFORMATION,INCLUDING BUT NOT LIMITED TO ITS CONDITION,QUALITY, PERFORMANCE, MERCHANTABILITY ORFITNESS FOR PURPOSE. Microchip disclaims all liabilityarising from this information and its use. Use of Microchipdevices in life support and/or safety applications is entirely atthe buyer’s risk, and the buyer agrees to defend, indemnify andhold harmless Microchip from any and all damages, claims,suits, or expenses resulting from such use. No licenses areconveyed, implicitly or otherwise, under any Microchipintellectual property rights.

© 2011 Microchip Technology Inc.

Trademarks

The Microchip name and logo, the Microchip logo, dsPIC, KEELOQ, KEELOQ logo, MPLAB, PIC, PICmicro, PICSTART, PIC32 logo, rfPIC and UNI/O are registered trademarks of Microchip Technology Incorporated in the U.S.A. and other countries.

FilterLab, Hampshire, HI-TECH C, Linear Active Thermistor, MXDEV, MXLAB, SEEVAL and The Embedded Control Solutions Company are registered trademarks of Microchip Technology Incorporated in the U.S.A.

Analog-for-the-Digital Age, Application Maestro, chipKIT, chipKIT logo, CodeGuard, dsPICDEM, dsPICDEM.net, dsPICworks, dsSPEAK, ECAN, ECONOMONITOR, FanSense, HI-TIDE, In-Circuit Serial Programming, ICSP, Mindi, MiWi, MPASM, MPLAB Certified logo, MPLIB, MPLINK, mTouch, Omniscient Code Generation, PICC, PICC-18, PICDEM, PICDEM.net, PICkit, PICtail, REAL ICE, rfLAB, Select Mode, Total Endurance, TSHARC, UniWinDriver, WiperLock and ZENA are trademarks of Microchip Technology Incorporated in the U.S.A. and other countries.

SQTP is a service mark of Microchip Technology Incorporated in the U.S.A.

All other trademarks mentioned herein are property of their respective companies.

© 2011, Microchip Technology Incorporated, Printed in the U.S.A., All Rights Reserved.

Printed on recycled paper.

ISBN: 978-1-61341-685-3

DS25073A-page 49

Microchip received ISO/TS-16949:2009 certification for its worldwide headquarters, design and wafer fabrication facilities in Chandler and Tempe, Arizona; Gresham, Oregon and design centers in California and India. The Company’s quality system processes and procedures are for its PIC® MCUs and dsPIC® DSCs, KEELOQ® code hopping devices, Serial EEPROMs, microperipherals, nonvolatile memory and analog products. In addition, Microchip’s quality system for the design and manufacture of development systems is ISO 9001:2000 certified.


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Documents

MCP6N11 - 500 kHz, 800 uA Instrumentation Amplifier