Continuous phase modulation for broadband satellite communications: design and trade-offs

INTERNATIONAL JOURNAL OF SATELLITE COMMUNICATIONS AND NETWORKINGInt. J. Satell. Commun. Network. (2013)Published online in Wiley Online Library (wileyonlinelibrary.com). DOI: 10.1002/sat.1029

Continuous phase modulation for broadband satellitecommunications: design and trade-offs

B. F. Beidas1, S. Cioni2, U. De Bie3, A. Ginesi4, R. Iyer-Seshadri1, P. Kim5, L. N. Lee1,D. Oh5, A. Noerpel1, M. Papaleo2 and A. Vanelli-Coralli2,*,†

1Hughes Network Systems, Germantown, MD, USA2University of Bologna, Bologna, Italy

3Newtec Technology, Sint-Niklaas, Belgium4European Space Agency, Noordwijk, the Netherlands

5Electronics and Telecommunications Research Institute, Daejeon, South Korea

SUMMARY

In this paper, we introduce the continuous phase modulations mode of the second-generation broadband satellitecommunications return link Digital Video Broadcast Interactive Satellite System standard. Starting from a detaileddiscussion about the impairments faced in a typical broadband satellite system user set-up, we introduce thewaveform selection rationale, the design trade-offs, and the waveform performance under realistic systemassumptions. We show that the designed continuous phase modulations mode exhibits an excellent trade-offbetween performance and robustness to nonlinear distortions at low and medium spectral efficiency thussuggesting its adoption for the consumer market profile of the Digital Video Broadcast Interactive Satellite System.Copyright © 2013 John Wiley & Sons, Ltd.

Received 25 October 2012; Accepted 24 January 2013

KEY WORDS: continuous phase modulation; broadband communications; satellite communications

1. INTRODUCTION

In the definition of the second-generation broadband satellite communications return link Digital VideoBroadcast-Interactive Satellite System (DVB-RCS2) [1, 2] air interface, one of the most criticalchallenges that had to be addressed by the DVB-RCS Technical Module, was the design of a signalwaveform with optimized performance and minimized impact on terminal and system costs. Spectralefficiency and robustness to satellite channel impairments in realistic operating conditions were consid-ered key performance metrics. In particular, a key requirement for the return link design of a broadbandsatellite communication system (i.e., user to satellite) is the necessity of maximizing the waveformresilience to nonlinear distortions, introduced by the high power amplifier at the user terminal thusallowing the terminal Outdoor Unit (ODU) Solid State Power Amplifier (SSPA) to be driven closeto saturation. The possibility of driving the ODU SSPA close to saturation entails in fact not onlyperformance advantages because of the absence of the link budget losses introduced by the power backoff and the nonlinear distortions but also the higher robustness of the end to end performance withrespect to ODU power instabilities. In such a framework, Continuous Phase Modulation (CPM)schemes [3–5] that are known to be resilient to nonlinear distortion become natural candidates forDVB-RCS2 waveform. However, CPM also shows a lower spectral efficiency with respect totraditional linear modulations schemes, for example, phase-shift keying and quadrature amplitude

*Correspondence to: Alessandro Vanelli-Coralli, DEI, University of Bologna, viale Risorgimento, 2, 40136 Bologna (BO), Italy.†E-mail: [email protected]. Cioni is now with the European Space Agency, Noordwijk, the Netherlands.M. Papaleo is now with Qualcomm Inc., San Diego, CA, USA.

Copyright © 2013 John Wiley & Sons, Ltd.

B. F. BEIDAS ET AL.

modulation that are not straightforwardly compensated by the reduced link budget loss. However,when dealing with the return link of a large consumer market, ODU power instabilities become animportant element of the system design that may not have been considered in previous waveformdesign. As we will show in the next section, the ODU power instabilities play a significant role inthe overall system performance justifying the selection of the CPM mode for the consumer profile inthe DVB-RCS2 standard. The rest of the paper is organized as follows: Section 2 discusses theODU instabilities impact and sets the rationale for the selection and design of the DVB-RCS2-CPMmode. Section 3 introduces the CPM mode. Section 4 summarizes the channel impairments used forperformance assessment. Section 5 contains extensive simulations that evaluate the performance ofCPM systems under various scenarios. Finally, Section 6 contains paper conclusions.

2. SYSTEM INSTABILITIES

Figure 1 shows an arrangement of an ODU-SSPA architecture typically used when amplifying linearmodulations. In this figure, the input signal is from an indoor unit and is typically at an intermediatefrequency at L-band. In this architecture, a control loop is implemented around the SSPA in order tostabilize as much as possible the SSPA operating point. Indeed, the stability of the output power ofthe SSPA is affected by both the outdoor environment, for example, aging and temperature changesand the part number production instabilities. The consequences are variations of the SSPA AM/AMand AM/PM characteristics including the 1 dB compression point as well as the maximum power.Another effect to be included is the variation of the SSPA input power because of the different interfacility link cable attenuation at different burst carrier intermediate frequencies especially when theterminal is supposed to hop over a large bandwidth (e.g. 500MHz). In order to compensate for sucheffects, a control loop, as shown in Figure 1, may be built around the SSPA unit within the ODU.In such a scheme, a power detector is used within the feedback loop to estimate the output power ofthe SSPA and compare it with a reference. Any difference of the output power with respect to thereference would generate a signal at the differential amplifier output which would then act on the SSPAdriver so to compensate.

However, although this control loop tends to reduce the SSPA power instabilities, the compensationis not perfect. Variations can also result from instabilities of the detector versus temperature andits power estimation errors which depend on the frequency, modulation, and symbol rate of thetransmitted burst. Overall, a total maximum instability within the range of �1.5 dB can be expectedin the output power. Other schemes for controlling the output power of the terminal SSPA may exist.In particular, the control loop may be implemented with a larger loop with the power estimate of thedetector being sent to the ODU through the inter facility link for adjustment of the driver input power.The DISEqC protocol may be used for this purpose. The output power measurements may be restrictedto quadrature phase-shift keying bursts in order to prevent the sensitivity effect of the power estimatewith respect to the modulation. As a result, this scheme may have less total instability than thearchitecture shown in Figure 1. The SSPA power instabilities give rise to variations of the output power

Figure 1. Typical terminal solid state power amplifier (SSPA) control loop block diagram.

Copyright © 2013 John Wiley & Sons, Ltd. Int. J. Satell. Commun. Network. (2013)DOI: 10.1002/sat

CONTINUOUS PHASE MODULATION FOR BROADBAND SATELLITE COMMUNICATIONS

back off (OBO) of the amplifier, which in the case of the scheme of Figure 1 may be in the order of�1.5 dB (see Figure 2).

In the case of linear modulations, such OBO variations have the consequence of changing boththe adjacent channel interference power as well as the level on nonlinear distortions for the terminalend-to-end link. Several dBs of variation may be expected for the latter depending on the modulationorder and the position of the nominal working point (more than 10 dB are expected for 16-quadratureamplitude modulation in the nonlinear region). Figure 3 shows the dependency versus OBO of thesignal side lobes for an 8-phase shift keying signal. In this example, the model used for the SSPAAM/AM characteristics is the Rapp model [6] with smoothness factor set to three. As shown, for a totalOBO delta of about 2 dB, the side lobes level in the adjacent carrier may vary up to 15 dB.

Thus, if a conservative OBO nominal setting is not used, the impact of a given terminal SSPAinstability over the performance of the other terminals in the network may be significant. The impactof nonlinear distortion on linear modulations when using multicarrier mode operation is investigatedin [7] and [8] where methods of nonlinear compensation using Volterra-based techniques are alsointroduced. CPM signals, having a constant envelope, may operate the SSPA even in the region ofoversaturation, that is, well beyond the 1 dB compression point (although limited by the requirementof not overdriving the SSPA) so that any power instabilities would result in minor variations of theoutput power.

In summary, the CPM waveforms have the advantage of being much more resilient to ODU powerinstabilities and thus do not require operating the SSPA with a conservative nominal power back off.Linear modulations may require a more conservative operating point in order to limit the link

Figure 2. Operating point instabilities shown over the AM/AM characteristics of the solid state poweramplifier (SSPA).

Figure 3. 8-phase shift keying (8PSK) Signal sidelobes level at the output of the solid state power amplifier(SSPA) as function of different output power back off (OBO) values SSPA AM/AM model =Rapp Model 3.


B. F. BEIDAS ET AL.

performance degradation versus time and temperature for the terminal under consideration as well as tolimit the side-lobe interference to other terminals in the network operating on adjacent frequencies. Forlinear modulations, this conservative power back off represents a waste of resources and may requireoversizing the SSPA max power class.

3. SYSTEM MODEL

3.1. Transmitter

The transmitter model is shown in Figure 4. A vector of Nu information bits u= [u0,u1, . . .,uNu� 1] is passedthrough a binary forward error correction (FEC) encoder to produce the codeword b0 = [b00,b01, . . .,bNb� 1],where Nb is a function of the code rate Rc and Nu. For convolutional coded CPM (CC-CPM)systems, nonrecursive convolutional codes of constraint length K= 3 and K=4, with generatorpolynomials (5, 7) octal and (15, 17) octal, respectively, are employed for FEC with higher code ratesobtained by puncturing the rate 1/2 constituent codes [9]. The encoded bit sequence is fed to thebit-interleaver Π with permutation indices Π= [Π(0),Π(1), . . .,Π(Nb� 1)]. The interleaved bit sequenceis given by b ¼ b0Π 0ð Þ; b0Π 1ð Þ; . . . ; b0Π Nb�1ð Þ

� �.

For an alphabet size M, the interleaved bits are mapped to one of the M symbols of the alphabet{�1,� 3, . . .,� (M� 1)} to produce the vector a ¼ a0; a1; . . . ; aNs�1½ �, which represents the symbolsequence to be transmitted and Ns =Nb / log2M. The choice of the bit-to-symbol mapping, either Grayor natural is influenced by the choice of modulation parameters specifically by M and the modulationindex h [10].

The symbol sequence is used to generate the CPM signal phase f(t,a) such that [5]

f t; að Þ ¼ 2phX1i¼0

aiq t � iTsð Þ t≥0 (1)

where the modulation index h =mh/ph is a rational number, Ts is the symbol period, and q(t) is the CPMphase response (phase pulse) selected to ensure that the phase has no discontinuities. More specifically,

q tð Þ ¼

0 t ≤ 0

Z1

0

g tð Þdt 0≤t ≤ LTs

0:5 t ≥ LT

8>>>>>><>>>>>>:

(2)

where g(t) is the frequency pulse shape, nonzero in the interval 0≤ t≤LTs, and L is the width of thepulse shape alternatively known as the memory of the modulation. Although a wide variety of pulseshapes may be used, we focus on choices of g(t) that allow for a good trade-off between energyefficiency and spectral efficiency in a multicarrier environment in which adjacent carrier interference(ACI) can have a significant impact on performance. In particular, it is desirable for the signal powerspectral density (PSD) to have a narrow or compact main lobe allowing adjacent carriers to be spacedcloser and thereby increasing the spectral efficiency. Additionally, it is also advantageous to haverapidly decaying side-lobes to minimize the interference to neighboring carriers. The DVB-RCS2standard includes an innovative method for designing pulse shapes that is based on using a linearcombination of the raised-cosine (RC) and rectangular (REC) pulse shapes well suited for suchbandwidth-constrained scenarios called the weighted average pulse (AV). The averaged frequencypulse shape is

Figure 4. Block diagram of the transmitter.



gAV tð Þ ¼ aRCgRC tð Þ þ 1� aRCð ÞgREC tð Þ (3)

where 0≤ aRC≤ 1, gRC(t) and gREC(t) are the RC and REC pulse shapes, respectively [5]:

gRC tð Þ ¼1

2LTs1� cos

2ptLTs

� �0≤ t ≤ LTs

0 otherwise

8><>: (4)

gREC tð Þ ¼1

2LTs0≤ t ≤ LTs

0 otherwise

8<: (5)

As an example, Figure 5 shows the phase response qAV(t) obtained using gAV(t) with aRC = 0.75along with the phase response functions for gRC(t) and gREC(t) pulse shapes.

The CPM signals that use this new family of pulses can also be decomposed using the Laurentrepresentation [11, 12]. This Laurent decomposition is useful such that by retaining only the principalpulses at the receiver, complexity of the receiver can be reduced. These functions can be obtained byapplying the same linear combination that produce the phase response pulse qAV(t) but on the Laurentfunctions associated with the individual pulses gREC(t) and gRC(t). As an example, Figure 6 displays

Figure 5. Phase response for raised-cosine (RC), rectangular (REC), and average (AV), pulse shapes.

Figure 6. Laurent functions associated with proposed continuous phase modulation (CPM) phase pulse qAV(t).


B. F. BEIDAS ET AL.

the Laurent functions for M = 4, h = 1/4, and L= 2 for gRC(t), gREC(t), and gAV(t), with aRC = 0.75,pulses. It can be seen that the first (M-1) components are the principal components that dominate therepresentation. This will be utilized in Section 5 to reduce matched filtering operations and moreimportantly the number of trellis states used at the receiver.

Figure 7 shows the signal PSD for the same pulse shapes. It is observed that the CPM signal withthe weighted average pulse shape exhibits a more compact PSD compared with CPM using RC pulseshape and smaller side-lobes relative to CPM using REC pulse shape.

The baseband CPM signal can now be written as:

s t; að Þ ¼ffiffiffiffiffiffiffi2Es

Ts

re jf t;að Þ (6)

where Es is the energy per symbol.The modulated signal is transmitted through an AWGN channel. A multicarrier FDM-CPM system

is assumed in which the modulated carrier shares the channel with N-1 independent interfering carrierssuch that the composite signal at the output of the channel can be expressed as:

r tð Þ ¼XN�1ð Þ=2

k¼� N�1ð Þ=2s kð Þ t � xkTsa

kð Þ� �

exp j2p fk t � xkTsð Þf g þ n tð Þ (7)

where a(k) and s(k) denote the symbol sequence and baseband signal corresponding to user k,respectively, 0≤ xk≤ 1 represents the relative time offset of the kth user, fk is the corresponding centerfrequency, and n(t) is zero-mean AWGN with single-sided PSD of N0 [Watt/Hz].

3.2. Receiver

The block diagram of the receiver is shown in Figure 8. Single-user detection is implied such that noinformation is exchanged with the receiver of adjacent carriers. The CPM demodulator composed of amatched filter/ correlator bank formulated either on the basis of Rimoldi’s decomposition approach[13] or the Laurent representation of CPM signals [11, 12] followed by the soft-in soft-out (SISO)CPM detector. The SISO-CPM detector generates the maximum a posteriori probabilities for thetransmitted codebits using a trellis [13] or factor graph [14] describing the modulation. Theconvolutional decoder also performs SISO detection by executing the Bahl, Cocke, Jelinek and Raviv(BCJR) algorithm [15] on a 4-state trellis for the constraint length 3 code and an 8-state trellis for theconstraint length 4 code. A maximum of 30 iterations are performed between the SISO-CPM detectorand the SISO convolutional decoder during which extrinsic information is exchanged between them.The decoder then generates hard decisions on the information bits. At higher spectral efficiencies, alow-pass filter can be applied at the receiver front end to mitigate the ACI.

Figure 7. Power spectral density for continuous phase modulation (CPM) signals with M= 4, h= 1/4, and L= 2with raised-cosine (RC), rectangular (REC), and average (AV) aRC = 0.75 pulse shapes.


Figure 8. Block diagram of the receiver. CPM, continuous phase modulation; FEC, forward error correction.


3.3. Configurations at different spectral efficiencies

As in [16], the spectral efficiency in bit/s/Hz is defined as:

� ¼ Rclog2MΔf Ts

(8)

where Δf is the difference between the center frequencies of two adjacent carriers.In multicarrier, coded CPM systems, the energy and bandwidth efficiency is influenced by the

choice of M, h, pulse shape, memory, the FEC code and its rate, and the adjacent carrier spacing[16, 17]. As an example, increasing h, while keeping the remaining parameters the same, typicallyresults in increased energy efficiency, but at the expense of bandwidth. Similarly, increasing Mincreases the available bits per symbol but also increases the spectral occupancy. Lowering the coderate alone could increase the coding gain, but this is accompanied by a loss in the spectral efficiency.The spectral efficiency can also be increased by reducing the spacing between adjacent carriers; thishowever entails an increase in the ACI. Finally, the complexity of the CPM detector also increases withincreasing L and M. The combination of code and modulation parameters for CC-CPM at differentspectral efficiencies have been selected considering the above trade-offs and are listed in Table I.

4. EVALUATION SCENARIO

For the final acceptance in the new DVB-RCS2 standard, the presented air interface has been evaluatedfollowing an incremental approach: starting from the simple AWGN channel up to a realistic satellitechannel including nonlinear effects, adjacent channel interference, phase noise, and frequency errors.In other words, to quantify the impacts of these channel impairments, each of them has been addedone at a time during the evaluation process for the selection of the best waveform candidate. A subsetof all performed simulation results is discussed in the later text. Hereafter, each component and modelof the complete evaluation channel is characterized.

To aid with the frequency estimation, a pair of unique words is inserted each containing a group of32 symbols with a separation of 32 symbols as described in [2].

Table I. Convolutional coded-continuous phase modulation configurations.

b/s/Hz M h L aRC Bit-to-Symbol K Rc ΔfTs

0.5 4 2/5 2 0.98 Gray (5, 7) 1/2 2.00.75 4 1/3 2 0.75 Natural (5, 7) 1/2 1.33331.1 4 2/7 2 0.75 Gray (5, 7) 2/3 1.211.25 4 2/7 2 0.75 Gray (5, 7) 2/3 1.0671.5 4 1/4 2 0.75 Gray (15, 17) 4/5 1.06671.8 4 1/5 2 0.625 Gray (15, 17) 6/7 0.974


B. F. BEIDAS ET AL.

4.1. Adjacent channel interference

In the satellite return link, ACI plays an important role on the final physical layer and systemperformance. The main reasons for increasing the ACI contribution are: nonlinear effects (via side-lobesspectral regrowth) and noncompensated fading events.

Fortunately, the first impairment can be neglected in case of CPM waveforms because of theirinherent constant envelope property. On the other hand, nothing can be performed to compensatefor different fading conditions experienced by all users in the return link with the consequent powerunbalance among adjacent carriers.

Therefore, a maximum power unbalance of +3 dB (i.e., the reference carrier is surrounded by userswith stronger received power) is considered in the following analysis. A total of four interferingcarriers (two on each side of the reference user) have been considered in the performance simulationswith variable carrier spacing and assuming they all use the same code and modulation parameters.

4.2. Phase noise

The carrier phase noise has been simulated using the mask suggested in the DVB-RCS2 call fortechnologies [18] that it is also reported in Table II.

4.3. Carrier frequency instability

To correctly evaluate the CPM synchronization and detection performance, the residual carrierfrequency error after the coarse tracking loop has been considered in the final results. The suggestedvalue has been � 4 [KHz] with a uniform distribution.

4.4. Nonlinear distortion

Finally, although the proposed technique is resilient to nonlinear distortion, to perform a thoroughassessment nonlinear distortion effects introduced by the user terminal high power amplifier have beenalso implemented in the simulation set up through the Rapp model [6].

5.. PERFORMANCE

Simulations were conducted to evaluate various aspects of the coded CPM system. The effect ofmodulation index and pulse shape on performance is illustrated in Figure 9 which shows the packeterror rate as a function of the per-information bit signal-to-noise ratio (Eb /N0) in an AWGN channelwithout ACI. The CC-CPM parameters corresponding to 0.5 [b/s/Hz] and 0.75 [b/s/Hz] are consideredat an information block length of 1504 bits. It is seen that for the same M, L, code rate, and constraintlength, the CPM configuration with h = 2/5 and aRC = 0.98 performs better than that with h= 1/3 andaRC = 0.75. However, this performance gain comes at the expense of a larger signal bandwidth whichis why the choice of h= 1/3 and aRC = 0.75 yields a higher spectral efficiency by permitting a tightercarrier packing (1.333/Ts as compared with 2.0/Ts).

The impact of ACI on performance is illustrated in Figure 10 which shows the CC-CPM configu-rations corresponding to 1.1 [b/s/Hz] and 1.25 [b/s/Hz] with and without ACI. ACI is introduced byplacing adjacent carriers 1.21/Ts apart (at 1.1 [b/s/Hz]) and it results in approximately 0.2 dB

Table II. Carrier phase noise masks.

Frequency offset Single-sideband (SSB) Phase Noise

10Hz �16 dBc/Hz100Hz �54 dBc/Hz1 kHz �64 dBc/Hz10 kHz �74 dBc/Hz100 kHz �89 dBc/Hz>1MHz �106 dBc/Hz


Figure 9. Packet error rate (PER) of rate 1/2 convolutional coded-continuous phase modulation (CC-CPM) withM= 4, h= 2/5, L= 2, average (AV) (aRC = 0.98) and M= 4, h= 1/3, L= 2, AV (aRC = 0.75) in AWGN. The

information block-length is 1504 bits. RC, raised-cosine.

Figure 10. Packet error rate (PER) of rate 2/3 convolutional coded-continuous phase modulation (CC-CPM) withM= 4, h= 2/7, L= 2, average (AV) (aRC = 0.75) in AWGN and AWGN+ACI. The information block length is

1504 bits. ACI, adjacent carrier interference.


performance penalty relative to the no ACI case. The degradation is increased by about 0.75 dB whenthe carrier spacing is further reduced to 1.067/Ts at 1.25 [b/s/Hz].

Finally, Figure 11 shows packet error rate for CC-CPM at the different spectral efficiencies listed inTable I for information block lengths of 400 and 1504 bits in the presence of ACI. Attaining higherspectral efficiencies necessitates reducing the modulation index, reducing the carrier spacing, and usingweaker codes, thereby resulting in an increased Eb/N0 requirement as indicated in the figure. It is notedthat in generating the curves reported in Figures 9–11, the CPM matched filter/correlator band and theSISO detector are formulated according to the Rimoldi’s representation [13]. The bit interleaver forCC-CPM described in [2] has been applied, although comparable results were also obtainedusing an S-random interleaver [19] with a suitable spread value (e.g. S≥ 14). Additionally, perfectsynchronization is assumed at the receiver.

The performance of CC-CPM in the presence of phase noise, frequency error, and ACI has alsobeen evaluated and some selected results are reported in Figure 12. The spectral efficiency is0.5 [bit/s/Hz] and the information block length is 400 bits. The phase noise mask is the one specifiedin Table II [16], the � 4 [KHz] frequency error is applied to the center carrier, and a bandwidth of512 [KHz] is assumed. To handle the significant impairments introduced, we rely on the differentiallycoherent CPM detector from [14]. The reason for this is the large frequency uncertainty that is left inthe system even after the unique words are used for frequency estimation. This receiver is based on theLaurent representation and is also robust to phase noise impairment. Results indicate that performancedifference relative to perfect synchronization is within 0.25 dB. This gap in performance was alsoobserved at the high spectral efficiency level of 1.8 [b/s/Hz].


Figure 11. Packet error rate (PER) performance at different spectral efficiencies for convolutional coded-continuousphase modulation (CC-CPM) in AWGN+ACI. ACI, adjacent carrier interference.

Figure 12. Packet error rate (PER) performance at 0.5 [b/s/Hz] convolutional coded-continuous phase modulation(CC-CPM) with and without phase noise and frequency errors in the presence of adjacent carrier interference

(ACI). ACI is from four adjacent carriers, 3 dB stronger than center carrier. Information block length is 400 bits.The bandwidth is 512 [KHz]. PN, pseudo noise; FE, front end.

B. F. BEIDAS ET AL.

6. CONCLUSIONS

In the framework of the DVB-RCS2 standardization, CPM schemes have been thoroughly analyzedand their benefit in terms of resilience to satellite channel impairments assessed in order to evaluateCPM suitability for the return link of a broadband satellite communications system. The proposedCC-CPM scheme strikes a very good trade-off between performance and robustness in the low tomedium range of spectral efficiencies, and it is therefore an excellent solution for the DVB-RCS2system. For this reason, the proposed CC-CPM solution has been included in the DVB-RCS2specifications as an alternative mode to the linear modulation schemes.

Notably, the comparative analysis performed between linear and continuous phase modulationsrequired a deep understanding of the behavior of the user terminal architecture under real workingconditions. The analysis argues that the large variation in the output power back-off should be consid-ered in the design of a satellite communication system. This is particularly true to avoid that a designbased only on theoretical elements brings the system to work in unacceptable interference anddistortion conditions when a large number of users are considered, as for example, in the consumerbroadband market.

Finally, after the inclusion in the DVB-RCS2 specifications, the proposed CPM mode has beenassessed for use in satellite mesh networks [20–23]. Although the proposed CPM scheme may requiremore complexity at the receiver side, the benefits deriving from the use of a nonlinear distortionresilient waveform in a mesh system with a large number of terminals, along with the rapid decreasingcost of baseband computational resources with respect to radio-frequency front-end, make CPM an at-tractive solution for the mesh scenarios as well as hub-spoke scenarios.



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AUTHOR’S BIOGRAPHIES

Copyright © 2013 John Wile

Bassel F. Beidas received his master’s degree (with honors) from the California Institute ofTechnology, Pasadena, and his PhD degree from the University of Southern California, LosAngeles, both in electrical engineering. Dr. Beidas is currently Advisory Engineer with theAdvanced Development Group at Hughes, Germantown,MD. He is responsible for researchand development in advanced transmission technologies, which have been successfullyincorporated into several premier product lines in cellular and satellite communications.Previously, he held the position of Principal Engineer at Corvis Corporation, Columbia,MD, where he developed innovative signal processing algorithms for 40 Gbps ultra long-haul optical communications. His general research interests lie in the areas of signal classifi-cation, interference cancellation, adaptive signal processing, synchronization, and nonlinearsystems. He holds fifteen US patents on digital communications techniques and has severalpatents pending. Dr. Beidas is a member of Phi Kappa Phi, Eta Kappa Nu, Tau Beta Pi, and

his biography appears in The National Dean’s List. In addition, he received numerous awards fromHughes including the1997 Outstanding Achievement Award, the 1999 Special Award for Exceptional Contributions to Third-GenerationWireless Technology, the 2008 Engineering Excellence Award for Significant Contributions to Advanced TechnologyDevelopment, and the 2012 Certificate of Achievement for Excellence in System Transmission and Satellite Design.

y & Sons, Ltd. Int. J. Satell. Commun. Network. (2013)DOI: 10.1002/sat

B. F. BEIDAS ET AL.


Stefano Cioni received his Dr.Ing. degree in telecommunication engineering and PhDfrom University of Bologna, Italy, in 1998 and in 2002, respectively. In 2010, he joinedthe European Space Agency (Noordwijk), where he is currently a CommunicationsSystems Engineer within the Radio Frequency Systems, Payload, and Technology Di-
vision. Since 2002, he has been a Senior Researcher of the Advanced Research Centerfor Electronic Systems (ARCES) of the University of Bologna. During the summer of2006, he was a visiting researcher at the Agilent Labs SMRD, Belgium. During thesummer of 2007, he was a visiting researcher at the German Aerospace Center(DLR), Oberpfaffenhofen (Germany). From 2008 to 2010, he has been the Head ofDigital Transmission Systems in Mavigex S.r.l. (Italy). His research activities aremainly focused on the next generation wireless telecommunication systems, both theterrestrial and the satellite networks. In particular, his interests include synchronization
techniques, medium access control resource allocation algorithms, OFDM systems, and iterative decoding tech-niques joint to channel parameter estimation. Dr. Cioni co-authored more than 70 papers and scientific conferencecontributions and he is co-recipient of the Best Paper Award at IEEE ICT 2001 and at IEEE ASMS/SPSC 2012.

Ulrik De Bie entered the world of satellite telecommunication when he joined Newtec cy, Belgium as a systemengineer in 2001/2002. His research interests include DVB-RCS and DVB-RCS2 satellite scheduling and encap-sulation in the return link, ACM scheduling and encapsulation in DVB-S/S2 forward link, and control plane func-tions in satellite internet access systems. He is an active participant in the specification and engineering of satellitetelecommunication standards. He participates in DVB TM, and the ad hoc working groups TM-S2 (Wideband,DVB-CID, extensions), TM-GBS (GSE), and TM-RCS (RCS and RCS2). He received the Licentiaat Informaticadegree from the Katholieke Universiteit Leuven, Belgium, in 1998.

A. Ginesi was born in Parma, Italy, in November 1967. He received his Dr. Ing. (cumlaude) and PhD degrees in electronic engineering from University of Pisa, Italy, in 1993and 1998, respectively. In 1996–1997, he spent one year at Carleton University, Ot-tawa, Canada, doing research on digital transmissions for wireless applications. In1997, he joined Nortel Networks and in 2000 Catena Networks, both in Ottawa, Can-ada, where he worked on Digital Subscriber Loop (DSL) technologies and contributedto the definition of the second-generation ADSL standard. Since 2002, he joined ESAResearch and Technology Centre (ESTEC), Noordwijk, The Netherlands, where he iscurrently covering the position of the Head of the Communication-TT&C Systems &Techniques Section. His main current research interests lie in the area of advanced dig-ital satellite communication systems and techniques from theory to HW implementa-tion.

Rohit Iyer Seshadri received his master’s degree from Pennsylvania State University,University Park, PA in 2003 and his PhD degree from West Virginia University, Mor-gantown, WV in 2007, both in electrical engineering. Dr. Iyer Seshadri is currently aSenior Member of Technical Staff in the Advanced Development Group at Hughes,Germantown, MD. His current research interests lie in the areas of communication the-ory, applied information theory, wireless communication systems, and error correctioncoding. He holds two US patents and has three patents pending. He is the recipient ofthe 2010 Hughes Engineering Excellence Award for Significant Contributions to Ad-vanced Technology Development.




Pansoo Kim received his BS and MS degrees in the ECE from Sung Kyun Kwan Uni-versity, Korea in 2000 and 2002, respectively. Since 2002, he has worked for ETRI
(Electronic and Telecommunication Research Institute), Deajeon, Korea, as a seniormember of engineering staff in the field of system implementation of broadband satel-lite communication and broadcasting system. His main research interests are in digitalsatellite broadcasting/communication system and modem technology with associatedequalization, synchronization and iterative decoder design, and VLSI implementation
Lin-Nan Lee is Vice President, Advance Development at Hughes. He leads researchand technology development at Hughes, has made many significant contributions tothe design and development of Hughes satellite and wireless communications productsand technology. He has participated in several key industry standards activities, includ-ing turbo codes standardization in the 3GPP and 3GPP2 third generation wireless stan-dards and the low-density parity check (LDPC) codes for the second generation DigitalVideo Broadcast Satellite broadcast standard (DVB-S2), and the IEEE802.11.n stan-dards. Before joining Hughes, he worked for Communications Satellite Corporation(COMSAT), serving in various research and development positions in the COMSATLaboratory, and as Chief Scientist of COMSAT System Division. Prior to COMSAT,he was with the Linkabit Corporation, where he co-developed the world’s firstpacket-based satellite multiple access protocol with quality of service (QoS) provisions.

Lin-Nan received his BS degree from National Taiwan University, his MS and PhD from the University of NorteDame. He is a Fellow of IEEE.

Deock-Gil Oh received his BS, MS, and PhD degrees in electronics engineering fromSeoul National University, Korea in 1980, 1984, and 1996, respectively. Since 1982, hehas been with the department of satellite and wireless convergence technology,ETRI,Daejeon, Korea, where he is currently a principal member of Research Staff in theCommunication Satellite Center. His research interests are in digital satellite communi-cation/broadcasting systems and wireless communication systems

Anthony Noerpel received his Bachelor of Science degree from Rutgers University inMathematics and his Masters of Science degree in Electrical Engineering from NewJersey Institute of Technology. He worked for Bell Telephone laboratories from1977 until 1984 and for Bellcore from 1984 until 1995. In 1995, he joined the Ad-vanced Development Group of Hughes Network Systems in Germantown Maryland.Mr. Noerpel’s research interests have included antenna design, radiowave propagation,point-to-point microwave radio, personal communications, and satellite communica-tions. He holds thirty-two patents and has published several journal articles, conferencepapers, and book chapters. He is currently chairman of Telecommunications IndustryAssociation TR 34 Satellite Equipment & Systems Engineering Committee and a mem-ber of Sigma Xi.


B. F. BEIDAS ET AL.


Marco Papaleo was born in Soverato, Italy, in 1981. He received his BS and MS de-grees (summa cum laude) in Telecommunication Engineering from University of Bolo-gna, Italy, in 2003 and in 2006, respectively. In 2010, he received his PhD inInformation Technology from the Advanced Research Center on Electronic Systems
for Information and Communication Technologies "Ercole De Castro" (ARCES) atthe University of Bologna. In 2006, he was a visiting affiliate student at the UniversityCollege of London (UCL). In summer 2008, he was a visiting PhD student at the Insti-tute of Communications and Navigation of the German Aerospace Center (DLR), inMunich, Germany, where he was involved in the design and analysis of LDPCconvolutional codes. In 2009, he was a visiting graduate student at the Center for Mag-netic Recording Research of the University of California, San Diego (UCSD). SinceJanuary 2011, he joined the Corporate Research and Development department at
Qualcomm Technology Inc., San Diego, California, where is currently involved in the design of multimode mobilestation modems. His research activities are mainly focused on the next generation wireless telecommunication sys-tems, both the terrestrial and the satellite networks. In particular, his interests include design of power efficient mo-dems for top tier tablets and smartphones

Alessandro Vanelli-Coralli received his Dr. Ing. degree (cum laude) in ElectronicsEngineering and his PhD in Electronics and Computer Science from the Universityof Bologna (Italy) in 1991 and 1996, respectively. In 1996, he joined the Universityof Bologna, where he is currently an Associate Professor at the Department of Electri-cal, Electronic, and Information Engineering "Guglielmo Marconi" (DEI). Since 2001,he has also been with the Advanced Research Center for Electronic Systems (ARCES)of the University of Bologna. During 2003 and 2005, he was a Visiting Scientist atQualcomm Inc. (San Diego, CA), working in the Corporate R&D Department onMobile Communication Systems. Dr. Vanelli-Coralli participates in national and inter-national research projects on satellite mobile communication systems. He is a co-Leader of the R&D group of the Integral SatCom Initiative (ISI) technology platform,and Scientific Responsible for several European Space Agency and European Commis-

sion funded projects. His research interests are in the area of wireless communication systems, digital transmissiontechniques, and digital signal processing. Dr. Vanelli-Coralli has been appointed member of the Editorial Board ofthe Wiley InterScience Journal on Satellite Communications and Networks and has been guest co-editor for sev-eral special issues of international scientific journals. Dr. Vanelli-Coralli has served as General Chairman andTechnical Chairman of several scientific conferences. Dr. Vanelli-Coralli co-authored more than 140 papers andscientific conference contributions and he is co-recipient of several Best Paper Awards. He is an IEEE SeniorMember.


Documents

Continuous phase modulation for broadband satellite communications: design and trade-offs