BUET - Semantic Scholar · (BUET) and Dr. Md. Ziaur Rahman Khan,professor, Department of Electrical and Electronic Engineering, Bangladesh University of Engineering and Technology

1

DEVELOPMENT OF A TWO QUADRANT BUCK BOOST DC-DC

CONVERTER WITH CONTINUOUS INPUT CURRENT

By

SADI MD. SHIHAB

MASTER OF SCIENCE IN ELECTRICAL AND ELECTRONIC ENGINEERING

BUET

DEPARTMENT OF ELECTRICAL AND ELECTRONIC ENGINEERING BANGLADESH UNIVERSITY OF ENGINEERING AND TECHNOLOGY (BUET)

JULY2016

2

DEVELOPMENT OF A TWO QUADRANT BUCK BOOST DC-DC

CONVERTER WITH CONTINUOUS INPUT CURRENT

By

SADI MD. SHIHAB

A thesis submitted to the

Department of Electrical and Electronic Engineering in partial fulfillment for the degree of

Master of Science in Electrical and Electronic Engineering

DEPARTMENT OF ELECTRICAL AND ELECTRONIC ENGINEERING

3

4

Candidate’s Declaration

I hereby declare that this thesis has been prepared in partial fulfillment of the

requirement for the degree of Master of Science in Electrical and Electronic

Engineering at the Bangladesh University of Engineering and Technology

(BUET),Dhaka and has not been submitted anywhere else for any other degree.

Signature of Candidate

Sadi Md. Shihab Student No: 0411062104 EEE,BUET,Dhaka

5

Dedicated

to

My parents

6

Acknowledgement

All praises goes to Almighty for blessing me with the knowledge and ability to do the

present study. My indebt gratitude must be to the most benevolent and the most

merciful for everything what I have received from him.

It is the my pleasure to acknowledge my gratitude to my supervisor Dr. Mohammad

Ali Choudhury, Professor, Department of Electrical and Electronic Engineering,

Bangladesh University of Engineering and Technology (BUET), Dhaka, for his

continuous guidance, kind cooperation, valuable suggestions, and encouragement at

all stages of the study.

I would like to express my sincere thanks and regards to all of my faculty members of

the department especially the thesis examination committee members Professor Dr.

Quazi Deen Mohd. Khosru,Professor and Head of the Department of Electrical and

Electronic Engineering, Bangladesh University of Engineering and Technology

(BUET) and Dr. Md. Ziaur Rahman Khan,professor, Department of Electrical and

Electronic Engineering, Bangladesh University of Engineering and Technology

(BUET), Dhaka. For critically reviewing the manuscript and for valuable suggestions

for improvement of thesis.

I wish to convey my sincere thanks to all of my well-wishers for their constant

encouragement, sympathetic co-operation and mental support as well as backing at all

stages of my thesis work. Heartfelt appreciation goes to my family. My family is the

crypt of my all muse, ethics and values. My little effort to this study is just a reflection

of that.

Finally, I express my thanks to the librarian and all staffs of the Department of

Electrical and Electronic Engineering, BUET, for their cordial help and assistance.

Sadi Md. Shihab

7

July, 2016

Abstract

The Buck-Boost dc-dc converter has a series switch with the source and a series diode

with the load. The series switch makes the input current discontinuous. Filtering out

the high frequency switching component from the discontinuous input current of the

buck boost dc-dc converter is difficult. In a recent research the one quadrant buck-

boost has been modified with additional diode-capacitors to have continuous input

current which is easy to filter. There are applications where two quadrant dc-dc

converters with boost-buck voltage gain/attenuation are necessary. DC motors and

battery chargers are the examples of such applications. In this thesis the modified one

quadrant buck-boost dc-dc converter has further been topologically changed to

operate the converter in two quadrant mode and yet maintain the continuous input

current during whichever side is being used as the input of the converter. The

proposed two quadrant buck-boost dc-dc converter with continuous input current has

been investigated with R, R-L and R-L-Emf loads by interchanging the positions of

source and load of the converter. The simulation results indicate the proposed buck-

boost converter performs two quadrant operations (i.e.-ve voltage - ve current and -ve

voltage +ve current with the change of duty cycle of control pulse of the switches of

the dc-dc converter). In the proposed two quadrant buck-boost dc-dc converter

additional switch, diodes, capacitors and inductors are used as per requirement.

Number of switches used in the proposed circuit is two and the gate pulses provided

to the switches are compliment of each other.

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1 TABLE OF CONTENT

Page No.

Signature of Examiners iii

Declarations iv

Acknowledgement vi

Abstract vii

Chapter 1: Introduction 1

1.1 Buck Boost DC-DC converter with continuous input current 2

1.2Specific aims and possible outcomes 8

1.3 Thesis Outline 10

Chapter 2: Classifications of DC-DC Choppers 11

2.1 Types of DC-DC Converter 11

2.1.1 BUCK Converter 11

2.1.2 BOOST Converter 16

2.1.3 BUCK-BOOST Converter 21

2.1.4 Ĉuk converter 26

2.1.5 SEPIC Converter 28

2.2 DC Choppers Quadrant Operation 34

2.2.1 Multiple Quadrant Operation 34

2.2.2 The One Quadrant Chopper 36

2.2.3 The Two Quadrant Chopper 37

2.2.4 The Three Quadrant Chopper 38

2.2.5 The Four Quadrant Chopper 38

2.2.6 The One and Two Quadrant Chopper 39

2.2.7 The Three and Four Quadrant Chopper 40

2.2.8 The Combined (Four) Quadrant Chopper 41

9

Chapter 3: Two quadrant Buck Boost DC-DC converter with continuous

input current

3.1 Introduction 45

3.1.1 Conventional DC-DC Buck-Boost Converter 45

3.1.2 Typical Simulation Results of conventional one Quadrant Buck 47

-Boost DC-DC Converter

3.1.3 Modified DC-DC Buck Boost Converter with Continuous input51

Current

3.1.4 Voltage Gain Expression of modified Buck-Boost DC-DC 53

Converter

3.1.5 Typical Simulated results of Modified Continuous input current 56

Buck-Boost DC-DC Converter

3.1.6 Two quadrants Buck Boost DC-DC converter with continuous 60

current

3.1.7 Ideal Voltage Gain expressions of Two Quadrant Buck-Boost 62

DC-DC converter with continuous Input Current

3.1.8 Typical Simulation Results of Proposed Modified Two quadrants 65

Buck Boost DC-DC converter with continuous current

3.1.8.1 Proposed Buck-Boost DC-DC converter with R-Load 66

3.1.8.2 Proposed Buck-Boost DC-DC converter with R-Load 75

(Source and load interchanged in position)

3.1.8.3 Proposed Buck-Boost DC-DC converter with R-L Load 84

3.1.8.4 Proposed Buck-Boost DC-DC converter with R-L Load 93

(Source and load interchanged in position)

3.1.8.5 Proposed Buck-Boost DC-DC converter with R-L-Emf102

Load

Chapter 4: Conclusions 111

4.1 Findings, achievements 111

4.2 Conclusion 113

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4.3 Future Scope of Works 114

References 115

LIST OF TABLES Page No

Table 1.1 The Switches and Diode‟s Status of Four – Quadrant 42

Operation

Table 3.1 Operational output of Two Quadrant Chopper mode of 112

forward and reverse Converter of circuit of Figure 2.28





11

LIST OF FIGURES Figure 1.1 DC-DC Converter Family Tree 2

Figure 1.2 Basic DC-DC converter 3

Figure 1.3 DC-DC converter voltage waveform. 4

Figure 1.4 Pulse width modulation concept 4

Figure 1.5 Block Diagram of SMPS 6

Figure 1.6 Linear (dissipative) power conversion circuit. 6

Figure 1.7 Switch mode (non dissipative) power conversion circuit. 7

Figure 1.8 Typical switch mode power conversion circuit. 8

Figure 1.9 a) Buck dc-dc converter; (b) Equivalent circuit for the 12

switch closed; (c) Equivalent circuit for the switch open.

Figure 1.10 Buck converter waveforms: (a) Inductor voltage;(b) Inductor 13

current; c) Capacitor current.

Figure 1.11 a) Boost dc-dc converter; (b) Equivalent circuit for the 17

switch closed;(c) Equivalent circuit for the switch open.

Figure 1.12 Boost converter waveforms: (a) Inductor voltage; (b) 18

Inductorcurrent; (c) Diode Current (d) Capacitor Current.

Figure 1.13 a) Buck Boost dc-dc converter; (b) Equivalent circuit for 22

the switch closed;(c) Equivalent circuit for the switch open.

Figure 1.14 Buck Boost converter waveforms: (a) Inductor Current; 23

(b) Inductor voltage; c) Diode Current (d) Capacitor Current

Figure 1.15 The Ĉuk converter (a) Circuit; b) Equivalent circuit for the 26

switch closed; (c) Equivalent circuit for the switch open;

(d) Current in L1 for a large inductance.

Figure 1.16 (a) SEPIC Circuit; b) Circuit with the switch closed and 29

the diode off; (c) Circuit with the switch open and the diode on.

Figure 1.17 Currents in SEPIC Converter(a) L1; b) L2; (c) C1; (d) C2; 34

(e) switch; (f) diode

12

Figure 1.18 Four Quadrant Operation 35

Figure 1.19 The Quadrant One Chopper 36

Figure 1.20 The Quadrant Two Chopper. 37

Figure 1.21 The Quadrant Three Chopper. 38

Figure 1.22 The Quadrant Four Chopper. 39

Figure 1.23 The One and Two Quadrant Chopper Circuit Diagram. 40

Figure 1.24 The Three and Four Quadrant Chopper Circuit Diagram. 40

Figure 1.25 The Four Quadrant Chopper Circuit Diagram 41

Figure 1.26 Fundamental four quadrant chopper (center) showing 43

deviations of four subclass DC choppers, (a). First

quadrant choppers-I, (b). Second quadrant choppers-II,

(c). First and second quadrant choppers-I & II, (d). First

and Fourth quadrant choppers – I & IV, and

(e). Four quadrant choppers.

Figure 2.1 Conventional DC-DC Buck- Boost converter 45

Figure 2.2 Conventional DC- DC Buck Boost converter when the 46

switch is on

Figure 2.3 Conventional DC- DC Buck Boost converter when 47

the switch is off

Figure 2.4 Average Input voltage wave shape of conventional 47

DC- DC BuckBoost converter (Duty Cycle = 0.4)

Figure 2.5 Average Output voltage wave shape of conventional 48

DC- DC BuckBoostConverter (Duty Cycle =0.4)

Figure 2.6 Average Input Current wave shape of conventional 48

DC- DC BuckBoostConverter (Duty Cycle = 0.4)

Figure 2.7 Average Output Current wave shape of conventional 49

DC- DCBuck Boost converter(Duty Cycle = 0.4)

Figure 2.8 Average Input voltage wave shape of conventional 49

DC- DC Buck BoostConverter (Duty Cycle = 0.8)

Figure 2.9 Average Output voltage wave shape of conventional 50


Figure 2.10 Average Input Current wave shape of conventional 50

13


Figure 2.11 Average Output Current wave shape of conventional 51

DC- DC Buck Boost converter (Duty Cycle = 0.8)

Figure 2.12 Modified DC- DC Buck Boost converter 51

Figure 2.13 Modified DC- DC Buck Boost converter when the 52

switch is ON and capacitor charging


switch is ON


switch is OFF

Figure 2.16 Input voltage wave shape of Modified DC- DC 56

Buck Boost converter(Duty Cycle = 0.4)

Figure 2.17 Output voltage wave shape of Modified DC- DC 56

Buck Boost converter (Duty Cycle = 0.4)

Figure 2.18 Input Current wave shape of Modified DC- DC 57


Figure 2.19 Inductor Current wave shape of Modified DC- DC 57

Buck Boostconverter (Duty Cycle = 0.4)

Figure 2.20 Input voltage wave shape of Modified DC- DC 58


Figure 2.21 Output voltage wave shape of Modified DC- DC 58


Figure 2.22 Input Current wave shape of Modified DC- DC 59


Figure 2.23 Inductor Current wave shape of Modified DC- DC 59


Figure 2.24 Two quadrants Buck Boost DC-DC converter with 60

Continuous Input Current

Figure 2.25 Two Quadrant DC- DC Buck Boost converterwhen 61

the switch S1 is ON,S2 is OFF andcapacitor charging

Figure 2.26 Two Quadrant DC- DC Buck Boost converter when the 62

switch S1 is OFF,S2 is ON when Duty Cycle < 0.5

14

Figure 2.27 Two Quadrant DC- DC Buck Boost converterwhen the 62

switch S1 is OFF, S2is ON when Duty Cycle > 0.5

Figure 2.28 Simplified Two Quadrant DC- DC Buck Boost converter 63

when theswitchS1 is ON, S2 is OFF

Figure 2.29 Simplified Two Quadrant DC- DC Buck Boost converter 63

when the switch S1 is OFF, S2 is ON

Figure 2.30 Two quadrant DC- DC Buck Boost converter with Resistive 66

Load

Figure 2.31 Input Voltage wave shape of Two quadrant DC- DC 67


Figure 2.32 Average Input Voltage wave shape of Two quadrant 67


Figure 2.33 Input Current wave shape of Two quadrant DC- DC 68


Figure 2.34 Average Input Current wave shape of Two quadrant 68


Figure 2.35 Output Voltage wave shape of Two quadrant DC- DC 69


Figure 2.36 Average Output Voltage wave shape of Two quadrant 69


Figure 2.37 Output Current wave shape of Two quadrant DC- DC 70


Figure 2.38 Average Output Current wave shape of Two quadrant 70





DC- DC Buck Boost Converter (Duty Cycle = 0.8)

Figure 2.41 Input current wave shape of Two quadrant DC- DC 72




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Figure 2.43 Output Voltage wave shape of two quadrant DC- DC 73





Buck Boostconverter (Duty Cycle =0.8)


DC- DC Buck Boost converter (Duty Cycle =0.8)

Figure 2.47 Two quadrant DC- DC Buck Boost converter with R-Load 75

(Source andload interchanged in position)




DC- DC BuckBoost converter (Duty Cycle =0.4)














Buck Boost converter (Duty Cycle =0.8)





16











Figure 2.64 Two quadrant DC- DC Buck Boost converter with R-L 84

Load












DC- DC BuckBoost converter (Duty Cycle =0.4)









17












DC- DC Buck Boost converter (Duty Cycle =0.8

Figure 2.81 Proposed Buck-Boost DC-DC converter with R-L Load 93

(Source andload interchanged in position)














Buck Boost Converter (Duty Cycle = 0.4)





18















Figure 2.98 Two quadrant DC- DC Buck Boost converter with 102

R-L-Emf Load

















19














Buck BoostConverter (Duty Cycle = 0.8)



20

LIST OF Symbols and Abbreviations

Abbreviations

Vin = The input voltage

Vo = Output voltage

VL = Inductor voltage

Ic = Capacitor Current

Ps= Power supply

Po= Output Power

Imin = Minimum Current

Imax = Maximum Current

TON = Turn on time.

TOFF = Turn off time.

T = TON + TOFF.= Time period

D = Duty cycle = TON / T.

f = Switching frequency

L = Inductor

C = Filter capacitance

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Chapter-1

INTRODUCTION

1.1 INTRODUCTION

The primary task of power electronics is to process and control the flow of electric

energy by supplying voltages and currents in a form that is suited for user loads.

Power quality, the quality of voltage and current is an important consideration in all

types of application. Power quality problems include transients, sags, swells, surges,

outages, harmonics, and impulses in a system. Among these voltage sags have

negative impact on industrial, residential, transportation, aerospace,

telecommunications productivity. It is necessary that some converters are to be used

to improve the quality of power supply. Circuits composed of power semiconductor

devices are make it possible to use variety controllersraise power quality to meet the

requirements.

In DC– DC converter, the input voltage is converted to a dc output voltage having a

larger or smaller magnitude, with same or opposite polarity having non-

isolated/isolation input/output. The average value of a chopper‟s output voltage can be

modified between zero and the full voltage, using the “Pulse Width Modulation

(PWM)” of constant frequency pulses. There are schemes of chopper circuits

operating in one to four quadrants. Advantages of PWM converters include low

component count, high efficiency, constant frequency operation, relatively simple

control and commercial availability of integrated circuit controllers, and ability to

achieve high conversion ratios for both step-down and step-up application. A

disadvantage of PWM dc-dc converters is that PWM rectangular voltage and current

waveformscause turn-on and turn-off losses in semiconductor devices, which limit

their operating frequency. Rectangular waveforms also generate EMI.

Several types of converter are available which operate in single or two quadrants.

There are more than 500 topologies of DC/DC converters. A common DC/DC

converter family tree is shown in Figure 1.1[1].

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Figure 1.1: - DC/DC converter family Tree

Figure 1.1 DC-DC Converter Family Tree [1]

1.1 Buck Boost DC-DC converter withcontinuous input current:

The conventional Buck-Boost dc-dc converter has discontinuous input current (supply

or source current) without the input current filter. The term continuous input current is

not the same as continuous inductor current operation of a dc-dc converter. In Boost

and Ĉuk dc-dc converters the input inductor current is the same as the supply/source

current. In Buck and Buck-Boost converters they are different entities. Buck-Boost

dc-dc converters being discontinuous input/supply current type, their use is limited

because input filters requirement in them are large. A modified Buck-Boost converter

23

(also may be considered as modified Ĉuk) dc-dc converter is available in literature

whose input (supply side) current is continuous in nature [2-9]. The Buck-Boost dc-dc

converter [8] operates in one quadrant like the conventional Buck-Boost dc-dc

converter, but in many applications dc-dc converters of two or four quadrant

operations are necessary. The two quadrant dc-dc converters are building blocks of

four quadrant dc-dc converters and inverters (dc-ac converters) [4-8]. Since Buck-

Boost dc-dc converter has buck and boost voltage gain capability in one stage, it

would be advantageous if a two quadrant version of Buck-Boost dc-dc converter can

be developed.

The output voltage in DC-DC converters is generally controlled by using a switching

concept, as illustrated in Figure 1.2. Early DC-DC converters were known as

choppers with silicon-controlled rectifiers (SCRs) used as the switching device.

Modern DC-DC converters also known as switch mode power supplies (SMPS)

employ insulated gate bipolar transistors (IGBTs) and metal oxide silicon field effect

transistors (MOSFETs) as switching device.

A switch mode power supply may have several functions [10]:

1. Step-down an unregulated DC input voltage to produce a regulated DC output

voltage using a buck or step-down converter,

2. Step-up an unregulated DC input supply to produce a regulated DC output voltage

using a step-up converter,

3. Step-down and then step-up an unregulated DC input voltage to produce a

regulated DC output voltage using a buck–boost converter,

Figure 1.2 Basic DC-DC converter.

24

Figure 1.3 DC-DC converter voltage waveform.

Figure 1.4 Pulse width modulation concept

4. Invert the DC input voltage if necessary and

5. Produce multiple DC outputs using a combination of SMPS topologies and

multiple transformer secondary operating at high frequency.

The regulation of the average output voltage in a DC-DC converter is a function of

the on-time Ton of the switch, the pulse width, and the switching frequency fs as

illustrated in Figure 1.3 [12]. Pulse width modulation (PWM) is the most widely used

method of controlling the output voltage. The PWM concept is illustrated in Figure

1.4. The output voltage control depends on the duty ratio D.

The duty ratio is defined as,

based on the on-time ton of the switch and the switching period Ts. PWM switching

involves comparing the level of a control voltage Vcontrol to the level of a repetitive

25

waveform as illustrated in Figure 1.4 [12]. The on-time of the switch is defined as the

portion of the switching period, where, the value of the repetitive waveform is less

than the control voltage. The switching period (switching frequency) remains constant

while the control voltage level is adjusted to change the on-time and therefore the

duty ratio of the switch. The switching frequency is usually chosen above 20 kHz so

the noise is outside the audio range [10, 11]. DC-DC converters operate in one of two

modes depending on the characteristics of the inductor current [11, 12]:

1. Continuous conduction and

2. Discontinuous conduction mode.

The continuous-conduction mode is defined by continuous inductor current (above

zero) over the entire switching period, whereas the discontinuous conduction mode is

defined by discontinuous inductor current, zero during any portion of the switching

period

A simple DC-DC SMPS consists of a rectifier fed directly from line voltage, a filter

and a static switch. The SMPS is switched by control circuitry at a very high

frequency to step-down or step-up dc voltage by on/off ratio (duty cycle) control. The

filter and the feedback circuit are the other components of a DC-DC SMPS. Figure-

1.5 shows the block diagram of a DC-DC SMPS.

Main components of a dc-dc SMPS are:

1. Power circuit,

2. Control circuit and

3. Magnetic circuit.

The control circuit of an SMPS generates high frequency gate pulses for the switching

device to control the dc. Switching is performed in multiple pulse width modulation

(PWM) fashion according to feedback error signal from the load to serve two

purposes,

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1. Produce high frequency switching signal.

2. Control on/off period of switching signal to maintain constant voltage

across the load.

High frequency switching reduces filter requirements at the input/output sides of the

converter. Simplest PWM control uses multiple pulse modulations generated by

comparing a dc with a high frequency carrier triangular wave.

Switching regulators are commonly available as integrated circuits. The designer can

select the switching frequency by choosing the value of RC to set oscillator

frequency. As a rule of thumb to maximize the efficiency, the oscillator period should

be about 100 times longer than the transistor switching time; for example, if a

transistor has a switching time of 0.5 µs, the

oscillator period would be 50 µs, which gives the maximum oscillator frequency of 20

KHz. The limitation is due to the switching loss in the transistor. The transistor

switching loss increases with the switching frequency and as a result the efficiency

decreases. In addition, the core loss of inductor limits the high frequency operation.

R2 +

_ Vin Vout

R1 V

Vin

Vout

Figure1.6: Linear (dissipative) power conversion circuit. t

Figure 1.5: Block Diagram of SMPS.

Rectifie

r

Filter Switc

h

Filter AC Source Load

Gate Signal

Generator

Feedback Control

Circuit

Reference

voltage

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Figure-1.6 illustrates the circuit of a linear power conversion. Here power is

controlled by a series linear element; either a resister or a transistor is used in the

linear mode. The total load current passes through the series linear element. In this

circuit greater the difference between the input and the output voltage, more is the

power lost in the controlling device. The linear power conversion is dissipative and

hence is inefficient. The efficiency range is typically 30 to 60% for linear regulators.

The circuit of Figure1.7 illustrates basic principle of a dc-dc switch mode power

conversion. The controlling device is a switch. By controlling the ratio of the time

intervals spent in on and off positions (defined as duty ratio), the power flow to the

load can be controlled in an efficient way. Ideally this method is 100% efficient. In

practice, the efficiency is reduced as the switch is non-ideal and losses occur in power

circuits.

The dc voltage to the load can be controlled by controlling the duty cycle of the

rectangular waveform supplied to the base or gate of the switching device. When the

switch is fully on, it has only a small saturation voltage across it. In the off condition

the current through the device is zero.

The output of the switch mode power conversion control (Figure-1.7) is not pure dc.

This type of output is applicable in cases such as oven heating without proper

filtration. If constant dc is required, then output of SMPS has to be smoothed by a

low-pass filter. Switches are required as basic components for efficient electric power

conversion and control.Inductors and capacitors are used to smooth the pulsating dc

originating from the switching action.

t

V

Vin

Vout

BJT

R +

_ Vin Vout Diode

Figure 1.7: Switch mode (non dissipative) power conversion circuit.

28

Although the conversion would be 100% efficient in the ideal case of lossless

components and circuit (Figure-1.8), in practice all components are lossy. Thus,

efficiency is reduced. Hence, one of the prime objectives in switch mode power

conversion is to realize conversion with the least number of components having better

efficiency and reliability.

1.2 SPECIFIC AIMS AND POSSIBLE OUTCOMES OF THE THESIS WORK:

The objectives of the research of this thesis are as follows,

a) A new Buck-Boost two quadrant DC-DC converter will be proposed

which will have continuity of input current(source current) unlike the

conventional Buck-Boost DC-DC converter,

b) The proposed new topology of Buck-Boost DC-DC converter will be

based on modifying the topology of the dc-dc converter proposed in [10]

which will operate in two quadrants of V-I plane,

c) Appropriate ideal voltage and current gain relationship of the proposed

Buck-Boost DC-DC converter based on volt-amp balance of the inductor

( not the input filter inductor current) will be developed,

d) By simulation study it will be proved that the proposed circuit works in

two quadrant of V-I plane,

e) Compare the input filter requirement of the proposed dc-dc converter with

that of the conventional two quadrant Buck-Boost dc-dc converter and to

prove that the proposed converter need smaller filters and

C Diode

R +

_ Vin

Vout

V

Vin

Vout

Figure-1.8: Typical switch mode power conversion circuit.

t

29

f) Study the performance of the proposed dc-dc converter by simulation

with R, R-L and R-L- Emf loads.

The Expected outcomes of proposed research are

a) A new Two Quadrant Buck-Boost DC-DC converter with inherent

continuous input current will be obtained,

b) The proposed two Quadrant dc-dc converter will require less input

current filter as a result of continuous input current and

c) The proposed two quadrant Buck-Boost converter will be useful in dc

drives requiring regenerative braking or energy recovery capability.

Two quadrant Buck-Boost converters will be proposed based on conventional and

modified [10] Buck-Boost DC-DC converter circuits. Appropriate ideal voltage and

current gain relationships of the two converters will be derived. Validity of the gain

relationship will be verified by Pspice simulation. Deviation from practical gain

relationship will be studied by introducing internal inherent resistance of the

inductance/s, voltage drops in the devices and assuming constant switching losses in

the device in the circuits. Proof of the two quadrant operations of the converter will be

given logically and by simulation study. Comparison of the two quadrant Buck-Boost

converter made of conventional and modified one quadrant [10] dc-dc converter will

be carried out on the basis of input filter requirement and the efficiency of the

converter on the same operating conditions and load. The performance of the

proposed dc-dc converter will be studied by simulation for R, R-L and R-L-Emf

loads.

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1.3 Thesis Outline

Chapter 1 and 2introduces the thesis topic. Overview of DC Choppers is provided.

Also Discussion has been made about DC choppers with different Quadrant operation

scenario with Continuous and discontinuous current mode. This chapter contains the

objective and outline of the thesis.

Chapter 3presents the conventional DC-DC Buck Boost converter with

discontinuous input current anddevelops the modified buck-boost dc-dc converter

with one quadrant continuous input current. Based on continuous input current a two

quadrant buck-boost DC-DC converter is developed and studied in chapter-2. Details

performance study, analysis and observations of the proposedtwo quadrant Buck

Boost DC-DC converter with continuous input current are the main focus of this

chapter.

Chapter 4draws the conclusion of this work. This chapter puts achievement and

forward suggestionsfor future scopes of works related to this thesis.

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Chapter-2

Classification of DC-DC Converter

2.1 TYPES OF DC-DC CONVERTERS

There are five basic topologies of switching regulators:

a. Buck converter,

b. Boost converter,

c. Buck-Boost converter,

d. Ĉuk converter,

e. SEPIC Converter and

2.1.1 BUCK CONVERTER [4, 6, 8-9, 28-32]

In Buck converters, output voltage is regulated and is less than the input voltage,

hence the name "Buck". Buck converters and dc-dc converters in general, have the

following properties when operating in the steady state:

1. The inductor current is periodic.

iL(t+T)=iL(t) (1-1)

2. The average inductor voltage is zero

VL=

∫ ( )

=0 (1-2)

3. The average capacitor current is zero

Ic=

∫ ( )

=0 (1-3)

4. The power supplied by the source is the same as the power delivered to the

load. For Non ideal conditions source also supplies losses.

Ps=Po Ideal

Ps=Po+Losses nonideal (1-4)

Analysis of the buck converter of Fig. 1.9(a) begins by making these assumptions:

1. The circuit is operating in the steady state,

2. The inductor current is continuous (always positive),

32

3. The capacitor is very large, and the output voltage is held constant at

voltageVo. This restriction will be relaxed later to show the effects of finite

Capacitance.

4. The switching period is T; the switch is closed for time DT and open for

time(1-D)T.

5. The components are ideal.

The key to the analysis for determining the output Vois to examine the inductor.

Current and inductor voltage first for the switch closed and then for the switchopen.

The net change in inductor current over one period must be zero for steady

stateoperation. The average inductor voltage is zero.

Analysis for the Switch Closed When the switch is closed in the buck

convertercircuit of Fig. 1.9(a), the diode is reverse-biased and Fig. 1.9(b) is an

equivalent circuit.

The voltage across the inductor is

vL=Vs-Vo=L

Figure1.9: a) Buck dc-dc converter; (b) Equivalent circuit for the switch closed;

(c) Equivalent circuit for the switch open.

33

Rearranging,

=

switch closed

Since the derivative of the current is a positive constant, the current increases linearly

as shown in Fig. 1.9(b). The change in current while the switch is closed is computed

by modifying the preceding equation.

=

=

=

( )Closed=(

) DT (1-5)

Analysis for the Switch Open When the switch is open, the diode becomes forward-

biased to carry the inductor current and the equivalent circuit of Fig. 1.9(c) applies.

The voltage across the inductor when the switch is open is

vL = -Vo = L

Rearranging,

=

switch open

Figure1.10: Buck converter waveforms: (a) Inductor voltage ;( b) Inductor current;

(c) Capacitor current.

34

The derivative of current in the inductor is a negative constant, and the current

decreases linearly as shown in Fig. 1.10(b). The change in inductor current when the

switch is open is

=

( ) =

( )Open = - (

)(1-D) (1-6)

Steady-state operation requires that the inductor current at the end of the switching

cycle be the same as that at the beginning, meaning that the net change in inductor

current over one period is zero. This requires

( )Closed + ( )open=0

Using Equations (1.5) & (1.6)

(

) DT- (

)(1-D) T = 0

Solving for VO,

VO= VsD (1-7)

An alternative derivation of the output voltage is based on the inductor voltage, as

shown in Fig. 1.10(a). Since the average inductor voltage is zero for periodic

operation,

VL =(Vs-Vo) DT-(-VO) (1-D) T=0

Solving the preceding equation for VOyields the same result as Eq. (1-7), VO=Vs D.

Output voltage depends on only the input and the duty ratio D. If the input voltage

fluctuates, the output voltage can be regulated by adjusting the duty ratio

appropriately. A feedback loop is required to sample the output voltage, compare it to

a reference, and set the duty ratio of the switch accordingly.

The average inductor current must be the same as the average current in the load

resistor, since the average capacitor current must be zero for steady-state operation:

35

IL=IR=

(1-8)

Since the change in inductor current is known from Eqs. (1-5) and (1-6), the

maximum and Minimum values of the inductor current are computed as

I max =IL +

=

+

[(

)(1-D) T] =VO (

+

) (1-9)

I min =IL -

=

-

[(

)(1-D) T] =VO (

-

) (1-10)

Where, f = 1/T is the switching frequency.

For the preceding analysis to be valid, continuous current in the inductor must be

verified. An easy check for continuous current is to calculate the minimum inductor

current from Eq. (1-10). Since the minimum value of inductor current must be

positive for continuous current, a negative minimum calculated from Eq. (1-10) is not

allowed due to the diode and indicates discontinuous current. The circuit will operate

for discontinuous inductor current, but the preceding analysis is not valid.

Discontinuous-current operation is discussed later in this chapter.

Equation (1-10) can be used to determine the combination of L and f that will result in

continuous current. Since Imin= 0 is the boundary between continuous and

discontinuous current,

I min =0=VO (

-

)

(Lf) min= ( )

(1-11)

If the desired switching frequency is established,

Lmin= ( )

for continuous current (1-12)

Where,Lmin is the minimum inductance required for continuous current. In practice, a

value of inductance greater thanLmin is desirable to ensure continuous current.

36

In the design of a buck converter, the peak-to-peak variation in the inductor current is

often used as a design criterion. Equation (1-5) can be combined with Eq. (1-7) to

determine the value of inductance for a specified peak-to-peak inductor current for

continuous-current operation:

=(

) DT = (

) D = ( )

(1-13)

Or L= (

) D = ( )

(1-14)

Since the converter components are assumed to be ideal, the power supplied by the

source must be the same as the power absorbed by the load resistor.

Ps=Po

VsIs=VoIo (1-15)

Or

=

The preceding relationship is similar to the voltage-current relationship for a

transformer in ac applications. Therefore, the buck converter circuit is equivalent to a

dc transformer.

2.1.2 BOOST CONVERTER [4, 6, 8-9, 24-30]

The boost converter is shown in Fig. 1.11. It is called a boost converter because the

output voltage is larger than the input.

Analysis assumes the following:

1. The circuit is operating in the steady state.

2. The inductor current is continuous (always positive).


voltage Vo.

4. The switching period is T; the switch is closed for time DTand open for

time (1-D) T.


37

The analysis proceeds by examining the inductor voltage and current for the

switch closed and again for the switch open.

Analysis for the Switch Closed When the switch is closed, the diode is reverse

biased. Kirchhoff‟s voltage law around the path containing the source, inductor and

closed switch is

vL=Vs=L

or

=

switch closed (1-16)

The rate of change of current is constant, so the current increases linearly while the

switch is closed, as shown in Fig. 6-9b. The change in inductor current is computed

from

=

=

Figure 1.11: a) Boost dc-dc converter; (b) Equivalent circuit for the switch closed;

(c) Equivalent circuit for the switch open.

38

Solving for for the switch closed,

( )Closed=

(1-17)

Analysis for the Switch Open When the switch is opened, the inductor current

cannot change instantaneously, so the diode becomes forward-biased to provide a path

for inductor current. Assuming that the output voltage Vois a constant, the voltage

across the inductor is

vL = Vs-Vo = L

Rearranging,

=

switch open

The rate of change of inductor current is constant, so the current changes linearly

while the switch is open. The change in inductor current while the switch is open is

Figure 1.12: Boost converter waveforms: (a) Inductor voltage; (b) Inductor current;

(c) Diode Current (d) Capacitor Current.

39

=

( ) =

Solving for

( )Open= (

)(1-D) T (1-18)

For steady-state operation, the net change in inductor current is zero. Using Eqns. (1-

17) and (1-18),

( )Closed + ( )open =0

Using Equations (1.5) and (1.6)

+ (

)(1-D) T= 0

Solving for VO,

VO=

(1-19)

Also, the average inductor voltage is zero. Expressing the average inductor voltage

over one switching period,

VL =VsD+ (Vs-Vo) (1-D)=0

Solving the preceding equation for VOyields the same result as Eq. (1-19), VO=

.

Equation (1-19) shows that if the switch is always (D is zero), the output voltage is

the same as the input. As the duty ratio is increased, the denominator of Eqn. (1-19)

becomes smaller than 1, resulting in a larger output voltage. The boost converter

produces an output voltage that is greater than or equal to the input voltage.

The output voltage cannot be less than the input, as was with the case of buck

converter.

As the duty ratio of the switch approaches 1, the output voltage goes to infinity

according to Eqn. (1-19). Eq. (1-19) is based on ideal components and operation

where, switching and conduction losses are assumed to be zero. Real components that

40

have losses will cause ideal voltage gain ratio to deviate. Figure 1-10 shows typical

voltage and current waveforms of the boost converter.

The average current in the inductor is determined by recognizing that the average

power supplied by the source must be the same as the average power absorbed by the

load resistor. Output power is,

Po=

=VoIo

And input power is Vs Is =Vs IL. Equating input and output powers and using Eqn. (1-

19),

Vs IL =

=

=

( ) (1-19)

By solving for average inductor current and after manipulation and simplifications,

ILcan be expressed as,

IL =

( ) =

=

(1-20)

Maximum and minimum inductor currents are determined by using the average value

and the change in current from Eqn. (1-17).

I max =IL +

=

( ) +

(1-21)

I min =IL -

=

( ) -

(1-22)

Equation (1-19) was developed with the assumption that the inductor current is

continuous, meaning that it is always positive. A condition necessary for continuous

inductor current is for Iminto be positive. Therefore, the boundary between continuous

and discontinuous inductor current is determined from

I min =IL -

=

( ) -

= 0

Or

( ) =

=

41

Where, f = 1/T is the switching frequency.

The minimum combination of inductance and switching frequency for continuous

current in the boost converter is therefore,

(Lf) min = ( )

(1-23)

Or Lmin = ( )

(1-24)

A boost converter designed for continuous-current operation will have an inductor

value greater than Lmin.

From a design perspective, it is useful to express L in terms of a desired ,

L=

=

(1-25)

2.1.3 BUCK-BOOST CONVERTER [3-5, 9, 12-20]

Another basic switched-mode converter is the buck-boost converter shown in Figure

1-13. The output voltage of the buck-boost converter can either be higher or lower

than the input voltage.

Analysis assumes the following:

1. The circuit is operating in the steady state,

2. The inductor current is continuous,


voltage VO,

4. The switching period is T; the switch is closed for time DTand open for

time (1-D) T and


The analysis proceeds by examining the inductor voltage and current for the

switch closed and again for the switch open.

Analysis for the Switch Closed When the switch is closed, the voltage across the

inductor is

vL=Vs=L

or

=

switch closed

42

The rate of change of current is constant, indicating a linearly increasing inductor

current. The preceding equation can be expressed as

=

=

Solving for for the switch closed,

( )Closed=

(1-26)

Figure-1.13: a) Buck Boost dc-dc converter; (b) Equivalent circuit for the switch

closed;(c) Equivalent circuit for the switch open.

43

Analysis for the Switch Open When the switch is open, the current in the inductor

cannot change instantaneously, resulting in a forward-biased diode and current into

the resistor and capacitor. In this condition, the voltage across the inductor is,

vL = Vo = L

Rearranging,

=

switch open

The rate of change of inductor current is a constant, currentchange linearly while the

switch is open. The change in inductor current while the switch is open is,

=

( ) =

Solving for

( )Open= (

)(1-D) T (1-27)

Figure-1.14: Buck Boost converter waveforms: (a) Inductor Current; (b) Inductor

voltage; c) Diode Current (d) Capacitor Current

44

For steady-state operation, the net change in inductor current must be zero. Using Equ

(1-26) and (1-27),

( )Closed + ( )open =0

+ (

)(1-D) T= 0

Solving for VO,

VO= -Vs(

) (1-28)

The required duty ratio for specified input and output voltages can be expressed as,

D=

(1-29)

Also, the average inductor voltage must be zero for periodic operation. Expressing the

average inductor voltage over one switching period,

VL= VsD +VO (1-D)=0

Solving the equation for VOyields the result as Eqn. (1-28),

VO=

Equation (1-28) shows that the output voltage has opposite polarity from the source

voltage. Output voltage magnitude of the buck-boost converter can be less than that of

the source or greater than the source, depending on the duty ratio of the switch. If D ≥

0.5, the output voltage is larger than the input; and if D≤ 0.5, the output is smaller

than the input. Therefore, this circuit combines the capabilities of the buck and boost

converters. Polarity reversal on the output may be a disadvantage in some

applications.

Source of buck-boost converter is never connected directly to the load. Energy is

stored in the inductor when the switch is closed and transferred to the load when the

switch is open. Hence, the buck-boost converter is also referred to as an indirect

converter.

45

Power absorbed by the load must be the same as that supplied by the source, where,

the average current in the inductor is determined by recognizing that the average

power supplied by the source must be same as the average power absorbed by the

load resistor. Output power is,

Po=

=VsIs where, Ps=VsIs

Average source current is related to average inductor current by,

Is=ILD

Results in

=VsILD

Substituting for VOusing Eqn. (1-28) and solving for IL, we find

IL =

=

=

( ) (1-30)

Maximum and minimum inductor currents are determined by using the average value

and the change in current from Eqn. (1-26).

I max =IL +

=

( ) +

(1-31)

I min =IL -

=

( ) -

(1-32)

For continuous current, the inductor current must remain positive. To determine the

boundary between continuous and discontinuous current, Iminis set to zero in Eqn. (1-

32), resulting in

(Lf) min = ( )

(1-33)

Or Lmin = ( )

(1-34)

Where, f is the switching frequency.

46

2.1.4 ĈuK CONVERTER[4, 6, 8, 25-31]

The circuit diagram of a Ĉukdc-dc converter is shown in Figure 6-13. Output voltage

magnitude can be either larger or smaller than that of the input, and there is a polarity

reversal on the output. The inductor on the input acts as a filter for the dc supply to

prevent large harmonic contents. Unlike the previous converter topologies, where,

energy transfer is associated with the inductor, energy transfer for the Ĉuk converter

depends on the capacitor C1. The analysis begins with these assumptions:

1. Both inductors are very large and the currents in them are constant,

2. Both capacitors are very large and the voltages across them are constant,

3. The circuit is operating in steady state, meaning that voltage and current

Waveforms are periodic,

4. For a duty ratio of D, the switch is closed for time DT and open for (1-D) T.

5. The switch and the diode are ideal.

Figure-1.15: The Ĉuk converter (a) Circuit; b) Equivalent circuit for the switch

closed; (c) Equivalent circuit for the switch open; (d) Current in L1 for a large

inductance.

47

The average voltage across C1 is computed from Kirchhoff‟s voltage law around the

outermost loop. The average voltage across the inductors is zero for steady state

operation, resulting in

VC1=Vs-Vo

With the switch closed, the diode is off and the current in capacitor C1 is,

(iC1)closed= - IL2 (1-35)

With the switch open, the currents in L1 and L2 force the diode on. The current in

capacitor C1 is,

(iC1)open= IL1 (1-36)

The power absorbed by the load is equal to the power supplied by the source:

-VoIL2= VsIL1 (1-37)

For periodic operation, the average capacitor current is zero. With the switch on for

time DT and off for (1-D)T,

[(iC1)closed]DT+ [(iC1)open](1-D)T=0

Substituting usingEqs. (1-35) and (1-36),

- IL2DT+ IL1(1-D)T=0

Or

= -

(1-38)

Next, the average power supplied by the source must be the same as the average

power absorbed by the load,

Ps=Po

VsIL1= -VoIL2

= -

(1-39)

Combining Eqs. (1-38) and (1-39), the relationship between the output and input

voltages is

Vo= -Vs (

) (1-40)

The negative sign indicates a polarity reversal between output and input. The

components on the output (L2, C2, and R) are in the same configuration as the buck

converter and that the inductor current has the same form as for the buck converter.

Therefore, the ripple, or variation in output voltage, is the same as for the buck

converter.

48

=

(1-41)

The output ripple voltage will be affected by the equivalence series resistance of the

capacitor as it was in the convertors discussed previously.

The ripple in C1 can be estimated by computing the change in vC1 in the interval when

the switch is open and the currents iL1 and iC1 are the same. Assuming the current in L1

to be constant at a level IL1 and using Eqs. (1-39) and (1-40), we have

c1≈

∫ ( )

=

(1-D)T =

(

)

Or c1≈

(1-42)

The fluctuations in inductor currents can be computed by examining the inductor

voltages while the switch is closed. The voltage across L1 with the switch closed is

vL1=Vs=L1

(1-43)

In the time interval DT when the switch is closed, the change in inductor current is

=

Or iL1=

=

(1-44)

For inductor L2, the voltage across it when the switch is closed is

vL2 =Vo+ (Vs-Vo) =Vs = L2

(1-45)

The change in iL2 is then

iL2=

=

(1-46)

For continuous inductor currents, the average current must be greater than one-half

the change in current. Minimum inductor sizes for continuous current are

L1, min=( )

L2, min =

( )

(1-47)

2.1.5 SEPIC Converter [8, 12, 17-20]

A converter similar to the Ĉuk is the single-ended primary inductance

converter(SEPIC), as shown in Figure1-14. The SEPIC can produce an output voltage

thatis either greater or less than the input but with no polarity reversal. To derive the

relationship between input and output voltages, these initialassumptions are made:

1. Both inductors are very large and the currents in them are constant.

2. Both capacitors are very large and the voltages across them are constant.

49

3. The circuit is operating in the steady state, meaning that voltage and current

Waveforms are periodic.

4. For a duty ratio of D, the switch is closed for time DT and open for (1 - D)T.

5. The switch and the diode are ideal.

The inductor current and capacitor voltage restrictions will be removed laterto

investigate the fluctuations in currents and voltages. The inductor currents areassumed

to be continuous in this analysis. Other observations are that the averageinductor

Figure-1.16: (a) SEPICCircuit;b) Circuit with the switchclosed and the diode off; (c)

Circuit with the switch open andthe diode on.

50

voltages are zero and that the average capacitor currents are zero forsteady-state

operation.

Kirchhoff‟s voltage law around the path containing Vs, L1, C1, and L2 gives

- Vs+vL1+vc1-vL2=0

Using the average of these voltages,

- Vs+0+Vc1-0=0

Showing that the average voltage across the capacitor C1 is

Vc1=Vs (1-48)

When the switch is closed, the diode is off, and the circuit is as shown in Fig. 1-14b.

The voltage across L1 for the interval DT is

vL1=Vs (1-49)

When the switch is open, the diode is on, and the circuit is as shown in Fig. 1-

14c.Kirchhoff‟s voltage law around the outermost path gives

- Vs+vL1+vc1+Vo =0 (1-50)

Assuming that the voltage across C1 remains constant at its average value of Vs[Eq.

(1-48)],

- Vs+vL1+Vs+Vo =0 (1-51)

Or vL1= -Vo (1-52)

for the interval (1-D)T. Since the average voltage across an inductor is zero

forperiodic operation, Eqs. (1-49) and (1-52) are combined to get

(vL1,sw closed)(DT)+(vL1,swopen)(1-D)T=0

Vs(DT)-Vo(1-D)T=0

WhereDis the duty ratio of the switch. The result is

51

Vo=Vs(

) (1-53)

which can be expressed as

D=

(1-54)

This result is similar to that of the buck-boost and Ĉuk converter equations, withthe

important distinction that there is no polarity reversal between input and

outputvoltages. The ability to have an output voltage greater or less than the inputwith

no polarity reversal makes this converter suitable for many applications.

Assuming no losses in the converter, the power supplied by the source is the

same as the power absorbed by the load.

Ps=Po

Power supplied by the dc source is voltage times the average current, and thesource

current is the same as the current in L1.

Ps=VsIs=VsIL1

Output power can be expressed as

Po=Vo Io

Resulting in

VsIL1=Vo Io

Solving for average inductor current, this is also the average source current,

IL1=Is=

=

(1-55)

The variation in iL1 when the switch is closed is found from

vL1=Vs=L1(

)=L1(

)=L1(

) (1-56)

Solving for iL1,

52

iL1=

(1-57)

For L2, the average current is determined from Kirchhoff‟s current law at thenode

where C1, L2, and the diode are connected.

iL2=iD-iC1

Diode current is

iD=iC2+Io

Which makes

iL2=iC2+Io-iC1

The average current in each capacitor is zero, so the average current in L2 is

IL2=Io (1-58)

The variation in iL2 is determined from the circuit when the switch is closed.Using

Kirchhoff‟s voltage law around the path of the closed switch, C1, and L2with the

voltage across C1 assumed to be constant Vs, gives

vL2=vC1=Vs=L2(

)=L2(

)=L2(

)

Solving for iL2,

iL2=

(1-59)

Applications of Kirchhoff‟s current law show that the diode and switch currentsare

{

(1-60)

{

Current waveforms are shown in Fig. 1-15.Kirchhoff‟s voltage law applied to the

circuit of Fig. 1-14c, assuming novoltage ripple across the capacitors, shows that the

voltage across the switchwhen it is open is Vs+ Vo. From Fig. 1-14b, the maximum

53

reverse bias voltageacross the diode when it is off is also Vs+ Vo.The output stage

consisting of the diode, C2, and the load resistor is the sameas in the boost converter,

so the output ripple voltage is

Vo= VC 2=

(1-61)

Solving for C2,

C2=

(

)

(1-62)

The voltage variation in C1 is determined form the circuit with the switch closed (Fig.

1-14b). Capacitor current iC1 is the opposite of iL2, which has previouslybeen

determined to have an average value of Io. From the definition ofcapacitance and

considering the magnitude of charge,

VC1=

=

=

Replacing Iowith Vo/R,

VC1=

(1-63)

Solving for C1,

C1=

(

)

(1-64)

The effect of equivalent series resistance of the capacitors on voltage variation is

usually significant, and the treatment is the same as with the converters discussed

previously.

54

2.2DC CHOPPERS QUADRANT OPERATION [7, 41-38]:

Basic SMPS circuits are single quadrant choppers that operate athigh frequency.

Choppers are circuits that convert fixed DC voltage to constant or variable DC

voltage.

2.2.1 MULTIPLE QUADRANT OPERATION

A DC motor can run in forward running or reverse running. During the forward

starting process its armature voltage and armature current are both positive. We

usually call this forward motoring operation or quadrant I operation. During the

Figure-1.17: Currents in SEPIC Converter(a) L1; b) L2; (c) C1; (d) C2; (e) switch; (f)

diode

55

forward braking process its armature voltage is still positive and its armature current

is negative. This state is called the forward regenerating operation or quadrant II

operation. Analogously, during the reverse starting process the DC motor armature

voltage and current are both negative. This reverse motoring operation is called the

quadrant III operation.

Figure-1.18: Four Quadrant Operation.

During reverse braking process its armature voltage is still negative and its armature

current is positive. This state is called the reverse regenerating operation quadrant IV

operation. Referring to the DC motor operation states, the classifications of DC-DC

choppers according to Vo-Io position in X-Y co-ordinates are as follows:

Quadrant I operation: forward motoring, voltage is positive, current is positive;

(+Vo +Io).

Quadrant II operation: forward regenerating, voltage is positive, current is negative;

(+Vo -Io).

Quadrant III operation: reverse motoring, voltage is negative, current is negative; (-

Vo -Io).

Quadrant IV operation: reverse regenerating, voltage is negative, current is positive;

(-Vo +Io).

56

The operation status is shown in the Figure 1.14. Choppers can convert a fixed DC

voltage into various other voltages. The corresponding chopper is usually named

according to its quadrant operation of the chopper, e.g., the first quadrant chopper or

“A”-type chopper. In the following description we use the symbols VIN as the fixed

voltage, Vp the chopped voltage, and VO the output voltage.

2.2.2 THE QUADRANT-ONE CHOPPER.

The one-quadrant chopper is also called “A”-type chopper and its circuit diagram is

shown in Figure 1.15a and corresponding waveforms are shown in Figure 1.15b. The

switch S can be a semiconductor devices such a BJT or an IGBT or a MOSFET.

Assuming all parts are ideal components, the output voltage is calculated by the

formula, (1.47)

Vo = ininon kVV

t

T (1.65)

Figure 1.19: The Quadrant One Chopper.

Where, T is the repeating period T = 1/f, f is the chopping frequency, ton is the switch-

on time, k is the conduction duty cycle k = ton/T.

57

2.2.3 THE QUADRANT- TWO CHOPPER.

The two-quadrant chopper is the called “B”-type chopper and the circuit diagram and

corresponding waveforms are shown in Figure 1.16a and b. The output voltage can be

calculated by the formula (1.48)

Vo = ininoff VkV

t)1(

T (1.66)

Figure 1.20: The Quadrant Two Chopper.

Where, T is the repeating period T = 1/f, f is the chopping frequency, toff is the switch-

off time toff = T – ton, and k is the conduction duty cycle k = ton/T.

58

2.2.4 THE QUADRANT- THREE CHOPPER

The three-quadrant chopper and corresponding waveforms are shown in Figure 1.17a

and b. All voltage polarity is defined in the Figure. The output voltage (absolute

value) can be calculated by the formula (1.67)

VO = ininon kVV

t

T (1.67)

Figure 1.21: The Quadrant Three Chopper.

Where, ton is the switch-on time, and k is the conduction duty cycle k = ton/T.

2.2.5 THE QUADRANT-FOUR CHOPPER

The four-quadrant chopper and corresponding waveforms are shown in Figure 1.18a

and b. All voltage polarity is defined in the figure. The output voltage (absolute value)

can be calculated by the formula (1.68).

59

Vo = ininoff VkV

t)1(

T (1.68)

Figure 1.22: The Quadrant Four Chopper.

Where, toff is the switch-off time toff = T – ton, time, and k is the conduction duty cycle

k = ton/T.

2.2.6 TWO QUADRANT CHOPPER (Operates in Quadrant I and II)

Theone and two quadrant chopper is shown in Figure 1.19. Two quadrant operation is

usually requested in the system with two voltage sources V1 and V2. Assume that the

condition V1 > V2, and the inductor L is an ideal component. During quadrant I

operation, S1 and D2 work, and S2 and D1 are idle. In a same manner, during

quadrant II operation, S2 and D1 work, and S1 and D2 are idle. The relation between

the two voltage sources can be calculated by the formula, (1.69)

60

1

12 )1( Vk

kVV (1.69)

Figure 1.23: The One and Two Quadrant Chopper Circuit Diagram.

2.2.7 THREE AND FOUR QUADRANT CHOPPERs

Three and four quadrant chopper is shown in Figure 1.20. Two quadrant operation is

usually requested in the system with two voltage sources V1 and V2. Both voltage

polarities are defined in the Figure. Absolute values of V1 and V2 are used in analysis

and calculation. Assuming the condition V1 > V2, the inductor L is ideal component.

During quadrant I operation, S1 and D2 work, and S2 and D1 are idle. Similarly,

during quadrant II operation, S2 and D1 work, and S1 and D2 are idle. The relation

between the two voltage sources can be calculated by the formula (1.70).

1

12 )1( Vk

kVV (1.70)

Figure 1.24: The Three and Four Quadrant Chopper Circuit Diagram.

QI Operation

QII Operation

QIII Operation

QIV Operation

61

2.2.8 THE FOUR-QUADRANT CHOPPER

The four-quadrant chopper is shown in Figure 1.21. The input voltage is positive;

output voltage can be either positive or negative. The switches and diode status for the

operation are shown in Table 1.1. The output voltage can be calculated by the formula

(1.71).

1

1

1

1

2

)1(

)1(

VkkV

VkkV

V

(1.71)

Figure 1.25: The Four Quadrant Chopper Circuit Diagram.

QI Operation

QII Operation

QIII Operation

QIV Operation

QIII Operation

62

Table-1.1:

Four quadrant chopper is a chopper composed of two ½ H Bridge and the other

choppers are the subclass of four quadrant choppers. DC-DC choppers according to

their V-I quadrants of operation are also shown in Figure 1.21 as follows:

In the parts of Figure: 1.22, the subscript of the active switches or switches and diodes

specify in which quadrants operation is possible. For example, the chopper in Figure

1.22d, uses switches T1 and T3, so can only operate in the one (+Io,+Vo) and three (-

Io,-Vo) quadrants.

The quadrant-one chopper in Figure: 1. 22a, (and Figure: 1. 22c) produces a positive

voltage across the load since the freewheeling diode D1 prevents a negative output

voltage. Also delivers current from the dc source to the load through the

unidirectional switch T1. So It is a single quadrant chopper and only operates in the

quadrant-one (+Io,+Vo).

63

Figure: 1.26. Fundamental four quadrant chopper (centre) showing deviations of four

subclass DC choppers,(a). First quadrant choppers-I, (b). Second quadrant choppers-

II, (c). First and second quadrant choppers-I & II, (d). First and Fourth quadrant

choppers – I & IV, and (e). Four quadrant choppers.

The quadrant-two chopper, (-Io,+Vo), in Figure: 1. 22 b, is a voltage boost circuit and

current flows from the load to the supply, Vs. The switch T2 is turned on to build-up

the inductive load current. When the switch is turned off current is forced to flow

64

through diode D2 into the dc supply. The two current paths (when the switches on and

when it is off) are shown in Figure: 1. 22b.

In the two-quadrant chopper, quadrants I and II chopper, (±Io,+Vo), Figure: 1. 22c,

the load voltage is clamped between 0V and Vs, because of the freewheel diodes D1

and D2. Because this chopper is a combination of the quadrant-one chopper in Figure;

1.22a and the quadrant-two chopper in Figure: 1.22b, it combines the characteristics

of both. Bidirectional load current is possible but the average output voltage is always

positive. Energy can be regenerated into the supply Vs due to the load inductive

stored energy which maintains current flow from the back emf source in the load.

The two-quadrant chopper, quadrants I and IV chopper, (+Io,±Vo), Figure; 1.22d,

produces a positive voltage, negative voltage or zero volts across the load, depending

on the duty cycle of the switches and the switching sequence. When both switches are

switched simultaneously, an on-state duty cycle of less than 50% (δ < ½) results in a

negative average load voltage, while δ > ½ produces a positive average load voltage.

Since Vo is reversible, the power flow direction is reversible, for the shown current io.

Zero voltage loops are created when one of the two switches is turned off.

The four-quadrant chopper in the centre of Figure: 1.22e and Figure 1.21, combines

all the properties of the four subclass choppers. It uses four switches and is capable of

producing positive or negative voltages across the load, and deliver current to the load

in either direction, (±Io,±Vo).

65

CHAPTER- 3

Two quadrants Buck Boost DC-DC converter with continuous input

current

3.1Introduction

In conventional Buck-Boost DC-DC converters, input current is discontinuous.

Discontinuity of input current creates problem of input current filtering. With AC

input from transformers or generators, discontinuous input current has the problem of

appearance of high voltage across the switch. DC-DC Buck-Boost conversion may be

done by modified Buck-Boost circuit which has continuous input current. In this

chapter conventional and modified DC-DC Buck-Boost converter with continuous

input current operation is briefly described. Finally the operation of two quadrants

Buck Boost DC-DC converter with continuous input current is described with R, R-L

and R-L- Emf loads.

3.1.1 Conventional DC-DC Buck-Boost Converter

Buck converters can step-down and Boost converters can step-up dc voltages

individually. The Buck-Boost converter in which the inductor is grounded can

perform either of these two conversions. The output voltage polarity is opposite to

input voltage and as a result the converter is known as an inverting converter. The

circuit of a conventionalBuck-Boost converter is shown in Figure 2.1.

Figure. 2.1 Conventional DC-DC Buck- Boost converter

VOFF=0 VAMPL=24 FREQ=50

TD = 0 TF = .001m

PW = .08m PER = .2m V1 = 0

TR = .001m V2 = 20

66

The ideal voltage and current gain equations are as follows:

= -

(2.1)

=

(2.2)

Where, D is the duty cycle defined as D=Ton/T,Ton is the ON time,Toff is the OFF

timeand T is the period of the controlling signal of the switch.

From equations 2.1 and 2.2 it is evident that either a step up (D>0.5) or a step

down(D<0.5)conversion can be achieved with the same converter.For D=1,the gain

becomes infinitebut practically a finite voltage results due to inductor‟s

parasiticresistance, switching losses and voltage drops across the switch/diode.

The operations of the conventional DC-DC Buck Boost converter can be explained

withthe help of the following Figures:

When the Switch is ON

When the switch is ON diode (D) is reverse biased and current flows through inductor

in clockwise direction from sources,

Figure 2.2: Conventional DC- DC Buck Boost converter whenthe switch is ON

When the Switch is OFF

When the switch is OFF the diode is forward biased and the inductor current

flowsthrough the load resistance in anticlockwise direction as shown in Figure 2.3. As

the switch is open so no current flows from source circuit hence discontinuous input

current mode operation occurs.

S D

67

Figure 2.3: Conventional DC- DC Buck Boost converter when the switch is OFF

3.1.2 Typical Simulation Results of conventional one Quadrant Buck-Boost DC-

DC Converter

The circuit simulation for Buck and Boost operations of conventional DC-DC Buck-

Boost converter for different duty cycles are shown through Figures 2.4 to 2.9. It is

seenthat input current is always discontinuous and output voltage is inverted.

Figure 2.4: Average Input voltage wave shape of conventional DC- DC Buck Boost

converter (Duty Cycle = 0.4)

Time 80ms 90ms 100ms 110ms 120ms 130ms 0V

10V

20V

25V

AVG(V(R3:2)-V(Vin:-))

S D

68

Figure 2.5: Average Output voltage wave shape of conventional DC- DC Buck Boost


Figure 2.6: Average Input Current wave shape of conventional DC- DC Buck

Boostconverter (Duty Cycle = 0.4)

Time 80ms 90ms 100ms 110ms 120ms 130ms 0A


-10V

-20V

AVG(V(Vout))

70mA

60mA

40mA

20mA

AVG(-I(Vin))

69

Figure 2.7: AverageOutput Current wave shape of conventional DC- DC Buck Boost


Figure 2.8: Input voltage wave shape of conventional DC- DC Buck Boost

converter(Duty Cycle=0.8)

Time 80ms 90ms 100ms 110ms 120ms 130ms

AVG(V(R3:2) - V(Vin:-))

0V

10V

20V

30V


AVG(I(R1)) 0A

90mA

80mA

60mA

40mA

20mA

70

Figure 2.9: Output voltage wave shape of conventional DC- DC Buck Boost converter

(Duty Cycle=0.8)

Figure 2.10: Average Input Current wave shape of conventional DC- DC Buck Boost

Converter (Duty Cycle=0.8)


AVG(-I(Vin)) 0A

0.2A

0.4A

0.6A

0.8A

1.0A


-20V

-40V

-60V

-75V

AVG(V(Vout))

71

Figure 2.11: Average Output Current wave shape of conventional DC- DC Buck

Boostconverter(Duty Cycle=0.8)

3.1.3 Modified DC-DC Buck Boost Converter:

The topological difference of modified Buck Boost DC-DC converter is the

capacitor‟s connection which is connected from the diode to the positive side of input

voltage instead of been connected from the diode to the negative side of the input

voltage. The input current is continuous. The circuit arrangement of thistype of

converter is shown in Figure 2.12

Figure 2.12: Modified DC- DC Buck Boost converter


500mA

400mA

300mA

200mA

100mA

AVG(I(R1))

72

The operations of the modified converter during the period of switch ON/OFF are

shown through Figures 2.13 to 2.15.

When the Switch is ON and Capacitor Charging

When the switch is ON, Current flows through capacitor and inductor which occurred

capacitor charging and energized inductor.During switch ON period no current flows

through load. Current flows through the switch and capacitor which is shown in

Figure 2.13

Figure. 2.13: Modified DC- DC Buck Boost converter when the switch is ON and

capacitor charging

When the Switch is ON and Capacitor is Open

When the capacitor is fully charged, it is open and both the diodes are reverse

biased.Input current flows through the switch and inductor as shown in Figure 2.14.

Figure 2.14: Modified DC- DC Buck Boost converter whenthe switch is ON

D

D

D

S

S D

C1

+

73

When the Switch is OFF

When the switch is OFF, both the diodes are active and the inductor current flows

through load and the capacitordischarges. In traditional Buck Boost converter no input

current flows during switch off period but in modified Buck Boost converter we

observe that input current always flows which ensures continuity of input current

resulting continuous input current operation. The circuit diagram is shown in Figure

2.15

Figure 2.15: Modified DC- DC Buck Boost converterwhen the switch is OFF

3.1.4 Voltage Gain Expression of modified Buck-Boost DC-DC Converter

Figure 2.13 to 2.23 shows the equivalent circuit according with the switching state,

and some important waveforms in continuous conduction mode CCM.

Defining d as the duty cycle, the time when the switch is on over the total switching

period Tsand by using the small ripple approximation [6], the average voltage across

the inductor in steady state can be expressed as:

<vL(t) >=DVi+(1-D)(Vi-Vc) (2.3)

Note than the DC-component of variables d, vCand viare written as D, and VCand Vi

respectively, during the steady state, this average voltage as the average current in the

capacitor are equal zero, and then the voltage in the capacitor can be expressed as

DVi+(1-D)(Vi-Vc)=0

D

D S

+

74

Vc=Vin

(2.4)

The voltage in the capacitor is the same as in a traditional boost converter, but in this

case the output voltage is not given only by the capacitors voltage but also by the

input voltage, because the input voltage is in series with the capacitor voltage, and

then, considering the polarity signs defined inFigures 2.11, 2.12 and Figure2.13, the

output voltage can be expressed as:

Vo=VC-Vi=Vc=Vin

-Vi=Vin

(2.5)

The modified Buck Boost converter has the same conversion ratio as the traditional

buck-boost converter. The main advantage of the proposed converter can be seen in

Figures2.11 to 2.13, the input voltage is connected to the reference node with the

inductor and the load, both the inductor and the load drain a continuous current and

the input current becomes continuous.

By using the small ripple approximation, the average current in the capacitor can be

expressed as:

<ic(t) >=D(-

)+(1-D)(IL-

)

<ic(t) >= -

+ (1-D) IL (2.6)

During the steady state, this average current is equal zero, and then the current in the

inductor can be expressed as:

IL=

( ) (2.7)

By substituting (2.4) in (2.7) the DC-current in the inductor is expressed as:

IL=

( ) (Vi

-Vi) =

( ) (

-1)

IL=

( ) (2.8)

75

The switch and diode voltage and current stress can be calculated with a similar

procedure.

When the switch is open it blocks the capacitors voltage given by (2.4) which is the

same voltage for a switch in a traditional buck-boost converter and in the Ĉuk

converter, the current in the switch can be averaged from the switching states (Figure

2) and expressed as:

<is(t) >= DIL=

(

) (2.9)

As the inductor current in the input-series buck-boost converter is the same as in the

traditional buck-boost converter for converters rated to the same voltage and output

power, the current in the switch is the same and we can say the switch is identical in

the proposed topology than in the traditional buck-boost converter.

When the diode is open it blocks the voltage in the capacitor expressed in (2.11 and

2.12) and the same as in the traditional buck-boost converter, the average current can

be expressed as:

<iD(t) >= (1-D)IL=(1-D)

( ) =

(2.10)

This is also the same as in the traditional buck-boost converter. The steady state

analysis is then resumed in equations (2.4), (2.5), (2.8), (2.9) and (2.10), it can be seen

the proposed converter has the same inductor, transistor and diode as the traditional

buck-boost converter.

From the equations, the main disadvantage of the proposed topology can be seen; the

capacitor has the same voltage as the traditional boost converter, which is higher than

in the traditional buck-boost converter, so far this is the only disadvantage of the

discussed topology, this disadvantage is also present in one of the capacitors of the

Ĉuk and in the SEPIC converter which are the other buck-boost converters with

continuous input current.

76

3.1.5 Typical Simulated results of Modified Continuous input current Buck-

Boost DC-DC Converter

The circuit simulation for Buck and Boost operations of modified DC- DC Buck

Boostconverter for different duty cycles are shown through Figures 2.16 to2.23. It is

seen thatinput current is continuous and output voltage is inverted. It is also observed

thatat duty cycle less than 0.5 output voltage is step down whereas at duty cycle more

than0.5 the output voltage is step up.

Figure 2.16: Average Input voltage wave shape of Modified DC- DC Buck

Boostconverter (Duty Cycle=0.4)

Figure 2.17: AverageOutput voltage wave shape of Modified DC- DC Buck



-5V

-10V

-15V


10V

20V

25V


AVG(V(Vout))

77

Figure 2.18:Average Input Current wave shape of Modified DC- DC Buck


Figure 2.19: AverageOutput Current wave shape of Modified DC- DC Buck



50mA


20mA

40mA

80mA

60mA

100mA

AVG(-I(Vin))

140mA

100mA

AVG(I(R1))

78

Figure 2.20: AverageInput voltage wave shape of Modified DC- DC Buck


Figure 2.21: AverageOutput voltage wave shape of Modified DC- DC Buck



AVG(V(Vout))

0V

-20V

-40V

-60V

-80V

-90V


10V

20V

24V


79

Figure 2.22: AverageInput Current wave shape of Modified DC- DC Buck


Figure 2.23: AverageOutput Current wave shape of Modified DC- DC Buck



0A


AVG(-I(Vin))

0A

1.0A

2.0A

3.0A

4.0A

900mA 800mA

600mA

400mA

200mA

AVG(I(R1))

80

3.1.6Two quadrants Buck Boost DC-DC converter with continuous current:

The topological difference of Two Quadrant Buck Boost DC-DC converter is the

using two switches with diodes across them. The input current is continuous. The

circuit arrangement of thistype of converter is shown in Figure 2.24

Figure 2.24: Two quadrants Buck Boost DC-DC converter with Continuous Input

Current

The operations of the Two Quadrant Buck Boost Converter during the period of

switch ON/OFF are shown through Figures 2.25 to 2.27

When Switch S1ON, Switch S2 OFF

When the switch S1 is ON, Current flows through capacitor C1 and inductor L3

which energized inductor. During switch S1 ON period. Current flows through the

switch and capacitor which is shown in Figure 2.25


TD = 0 TF = .001m PW = .08m

PER = .2m V1 = 20 TR = .001m

V2 = 0

TD = 0 TF = .001m PW = .08m

PER = .2m V1 = 0 TR = .001m

V2 = 20


81

Figure 2.25: Two Quadrant DC- DC Buck Boost converterwhen the switch S1 is

ON,S2 is OFF andcapacitor charging

When Switch S1 OFF, Switch S2 ON

When the switch S1 is OFF and switch S2 is ONthe diodes D2 isinactive and the

inductor current flows through load. In traditional Buck Boost converter no input

current flows during switch off period but in modified Buck Boost converter we

observed that input current always flows which maintains continuity of input current

causing continuous input current operation. When duty cycle is less than 0.5current

flows through load in anti-clockwise direction and the circuit diagram as shown is

Figure 2.26. But when duty cycle is greater than0.5 the current flows through load is

clockwise direction which shown in Figure 2.27. Hence it‟s two quadrant operations.

82

Figure 2.26: Two Quadrant DC- DC Buck Boost converter when the switch S1 is

OFF,S2 is ON when Duty Cycle < 0.5

Figure 2.27: Two Quadrant DC- DC Buck Boost converterwhen the switch S1 is OFF,

S2is ON when Duty Cycle > 0.5

3.1.7 Ideal Voltage Gain expressions of Two Quadrant Buck-Boost DC-DC

converter with continuous Input Current:

For equation derivation we simplified the above circuit withinFigure 2.28 to 2.29

shows the equivalent circuit according with the switching state.

83

Figure 2.28: Simplified Two Quadrant DC- DC Buck Boost converterwhen the

switchS1 is ON, S2 is OFF

Figure 2.29: Simplified Two Quadrant DC- DC Buck Boost converterwhen the

switchS1 is OFF, S2 is ON

Defining d as the duty cycle, the time when the switch is on over the total switching

period Tsand by using the small ripple approximation [6], the average voltage across

the inductor in steady state can be expressed as:

<vL(t) >=DVin+(1-D)(Vin-Vc) (2.11)

The DC-component of variables d, vCand viare written in term ofD, and VCand Vi

during the steady state.Voltage on the capacitor can be expressed as:

DVin+(1-D)(Vin-Vc)=0

Vc=Vin

(2.12)

The voltage of the capacitor is the same as in a traditional boost converter, but in this

case the output voltage is not given only by the capacitors voltage but also by the

input voltage, because the input voltage is in series with the capacitor voltage, and

84

then, considering the polarity signs defined in Figure 2.11, 2.12 and Figure 2.13, the

output voltage can be expressed as:

Vo=VC-Vi=Vc=Vin

-Vi=Vin

(2.13)

The modified Buck Boost converter has the same conversion ratio as the traditional

buck-boost converter. The advantage of the proposed converter can be seen in Figures

2.11 to 2.13, the input voltage is connected to the reference node with the inductor

and the load, both the inductor and the load have continuous current and the input

current is continuous.

By using the small ripple approximation, the average current in the capacitor can be

expressed as:

<ic(t) >=D(-

)+(1-D)(IL-

)

<ic(t) >=-

+ (1-D) IL (2.14)

During the steady state, this average current is equal zero, and then the current in the

inductor can be expressed as:

IL=

( ) (2.16)

By substituting (2.12) in (2.16) the DC-current in the inductor is expressed as:

IL=

( ) (Vi

-Vi) =

( ) (

-1)

IL=

( ) (2.17)

The switch and diode voltage and current stress can be calculated with a similar

procedure.

When the switch is open it blocks the capacitors voltage given by (2.12) which is the

same voltage of a switch in a traditional buck-boost converter and in the Ĉuk

converter, the current in the switch can be averaged from the switching states (Figure

2) and expressed as:

85

<is(t) >= DIL=

(

) (2.18)

As the inductor current in the input-series buck-boost converter is the same as in the

traditional buck-boost converter for converters rated of the same voltage and output

power, the current in the switch is the same and we can say the switch is identical in

the proposed topology as in the traditional buck-boost converter.

When the diode is open it blocks the voltage in the capacitor expressed in (2.12 and

2.13) and the same as in the traditional buck-boost converter, the average current can

be expressed as:

<iD(t) >= (1-D)IL=(1-D)

( ) =

(2.19)

This is also the same as in the traditional buck-boost converter. The steady state

analysis is then resumed in equations (2.12), (213), (2.17), (2.18) and (2.19), it can be

seen the proposed converter has the same inductor, transistor and diode currentas the

traditional buck-boost converter.

From the equations, the disadvantage of the proposed topology can be seen as the

capacitor has the same voltage as the traditional boost converter, which is higher than

in the traditional buck-boost converter. This disadvantage is also presentin the Ĉuk

and SEPIC converter.

3.1.8Typical Simulation Results of Proposed Modified Two quadrants Buck-

Boost DC-DC converter with continuous current

The proposed continuous input current two quadrant Buck-Boost DC-DC converter

has been studied for R, R-L and R-L-Emf loads. Some typical simulation results of

the study are presented in the following subsections.

86

3.1.8.1 Proposed Buck-Boost DC-DC converter with R-Load

Figure 2.30 shows the simulation circuit of the proposed continuous input current

based two quadrant Buck-Boost converter with resistive load. In the simulation circuit

input source voltage is 20Vdc and initial load is 100Ω. The input voltage waveform

during switched control for duty cycle D = 0.4 is shown in Figure 2.31 with

corresponding average input voltage of 20V shown in Figure 2.32. The time variation

of the input current is shown in Figure 2.33 which is continuous in nature having both

positive and negative current of the source. The average current is dc as shown in

Figure 2.34 and the current is flowing out of the source the instantaneous output

voltage, average output voltage, instantaneous output current ( of the R-Load) and the

average output current ( of the R-Load) are shown in Figures 2.35 to 2.38

respectively. The output voltage is – ve and output current is+ve indicating quadrant

one of the converter which it should be for R-Load. Figures2.39 to 2.46 shows the

simulation results of the proposed DC-DC converter of Figure 2.30 for duty cycle of

0.8 operation. The result being increase in all voltages and currents in their respective

direction. Increase of duty cycle causes inverted dc output voltage to increase which

causes flow of large negative current in the load due to inverted voltage and as the

output increases current increases to maintain power balance of the circuit.

Figure 2.30: Two quadrant DC- DC Buck Boost converter with Resistive Load

TD = 0 TF = .001m PW = .08m PER = .2m V1 = 0 TR = .001m V2 = 20

TD = 0 TF = .001m PW = .08m PER = .2m V1 = 20 TR = .001m

V2 = 0


87

Figure 2.31: Input Voltage wave shape of Two quadrant DC- DC Buck Boost

converter (Duty Cycle=0.4)

Figure 2.32: Average Input Voltage wave shape of Two quadrant DC- DC Buck

Boost converter (Duty Cycle=0.4)

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

AVG(V(R1:2,Vin:-))

0V

10V

20V

24V

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

V(R1:2,Vin:-)

-80V

-40V

0V

40V

80V

120V

160V

88

Figure 2.33: Input Current wave shape of Two quadrant DC- DC Buck Boost


Figure 2.34: Average Input Current wave shape of Two quadrant DC- DC BuckBoost


Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

AVG(I(R1))

0A

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

0A

400mA

200mA

-200mA

-400mA

I(R1)

80mA

60mA

40mA

20mA

89

Figure. 2.35: Output Voltage wave shape of Two quadrant DC- DC Buck


Figure 2.36: Average Output Voltage wave shape of Two quadrant DC- DC Buck


Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

0V

-4.0V

-8.0V

-10.0V

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

V(R5:2,R5:1)

-60V

-40V

-20V

-0V

20V

AVG(V(R5:2,R5:1))

90

Figure 2.37: Output Current wave shape of Two quadrant DC- DC Buck


Figure 2.38: Average Output Current wave shape of Two quadrant DC- DC Buck


Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

AVG(I(R5))

0A

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

I(R5)

0A

600mA

400mA

200mA

-200mA

80mA

60mA

40mA

20mA

91

Figure 2.39: Input Voltage wave shape of Two quadrant DC- DC Buck



BoostConverter (Duty Cycle=0.8)


AVG(V(R1:2,Vin:-))

0V

10V

20V

24V


V(R1:2,Vin:-)

-200V

-100V

0V

100V

200V

300V

400V

92

Figure 2.41: Input current wave shape of Two quadrant DC- DC Buck Boostconverter

(Duty Cycle = 0.8)

Figure 2.42: Average Input Current wave shape of Two quadrant DC- DC Buck Boost



0A


-0.8A

-0.4A

0A

0.4A

0.8A

1.2A

-I(Vin)

600mA

400mA

200mA

AVG(-I(Vin))

93

Figure 2.43: Output Voltage wave shape of Two quadrant DC- DC Buck Boost



Boost converter (Duty Cycle = 0.8)


AVG(V(R5:2,R5:1))

0V

-4V

-8V

-12V

-16V


V(R5:2,R5:1)

-200V

-160V

-120V

-80V

-40V

-0V

40V

94

Figure 2.45: Output Current wave shape of Two quadrant DC- DC Buck Boost





0A

50mA


I(R5)

0A

0.5A

1.0A

1.5A

2.0A

-0.4A

150mA

100mA

AVG(I(R5))

95

3.1.8.2 Proposed Buck-Boost DC-DC converter with R-Load(Source and load

interchanged in position)


based two quadrant Buck-Boost converter with R-load. In figure 2.47 the source and

R-Load position of Figure 2.30 has been interchanged so as to show that the proposed

circuit operates in Quadrant-2 with R-load(in bidirectional direction). The input

voltage waveform during switched control for duty cycle D = 0.4 is shown in Figure

2.48 with corresponding average input voltage of 20V shown in Figure 2.49. The time

variation of the input current is shown in Figure 2.50 which is continuous in nature

having both positive and negative current of the source. The average input current is

dc as shown in Figure 2.51 and the current is flowing out of the source. The

instantaneous output voltage, average output voltage, instantaneous of output current (

of the R-Load) and the average output current ( of the R-Load) are shown in Figures

2.52 to 2.55 respectively. The output voltage is (+ ve)and output current is (– ve)

indicating quadrant two of the converter which it should be for R-Load. Figures 2.56

to 2.63 shows the simulation results of the proposed DC-DC converter of figure 2.47

for duty cycle of 0.8 operation. The result being increase in all voltages and currents

in their respective direction. Increase of duty cycle causes inverted dc output voltage

to increase which causes flow of negative current in the load due to inverted voltage

and as the output increases current increases to maintain power balance of the circuit.

Figure 2.47: Two quadrant DC- DC Buck Boost converter withR-Load (Source and

load interchanged in position)



VOFF=0 VAMPL=20

FREQ=50

96






0V

10V

20V

24V


-80V

-40V

0V

40V

80V

120V

160V

V(Vin:+,R5:2)

AVG(V(Vin:+,R5:2))

97






AVG(-I(Vin))

0A


0A

400mA

200mA

-200mA

-400mA

-I(Vin)

80mA

60mA

40mA

20mA

98






0V

4.0V

8.0V

10.0V


-20V

0V

20V

40V

60V

V(R1:2,R1:1)

AVG(V(R1:2,R1:1))

99






0A


I(R1)

0A

200mA

-200mA

-400mA

-600mA

-80mA

-60mA

-40mA

-20mA

AVG(I(R1))

100






0V

10V

20V

24V


-200V

0V

200V

400V

600V

V(Vin:+,R5:2)

AVG(V(Vin:+,R5:2))

101






0A


-I(Vin)

-0.4A

0A

0.4A

0.8A

1.2A

-0.8A

700mA

600mA

400mA

200mA

AVG(-I(Vin))

102






AVG(V(R1:2,R1:1))

0V

5V

10V

15V


0V

50V

150V

200V

-40V

100V

V(R1:2,R1:1)

103






AVG(I(R1))

0A

-50mA


-1.6A

-0.8A

-0.0A

-2.0A

0.4A

I(R1)

-150mA

-100mA

104

3.1.8.3Proposed Buck-Boost DC-DC converter with R-L Load


based two quadrant Buck-Boost converter with R-L load. In the simulation circuit

input source voltage is 20Vdc and initial load is 50Ω. The input voltage waveform

during switched control for duty cycle D = 0.4 is shown in Figure 2.65 with

corresponding average input voltage of 20V shown in Figure 2.66. The time variation

of the input current is shown in Figure 2.67 which is continuous in nature having both

positive and negative current of the source. The average input current is dc as shown

in Figure 2.68 and the current is flowing out of the source the instantaneous output

voltage, average output voltage, instantaneous output current ( of the R-L Load) and

the average output current ( of the R-L Load) are shown in Figures 2.69 to 2.72

respectively. The output voltage is – veand output current +ve indicating quadrant one

of the converter which it should be for R-L Load. Figures 2.73 to 2.80 shows the

simulation results of the proposed DC-DC converter of figure 2.64 for duty cycle of

0.8 operation. The result being increase in all voltages and currents in their respective

direction. Increase of duty cycle causes inverted dc output voltage to increase which

causes flow of negative current in the load due to inverted voltage and as the output

increases current increases to maintain power balance of the circuit.

Fig. 2.64: Two quadrant DC- DC Buck Boost converter with R-L Load



VOFF=0 VAMPL=20

FREQ=50

105



Figure 2.66:Average Input Voltage wave shape of Two quadrant DC- DC Buck Boost


Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

0V

10V

20V

24V

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

-50V

0V

50V

100V

150V

V(R1:2,Vin:-)

AVG(V(R1:2,Vin:-))

106





Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

0A

40mA

80mA

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

-0mA

200mA

400mA

-200mA

-400mA

-I(Vin)

120mA

AVG(-I(Vin))

107





Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

0V

-4.0V

-8.0V

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

-50V

-0V

50V

-100V

-150V

V(L3:1,R5:1)

-10.0V

AVG(V(L3:1,R5:1))

108





Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

AVG(I(R5))

0A

50mA

100mA

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

0A

200mA

400mA

600mA

800mA

-200mA

I(R5)

150mA

109





Time 120ms 130ms 140ms 150ms 160ms 170ms 180ms 190ms 200ms

0V

10V

20V

24V


-200V

-100V

0V

100V

200V

300V

(V(R1:2,Vin:-))

AVG(V(R1:2,Vin:-))

110






AVG(-I(Vin))

0A


-1.0A

-0.5A

0A

0.5A

1.0A

1.5A

-1.5A

-I(Vin)

750mA

250mA

500mA

950mA

111






0V

-5V

-10V

-15V


V(L3:1,R5:1)

-300V

-200V

-100V

0V

100V

AVG(V(L3:1,R5:1))

112






AVG(I(R5))

0A


0A

1.0A

2.0A

-0.5A

2.5A

I(R5)

300mA

200mA

100mA

113

3.1.8.4 Proposed Buck-Boost DC-DC converter with R-L Load(Source and load



based two quadrant Buck-Boost converter with R-L load. In figure 2.81 the source

and R-L load position of Figure 2.64 has been interchanged so as to show that the

proposed circuit operates in Quadrant-2 with R-L load (operates in bidirectional

direction). The input voltage waveform during switched control for duty cycle D = 0.4

is shown in Figure 2.82 with corresponding average input voltage of 20V shown in

Figure 2.83. The time variation of the input current is shown in Figure 2.84 which is

continuous in nature having both positive and negative current of the source. The

average input current is dc as shown in Figure 2.85 and the current is flowing out of

the source the instantaneous output voltage, average output voltage, instantaneous of

output current ( of the R-L Load) and the average output current ( of the R-L Load)

are shown in Figures 2.86 to 2.89 respectively. The output voltage is (+ ve) and

output current is (– ve) indicating quadrant two of the converter which it should be for

R-L Load. Figures 2.90 to 2.97 shows the simulation results of the proposed DC-DC

converter of figure 2.47 for duty cycle of 0.8 operation. The result being increase in

all voltages and currents in their respective direction. Increase of duty cycle causes

inverted dc output voltage to increase which causes flow of negative current in the

load due to inverted voltage and as the output increases current increases to maintain

power balance of the circuit.

Figure 2.81: Proposed Buck-Boost DC-DC converter with R-L Load(Source and load




VOFF=0 VAMPL=20

FREQ=50

114

Figure 2.82: Input Voltagewave shape of Two quadrant DC- DC Buck Boost





0V

4V

8V

12V

16V


-20V

0V

20V

40V

60V

V(Vin:+,L3:1)

80V

AVG(V(Vin:+,L3:1))

115

Figure 2.84: Input Currentwave shape of Two quadrant DC- DC Buck Boost





0A


0A

100mA

200mA

-100mA

-200mA

-I(Vin)

AVG(-I(Vin))

20mA

40mA

60mA

70mA

116

Figure 2.86: Output Voltagewave shape of Two quadrant DC- DC Buck Boost





0V

0.25V

0.50V

0.75V

1.00V


V(L1:1,R1:1)

-50V

0V

25V

50V

-25V

AVG(V(L1:1,R1:1))

117

Figure 2.88: Output Currentwave shape of Two quadrant DC- DC Buck Boost

Converter (Duty Cycle = 0.4)


Boost Converter (Duty Cycle = 0.4)


0A

-20mA

-40mA

-60mA

-80mA


0mA

100mA

-100mA

-200mA

-300mA

I(R1)

AVG(I(R1))

-100mA

118

Figure 2.90: Input Voltagewave shape of Two quadrant DC- DC Buck Boost





0V

2.0V

4.0V

6.0V


-20V

0V

20V

40V

60V

80V

V(Vin:+,L3:1)

AVG(V(Vin:+,L3:1))

119






0A

40mA

80mA


-I(Vin)

0A

100mA

200mA

-200mA

-100mA

160mA

120mA

AVG(-I(Vin))

120






AVG(V(L1:1,R1:1))

0V


-20V

0V

20V

40V

60V

V(L1:1,R1:1)

100mV

200mV

300mV

400mV

450mV

121






0A

-20mA

-40mA

-50mA


I(R1)

0mA

100mA

-400mA

-300mA

-200mA

-100mA

AVG(I(R1))

122

3.1.8.5Proposed Buck-Boost DC-DC converter with R-L-Emf Load


based two quadrant Buck-Boost converter with R-L-Emf load. In the simulation

circuit input source voltage is 20Vdc and initial load is 100Ω. The input voltage

waveform during switched control for duty cycle D = 0.4 is shown in Figure 2.99 with

corresponding average input voltage of 20V shown in Figure 2.100. The time

variation of the input current is shown in Figure 2.101 which is continuous in nature

having both positive and negative current of the source. The average input current is

dc as shown in Figure 2.102 and the current is flowing out of the source the

instantaneous output voltage, average output voltage, instantaneous output current ( of

the R-L-Emf Load) and the average output current ( of the R-L-Emf Load) are shown

in Figures 2.103 to 2.106 respectively. The output voltage and output current both –

ve indicating quadrant one of the converter which it should be for R-L-Emf Load.

Figures 2.107 to 2.114 shows the simulation results of the proposed DC-DC converter

of figure 2.98 for duty cycle of 0.8 operation. The result being increase in all voltages.

Increases dutycycle causes inverted change of output current. At Duty cycle 0.8

output voltage is negative but output current is positive which ensure quadrant II

operation. In this scenario same circuit is operated in different quadrant operation

based on duty cycle.

Figure 2.98: Two quadrant DC- DC Buck Boost converter with R-L-Emf Load



VOFF=0 VAMPL=20

FREQ=50

VOFF=0 VAMPL=20

FREQ=50

123





Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

AVG(V(R1:2,Vin:-))

0V

10V

20V

24V

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

V(R1:2,Vin:-)

-50V

0V

50V

100V

150V

124



Figure 2.102: Average Input Current wave shape of Two quadrant DC- DC Buck


Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

AVG(-I(Vin))

0A

-200uA

-400uA

-550uA

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

-I(Vin)

0A

200mA

300mA

-200mA

-300mA

125





Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

AVG(V(R5:2,Vbatt:+))

0V

-5V

-10V

-15V

-20V

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

-50V

-40V

-30V

-20V

-10V

0V

V(R5:2,Vbatt:+)

126





Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

AVG(I(R5))

0A

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

I(R5)

0A

300mA

-200mA

-40mA

-50mA

-20mA

127





Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

AVG(V(R1:2,Vin:-))

0V

10V

20V

24V

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

V(R1:2,Vin:-)

-200V

0V

200V

400V

128



Figure 2.110: Average Input Current wave shape of Two quadrant DC- DC Buck


Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

AVG(-I(Vin))

0A

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

-I(Vin)

-1.0A

-0.5A

0A

0.5A

1.0A

500mA

400mA

200mA

129





Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

AVG(V(R5:2,Vbatt:+))

0V

-10V

-20V

-30V

-35V

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

V(R5:2,Vbatt:+)

-200V

-150V

-100V

-50V

-0V

130





Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

AVG(I(R5))

0A

50mA

Time 0.90s 0.92s 0.94s 0.96s 0.98s 1.00s

I(R5)

-0.5A

0A

0.5A

1.0A

1.5A

120mA

100mA

131

Chapter-4

CONCLUSIONS

4.1 FINDINGS, ACHEIVEMENTS

The advances in the power semiconductor devices have led to the increase in the use

of power electronic converters in various applications such as heating, lighting,

ventilating and air conditioning applications, large dc and ac adjustable speed drives,

uninterruptible power supplies, high voltage DC systems, utility interfaces with non-

conventional energy sources such as solar photovoltaic systems, battery energy

storage systems, in process technology such as electroplating, welding units, battery

charging for electric vehicles and power supply for telecommunications systems etc..

Switching DC-DC converters have become part of electronic equipment‟s to provide

regulated dc of desired voltages at low cost and high efficiency. These converters

have advantages over their counterpart the linear power supplies. Main advantage

istheir light weight and small size due to high frequency operation. These converters

have high efficiency because the regulating devices in them work as switches

ensuring low device loss. Their output voltage can be controlled for a wide range of

input voltage fluctuation by changing the duty cycle of the switching signals.

The dc/dc converters are widely used in industrial applications and computer

hardware circuits. Four common types of switch mode converters are used in DC to

DC conversion. They are buck, boost, buck-boost and ĈUK converters.

In this research Spice simulation have been carried out for a modified 2-Q Buck Boost

Converter considering R, R-L and Battery loads. During R and R-L loads (load and

source have been interchanged to study multiquadrant (2Q) operation. The proposed

circuit has capability to work in two quadrant mode. When we consider Battery load,

same circuit operates in two quadrant mode based on duty cycle variation. When duty

cycle is less than 0.5 output current is from load to source and when duty cycle is

greater than 0.5 output current is from source to load.

132

By the changing the duty cycle from 0.4 to 0.8, it was found that the necessary

characteristics of operational output of two quadrant chopper mode of forward and

reverse Converters (Figure 2.22) as shown in Tables 3.1,3.2 and 3.3 have been

achieved.

Table 3.1: Operational output of Two Quadrant Chopper mode of forward and reverse

Converter of circuit of Figure 2.28

Sl

no. Operational mode

Load

Voltage

Output Current,

IL Quadrant

01 Operation -1 (Figures

2.29 - 2.45) Negative Positive IV

02 Operation -2 (Figure

2.46 - 2.63) Positive Negative II



Sl

no. Operational mode

Load

Voltage

Output

Current, IL Quadrant


2.65 - 2.81) Negative Positive IV


2.82 - 2.99) Positive Negative II

133



Sl

no. Operational mode Load Voltage

Output

Current, IL Quadrant


2.101 - 2.108) Negative Negative III


2.109 - 2.11) Negative Positive II

4.2 CONCLUSION

Buck-Boost dc-dc converter topology has minimum component count in its class to

have ideal voltage gain relationship

=

. The other two dc-dc converters which

have Vo=

relationship are the Ĉuk and SEPIC converters. Ĉuk and SEPIC

converters have extra coupling capacitor and output inductor between input and

output of the circuit. Though the Buck-Boost dc-dc converter has ideal

=

relationship, it has discontinuous input current. To make Buck-Boost dc-dc

converters input current continuous, topology change has been proposed in the past in

literature. The one quadrant continuous input current Buck-Boost dc-dc converter

proposed in the literature is modified for two quadrant operation in this research. It

has been established by the study that the proposed quadrant Buck-Boost dc-dc

converter with continuous input current works as suggested. The proposed circuit

don‟t have extra output inductor for buck operation as used in Ĉuk abs SEPIC dc-dc

converters. The circuit uses a coupling capacitor for making input current continuous

but it is not a coupling capacitor in true sense as used in Ĉuk and SEPIC dc-dc

converter. The proposed Buck-Boost circuit may have applications in renewable

energy conversion for battery charging and in electric drives of Electric vehicles.

134

4.3 FUTURE SCOPE OF WORK

The contributions of this thesis indicate the opportunities of extending this work in

future to meet other goals.

1. Only Spice simulation is performed in this study. The proposed new TWO

QUADRANT BUCK BOOST DC-DC COVERTER WITH CONTINUOUS

INPUT CURRENT may be implemented practically to investigate its actual

potential. Such practical implementation would give an insight regarding the

cost effectiveness of the proposed scheme compared to the existing schemes

for the similar purpose.

2. The PWM module has been used to generate gating signals for switching the

proposed converter switches at varying duty cycles. Investigation can be made

to improve the quality of the gating signals at different duty cycle.

135

REFERENCES:

[1] R. P. Severns and G. Bloom, "Modern DC-to-DC Switch mode Power

Converter Circuits.", New York: Van Nostrand, 1985.

[2] G.Ch.Ioannidis and S. N. Manias “AC-DC and DC-DC Converters For DC

Motor Drives: Review of Basic Topologies” Proceedings of the 2013

International Conference on Electronics and Communication Systems, PP 96-

103,2013.

[3] J.C.Mayo-Maldonado and Rivera Barrios “Dynamic Modeling and Current

Mode Control of a Continuous Input Current Buck-Boost DC-DC Converter”

Proceedings of the World Congress on Engineering and Computer Science

2011 Vol I VCECS 2011, San Francisco USAPP,2011.

[4] M. H. Rashid Hand book-“Power Electronics Handbook” Academic Press,

Chapter-13 Page 211-223,2001.

[5] L. TimothySkvarenina -“The Power Electronics Handbook” @2002 CRC Press

LLC, Chapter-2,2002.

[6] Ned Mohan -“First Course on Power Electronics And Drives” Year 2003

edition Published by MNPERE, Chapter-3 Page 3-1 to 3-32,2003.

[7] M. H.Rashid- “Power Electronics: Circuits, Devices, and Applications”

3rdEdition by Pearson Education, Inc.Chapter-5 PP 166-220,2004.

[8] D. W. Hart, "Introduction to Power Electronics." Upper Saddle River, NJ:

Prentice Hall, 1997.

[9] MohammadMazharul Islam“Analysis of a New Four Quadrant Switch Mode

Dc-Dc Converter”M.Sc thesis EEE,BUET,2011.

136

[10] J. C. Rosas-Caro, J. C. Mayo-Maldonado, J. E. Valdez-Resendiz, R. Salas-

Cabrera, H. Cisneros-Villegas, R. Castillo-Ibarra, J. G. Gonzalez-

Hernandez“Design-Oriented Analysis and Modeling of a Single-Inductor

Continuous Input-Current Buck-Boost DC-DC Converter”, WCECS 2011, San

Francisco, USA,2011.

[11] R.Venkat“Switch Mode Power Supply”, University of Technology, Sydney,

Australia, available at http://www.ee.uts.edu.au/~venkat/pe_html

/pe07_nc8.htm,2001.

[12] J. P.Agrawal, “Power Electronics Systems: Theory and Design”, Prentice-Hall,

Upper Saddle River, NJ, chap. 6, 2001.

[13] Barry W. Williams "Principle and elements of Power Electronics-Devices,

Drivers, Applications and passive Components" Chapter 14,Page-613, 2006.

[14] P. T. Krein "Elements of Power Electronics", New York: Oxford University

Press, 1998.

[15] R. W. Erickson and D. Maksimovi´c, "Fundamentals of Power Electronics",

2nd edn. Norwall, MA: Kluwer Academic, 2001.

[16] W. Shepherd and L. Zhang, "Power Converter Circuits", New York: Marcel

Dekker, 2004.

[17] S. Ang and A. Oliva, "Power-Switching Converters", 2nd edn. Boca Raton, FL:

CRC/Taylor & Francis, 2004.

http://www.ee.uts.edu.au/

137

[18] F L Luo, and L. Jin, “Two-quadrant DC/DC Soft-switching Converter,”

Proceedings of IEEE Power Electronics Specialists Conference, vol. 1, pp.173-

178, 2000.

[19] S. Waffler and J.W. Kolar, “Comparative Evaluation of Soft-Switching

Concepts for Bi-directional Buck Boost DC-DC Converters,” International

Power Electronics Conference (IPEC), 21-24 June, 2010.

[20] PremanandaPany, R.K. Singh, R.K. Tripathi, “Bidirectional DC-DC converter

fed drive for electric vehicle system”, International Journal of Engineering,

Science and Technology, Vol. 3, No. 3, pp. 101-110, 2011.

[21] S. Canter and R. J. Lenk, „„Buck-boost dc–dc switching power conversion,‟‟

U.S. Patent 5359280, Feb.4, 1994.

[22] K. Rajashekara, “Power converter and control strategies for fuel cell vehicles,”

in Proc. IEEE IECON, Bologna, Italy, vol.3, pp.2865-2870,2003.

[23] A.D. Swingler and W.G. Dunford, “Development of a bi-directional dc/dc

converter for inverter/charger applications with consideration paid to large

signal operation and quasi-linear digital control,” in Proc. IEEE PESC, Cairns,

Queensland, Australia, Volume 2, pp. 961 – 966,2002.

[24] M. Jain, P.K. Jain, and M. Daniele, “Analysis of a bi-directional dc-dc

converter topology for low power application,” in Conf. Rec. of IEEE 1997

CanadianConference on Electrical and Computer Engineering, St. John‟s, NF,

Volume 2 , pp. 548 – 551,1997.

[25] R.W. De Doncker, D.M. Divan, and M.H. Kheraluwala, “A three-phase

softswitched high power density dc-to-dc converter for high power

applications,” IEEETran. Ind. Appl., Vol. 27, No. 1, pp.63-73,1991.

138

[26] M.H. Kheraluwala, and D.M. Divan, “Design considerations for high power

density DC-DC converters,” in Proc. High Frequency Power Conversion

(HFPC, pp.324-335,1990.

[27] F. Caricchi, F. Crescimbini, G. Noia, and D. Pirolo, “Experimental study of a

bidirectional dc-dc converter for the dc link voltage control and the

regenerative braking in PM motor drives devoted to electrical vehicles ,” in

Proc. IEEE APEC, Orlando, FL, pp. 381 – 386,1994.

[28] H.J. Chiu, H. M. Huang, L. W. Lin, and M. H. Tseng, “A multiple-input dc/dc

converter for renewable energy systems,” in Proc. IEEE ICIT, Hong Kong,

China, pp. 1304 – 1308,2005.

[29] H. Chung, W. Chow, S. Hui, and S. Lee, “Development of a Switched-

Capacitor DC-DC Converter with Bidirectional Power Flow,” IEEE Trans. on

Circuits andSystems- I: Fundamental Theory and Applications, vol. 47, no. 9,

pp. 1383-1389, Sep. 2000.

[30] S. MakowskiMarek, and DraganMaksimovic, “Performance Limits of

Switched- Capacitor DC-DC Converters,” IEEE/PESC, vol. 2, pp. 1215-1221,

June 1995.

[31] S. V. Cheong and H. Chung, “Inductorless DC-to-DC Converter with High

Power Density,” IEEE Trans. of Industrial Electronics, vol. 41, no. 2, pp. 208-

15,1994.

[32] TohruUmeno, Kiyoshi Takahashi, Ichirou Oota, Fumio Ueno, and Takahiro

Inoue, “New Switched-Capacitor DC-DC Converter With Low Input Current

and Its Hybridization,” Proceedings of the 33rd Midwest Symposium on

Circuits and Systems, vol. 2, pp. 1091-1094,1990.

139

[33] Ju Won Baek, MyungHyoRyoo, Tae-Jin Kim, Dong-WookYoo, and Jong-Soo

Kim, “High Boost Converter with Voltage Multiplier,” IEEE/IECON, pp. 567-

572, 2005.

[34] Danwei Liu, and Hui Li, “A Novel Multiple-Input ZVS Bidirectional DC-DC

Converter,” IEEE/IECON, pp. 579-584, Nov. 2005.

[35] M. TolbertLeon, A. PetersonWilliam, P. WhiteCliff, J. TheissTimothy, and B.

ScudiereMatthew, “A Bi-Directional DC-DC Converter with Minimum Energy

Storage Elements,” IEEE/IAS, vol. 3, pp. 1572-1577, Oct. 2002.

[36] R. D. Middlebrook, “Transformerless DC-to-DC Converters with Large

Conversion Ratios,” IEEE Trans. of Power Electronics, vol. 3, no. 4, pp. 484-

488,1998.

[37] R. W. Erickson and D. Maksimovic, “Fundamental of power electronics,”

Kluwer Academic Publishers, 2001.

[38] J. G. Kassakian, M. F. Schlecht and G. C. Verghese, Principles of Power

Electronics, Addison-Wesley, 1991.

[39] Robert Erickson, DraganMaksimovic, Fundamentals of Power Electronics, 2nd

edition, Kluwer Academic Publishers, 2001.

[40] W. Harris, K. Ngo, “Power Switched-Capacitor DC-DC Converter, Analysis

and Design,” IEEE Transactions on Aerospace and Electronic Systems, vol. 33,

no. 2, pp. 386-395, April 1997.

[41] S. Cûk and R. D. Middlebrook, "Advances in Switched Mode Power

Conversion," IEEE Transactions on Industrial Electronics, Vol. IE 30. No. 1,

pp.10-29,1983.

140

[42] Ned Mohan, M. UndelandTore and P.RobbinsWilliam, “Power Electronics-

Converters, Applications, and Design, " John Wiley and Sons Inc., Second

ed.,pp. 161-195 & 669-695,1995.

[43] “Loss Recovery,” IEEE Transactions on Power Electronics, 1994.

[44] S. B. Dewan and A. Straughen, “Power Semiconductor Circuits,” Wiley-

Interscience, 1975.

[45] Slobodan Cûk, "Basics of Switched Mode Power Conversion Topologies,

Magnetics, and Control,” Modern Power Electronics: Evaluation, Technology,

and applications, Edited by B.K. Bose, IEEE Press, pp.265-296,1992.

[46] P.C. Sen, "Power Electronics,” Tata McGraw-Hill Publishing Company Ltd.,

India, pp. 588-614.1987.

[47] E. J. Cham and W. R. Roberts, “Current Regulators for Large Rectifier Power

Supplies Used on Electrochemical Processing Lines,” IEEE Trans. on Ind. and

Gen. Application IGA-4 6, pp. 609-618,1968.

[48] S. Cûk, "Survey of Switched Mode Power Supplies,” IEEE International

Conference on Power Electronics and Variable Speed Drives, London, pp. 83-

94,1985.

Documents

BUET - Semantic Scholar · (BUET) and Dr. Md. Ziaur Rahman Khan,professor, Department of Electrical and Electronic Engineering, Bangladesh University of Engineering and Technology