silica fibre optical amplifier was used as an auxiliary pump source. The signal source was unchanged. The output from all three sources was passed through the optical isolator and into the fibre as before. 91mW of 1.48pm pump power was launched of which 82mW was absorbed. The launched signal was again held constant at -31dBm but this time, for each wavelength step, the input auxiliary pump power was varied between -30 and + 1 dBm. The gain and noise figure were obtained as before.

t

I . - 1570 l;80 I590 1600 1610 wavelength ,nm

Fig. 2 Gain spectra of I 4 p m amplifier pumped singly with 87mW of pump power at 1.555pm only, and dual pumped with 82mW at I.48pm with auxiliary powers of + I , -8 , - 17 and -21 dBm 0 1.555pm, single pump V 1.48 pm, 1 dBm auxiliary pump 0 1.48 pm, - 8 dBm auxiliary pump W 1.48 pm, - 17dBm auxiliary pump A 1.48 pm, -21 dBm auxiliary pump

Results: Fig. 2 shows the gain spectrum of the 1.55pm pumped amplifier and the evolution of the gain spectra of the dual pumped amplifier as a function of auxiliary input level. The singly pumped amplifier spectrum corresponds to that of a dual pumped amplifier with an auxiliary input power of approximately -19dBm. A maximum gain of 31dB is achieved in the dual pumped case with an auxiliary power level of -9dBm whereas for an auxiliary input power of - 17dBm a particularly flat spectrum is obtained showing less than 1 dB gain variation between 1.57 and 1.6pm.

The noise figure was found to vary by less than 1 dB with auxiliary input level over the range measured. The noise figure spectra of the singly pumped amplifier and the dual pumped amplifier with an auxiliary level of -17dBm are shown in Fig. 3. The lines are the minimum noise figures for amplifiers pumped at 1.48 and 1.555pm as predicted by eqn. 2. The noise performance of the dual pumped amplifier corresponds well to the minimum expected from a 1.48 pm pumped ampli- fier in the absence of ASE: a 5dB improvement is achieved with respect to the measured and expected noise figures for an amplifier pumped at 1,555~111 only. The increase in noise figure beyond a 1,602pm is believed to be caused by excited state absorption from the 4113,2 to 419,2 levels [SI.

t

Conclusions: A configuration for the 1.6 pm amplifier combin- ing high gains with good noise performance has been demon- strated. Using 82mW of pump power at 1.48pm together with an auxiliary pump of -9dBm at 1.555pm, a maximum gain of 31 dB at 1.57 pm has been achieved. With adjustment of the auxiliary power level, a gain of 24dB with less than 1 dB variation is obtained over a bandwidth of 30 nm. The noise figure of this dual pumped amplifier approaches the minimum possible for a 1.48pm pumped device, remaining below 5dB over a bandwidth of 35nm. This represents an improvement in noise figure of up to 5 dB over a similar using a single 1.555 pm pump.

17th August I992 J. F. Massicott, R. Wyatt and B. J. Ainslie (British Telecom Labor- atories, Martlesham Heath, Ipswich IPS 7RE, United Kingdom)

References ATKINS, C. G., MASSICOTT, J . F., ARMITAGE, J. R., WYATT, R., AINSLIE, B. J., and CRAIG-RYAN, s. P.: ‘High-gain, broad spectral bandwidth erbium-doped fibre amplifier pumped near 1.5 pm’, Electron. Lett., 1989,14, pp. 910-911 MASSICOTT, I., ARMITAGE, I . R., W Y A T ~ , R., AINSLIE, B. I., and CRAIG- RYAN, s. P.: ‘High gain, broadband, 1.68111 Er3+ doped silica fibre amplifier’, Electron. Lett., 1990,20, pp. 164-1646 GILES, C. R., OESURVIRE, E., ZYSKIND, I. L., and SIMPSON, I. R.: ‘Noise performance of erbium-doped fibre amplifier pumped at 1,49pm, and application to signal preamplification at 1.8 Gbit/s’, IEEE Photonics Technol. Lett., 1989, 11, pp. 367-369

M., TACHIKAWA, Y., and SUGITA, E. : ‘Noise characteristics of Er3+- doped fibre amplifiers pumped by 0.98 and 1.48pm laser diodes’, IEEE Photonics Technol. Lett., 1990,3, pp. 205-207 LAMING, R. I., and PAYNE, D. N.: ‘Noise characteristics of erbium- doped fibre amplifier pumped at 980nm’, IEEE Photonics Technol. Lett., 1990,6, pp. 418-421 MINISCALCO, w. J., THOMPSON, B. A., DAKSS, M. L., ZEMON, s. A., and ANDREWS, L. J . : ‘The measurement and analysis of cross sections for rare-earthdoped glasses’. SPIE Fibre Laser Sources and Amplifiers 111, 1991, Vol. 1581, pp. 80-90 OLSHANSKY, R . : ‘Noise figure for erbium-doped optical amplifiers’, Electron. Lett., 1988,22, pp. 1363-1365 MASSICOTI, I. F., WYATT, R., AINSLLE, B. I., and CRAIG-RYAN, s. P.: ‘Efficient, high power, high gain, Er3+ doped silica fibre amplifier’, Electron. Lett., 1990, 14, pp. 1038-1039 WYATT, R.: ‘Spectroscopy of rare earth doped fibres’. SPIE Fibre Laser Sources and Amplifiers, 1989, Vol. 1171

YAMADA, M., SHLMIZU, M., OKAYASU, M., TAKESHITA, T., HORIGUCHI,

HIGH-T, BANDPASS FILTER USING MINIATURISED MICROSTRIP HAIRPIN RESONATORS

A. Enokihara, K. Setsune, K. Wasa, M. Sagawa and M. Makimoto

Indexing terms: Superconductors, Superconducting devices, Microwave filters

Superconducting bandpass filters of 4.9GHz with 0.5% rela- tive bandwidth were designed using miniaturised microstrip hairpin resonators and fabricated with GdBa,Cu,O, thin films prepared on MgO. The passband insertion loss was 0.5 dB at 20 K and the filter configuration was defined in an area less than 5 x 5 mm2.

- 1570 I580 1590 1600 1610

P E wavelength, n m

Fig. 3 Noise figure for the single and dual pumped I 6 p m amplifier, with theoretical minimum noisefigures for pump wavelengths of 1.48pm and 1,555 pm

0 singlepump W dualpump

1.48 pm ~ 1.555pm _ - - -

Introduction: High critical temperature (l;) thin films are used in narrowband microwave filters to realise low insertion-loss performance, because of their extremely low surface resistance [l-31. Most of the high-l; filters demonstrated use the con- ventional parallel-coupled-line structure. The configuration of such filters is however too large to define in an area of about 1 cm2 when designed with a centre frequency less than several gigahertz on MgO crystals, on which high-T, films of high

ELECTRONICS LETTERS 24th September 1992 Vol. 28 No. 2 0 1925

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quality can be prepared. With regard to a cryogenic environ- ment, the large device size is directly related to the heavy refrigeration burden. Developing high-T, filters with miniatur- ised configuration is, therefore, very important in reducing the size and operating power of the refrigeration systems and in integrating the filters with other circuit elements onto a single substrate.

In this work, we designed and fabricated narrowband high-T, filters using miniaturised microstrip hairpin reson- ators, which have a suitable structure to reduce the size of the filter configuration.

Miniaturised hairpin resonator [4] : The miniaturised hairpin resonator consists of two sections, a symmetrical parallel coupled line and a single line. One end of the coupled line is open and the other end is connected in parallel by the single line. The resonator length decreases compared with that of the conventional half-wavelength resonator when Z, Z , < Z:, where Z, and Z , are the characteristic impedances of the odd and even modes in the coupled line section, rapectively, and Z , is that in the single line section.

m

U;

U

0 - c 5 - e % L -

c_

1

0-

-

b 11)6111 5"

Fig. 1 Miniaturised hairpin resonntor filter and conuentionnl parallel- coupled-linefilter with snme electricol speci$cations

a Hairpin b Conventional parallel-coupled-line

Filter design and fabrication: We designed a two-stage bandpass filter with centre frequency of 4.9 GHz and relative bandwidth of 0.5% using the miniaturised microstrip hairpin resonators on a 0.5" thick MgO substrate (E, = 9.2). A schematic diagram of the filter circuit is shown in Fig. la. The resonators have a coupled line of 60pm gap spacing. Input and output coupling is achieved by tapping at the single line section. The conventional parallelcoupled-line filter with the same electrical specifications is presented in Fig. lb. The hairpin resonator filter is defined in an area less than 5 x 5mm2, which is around one third of that of the conven- tional one.

The filters were fabricated with GdBa,Cu,O, supercon- ducting thin films of 680nm thickness. The films were pre-

I I I I I I I I I I I

I I I I I I I I I I I 4 5 5 0

1136121 frequency, GHz

Fig. 2 Frequency responses of attenuation forfilters at 20 K (i) GBCO superconducting filter (ii) gold filter

1926

--

pared on (100) MgO crystals by RF magnetron sputtering. The as-deposited films have a flat surface and T, of around 82 K. The usual photolithographic technique with argon ion- beam etching was employed for the film patterning. A gold thin film 800nm thick was deposited on the reverse side of the substrate as the ground plane. The filter of the same configu- ration was also fabricated with an 800nm-thick gold thin film as a comparison.

Results and discussions: Fig. 2 shows the frequency responses of the filters at 20K. The passband insertion loss Lo of the GBCO superconducting filter is 0.5 dB, which is 1.9 dB lower than that of the gold filter. The temperature dependence of Lo is shown in Fig. 3, where the broken lines indicate Lo esti- mated from unloaded quality factors Q,, which were mea- sured with the hairpin resonators of the same configuration to those in the filter structure. The Q, values of the GBCO and the gold resonators were 1550 and 330 at 20K, respectively. The measured Lo agree well with those estimated from the Q, for both the filters, which means that the low Lo of the GBCO filter was obtained from the high-Q performance due to the low conductor-loss property of the high-T, materials.

t 1 1

10 0 50

temperature. K

Fig. 3 Temperature dependence of passband insertion loss for GBCO filter and goldfilter

(i) GBCO filter (ii) gold filter _ _ _ _ insertion loss calculated from unloaded quality factors of resonators

Conclusion: Both small size and low insertion loss could be realised for narrowband filters by using the miniaturised microstrip hairpin resonators and fabricating with GdBa,Cu,O, high-T, thin films. The configuration of the filter is defined in an area less than one third of that of a conven- tional filter with the same electrical specifications. The size of the high-T, filter including the cryogenic refrigerator can con- siderably decrease with a reduction in the refrigeration burden.

24th August I992 A. Enokihara, K. Setsune, K. Wasa, M. Sagawa. and M. Makiioto' (Central Research Loboratories, Matsushito Electric Industrial Co., Ltd., Moriguchi, Osaka 570, Jopan

* Also with Information and Communication Research Center, Mat- sushita Electric Industrial Co., Ltd., Higashimita, Tama-ku, Kawasaki 214, Japan

References 1 LYONS, W. G., BONETII, R. R., WILLIAMS, A. E., MANKIWCH, P. M.,

O'MALLEY, M. L., HAMM, I. M., ANDERSON, A. c., WITHERS, R. s., MEU- L ~ E R G , A., and HOWARD, R. E.: 'High-T, superconductive micro- wave filters', IEEE Trans., 1991, MAG27, (2). pp. 2537-2539

~ E S , M. E.: 'Microwave devices using YBa,Cu,O,_, films made 2 NEWMAN, H. S., CHRISEY, D. B., HORWTZ, 1. S., WEAVER, B. 0.. and

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by laser deposition', IEEE Trans., 1991, MAGZ7, (2), pp. 2540- 2543

N A W , T., and NAWO, K.: '13.3GHz YBCO microstrip bandpass filter', Electron. Lett., 1992,28, (4), pp. 355-357

4 SAGAWA, M., TAKAHASHI, K., and "om, M.: 'Miniaturized hairpin resonator tillers and their application to receiver front-end MIC's', IEEE Trans., 1989, M - 3 7 , (12), pp. 199-1997

3 SUGINOSHITA, F., IMAI, K., YAZAWAN, N., SUZUKI, K., N J I O , S., TAKE-

INTERPOLATION USING THE DISCRETE COSINE TRANSFORM

J. I. Agbinya

Indexing term: Signal processing, Transforms, Algorithms

Zooming and zero padding interpolation algorithms based on the conventional discrete cosine transform (DCX) arc pre- sented. The DCT algorithms arc shown to be more accurate and outperform algorithms based on the Wang modified DCT-I which was proposed to replace the DCT for inter- polation purposes.

Introduction: The discrete cosine transform [l-31 is widely used in speech coding and image compression, because it is nearly optimal compared with the Karhunen-Loeve trans- form. For example the recommended CCITT H.261 coding algorithm for moving pictures uses the DCT after the motion compensation stage in the encoder. Before encoding with the DCT, motion vectors are derived using linear interpolation algorithms based on nearest neighbour pixels to improve the efficiency of the prediction of pixel values. The motion vectors are used to provide offsets into both path and future picture reference frames [SI. To effciently use the encoder, it is desir- able to use the DCT in the motion compensation stage for interpolation to predict pixels required for generating the motion vectors and for subsequent encoding. We present a discrete interpolation algorithm based on the DCT suitable for predicting samples between any two neighbouring samples of a sequence of length N an integer power of two. The results presented here are contrary to the assertions of Wang [4 ] that the DCT in its conventional form is unsuitable for inter- polation applications unless the definition is modified. Here the DCT is not modified. We give the DCT interpolation algorithm using a zooming algorithm originally used with the discrete Hartley tranform [SI,* by inserting zeros between the (Nyquist point of the parent sequence) N / 2 and P x N - N / 2 sample position in the interpolated sequence. The second method, applicable to real signals involves padding with zeros the transform coefficients after N , thereby extending the sequence to P x N samples. Thus the factor by which the sampling rate is increased is P , and P - 1 samples are inter- polated between any two consecutive samples of the parent sequence. Interpolation using the DCT: The discrete cosine transform of a sequence x(n) is defined over N samples as [ 11

k = 0 , 1 , 2 ,..., N - t (1)

where

k = 1, 2, . . . , N - 1 [ O otherwise

* AGBINVA, J. 1.: 'Multidimensional interpolation using discrete Hartley transform', submitted to IEEE Trans. Sign. Process., 1992

The inverse LXT is

N- 1

k = O

n = 0, 1, ..., N - 1 (3)

The modified DCT-I defined in Reference 4 is given by the expression

Wang did not define M, and we assume M = 2N in eqn. 4a and M = 2 PN in eqn. 8, and the multiplying constants k; are

In the sequel define the length of the interpolated sequence to be S = P x N, and the coefficients Y(&) of the DCT (DCT-I) transform of the interpolated sequence fin) using the DCT coefficients X(k) .

(i) Case I: Zooming algorithm: Define Y ( k ) or Yl(k) as

k = 0, 1 , 2 , ___, N / 2 - 1

N 2

k = - + 1, .. ., P N - N / 2 - 1 Y ( k ) = 0

0.5X( N / 2 ) k P N - N / 2

X ( k - P N + N ) N 2

k P N -- + 1, .._, P N - 1

( 5 )

The coefficients at NI2 and PN - N / 2 are reduced by half to minimise leakage errors at the data boundaries. To use this algorithm, we modify the DCT multiplying factor c(k) as follows :

i 1 k = O

= { J ( 2 ) k = 1 , 2 , . . . , P N - 1

This modification of the multiplier ensures performance. The inverse DCT of the interpolated sequence is therefore given by the expression

n = 0, 1 , . . . , S - 1 (7)

The inverse modified DCT-I for the interpolated sequence yl(n) is given by the expression

mnn yl(n) = 1 Y ( m ) cos -

II= 1 M PN- I

(ii) Case 2: Zero padding algorithm: We assign transform coef- ficients to Y ( k ) as

(9) {F k = O , l , ..., N

Y ( k ) = k = N + 1, ..., P N - 1

where k = 0, 1, 2, .. ., S - 1, and the inverse DCT and DCT-I are used to form the interpolated time series fin) and yl(n), respectively. Effectively the DCT coefficients are scaled with c l ( k ) = J(2) .

Performance of cosine transform interpolation algorithms: To fully appreciate the effectiveness of transform domain inter- polation algorithms, it is important to test them with short

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