sample of methodology of project

8/13/2019 sample of methodology of project

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CHAPTER 3

METHODOLOGY

3.1. Methodology overview

The aim of the project is to design defected ground structure (DGS) to reduce the

mutual coupling between microstrip array elements to design miniaturized microstrip

array antenna with very high gain to meet the demand of the application require long

distance communications. This chapter consists of the design methodology used thought

out this work.

3.2. flow chart of design methodology

The implementation of this project includes two parts, software design and

hardware design. The approaches of the project design are represented in the flow chart

in figure 3.1. The software simulation includes the designing of two-element microstrip

array antenna with and without DGS. Simulations will be done in CST Microwave

Studio software.


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Figure 3.1Flow chart of antenna design.


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3.3. Microstrip Patch Antenna Design

The three important parameters for microstrip antenna design are the resonant

frequency (fo), the dielectric constant (r) of the substrate and its height (h). Selecting

the frequency of operation depend on the application requirement, while the dielectric

constant of the substrate depend on the substrate material. The substrate material

provides mechanical support for the radiating patch elements.

The procedure is as follows.

Step 1:Calculate the width (W) of the patch. For an efficient radiator, a practical width

that leads to good radiation efficiency is given by [8]

wherecis the speed of light in free space which is 3x10 m/s.Step 2: Determine the effective dielectric constant of the microstrip antenna using

formula 2.8.

Step 3: Once wis found, determine the extension of the length Ldue to the fringing

effects, this can be computed by using formula 2.9.

Step 4: Compute the actual length of the patch that can be determined by

3.4. Choosing for Feeding Technique

There are three different feeding techniques for microstrip patch antenna as discussed in

Chapter 2; in this work Microstrip feed line is chosen. Microstrip line feed is a feeding

method where a conducting strip is connected to the patch directly from the edge as


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shown in figure 2.3. Therefore, the width of the feed line plays important role in

microstrip antenna design, and it must be computed in a mathematical model.

The characteristic impedance of the microstrip line can be calculated as

( )

* +

or design purpose there is a relation that allows us to compute the ratio of the line width

to the height of the substrate () based on a given characteristic impedance anddielectric constant of the substrate.

[ }]

where

( )


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3.5. Impedance matchingThe input impedance of Microstrip patch antenna is a vital parameter in deciding

the amount of input power delivered to the antenna, thus, reducing the coupling effect

of the RF signal to the nearly circuits. The calculation of an exact 50 ohms input

impedance of a Microstrip patch antenna becomes extremely difficult when the antenna

size is drastically small.

At the edges of the patch, the impedance is generally higher than 50 ohm that ranges

from 150 to 300. To avoid impedance mismatch, between the patch and feed line,

there are two methods that can be used for impedance matching microstrip patch

antenna.

3.5.1 Inset Feed

This method of the impedance matching is to extend the microstrip line into the center

of the patch. Since the input impedance is smaller at points away from the edges (e.g.

center of the patch), this is achieved by properly controlling the inset position. Hence

this is an easy feeding scheme, since it provides ease of fabrication and simplicity in

modeling as well as impedance matching. However as the thickness of the dielectric

substrate being used, increases, surface waves and spurious feed radiation also

increases, which hampers the bandwidth of the antenna. The impedance of the patch is

given by

where, G1 and G12 are self and mutual conductances expressed in section 2.3.1.

The impedance of the patch is also related to the electrical dimensions of the patch and

dielectric constant of the substrate, which is given by

( )


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The input impedance related to the length of the inset is given by

where y0is the inset length from slot at the feeding edge of patch, L is the length of the

patch. Therefore, this technique can be used effectively to match patch antenna to a 50-

microstrip-line feed.

where is the input impedance at the leading radiating edge of the patch and is the desired input impedance (50 ).3.5.2 Quarter-wave transformers

Sections of quarter-wave transformers can be used to transform from large input

impedance to 50 ohm line, this is shown in figure 2.4. A quarter-wave transformer uses

a section of line of characteristic impedance of long.

To have a matching condition, we want the resonant input impedance of the patch ( )equal to the line impedance (), this can be achieved by using this equation

In microstrip patch antennas, the total input admittance () is real. Therefore, theresonant input impedance is also real, or


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3.6 Array Configuration and Design

Theproposed antenna configurationis shown in Fig. 3.4. To visualize mutual coupling

between the elements of the array in figure 3.4, a two element of microstrip antenna

array with separate feed lines was introduced as shown in Fig.3.2. The figure illustrates

the layout of the array that operates at a frequency of 2.4GHz. Dimensions of all the

parameters are tabulated in table 3.1. Each patch is excited on its symmetrical axis by a

50 microstrip with an inset of 11.3 mm to match the feed line to the patch.

Table 3.1 Dimensions of the antenna parameters

Symbol Parameters Values (mm)L Length of the patch 28.9

W Width of the patch 31.0

d Distance between centers of patches 28.13

Y0 Inset feed line 11.3

Wf Width of the microstrip line 3.10

Fig. 3.2 Geometry of two element microstrip antenna array


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CST Microwave Studio was used to simulate the E-plane coupled elements in the array.

The E-plane coupled microstrip patch antenna arrays suffer from strong mutual

coupling because of surface waves. Due to the capability of DGS to suppress surface

waves, a two T-shaped DGS placed back to back were inserted between the two antenna

elements in order to reduce the mutual coupling as shown in Fig. 3.3. Dimensions of all

the DGS parameters are given in table 3.2

Fig. 3.3 Geometry of two element microstrip antenna array on a defected ground plane

Table 3.2 Dimensions of the DGS parameters

Symbol Parameters Values (mm)

a DGS head width 4.2

b DGS head length 4.9

c DGS overlap 8.7

d DGS slot length 26.3

x DGS slots separation 10.8

s DGS slot width 2.0


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Fig. 3.4 Proposed microstrip antenna array with corporate feed

3.7 DGS Configuration and Response

In this project, a new technique of DGS has been proposed to reduce the mutual

coupling between elements of an antenna array by introducing two T-shaped DGS

placed back to back between elements. The presence of the two slots improves the

isolation between array elements and increases the stop band.

(a)


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(b)Figure 1 DGS configuration. (a) Microstrip transmission line with DGS on the ground-

plane. (b) Simulated S-parameters response.

From Fig. 1b, it is observed that DGS exhibits Bandstop response with attenuation pole

at different frequencies depending on the dimensions of the DGS. In this design the

DGS dimensions are optimized by simulation to achieve attenuation pole at design

frequency 2.4 GHz.

The presence of the DGS section operating at below its pole frequency increases the

effective inductance of a microstrip line. The cutoff frequency is mainly dependent on

the etched slot head area in the ground plane. There is also attenuation pole location,

which is due to the etched slot width of the slot. Actually, it is well known that an

attenuation pole can be generated by combination of the inductance and capacitance

elements. Thus, the DGS section is fully described by the etched slot width, length and

head area in case of slots with certain head shape.

3.7.1 Modeling and parameter extractionA parallel LC circuit can represent the equivalent circuit of the DGS as shown from its

response. From the application point of view, the DGS section can serve as replacement

for a parallel LC resonator circuit in many applications. To apply the DGS section to a

-30

-25

-20

-15

-10

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0

2 2.2 2.4 2.6 2.8 3

MagnitudeindB

Frequency (GHz)

S11 for d=24 mm

S12 for d=24mm

S11 for d=26 mm

S12 for d=26mm

S11 for d=28 mm

S12 for d=28mm


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practical circuit design example, it is necessary to extract the equivalent circuit

parameters.

As an example of the parameter extraction procedure, Fig. 1a shows the geometric

configuration of T-shaped DGS etched on the ground plane of 50 ohm microstrip line

with width WL=3.3mm, FR4 substrate with dielectric constant of 4.4 and thickness of

1.59 mm. Simulated S parameters response of the DGS is shown in Fig 1b. There is an

attenuation pole near 2.4 GHz in the field simulation result. In order to explain the

cutoff and attenuation pole characteristic of the proposed DGS section simultaneously,

the equivalent circuit should exhibit performances of low-pass and band-stop filter at

the same time. Thus, the simple circuit shown in Fig. 3 can explain the phenomenon for

the proposed DGS section. The circuit parameters for the derived equivalent circuit can

be extracted from the simulation result.

The simulation result of the proposed DGS unit section can be matched to the one-pole

Butterworth-type low-pass response. The series reactance value shown in Fig. 3 can be

easily calculated by using the prototype element value of the one-pole Butterworth

response.

Figure 3LC equivalent circuit: (a) equivalent circuit of the DGS circuit, where the

dotted box shows the DGS Section, (b) Butterworth-type one-pole prototype low-pass

filtercircuit.


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The prototype element value is given by various references. [14], [15]. The parallel

capacitance value for the given DGS unit dimension can be extracted from the

attenuation pole location, which is a parallel LC resonance frequency and prototype

low-pass filter characteristic by using the following procedures. The reactance value of

the proposed DGS unit can be expressed as follows:

( ) where, 0 is the resonance angular frequency of the parallel LC resonator,which

is corresponding to attenuation pole location in Fig. 1b. The series inductance of the

Butterworth low-pass filter, shown in Fig. 3b, can be derived as follows:

where 'denotes the normalized angular frequency, Z0 denotes the impedance level of

the in/out terminated ports, and is given by the prototype value of the Butterworth-type

low-pass filter. In order to have the low-pass filter characteristics, the equivalent circuit

of proposed DGS unit section, shown in Fig. 3a, should be equal to the prototype low-

pass filter, shown in Fig. 3b, at a certain frequency. The equality at the cutoff frequency

of the low-pass filter is given by the following:

| |

From above equality, the series capacitance of the equivalent circuit, shown in Fig. 5,

can be obtained as follows:


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Once the capacitance value of the equivalent circuit is extracted, the series equivalent

inductance for the given DGS unit section can be calculated by the following:

where f0and fc are resonance (attenuation pole) and cutofffrequency which can beobtained from EM simulation results. The characteristics of most of DGS are similar to

dumbbell DGS, so they could be discussed by one-pole Butterworth low-pass filter too.

Furthermore, radiation effects are more or less neglected.DGS unit can be modeledmost efficiently by a parallel R, L, and C resonant circuit connected to transmissionliens at its both sides as shown in Fig. 6. This resistance corresponds to the radiation,

conductor and dielectric losses in the defect. From EM simulations or measurements

for a given DGS, the equivalent R, L, and C values are obtained from the expression in

[27].

Figure 4RLC equivalent circuit for unit DGS

{

|| ( )


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The size of DGS is determined by accurate curve-fitting results for equivalent-circuit

elements to correspond exactly to the required inductance. Fig. 3.4 shows the design

process of the DGS section.

Figure 3.4DGS design procedure.

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sample of methodology of project