RF MEMS, BST, and GaAs Varactor System-Level …kentesar/Kamran_J3.pdf · RF MEMS, BST, and GaAs Varactor System-Level Response in Complex Modulation Systems Kamran Entesari,1 Gabriel

RF MEMS, BST, and GaAs Varactor System-LevelResponse in Complex Modulation Systems

Kamran Entesari,1 Gabriel M. Rebeiz2

1Department of Electrical and Computer Engineering, Texas A&M University,College Station, TX 778432Electrical and Computer Engineering Department, The University of California at San Diego,La Jolla, CA 92037

Received 19 January 2007; accepted 23 April 2007

ABSTRACT: This article presents the response of RF microelectromechanical systems (RF

MEMS), barium strontium titanate (BST), and gallium arsenide (GaAs)-based tunable filters

and reconfigurable matching networks to a wideband code-division-multiple-access signal cen-

tered at 1.95 GHz. The RF MEMS tunable filter and impedance tuner result in very low inter-

modulation distortion and spectral regrowth compared to their BST and GaAs counterparts.

The linearity of the BST and GaAs tunable networks improves considerably by using a series

combination of BST and GaAs varactors, but the RF MEMS-based networks still show the best

linearity of all three technologies. Also, it is shown that the reconfigurable networks, tuned

with capacitive RF MEMS can handle up to 1 W of RF power with no self-actuation. VVC 2007

Wiley Periodicals, Inc. Int J RF and Microwave CAE 00: 000–000, 2007.

Keywords: reconfigurable networks; RF microelectromechanical systems (RF MEMS); wideband

code-division-multiple-access (WCDMA); spectral regrowth; linearity

I. INTRODUCTION

One of the important applications of RF microelec-

tromechanical system (RF MEMS) devices is in

reconfigurable networks such as tunable filters and

impedance tuners [1–3], and the advantage of RF

MEMS devices is their excellent linearity [4]. From a

system-level point of view, linearity is a major issue

in wireless communications, especially for systems

which employ orthogonal types of modulation techni-

ques such as wideband code-division-multiple-access

(WCDMA) [5, 6].

The presence of two tones at the input of a non-

linear circuit generates third-order intermodulation

tones, and as a consequence, a WCDMA signal

results in spectral regrowth at the output port and a

nondesirable adjacent channel power leakage phe-

nomena (Fig. 1). The reason is that the abrupt phase

transitions in the WCDMA signal appear as ampli-

tude modulation in practice [7, 8], and therefore, the

WCDMA signal is very sensitive to circuit nonlinear-

ities in the RF front-end [5, 6]. It is therefore critical

to obtain RF reconfigurable circuits as linear as possi-

ble to avoid the adjacent channel interference in

WCDMA transceivers.

In this paper, two different multifunctional circuits

centered at 1.95 GHz, a tunable filter, and a tunable

matching network are designed using RF MEMS,

barium strontium titanate (BST), and gallium

arsenide (GaAs) varactors for a WCDMA front-end

centered at 1.95 GHz to investigate and compare their

response to a WCDMA input signal. Also, to improve

the linearity of the BST tunable networks, BST array-

Correspondence to: K. Entesari; e-mail: [email protected] 10.1002/mmce.20275Published online in Wiley InterScience (www.interscience.

wiley.com).

VVC 2007 Wiley Periodicals, Inc.

1

based varactors are employed instead of a single var-

actor and their response is compared with other tuna-

ble networks.

II. TUNABLE FILTERS IN WCDMATRANSCEIVERS

A. Tunable Filter Topology

Figure 2 shows the schematic of a two-pole band-

pass tunable filter suitable for wideband tuning at RF

frequencies [3, 9]. The filter is designed as a 3% Che-

byshev filter, with a center frequency of 1.95 GHz

(Table I). The inductors have a resistance of 0.15 Oresulting in a Q ¼ 200 at 1.95 GHz and a filter inser-

tion loss of 1.5 dB in the passband (the capacitors are

assumed lossless). By changing CR from 1.7 to 3.1

pF and keeping CM fixed, one can achieve a tuning

range of 1.7–2.2 GHz with a resonator match better

Figure 1. The effect of circuit nonlinearity on the spectral regrowth and adjacent channel inter-

ference of a WCDMA signal.

Figure 2. Schematic of a two-pole bandpass tunable fil-

ter at 1.95 GHz (CM: Matching capacitor, CR, LR: Resona-

tor capacitor and inductor).

TABLE I. Tunable Filter Element Values at 1.95

GHz for a 3% Chebyshev Design

Element Values

CR (pF) 2.3

CM (pF) 0.42

LR (nH) 2.48

R (O) 0.15

k 0.04

Figure 3. The simulated (a) insertion loss and (b) return

loss of the tunable filter covering the 1.7–2.2 GHz tuning

range (Qu ¼ 200 at 1.95 GHz, IL ¼ 1.0–1.8 dB). Simula-

tion is done using ADS [10].

TABLE II. Tunable Filter Characteristics for

Different Center Frequencies (Qu ¼ 200 at 1.95 GHz)

Frequency

(GHz) CR (pF) BW (%) IL (dB) BW (MHz)

1.7 3.1 3.2 1.8 54

1.8 2.7 3.3 1.7 60

1.95 2.3 3.0 1.5 58

2.1 2.0 3.4 1.2 71

2.2 1.7 3.5 1.0 77

2 Entesari and Rebeiz

International Journal of RF and Microwave Computer-Aided Engineering DOI 10.1002/mmce

than �13 dB (Fig. 3). Table II presents the band-

width, insertion loss, and the CR capacitor values for

the tunable filter at different frequencies. It can be

seen that the filter is slightly undercoupled at 1.7

GHz and slightly overcoupled at 2.2 GHz. A near-

perfect tunable filter response can be achieved if CM

is changed between 0.4 and 0.44 pF (not done for

simplicity).

To implement the tunable filter, CR is substituted

by RF MEMS switched capacitors, BST, and GaAs

varactors. Figures 4a and 4b show the side and top

views of the RF MEMS shunt capacitive switch,

respectively, and Figure 4c presents the electrical

model of the switch [1]. Table III provides the infor-

mation regarding physical dimensions and electrome-

chanical parameters of the switch.

The DC biasing electrode and the RF signal path

are identical in this device and result in the worst case

IIP3 compared to other types of capacitive switches. If

the DC biasing electrode is separated from the RF sig-

nal path, the RF MEMS switched capacitor shows bet-

ter linearity and higher IIP3 compared to the standard

design, but this is not considered here [1].

Figure 5a shows the C-V curve for a RF MEMS

capacitive switch [1]. The switch remains in the upstate

position for a bias voltage between �26 and 26 V with a

capacitance of 110 fF (0 V) � 148 fF (24 V). Above 26

V, the capacitive switch actuates to the downstate posi-

tion and results in a capacitance of 3.5 pF (capacitance

ratio of 30). The CAD-based model of a nonlinear

MEMS capacitor is shown in Figure 6. This model is

composed of three blocks: (A) electrostatic force gener-

ation, (B) the mechanical bridge, and (C) the variable-

gap parallel plate capacitor. This model is used to create

a behavioral model in ADS [4, 10].

Figure 5b shows the measured C-V curve for a

typical ferroelectric varactor [11]. The ferroelectric

varactor has a symmetric C-V curve and acts as a

variable capacitor for both positive and negative vol-

tages across the varactor. It provides a capacitance ra-

tio of 3.4 and is biased at 6 V (Cvar ¼ 1.8 pF) to

achieve maximum capacitance swing. The CAD-

based model of a nonlinear BST varactor is based on

fitting the mathematical Eq. (1) to the measured BST

C-V curve. This expression is modeled in ADS for

simulation purposes as

Figure 4. (a) Side view, (b) top view, and (c) the electrical model of a shunt capacitive RF

MEMS switch.

TABLE III. Physical Dimensions and

Electromechanical Parameters of the RF MEMS

Capacitive Switch

Physical dimensions

Bridge length, L (lm) 280

Bridge width, W (lm) 130

Bridge thickness, t (lm) 0.8

Air gap, g (lm) 2.0

Electrode width, w (lm) 160

Dielectric (er)–Roughness ratio (e)a 7.0–0.6

Dielectric thickness (lm) 0.2

Electromechanical parameters

Spring constant, k (N/m) 52

Pulldown voltage, Vp (V) 26

Mechanical resonant freq., f0m (kHz) 76

Mechanical quality factor, Qm 1.0

CMEMS, up (pF)/CMEMS, down (pF) 0.11/3.5

Switch inductance, LS (pH) 10

Switch resistance, RS (O)F 0.6

a Roughness ratio: Percentage of contact area between bridge

layer and dielectric layer in the downstate position. This is

between 0.5 and 0.7, and depends on the roughness of the dielec-

tric layer.

System-Level Responses in Complex Modulation Systems 3


CðVÞ ¼ aV= bþ V=cj jmð Þð Þ þ d ð1Þ

where a ¼ 2.3 3 10–12, b ¼ 0.9, c ¼ 8.5, m ¼ 1.14, d¼ 1.2 3 10–12 are constants found from mathematical

curve fitting.

Figure 5c shows the measured C-V curve for a

typical reversed-biased GaAs varactor [12]. The

GaAs varactor provides a capacitor ratio of 4 and is

biased at 6 V (Cvar ¼ 1.8 pF) to achieve maximum

voltage swing and linearity. The CAD-based model

of a nonlinear GaAs varactor is implemented by

employing a reverse-biased P-N semiconductor junc-

tion model in ADS and is fitted to the measured

GaAs C-V curve using

CðVÞ ¼ Cjo 1 þ V=Vj

� �� M ð2Þ

Figure 5. Capacitance vs. biasing voltage for a (a) RF MEMS (b) BST, and (c) GaAs varactors.

Figure 6. A CAD-based nonlinear model for the RF MEMS capacitive switch [4].



where Cjo ¼ 4.0 pF is the zero-bias junction capacitance,

Vj ¼ 5 V is the junction potential and M¼ 1 is the grad-

ing coefficient. Also, a comparison among the switching

speed of three different varactors shows that the switch-

ing speed for the existing RF MEMS switched capaci-

tors, BST, and GaAs varactors is in the order of 3–10 ls

[1], 1 ns [11], and 0.1 ns [12], respectively.

The tunable filters are now implemented using the

RF MEMS, BST, and GaAs devices (Fig. 7). In the

case of the RF MEMS design, the varactor CR is

implemented as a 3-bit switched capacitor, where

each RF MEMS switched capacitor is a series combi-

nation of the RF MEMS switch and a fixed capacitor

[3]. Table IV presents the different capacitance val-

ues and center frequencies vs. bit combinations. The

3-bit tuning approach is a practical solution in RF

MEMS, since it is very hard to fabricate a reliable

high-capacitance ratio RF MEMS analog varactor

[1]. Figures 7b and 7c show the tunable filter using

GaAs and BST varactors, respectively. Both GaAs

and BST varactors are biased at 6 V to provide the

filter response at the middle of tuning band (CR ¼ 2.3

pF and f0 ¼ 1.95 GHz). Table IV also shows the

value of BST and GaAs varactors and their biasing

voltages for different filter center frequencies. To

investigate the filter response to the WCDMA input

signal, it is assumed that all three filter center fre-

quencies are at 1.95 GHz.

Figure 7. Tunable filters implemented using (a) RF MEMS, (b) BST, and (c) GaAs devices. The

RF chokes (RFC) and bias resistors assumed to be ideal have no effect on the filter transfer function.

TABLE IV. MEMS Switch Combinations, BST and GaAs Varactor Values, and Biasing

Voltages for Different Filter Center Frequencies

f0 (GHz)

MEMS Design BST Design GaAs Design

S1-S2-S3 CR (pF) CVar (pF) CR (pF) VBias (V) CVar (pF) CR (pF) VBias (V)

1.7 ON-ON-ON 3.1 3.0 3.1 2.0 3.0 3.1 1.8

1.8 OFF-ON-ON 2.7 2.3 2.7 4.0 2.3 2.7 3.8

1.95 OFF-OFF-ON 2.3 1.8 2.3 6.0 1.8 2.3 6.0

2.1 ON-OFF-OFF 2.0 1.4 2.0 9.0 1.4 2.0 9.8

2.2 OFF-OFF-OFF 1.7 1.2 1.7 13.0 1.2 1.7 11.6



Notice that for the BST and GaAs design, two

devices are used for CR to achieve reasonable values

for the tunable devices. Since the two devices are in

parallel, the resulting intermodulation products are

the same as a large device with twice the capacitance

value. A series combination of the BST (or GaAs)

varactor with a fixed capacitor is also used in the tun-

able filter topology to result in smaller voltage swing

across the BST (or GaAs) varactor and keep the de-

vice in the linear region as much as possible. This

will be discussed in Section B.

TABLE V. Voltage Swing Across a Single Varactor

Inside the Tunable Filter at node ‘‘a’’ for Different

Input Powers (f0 ¼ 1.95 GHz)

Pin (dBm) RF MEMS BST GaAs

10 3.2 2.4 2.4

20 10.0 �0.4 to 15a �0.4 to 15a

30 31.0 �17 to 36a �0.7 to 19b

a Distorted signal across Va.b GaAs design cannot handle >22 dBm. Voltage swing turns

on diodes completely.

Figure 8. Va(t) across the RF MEMS switch and the BST

and GaAs varactors for Pin ¼ 30 dBm at f0 ¼ 1.95 GHz.

Figure 9. Large-signal S21 for RF MEMS, BST, and GaAs tunable filter for different input

powers.

Figure 10. A tunable filter with (a) 3 3 1 and (b) 3 3 3 BST array varactors.



B. Simulation Results

Table V shows the voltage swing across the RF

MEMS, BST, and GaAs varactors inside the tunable

filter (Va in Fig. 7) for the 1.95 GHz single-tone exci-

tation, using harmonic balance simulation in ADS.

For Pin ¼ 30 dBm, Va ¼ 31 V peak (Vb ¼ 28.6 V due

to losses in the resonator) for the RF MEMS switch.

In this case, Va(r.m.s) ¼ 22 V, which is still lower than

the pulldown voltage of the switch (Vpulldown ¼ 26

V), and therefore the switch remains in the upstate

position, and self-biasing does not occur (as shown in

[1], If V(r.m.s) > Vpulldown, then the RF voltage can

pull down the switch even with no DC bias applied).

The voltage across the BST varactor is distorted for

Pin ¼ 30 dBm, since the signal experiences a huge

nonlinearity due to large capacitance variation of the

BST varactor.

For Pin ¼ 30 dBm, the voltage across the GaAs

varactor at nodes (a) and (b) is compressed and this is

due to the high voltage swing across the varactor

which pushes the varactor to the ohmic region

(Fig. 8). The large signal S-parameters of the filter

are simulated in ADS for different input power levels

(single tone excitation). The RF MEMS tunable filter

response does not change at all up to Pin ¼ 28 dBm

(Fig. 9). For Pin ¼ 30 dBm, the RF MEMS filter

response is slightly distorted due to the self-biasing

effect of the upstate MEMS bridges (Va(r.m.s) ¼ 22 V)

and the MEMS bridges are starting to pull down. For

the BST and GaAs tunable filter, the filter response is

distorted even at Pin ¼ 20 dBm because of the non-

linear behavior of the BST and GaAs varactors, andFigure 12. Large-signal S21 for BST (1 3 1), (3 3 1),

and (3 3 3) tunable filters for different input powers.

Figure 13. Simulated WCDMA spectrum at the output

of tunable filters for different input powers.

Figure 11. Simulated fundamental and IM3 output

power vs. input power for different BST topologies in the

2-pole tunable filter.



both designs cannot handle 30 dBm of input power

(Fig. 9).

To improve the power handling and linearity of

the BST tunable filter, each BST varactor is substi-

tuted by a 3 3 1 and a 3 3 3 array of similar BST

varactors (Fig. 10). In the 3 3 1 case, each BST de-

vice is three times larger than the nominal value of

Figure 5, while in the 3 3 3 case, the same device in

Figure 5 is used. This topology divides the RF volt-

age across each BST device by a factor of 3 while

maintaining the same capacitance CR.

A two-tone intermodulation test is done in ADS

using two sinusoidal signals at f1 ¼ 1.95 GHz and f2¼ f1 þ Df (Df ¼ 1 MHz), and the simulated funda-

mental and IM3 powers for the 1 3 1, 3 3 1, and 3

3 3 BST-based tunable filters are shown in Figure 11.

As expected, the IIP3 level increases from 23 dBm

for 1 3 1 BST topology to 33 dBm for 3 3 1 and 3

3 3 BST topology. Figure 12 presents the large-sig-

nal S-parameters for 1 3 1, 3 3 1, and 3 3 3 BST-

based tunable filter at Pin ¼ 20 and 30 dBm. The

filter response for 3 3 1 and 3 3 3 BST topologies

has no distortion at Pin ¼ 20 dBm, and the power

handling of the 3 3 1 and 3 3 3 BST topology is

improved to 26 dBm without any shift in the large-

signal S-parameters.

The single-tone source is now substituted with a

WCDMA source at 1.95 GHz. The circuit is simu-

lated in ADS using envelope-simulation technique,

and the WCDMA spectrum is extracted at the output

of the tunable filter (Fig. 13). For Pin ¼ 10 dBm and

Va ¼ 2.4 V, the GaAs and 1 3 1 BST filters are

pushed to their nonlinear region while the 3 3 1 and

3 3 3 BST topologies and RF MEMS filters are still

in their linear region. The WCDMA output spectrum

for GaAs and 1 3 1 BST tunable filters shows a spec-

tral regrowth of more than 40 dB compared with their

MEMS counterpart at the adjacent channel. For Pin ¼20 dBm, the BST-tunable filters show 50 dB (1 3 1)

and 20 dB (3 3 1, 3 3 3) of spectral regrowth com-

pared to the input signal. For Pin ¼ 30 dBm, the RF

MEMS filter remains in the linear region and still fol-

lows the input signal, while both GaAs and BST fil-

ters show extremely nonlinear behavior. The output

spectrum of the GaAs filter is not shown in Figure

13c, because the GaAs filter response is totally dis-

torted at this power level and does not show a filter

response any more.

Figure 14 presents the simulated IIP3 for different

filter topologies at different filter center frequencies.

The GaAs and BST filters (1 3 1, 3 3 1, 3 3 3)

show similar behaviors: At f0 ¼ 1.7 GHz, the GaAs

and BST varactors are biased at 1.8 V and 2 V,

respectively, and because of the steep C-V curves at

these bias voltages, this results in the worst IIP3 com-

pared with other center frequencies. Also, the IIP3 is

improved by 10 dBm for (3 3 1, 3 3 3) BST topolo-

gies, compared with the 1 3 1 BST topology for all

center frequencies. The IIP3 for the RF MEMS filter

is simulated for Df ¼ 100 kHz and Df ¼ 1 MHz at

different filter center frequencies, and it shows

20 dBm improvement at Df ¼ 1 MHz compared to Df¼ 100 kHz.

This is because the intermodulation component

follows the mechanical response of the bridge, and

the IM3 level drops by 40 dB/decade for Df > f0m

(f0m ¼ 76 kHz), which is in agreement with theory

[4]. At f0 ¼ 2.2 GHz, all the MEMS switches are in

the upstate position, and this state results in the worst

IM3 products and the worst IIP3 level. At f0 ¼ 1.7

GHz, all the switches are in the downstate position

and this results in the highest IIP3 compared to other

filter center frequencies. Note for Df > 1 MHz, the

IIP3 of the tunable filter is >55 dBm for all tuning

conditions (IIP3 > 75 dBm for Df > 10 MHz).

Figure 14. Simulated IIP3 at different filter center fre-

quencies for different filter topologies.

Figure 15. The schematic of a WCDMA transmitter

front-end including a linear power amplifier and a tunable

matching network.



Figure 16. ZLoad impedance loci at the output of the power amplifier for (a) CS ¼ CL ¼ 5.7

pF, (b) CS ¼ CL ¼ 4.0 pF, (c) CS ¼ CL ¼ 7.4 pF. L ¼ 1.15 nH for all cases. (Smith chart im-

pedance ¼ 4 O). ZAmp impedance loci at the load (Zout) for (d) CS ¼ CL ¼ 5.7 pF, (e) CS ¼ CL

¼ 4.0 pF, (f) CS ¼ CL ¼ 7.4 pF. L ¼ 1.15 nH for all cases (Smith chart impedance ¼ 50 O).



III. IMPEDANCE TUNER IN WCDMATRANSCEIVERS

A. General Matching Network Topology

Figure 15 presents a power amplifier with a reconfig-

urable matching network for wireless applications.

The power amplifier is assumed ideal with 20 dB of

gain and an input impedance of 50 O. The output im-

pedance, ZAmp, and the antenna impedance, ZAnt, are

assumed 4 and 50 O, respectively. To match the 4 Oto the 50 O impedance, a CS-L-CL p-network is used

(CS ¼ CL ¼ 5.7 pF, L ¼ 1.15 nH). In practice, ZAmp

and ZAnt are variable impedances, and a tunable

matching network is necessary to match ZAmp and

ZAnt for different impedance values.

The impedance tuner is designed for two cases:

1. ZAmp ¼ 4 O and ZAnt varies between 15 6

j50 and 100 6 j50.

Figure 16a shows ZLoad for ZAnt ¼ 15 6

j50 to 100 6 j50, for CS ¼ CL ¼ 5.7 pF, and

L ¼ 1.15 nH. All the impedance loci inside

the G ¼ 10 dB circle are matched to 4 O with

Figure 17. (a) RF MEMS, (b) BST, and (c) GaAs tunable matching networks.

TABLE VI. MEMS Switch Combinations, BST and GaAs Varactor Values, and Biasing

Voltages for Different Matching Capacitors

MEMS Design BST Design GaAs Design

S1-S2 CS,L (pF) CVar (pF) CS,L (pF) VBias (V) CVar (pF) CS,L (pF) VBias (V)

ON-ON 7.4 3.0 7.4 0.5 3.0 7.4 0.6

OFF-ON 5.7 2.5 5.7 3.3 2.5 5.7 3.0

OFF-OFF 4.0 1.8 4.0 15.5 1.8 4.0 12.5



a return loss better than 10 dB, but many of

the impedance loci are outside this circle. By

changing the capacitors to CS, CL ¼ 4.0 pF

(Fig. 16b) and 7.4 pF (Fig. 16c), the matching

range is extended to cover virtually as much

as possible antenna impedances.

2. ZAnt ¼ 50 O, and ZAmp varies between 2 6 j8and 8 6 j8.

Figure 16d shows the power amplifier output

impedance locations after the matching net-

work (Zout) for ZAmp ¼ 2 6 j8 to 8 6 j8, and

CS ¼ CL ¼ 5.7 pF, L ¼ 1.15 nH. All the imped-

ance loci inside the G¼ 10 dB circle are

matched to 50 O with a return loss better than

10 dB, but many of the impedance loci are out-

side this circle. To extend the matching range,

the capacitors CL and CS are changed while the

series inductor is fixed. Figures 16e and 16f

show Zout for CS ¼ CL ¼ 4.0 and 7.4 pF,

respectively.

B. RF MEMS, BST, and GaAsImpedance Tuners

CS and CL are substituted with the three different

types of varactors presented in Figure 5. Figure 17a

shows the RF MEMS impedance tuner, and each

varactor (CS, CL) is implemented as a 2-bit switched

capacitor.

Table VI presents the different capacitance values

vs. bit combinations. Figures 17b and 17c present the

impedance tuner using GaAs and BST varactors,

respectively. By changing the biasing voltage of the

GaAs or BST devices, the capacitors CS and CL vary

between 4.0 and 7.4 pF. Table VI also shows the

value of BST and GaAs varactors and their biasing

voltages for CS ¼ CL ¼ 4.0 pF, CS ¼ CL ¼ 5.7 pF,

and CS ¼ CL ¼ 7.4 pF. To investigate the effect of

the impedance tuner nonlinearity on the WCDMA

signal, we consider the nominal case where CS ¼ CL

¼ 5.7 pF (ZAnt ¼ 50 O, ZAmp ¼ 4 O).

TABLE VII. Voltage Swing Across the Varactors at Nodes ‘‘a’’ and ‘‘b’’ in Figure 17

Pout (dBm)

RF MEMS BST GaAs

Va (V) Vb (V) Va (V) Vb (V) Va (V) Vb (V)

10 0.3 1 0.2 0.75 0.2 0.75

20 0.9 3.2 0.7 2.5 0.7 2.5

30 2.7 9.2 2.3 �3.3 to 12.8a 2.8 �0.8 to 9.8b

a Distorted signal across Vb.b Voltage swing turns on GaAs diodes completely.

Figure 18. Impedance tuners with (a) 3 3 1 and (b) 3 3 3 BST array varactors.



C. Simulation Results

Table VII shows the simulated voltage swing across

the RF MEMS, BST, and GaAs varactors (using har-

monic balance simulation in ADS) inside the recon-

figurable network (Va and Vb in Fig. 17) for a 1.95-

GHz single-tone excitation for different output power

levels. Vb is larger than Va, because node ‘‘b’’ is a

higher impedance node. As a result, varactors con-

nected to node b are pushed to the nonlinear region

more strongly. For Pout ¼ 30 dBm, the voltage across

the GaAs varactor at node b is compressed due to the

high voltage swing across the varactor which pushes

the varactor to the ohmic region. The voltage across

the BST varactor at node b is also distorted due to the

nonlinearly of the BST varactor. For the RF MEMS

capacitive switch, Vb reaches to 9.2 V (Vb(r.m.s) ¼ 6.5

V), which is much lower than the pulldown voltage

of the switch (Vpulldown ¼ 26 V).

To improve the linearity of the BST impedance

tuner, each BST varactor is substituted by 3 3 1 and

3 3 3 arrays as was done previously (Fig. 18).

Table VIII presents the voltage swing across a single

BST device connected to node ‘‘a’’ or ‘‘b.’’ For Pout

¼ 30 dBm, the 3 3 1 and 3 3 3 topologies show no

distortion in Vb. The 3 3 1 and 3 3 3 topologies also

show a 10-dB improvement in IIP3 compared to the

1 3 1 topology (not shown).

The single-tone source is now substituted with a

WCDMA source at 1.95 GHz. The circuit is simu-

lated in ADS using envelope-simulation technique,

and the WCDMA spectrum is extracted at the output

of the impedance tuner (Fig. 19). For Pout ¼ 10 dBm,

all three tuners are in the linear region and the spec-

tral regrowth at the adjacent channel is negligible

(not shown). For Pout ¼ 20 dBm, the GaAs and 1 3

1 BST array tuners are pushed to their nonlinear

region while the 3 3 1, 3 3 3 BST topology and RF

MEMS tuner stay in the linear region. The output

spectrum for GaAs and 1 3 1 BST tuners shows a

spectral regrowth of 25 and 20 dB compared to the

input signal, respectively. For Pout ¼ 30 dBm, the

GaAs and 1 3 1 BST tuners show large spectral

regrowth (50–25 dB) while the RF MEMS tuner still

stays in the linear region with no spectral regrowth.

CONCLUSION

This paper presented the spectral regrowth for RF

MEMS, BST, and GaAs tunable filters and reconfig-

urable matching networks with WCDMA excitation.

The RF MEMS tunable filter and impedance tuner

have the best linearity and lowest spectral regrowth

compared with their GaAs and BST counterparts.

The (3 3 3, 3 3 1) BST topologies improve the line-

arity by 10 dB, but still show significant spectral

regrowth at 20 dBm (tunable filter) and 30 dBm (tun-

able matching network). As a result, RF MEMS front

ends are excellent candidates for complex modulation

reconfigurable systems and can handle 1 W of power

in narrowband tunable filters.

REFERENCES

1. G.M. Rebeiz, RF MEMS theory, design, and technol-

ogy, Wiley, New York, 2003.

TABLE VIII. Voltage Swing Across a Single Varactor

Inside BST Arrays at Nodes ‘‘a’’ and ‘‘b’’

Pout

(dBm)

BST (1 3 1) BST (3 3 3) BST (3 3 1)

Va (V) Vb (V) Va (V) Vb (V) Va (V) Vb (V)

20 0.7 2.5 0.2 0.8 0.2 0.8

30 2.3 �3.3 to

12.8

0.7 2.6 0.7 2.6

Figure 19. Simulated WCDMA spectrum at the output

of the RF MEMS, GaAs, and BST impedance tuners.



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BIOGRAPHIES

Kamran Entesari received the B.S.

degree in electrical engineering from

Sharif University of Technology, Tehran,

Iran, in 1995, the M.S. degree in electrical

engineering from Tehran Polytechnic Uni-

versity, Tehran, Iran, in 1999, and the

Ph.D. degree from the University of

Michigan, Ann Arbor, in 2005. In 2006,

he joined the Department of Electrical

and Computer Engineering at Texas A&M University where he is

currently an assistant professor. His research interests include

design of radio frequency/microwave/millimeter-wave integrated

circuits and systems, microelectromechanical systems (MEMS)

for microwave/millimeter-wave applications, related front-end

analog electronic circuits and antennas, microwave filters and pas-

sive components, and active and passive sensors. He is a member

of IEEE Microwave Theory and Techniques Society.

Gabriel M. Rebeiz is a professor of elec-

trical engineering at the University of Cal-

ifornia, San Diego. He received his Ph.D.

(1988) from the California Institute of

Technology. He has contributed to planar

mm-wave and THz antennas and imaging

arrays from 1988 to 1998, and to the de-

velopment of RF MEMS from 1996 to

present. He is the author of the book, RF

MEMS: Theory, Design and Technology, Wiley (2003). Prof.

Rebeiz’ group recently developed the fastest mm-wave SiGe

switch to date (70 ps), and 6–18 GHz and 30–50 GHz 8-element

phased array receivers and transmitters on a single chip, making

them the most complex RFICs ever built at this frequency range.

Prof. Rebeiz is an IEEE Fellow, an NSF Presidential Young In-

vestigator, an URSI Koga Gold Medal Recipient, an IEEE MTT

Distinguished Young Engineer (2003), and the 1998 Eta-Kappa-

Nu Professor of the Year and the 1998 Amoco Teaching Award

given to be the best undergraduate teacher at the University of

Michigan.



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RF MEMS, BST, and GaAs Varactor System-Level …kentesar/Kamran_J3.pdf · RF MEMS, BST, and GaAs Varactor System-Level Response in Complex Modulation Systems Kamran Entesari,1 Gabriel