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1Ultralow-power MEMS-based Radio for WBAN

Cours Highlight© C. Enz 2014

ÉCOLE POLYTECHNIQUE

FÉDÉRALE DE LAUSANNE

Cours Highlight

Ultralow-power MEMS-based Radio for WBAN

Christian Enz1, Aravind Heragu2, David Ruffieux3

1) ICLAB, IMT-STI-EPFL, Switzerland

2) SemTech, Switzerland

3) CSEM, Switzerland

ICLAB

Introduction

High-Q resonators

Co-design of IC with MEMS devices VCO

LNA

PA

MEMS-based radio architectures

Conclusion

Outline

© C. Enz | 2014 Ultralow-power MEMS-based Radio for WBAN Slide 1



Introduction

ICLAB

WBAN Require ULP Miniaturized Sensor Nodes

Wireless body area networks (WBAN) for health monitoring, connecting wearable devices and as smart user interface

The nodes feature sensing, processing, storing and wireless communication

They are usually battery powered or use remote powering

They require ultralow-power (ULP) and miniaturized wireless sensor nodes

Combination of CMOS system-on-chip (SoC), RF and LF MEMS in a system-in-package (SiP) to achieve a 2.4 GHz, <mW-level, <20 mm3 node


battery powered nodes

remote powered nodes

Introduction

ICLAB

2.4GHz

mW‐level

<4x4x1 mm3

WBAN

The WiserBAN EU Project – Conceptual view


Hearing aids

Cardiac implants

Cochlear implants

Insulin pumps

RF+IF+LF MEMS

RF & DSP SoC

RADIO IC65nm CMOS

RF & LF MEMS

WiserBANmicrosystem

Heterogenous SiP

Miniatureantenna

EU-FP7-WiserBAN Project at www.wiserban.eu



Introduction

ICLAB

MEMS-based Circuits for Wireless Connectivity

Use BAW resonator to build RF oscillators and filters

Silicon resonator for frequency reference and real-time-clock (RTC)

Hybrid integration (yield, reduced complexity)

0-level packaging (vacuum sealing & interconnects)


EU-FP7-Go4Time Project at www.go4time.eu

MEMSSi Resonator FBAR & ALN

CMOS SoC IC

(Radio, DSP, Memory,...)

Introduction

ICLAB

Simplified MEMS-based Transceiver Architecture

Front-end filters before the LNA Interferers and image rejection, relax linearity requirements, avoid impedance matching network

Front-end filters after the power amplifier (PA) Spurious filtering, avoid impedance matching network

Synthesizer Fixed low phase noise RF LO thanks to high Q

Merged Time & Frequency reference with LF silicon resonator (SiRes)


Digital B

aseb

and



ICLAB

Introduction

High-Q resonators


LNA

PA


Conclusion

Outline


High-Q Resonators

ICLAB

High-Q Resonators

On-chip inductors usually have low Q (Q typically smaller than 10 for f0=1GHz)

This affects phase noise (inversely proportional to Q2) and power consumption (inversely proportional to Q)

Resonators with higher Q are hence needed for frequency reference in frequency synthesis

There are different high-Q resonators depending on the frequency range Quartz crystals can be used up to about a few 10 MHz

Surface Acoustic Wave (SAW) resonators can be used from 100 MHz to 1 GHz

Bulk Acoustic Wave (BAW) resonators can be use from 1 GHz to 10 GHz

They all have rather large Q-factor and similar characteristics




High-Q Resonators

ICLAB M.-A. Dubois et al. ISSCC 2005.

Bulk Acoustic Wave (BAW) Devices

AlN piezo layer sandwiched between two electrodes

Thickness around 1 µm allows for f0≈1 to 7 GHz

Acoustic isolation from substrate for high-Q Thin-film Bulk Acoustic Resonator or FBAR (membrane resonators)

Solidly mounted resonator or SMR (resonators with acoustic reflector)

Coupling coefficients up to 7% and Q≈500 to 1000

Frequency trimming by loading top electrode (reduces res. freq.)


FBAR

AlN

electrodes

Si wafer

SMR

AlN

Si wafer

electrodes

acoustic

reflector

High-Q Resonators

ICLAB

Si Flexural mode MEMS Resonators


Z

X

Y

J. Baborowski, et al., Freq. Control Symp. 2007.

20B_32kHz_380

y = -0.000004x 3 - 0.034592x 2 + 0.071019 x - 0.480839R2 = 0.998416

5

-40-35-30-25-20-15-10

-50

-40 -30 -20 -10 0 10 20 30 40

T (°C) with Tref=25°C

f/f

(ppm

)

Description ValueFlexural mode Out-of-planeStructure Optimized for zero TCF Resonant frequency 25kHz to 30kHzQuality factor 10 MHz

in air 500 to 1000In vacuum 2000 to 3000

Coupling coefficient k2 1.1%Qvac*k2 FoM 20 to 30TCF 0.07 to -2 ppm/°C



High-Q Resonators

ICLAB

High-Q Resonator Equivalent Circuit

The general equivalent circuit of a high-Q resonator is given by


The access resistance Rs and inductance Ls can usually be neglected. Assuming that Qp1, and combining all the capacitances between nodes 1 and 2 into a single capacitance C0, the circuit reduces to the simplified one given below

high-Q resonator Rs Ls

CmRm Lm

RpCp

intrinsic resonator

C10 C20

1 2

C12

1

CmRm Lm

C02 10 20

0 1210 20

pC C

C C CC C

High-Q Resonators

ICLAB

High-Q Resonator Simplified Equivalent Circuit

The simplified equivalent circuit of a high-Q resonator is basically the connection of a motional impedance Zm and a parasitic parallel capacitance C0

Zm is made of a series RmLmCm resonant circuit that corresponds to the mechanical part of the resonator defining the mechanical resonant frequency m and quality factor Qm

The motional inductance Lm is proportional to the mass of the resonator whereas the motional capacitance Cm is proportional to the inverse of its stiffness. The motional resistance Rm represents the mechanical losses

The motional current Im flowing in the motional branch is proportional to the velocity of oscillation


1

1

mm m

m mm

m m m m

L C

LQ

R C R



High-Q Resonators

ICLAB

Resonance and Anti-resonance Frequencies

The imaginary part of the high-Q resonator impedance Zres is equal to zero at the resonance (or series resonance) and anti-resonance (or parallel resonance) frequencies s and p, respectively

Assuming that Qm1, s and p can be approximated by


11ands m p r r

m m

K kL C

Where K is defined as0

1andmCK k KC

Typical values: fm 1-10 GHz, K 5-6% , Qm 500-1000, M 25-60

1

1

mm m

m mm

m m m m

L C

LQ

R C R

High-Q Resonators

ICLAB

Coupling Factor

K is linked to the coupling factor which represents the ratio of the stored mechanical energy to the electrical energy and is given by


2 22 1

8 1 8since usuallyeff

Kk K K

K

The higher this coupling factor, the larger the tunability of a VCO or the bandwidth of a filter

The product M = K·Qm represents the ratio of the current in the motional and dielectric branch, which should be maximized

This is why M is often used as a factor of merit for piezoelectric resonators

0 0

1mm m

m m

CM K Q Q

C R C



High-Q Resonators

ICLAB

Mechanical Energy

The motional inductance Lm is proportional to the mass of the resonator whereas the motional capacitance Cm is proportional to the inverse of its stiffness. The motional resistance Rm represents the mechanical losses

The motional current Im flowing in the motional branch is proportional to the velocity of oscillation

The mechanical energy of oscillation is given by


2 2 2

22 22

m m m m m m mm m

m mm m

L I I Q R I QE P

C

The mechanical power dissipated in the resonator is given by2 2

22 2

m m mm

m m m

R I IP

Q C

High-Q Resonators

ICLAB

High-Q Resonator Oscillator – Motional Impedance

The high-Q resonator motional impedance is given by


1 11

1with and (mistuning)

mm m m m m m

m m m m

m m mm

m m m m m

Z R j L R j R jQj C C

LQ

R C R

In high-Q oscillators, is always close to m (or close to 0) and therefore the pulling p is always much smaller than unity

1m

m mp

Close to the series resonance m, Zm can then be approximated as

2 21 2m m m m m m m

m m m m

p pZ R j R j pQ R jX X

C C with

Hence2

2m

m mp



High-Q Resonators

ICLAB

Resonator Impedance


2

20

11

1

m m mres

m

p p p

jQ

Zj C C

jQ

0

1 1, , , , 1 , m

m m p m p mm m mm m

CQ k Q k Q k K K

R C CL C

2

0 0 0 0

1 1( ) ( )

1andr res m m p res p m

m m m m

M jk MZ Z R Z Z M R

C M j C M C K C

0.95 0.975 1 1.025 1.051

0.5

0

0.5

1

argZres f( )

Zp

2

f

fm m

fx

f p mf f

0.95 0.975 1 1.025 1.051

10

100

1 103

1 104

Zres f( )

Zp

Zr

f

fm m

fx

f

pZ

rZ

p mf f

High-Q Resonators

ICLAB

BAW Resonator 1st-order Model

The resonator impedance Zres is minimum at s


2( ) withp res p m mZ Z M R M K Q

Typical values:

fm 1-10 GHz

K 5-6%

Qm 500-1000

M 25-60

The resonator impedance Zres is maximum at p

( )s res s mZ Z R

0.95 0.975 1 1.025 1.051

10

100

1 103

1 104

Zres f( )

Zp

Zr

f

fm m

fx

f

pZ

sZ

p mf f

resZ

with Rm being a few , Zp is in the k range



High-Q Resonators

ICLAB

Qm 500 K 0.05 zmin 0.042 zmax 25.021 k 1.025

0.95 0.975 1 1.025 1.051

0.5

0

0.5

1

arg zn x( ) 2

x

Qm 500 K 0.05 zmin 0.042 zmax 25.021 k 1.025

0.95 0.975 1 1.025 1.050.01

0.1

1

10

100

zn x( )

zmax

zmin

x

Resonator Normalized Impedance


2

0 2

2

11

( ) ( )

1

mn m res

m

xx j

Qz x C C Z x

j x xj

k Q k

2 2min 0 max 02

1 1, ,

1m m r p m p

m

Kx z C C Z z C C Z k M k k M

MM

p mk f fmx f f mx f f

p mk f f

maxz

minz

High-Q Resonators

ICLAB

BAW Impedance


Parameters unloaded Units

Coupling coefficient K 5.04 %Intrinsic series resonance 2552 MHz

Parallel resonance f p 2605 MHzIntrinsic series resonance quality factor Q m

487

Parallel cap quality factor at f m Q p

381

Motional inductance L m 104.62 nH

Motional capacitance C m 37.19 fF

Motional resistance R m 3.44 Parallel capacitance C p 872.68 fF

Parallel resistance R p 1.06 Parasitic series resistance R s

1.01



Parallel-resonance Oscillator

ICLAB

BAW Lattice Filters RF bandwidth limited by K=Cm/C0

Out-of band rejection depends on cell matching

Proper in/out impedances required

Equivalent to LC/CL network in-band


2.35 109 2.45 109 2.55 109

104

-100

-50

0

50

100

103

102

101

100

ZS

ZP

ph(ZS)

f [Hz]

2.3 109 2.45 109 2.6 109

-20

-10

0

10

S21

f[Hz]

ICLAB

Introduction

High-Q resonators


LNA

PA


Conclusion

Outline





ICLAB

The Cross-coupled Pair for Parallel-resonance Oscillator The classical cross-coupled pair cannot be used with high-Q resonators, since

their dc impedance is extremely high making the cross-coupled pair to latch

The source of the cross-coupled pair has to be decoupled by a capacitor CS to make it stable at dc and show a negative transconductance at the oscillation frequency


M2M1

Xtal

CS

Ib Ib

M3 M4

Not appropriate

since it is a latch

Not appropriate

because of mismatches

Stable at dc thanks to

feedback transistors M3-M4

(Ruffieux oscillator) D. Ruffieux, ESSCIRC ’02.


ICLAB

Modes of Operation

The Ruffieux oscillator can operate in two modes: Relaxation mode independent of the high-Q resonator

Harmonic mode with a resonance frequency set by the high-Q resonator

Relaxation mode can be avoided by proper sizing of coupling capacitor CS

Some frequency tuning is provided by capacitor CD

Circuit can be analyzed in small-signal to extract the conditions for relaxation oscillation

Then the circuit impedance will be derived allowing to find the conditions for avoiding any relaxation and force harmonic oscillations will be derived from a small-signal analysis


M2M1

BAW

CS

Ib Ib

M3 M4

CD

D. Ruffieux, ESSCIRC ’02.




ICLAB

Small-signal Circuit Admittance

Assuming GSGms, the circuit admittance is the given by


Finally circuit reduces to

Sm

LL

in

inc

Csn

G

CsG

V

IY

2

1

2

Balanced operation

0L DC C C

Lm

V1

CDGm· V1

V2

Gm· V2

Gms· VS2Gms· VS1

VS2

Rm Cm

C0

CS

GL GL

VS1 GS GS

V1

CDGm· V1

V2

Gm· V2

Gms· VS2Gms· VS1

VS2

C0

CS

GL GL

VS1 GS GS

VinIin

−Gm/2 −CS/n

CL=C0+CD

Vin

GL/2

−GS/2n

Iin


ICLAB

Conditions for Relaxation Oscillation

This circuit can oscillate (without the high-Q resonator) because of the negative conductance and susceptance

The oscillation frequency and critical transconductance can be found by setting Yc=0, which can then be solved for 0 and Gmcrit. This leads to


1

1

1

2

Lrelaxmcrit

L

S

L

Lrelax

GG

Cn

Cα

C

Gwith

LS CnC or1

Relaxation mode is therefore avoided by choosing

Relaxation can only occur for

LS CnC or1




ICLAB

Harmonic Oscillation

If the losses due to GL are neglected, the circuit impedance is then given by


1

21/

1/1

2

22

L

S

S

ms

s

s

LmSLLS

Smc Cn

Cα

C

Gn

s

s

sCsGCCnsCC

sCGnZ andwith

The real and imaginary parts are then given by

22

2

222422

222

22

2

22222

2

1/

1/1

4

4

1/

2

4

2

s

s

LmSLLS

SLmSLc

smmSLLS

mSc

CGCnCCC

CCnGCnCX

GGCnCCC

GCR

For Gm=0 and Gm→∞ we get Rc=0 and

1

1)(

1)0(0

Lmcc

Lmcc C

GXXC

GXX and

Zc(Gm) forms a circle in the complex plane with a radius given by

12 LCradius


ICLAB

Im

Re

−Xc

−Rm

A

B

pGm

−Rc

Gm=0

Gm=Gmopt

Gm=Gmmax

Gm→∞

−Xm

2 1LC

2m m

m

pZ R j

C

1

LC

1 1 / 2

1LC

1

1LC

Gm=Gmcrit

Complex Plane Impedance Locus

Two intersections A and B for

It can be shown that, similarly to the basic Pierce oscillator, the only stable point is A


12 Lm CR

1For S

L

Cα

n C

S

L

Cα

n C




ICLAB

Oscillation Frequency

If the losses due to GL can be neglected and assuming 0 Gms/(2*CS), capacitance CS can be replaced by a short and the circuit reduces to the parallel circuit on the left, which is equivalent to the series circuit on the right


0L DC C C

2 2 22

22

2Re{ }

222

4 1Im{ }

22

for

for

m m mc c

LLm L

mLc c

L Lm L

G G GR Z ω

CCG C

GCX Z ω

C CG C

The complete circuit with the motional branch of the resonator is shown belowAt the resonance frequency, Rm is

compensated by −Rc and the

circuit reduces to

mmm

L

mm

L

mm

eqm

CL

C

C

C

C

CL

1

211

10


ICLAB

Oscillation Frequency

For high-Q resonator, Rm is small and the locus of Xm with p lies very close to the imaginary axis. The operation point is therefore close to the intersection of the circle with the imaginary axis corresponding to Gm=0 for which the circuit reactance is given by


Lmcc CGXX

1

)0(0

The oscillation frequency can therefore be approximated by solving the following equation for 0

mm

m

LmLmc CCC

p

CXX

0

0

0000

2121

0 0 0

0 0

1 11 1 1

2 2 2 2 2 21 1andm m

D Dm L L m L L

C C C CK K K Kp

C CC C C CC C

1

LC




ICLAB

Critical Transconductance

The critical transconductance is obtained from


2 202mcrit

c mL

GR R

C

Which leads to

Which shows that for a given frequency 0 and a given resonator M and C0, the only way to minimize the power consumption is to reduce the load capacitance CD

Assuming the transistors are biased in weak inversion, the power consumption is given by

222 2 0 0 0 00 2

2 22 1

2withL mD

mcrit L m mm m p m c

C C CCG C R M Q K

Q C M C Q p

2 2 withDD b DD T mcrit TkT

P V I V nU G Uq

where >1 is a factor insuring the start-up and maintenance of the oscillations

1 m mm

m m m m

LQ

R C R


ICLAB

Frequency Tunability

To implement a VCO, the oscillation frequency can be tuned by changing the capacitance value CD

The ability to tune the frequency is evaluated by deriving the relative variation of the resonance frequency due to a relative change in the load capacitance CD


0 0 0

20 0 max0

2 81

D D D

D DD

C C C CK K

C CC C

The tunability is maximum for CD=C0, which also corresponds to a good engineering trade-off for the power consumption

Since 0/0 is proportional to K, tunability is therefore ultimately limited by the K factor, which is usually rather low for high-Q resonators

This limitation has to be accounted for, but can be circumvented by new radio and synthesizer architectures

J. Chabloz, et al., ESSCIRC '07




ICLAB

Oscillation Amplitude and Bias Current

The relation between the actual amplitude and the bias current can be obtained in by calculating the transconductance for the fundamental and equating it to the critical transconductance

Unfortunately no closed form expression can be found, but a numerical evaluation leads to the normalized plot shown below


IbGmcrit

A=8limit case:

0

4

8

12

16

20

0 1 2 3 4 5 6 7 8 9 10Normalized bias current Ib/ Icritmin

AnUT

weakinversion

IC0=8 16 32 64

m m L

L S

Q C nC

C C

E. Vittoz, Springer 2010


ICLAB

−Gm

V2

2CL

V1Vn

GL

In1In2

Lm

VnL

Rm

−Gm

2CL

GL

Cm

Im+In

Linear Noise Analysis

The small-signal circuit of the parallel-resonance oscillator shown on the left is presented in the middle where it includes the noise sources of transistors M1, M2 and of the high-Q resonator (noise source from the top and bottom transistors have been neglected)


M2M1

BAW

CS

Ib Ib

M3 M4

CD

0 2ms

S

G

C




ICLAB

Vn

Lm

Vnm

Cm

CLVnc

Im+In

Vn

Lm

Vnm

RmCm

−Gm/2

CL

(In1−In2)/2

Im+In

Vn

Lm

Vnm

RmCm

Vnc

Im+In

−RcCL

Linear Noise Analysis

If the losses due to GL can be neglected, the circuit reduces to the left circuit which is equivalent to the middle circuit

When the circuit oscillates, the losses due to Rm are compensated by the negative resistance Rc provided by the circuit

The circuit then reduces to the right circuit


Neglecting

losses due to GL

Equivalent series

representation

Equivalent series

representation at resonance

0 2m

L

G

C


ICLAB

Linear Phase Noise Analysis

From the basic linear analysis, the phase noise L is approximately given by


20

22

1

m m m

kT

Q R I

L

22 2 00 2 2

2 12 ( )mcrit L m

osc m mcrit

kTG C R

V Q G

L

which can also be written in terms of the oscillation amplitude Vosc as

2 220 2 2 2 2

1( )m L osc

osc m m L

kTI C V

V R Q C

L

Introducing the critical transconductance, we get




ICLAB

Phase Noise and FoM

Recalling the power consumption in weak inversion


2

02

( ) 12 1 T DD

osc oscm

nU VPFoM

kT V VQ

L

2

02

1( ) 2 1 T DD

osc oscm

nU VP kT

V VQ

L

Figure of merit (actually demerit) to minimize can then be defined as

Does not depend on the reactive components, only depends on Qm, Vosc/VDD and on the noise factor (1+)

We find that the product phase noise x power consumption is actually independent of Gmcrit

2 2DD b DD T mcritP V I V nU G

J. Chabloz, et al., ESSCIRC '07


ICLAB

BAW and LC Comparison – Phase Noise

Comparing phase noise between LC and BAW oscillator assuming identical 0, Vosc, CL, and T


20

20

1 1LC

L LCosc

kT

C QV

L

LC-oscillator BAW oscillator

20

2 2 20

1 m mBAW

L mosc

kT C

C QV

L

The ratio is then given by

01

BAW LC m m LC mm LC L m

LC m L m L

Q C Q CQ Q C C

Q C Q C

since and

LL

The phase noise of the BAW oscillator is therefore much smaller than that of the LC oscillator

Assuming QLC=10, Qm=1000, K=5% and CD=C0 (CL=2C0) leads to

362

LC m LC

m L m

Q C Q KdB

Q C Q




ICLAB

BAW and LC Comparison – Power Dissipation

Comparing critical transconductances between LC and BAW oscillator assuming identical 0, Vosc, CL and assuming transistors are biased in weak inversion



2DD b DD T mcritP V I V nU G

202 L

mcrit BAWm m

CG

Q C

02 L

mcrit LCLC

CG

Q

The ratio is then given by

mcrit BAW LC L

mcrit LC m m

G Q C

G Q C

Since Qm>>QLC and CL>>Cm, there is no big difference in power consumption

Assuming QLC=10, Qm=1000, K=5% and CD=C0 (CL=2C0) leads to ≈0.4


ICLAB

BAW and LC Comparison – Tuning Range

Comparing tuning range between LC and BAW oscillator assuming identical 0, Vosc, CL



0 0

20 0

2 1

D D

DD

C C CK

CC C

0

0 max8

D

D

CK

C

Maximum for CD=C0

0

0

L

L

C

C

Much smaller for the BAW oscillator than for the LC oscillator

Assuming the same CL/CL= CD/CD and K=5% shows that tuning range of the BAW oscillator is about 160 times smaller than that of the LC oscillator




ICLAB

BAW and LC Comparison


Ratio Numerical values

Assuming:QLC=10, Qm=1000, K=5%

and CD=C0 (CL=2C0)

Phasenoise

· ≪ 1 36

Power dissipation

· 1 0.4

Frequency tuning 2

·⁄

1 ⁄ 8≪ 1

1160


ICLAB

BAW Quadrature VCO at 2.1 GHz


> 35 dB

S. Rai, B. Otis, ISSCC 2007.



ICLAB

Introduction

High-Q resonators


LNA

PA


Conclusion

Outline


MEMS-based LNA

ICLAB

Classical Heterodyne RF Front-End

Blocker-reject filter: low insertion loss, decrease linearity requirements

Image-reject filter: high selectivity


LNA

Blocker-rejectFilter

Image-rejectFilter

Off-chip On-chip Off-chip On-chip



MEMS-based LNA

ICLAB

LNA

High-QBAWFilter

Off-chip On-chip Off-chip On-chip

LNA

High-QBAWFilter

Off-chip On-chip

Front-End with High-Q Filter

High-Q BAW filter can provide both low insertion loss and high selectivity

No need for further image rejection


MEMS-based LNA

ICLAB

Filter-LNA interface

BAW resonators require correct termination impedances to work as specified

Impedance matching usually required for correct filter termination

Impedance matching network between BAW filter and LNA can be avoided by using the same BAW resonators within a selective LNA


BAWFilter

MatchingNetwork

LNALNA

BAW Filter +

Selective LNA



MEMS-based LNA

ICLAB

Proposed Selective LNA

Embed lattice filter into the active differential pair taking advantage of the balanced signals


J. Chabloz, et al., ISSCC 2006.

Cc CcIq Iq

Z1

Z2

Vin−

Z1

Z2

Vin+

Iout+ Iout−

MEMS-based LNA

ICLAB

LNA Simulated Gain


Image Band Signal Band

Frequency [GHz]

Effe

ctiv

e Vo

ltage

Gai

n [d

B]

-80

-60

-40

-20

0

20

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8

<30 dB




MEMS-based LNA

ICLAB

Complete BAW Selective LNA


Cc CcIq Iq

Z1

Z2

Z1

Z2

Z1

Z2 Z1

Z2

L1

Vout−

Vout+

Vin−

Vin+

L1

VDD

Lattice prefilter Selective LNA


MEMS-based LNA

ICLAB

LNA Simulated Gain


Image Band Signal Band

Frequency [GHz]

Effe

ctiv

e Vo

ltage

Gai

n [d

B]

-80

-60

-40

-20

0

20

2.1 2.2 2.3 2.4 2.5 2.6 2.7 2.8

> 50 dB




MEMS-based LNA

ICLAB

Layout Implementation and Chip Photograph



MEMS-based LNA

ICLAB

Measured RF Front-End Gain


-50

-40

-30

-20

-10

0

10

20

2200 2300 2400 2500 2600 2700

RF Frequency [MHz]

Volta

ge G

ain

[dB

]

Simulated

Simulated w extracted BAWsand losses

Measured

50 dB




MEMS-based LNA

ICLAB

Measurements Summary


Parameter Units

Supply voltage 1.2 VCurrent consumption 1.5 mAPower consumption 1.8 mW

Voltage gain 12 dBImage rejection > 50 dBRF bandwidth 100 MHz

Noise figure 11 dBInput 1dB compression -26 dBm

Input IP3 -16.1 dBm

SFDR (BW=300kHz, SNR=0dB) 92 dB

fRF = 2.5GHz, fLO = 2.41GHz


MEMS-based LNA

ICLAB

-50

-40

-30

-20

-10

0

10

20

2200 2300 2400 2500 2600 2700RF Frequency [MHz]

Volta

ge G

ain

[dB

]

BAW based LNA


Simulated w extracted BAWsand losses

Measured

Cc CcIq Iq

Z1

Z2

Z1

Z2

Z1

Z2 Z1

Z2

L1

Vout−

Vout+

Vin−

Vin+

L1

VDD

Lattice prefilter Selective LNA




ICLAB

Introduction

High-Q resonators


LNA

PA


Conclusion

Outline


MEMS-based PA

ICLAB

Transmitter: Class-E Power Amplifier Principle


M. Contaldo, et al., ECCTD 2009.



MEMS-based PA

ICLAB

Class-E BAW PA Equivalence

Take advantage of balanced nature of signals in differential structure and combine with lattice filter


Mason’s equivalence allows to combine the parallel branches of the BAW resonators and include them in the shunt capacitor Csh

The motional branch can then be used for the series branch of the class-E PA

1 1m pY Y Y

Loaded shunt device Unloaded series device

2 2m pY Y Y

MEMS-based PA

ICLAB

PPA & PA Implementation with BAW Filter Stage


M. Contaldo, et al., ISSCC 2010.



MEMS-based PA

ICLAB

Transmitter Layout and PCB Assembly


MEMS-based PA

ICLAB

Frequency Selectivity

Without frequency trimming the BAW filter is about 35MHz lower than needed

Relatively good match with simulated value


Measured performance:

VDD = 1.2 V

CF = 2.405 GHz

−1dB BW = 87 MHz

Ripple = 0.7 dB

@ 2.24 GHz = −23 dB

@ 2.56 GHz = −23 dB2.2 2.3 2.4 2.5 2.6

x 109

-25

-20

-15

-10

-5

0

5

10

Frequency [Hz]

Ou

tpu

t Po

we

r [d

Bm

]

SimulatedMeasured




MEMS-based PA

ICLAB

Output Power Control

By sweeping discretely the bias of the power stage

All curves include the BAW filter losses


Measured performance:

VDD = 1.2 V

Pout = −6 dBm to 8.4 dBm

max ηd = 24.2%

max ηov = 21%

0 5 10 15-10

-5

0

5

10

15

20

25

Input DC bias PA step

Effi

cie

ncy

[%] a

nd

Ou

tpu

t Po

we

r [d

Bm

]

Drain efficiencyOver. efficiencyOutput power


MEMS-based PA

ICLAB

Transmitter Layout

0.18µm CMOS technology

1.25 x 1.5 mm2

Integrated in a complete BAW-based transceiver

No external components in the Tx other than the BAW filter and the BALUN for test purposes




MEMS-based PA

ICLAB

Measurements Test-bench


MEMS-based PA

ICLAB

Unmodulated RF Signal

fBAW = 2.339 GHz

fIF= fBAW/32 = 73 MHz

fLC = 28/32 fBAW = 2.046 GHz

fRF = 2.412 GHz

PN, Integer Mode


102

103

104

105

106

107

-160

-140

-120

-100

-80

-60

-40

-20

Offset frequency [Hz]

Ph

ae

no

ise

[dB

c/H

z]

L = −136.6 dBc/Hz at 1 MHz offset



MEMS-based PA

ICLAB

Broadband Spectrum

Curve normalized to

Pout = 5.4 dBm

fBAW = 2.339 GHz

fIF = 73 MHz

Spurs attenuations

IF 2nd h. @ -48 dBc

-1, +3, -5 h. < -63 dBc

-3, +5, -7 h. < -50 dBc

DCO feedth.: -61 dBc


-5 -4 -3 -2 -1 0 1 2 3 4 5-80

-70

-60

-50

-40

-30

-20

-10

0

fBAW

+ n fIF

dB

c

MEMS-based PA

ICLAB

Modulated Spectra

1 Mb/s GFSK

BT modulation

-21.7 dBc, -21.4 dBc@ ±500 kHz

ACP 2: -42 dBm

ACP 3: -49 dBm

BT LE modulation

ACP 2: -41 dBm

ACP 3: -44 dBm


-4 -3 -2 -1 0 1 2 3 4 -80

-70

-60

-50

-40

-30

-20

-10

0

Frequency offset [MHz]

dB

c

BT mask

BT mi=0.34BT LE



MEMS-based PA

ICLAB

Power Consumption Breakdown


Block Cons. [mW]

Synthesis 11.11

BAW DCO 2.37

Dividers, ΣΔ 3.28

LC VCO 3.38

PLL div., PFD, CP 2.08

Selective TX 36.19

IF buffer 0.56

RC/CR 2.34

SSB mixer 3.68

PPA 3.82

PA 25.79

Chip in TX mode 47.3

55%

8%

8%

6%

23%

At Pout = 5.4 dBm

PA PPA

SSB mix RC/CR, Buf IF

Synthesis

MEMS-based PA

ICLAB

BAW based PA


2.2 2.3 2.4 2.5 2.6

x 109

-25

-20

-15

-10

-5

0

5

10

Frequency [Hz]

Ou

tpu

t Po

we

r [d

Bm

]

SimulatedMeasured




ICLAB

Introduction

High-Q resonators


LNA

PA


Conclusion

Outline


MEMS-based Receiver Architectures

ICLAB

N

BAW DCO

PLLAD

PLLLF

SiResRTC M

ADC

DIG

ITAL BASEBAND

LNA

PA

BAW

2.32GHz 80-160MHz

RX DATA

CHANNEL

SELECT

TX DATA

PROG SYS

CLOCK

2.4GHz

Wide-IF to Compensate BAW LO Lack of Tuning

IF signal obtained from BAW LO with fractional-N divider Quantization noise issue !

Quadrature needed for I & Q or SSB mixing

Wide bandwidth PLL for single point quasi direct modulation at up to Mbps No 20log(N) noise scaling, higher fREF


D.Ruffieux et al., JSSC, Jan. 2009, ESSCIRC 2010.




ICLAB

PLL-free Super-high IF Rx Architecture with BAW Filter

Channel selection directly done at RF (super-high-IF) by the proposed low powerBAW pseudo-lattice based Amplifier-Mixer-Filter (AMF) cell.

Quadrature down-conversion to baseband by sub-sampling.

BAW pseudo-lattice also provides bandpass anti-aliasing filtering

All clocks required for channel selection, down-conversion, ADC - generated byinteger division of the BAW DCO signal.


A. Heragu et al., ESSCIRC 2012, A. Heragu et al., RFIC 2012.


ICLAB

Measurements

Integrated in 0.18-µm CMOS process.

BAW resonators wire bonded to the chip.


Block Current (mA)

LNA 1.5

BAW DCO 1.21

AMF cell 1.08

IF amplifier 4x 0.1

Dividers, buffers, ADC, demodulator

1.75

Total 5.94

Power supply VDD = 1.8 V



ICLAB

Introduction

High-Q resonators


LNA

PA


Conclusion

Outline


ICLAB

WBAN require ULP and miniaturized nodes that feature sensing, processing, storage and wireless connectivity

Such nodes can be designed taking advantage of MEMS resonators to build an ULP MEMS-based radio

BAW resonators can be used in: the LNA to provide selectivity, relax linearity requirements and avoid any impedance

matching

a class-E PA for adding selectivity and avoid impedance matching while achieving good power efficiency

RF oscillator to significantly improve the phase noise

High-Q silicon resonators can be used in the frequency synthesis replacing the quartz crystal for reference frequency and RTC

The use of these MEMS devices requires new and optimized radio architecturesand dedicated circuits

A 2.4 GHz radio for ULP Bluetooth link has been demonstrated

Conclusion




ICLAB

© C. Enz |


Slide

74Ultra-low Power 2.4 GHz MEMS-based Radio (RFIC 2012)

Thank You for your Attention !

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