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Chapter 4
Digital Processing of ConDigital Processing of Continuous-Time Signalstinuous-Time Signals
§4.1 Introduction
Digital processing of a continuous-time signal involves the following basic steps:
(1) Conversion of the continuous-time signal into a discrete-time signal,
(2) Processing of the discrete-time signal,
(3) Conversion of the processed discrete- time signal back into a continuous-time signal
§4.1 Introduction
Conversion of a continuous-time signal into digital form is carried out by an analog-to- digital (A/D) converter
The reverse operation of converting a digital signal into a continuous-time signal is performed by a digital-to-analog (D/A) converter
§4.1 Introduction
Since the A/D conversion takes a finite amount of time, a sample-and-hold (S/H) circuit is used to ensure that the analog signal at the input of the A/D converter remains constant in amplitude until the conversion is complete to minimize the error in its representation
§4.1 Introduction
To prevent aliasing, an analog anti-aliasing filter is employed before the S/H circuit
To smooth the output signal of the D/A converter, which has a staircase-like waveform, an analog reconstruction filter is used
§4.1 Introduction
Since both the anti-aliasing filter and the reconstruction filter are analog lowpass filters, we review first the theory behind the design of such filters
Also, the most widely used IIR digital filter design method is based on the conversion of an analog lowpass prototype
Anti-Aliasing
filter
Digitalprocessor D/A
ReconstructionfilterA/DS/H
Complete block-diagram
§4.2 Sampling of Continuous-Time Signals
As indicated earlier, discrete-time signals in many applications are generated by sampling continuous-time signals
We have seen earlier that identical discrete- time signals may result from the sampling of more than one distinct continuous-time function
§4.2 Sampling of Continuous-Time Signals
In fact, there exists an infinite number of continuous-time signals, which when sampled lead to the same discrete-time signal
However, under certain conditions, it is possible to relate a unique continuous-time signal to a given discrete-time signal
§4.2 Sampling of Continuous-Time Signals
If these conditions hold, then it is possible to recover the original continuous-time signal from its sampled values
We next develop this correspondence and the associated conditions
§4.2.1 Effect of Sampling in the Frequency Domain
Let ga(t) be a continuous-time signal that is sampled uniformly at t = nT, generating the sequence g[n] where
g[n]=ga(nT), -∞ <n<∞
with T being the sampling period The reciprocal of T is called the sampling fre
quency FT , i.e.,
FT=1/T
§4.2.1 Effect of Sampling in the Frequency Domain
Now, the frequency-domain representation of ga(t) is given by its continuos-time Fourier transform(CTFT):
dtetgjG tjaa
)()(
nnjj engeG ][)(
The frequency-domain representation of g[n]
is given by its discrete-time Fourier transform (DTFT):
§4.2.1 Effect of Sampling in the Frequency Domain
To establish the relation between Ga(jΩ) and G(ejω) ,we treat the sampling operation mathematically as a multiplication of ga (t) by a periodic impulse train p(t):
n
nTttp )()(
)(
)()(
tp
tgtg p
§4.2.1 Effect of Sampling in the Frequency Domain
p(t) consists of a train of ideal impulses with a period T as shown below
n
aap nTtnTgtptgtg )()()()()(
The multiplication operation yields an impulse train:
§4.2.1 Effect of Sampling in the Frequency Domain
g p (t) is a continuous-time signal consisting of a train of uniformly spaced impulses with the impulse at t = nT weighted by the sampled value g a (nT) of g a (t) at the instant
§4.2.1 Effect of Sampling in the Frequency Domain
There are two different forms of Gp(jΩ): One form is given by the weighted sum of th
e CTFTs of δ(t-nT) :
n
nTjap enTgjG )()(
ktj
k TnTejk
TnTt
)(1
)(
where ΩT =2π/T and Φ(jΩ) is the CTFT of φ(t)
To derive the second form, we make use of the Poisson’s formula:
§4.2.1 Effect of Sampling in the Frequency Domain
For t=0ktj
k TnTejk
TnTt
)(1
)(
k Tnjk
TnT )(
1)(
Now, from the frequency shifting property of the CTFT, the CTFT of g a (t) e-jΨt is given by Ga
(j(Ω+Ψ))
reduces to
§4.2.1 Effect of Sampling in the Frequency Domain
Substituting φ(t)=ga(t)e-jΨt in
k Tnjk
TnT )(
1)(
k Tan
nTja kjG
TenTg ))((1)(
we arrive at
By replacing Ψ with Ω in the above equation we arrive at the alternative form of the CTFT G p (jΩ ) of g p (t)
§4.2.1 Effect of Sampling in the Frequency Domain
The alternative form of the CTFT of g p(t) is given by
kTaTp kjGjG )()( 1
Therefore, G p (jΩ ) is a periodic function of Ω consisting of a sum of shifted and scaled replicas of G a (jΩ ) , shifted by integer multiples of ΩT
and scaled by 1/T
§4.2.1 Effect of Sampling in the Frequency Domain
The term on the RHS of the previous equation for k=0 is the baseband portion of G p (jΩ ), and each of the remaining terms are the frequency translated portions of
G p (jΩ ) The frequency range
22TT
is called the baseband or Nyquist band
§4.2.1 Effect of Sampling in the Frequency Domain
Assume ga(t) is a band-limited signal with a CTFT Ga(jΩ) as shown below
Ga(jΩ)
Ωm-Ωm
Ω0
ΩT-ΩT 2Ω
T
3ΩT 0
P(jΩ)
Ω
1 … …
The spectrum P(jΩ) of p(t) having a sampling period T=2π/ ΩT is indicated below
§4.2.1 Effect of Sampling in the Frequency Domain
Two possible spectra of G p (jΩ ) are shown below
§4.2.1 Effect of Sampling in the Frequency Domain
It is evident from the top figure on the previous slide that if ΩT>2Ωm,there is no overlap between the shifted replicas of
Ga(jΩ) generating Gp(jΩ) On the other hand, as indicated by the figure
on the bottom, if ΩT<2Ωm ,there is an overlap of the spectra of the shifted replicas of Ga(jΩ) generating Gp(jΩ)
§4.2.1 Effect of Sampling in the Frequency Domain
Hr(jΩ)ga(t)gp(t)
p(t)
)(ˆ tga
If ΩT>2Ωm , ga(t) can be recovered exactly from gp(t) by passing it through an ideal lowpass filter Hr(jΩ) with a gain T and a cutoff frequency Ωc greater than Ωm and less than ΩT -Ωm
as shown below
§4.2.1 Effect of Sampling in the Frequency Domain
The spectra of the filter and pertinent signals are shown below
§4.2.1 Effect of Sampling in the Frequency Domain
On the other hand, if ΩT<2Ωm ,due to the overlap of the shifted replicas of Ga(jΩ) , the spectrum Ga(jΩ) cannot be separated by filtering to recover Ga(jΩ) because of the distortion caused by a part of the replicas immediately outside the baseband folded back or aliased into the baseband
§4.2.1 Effect of Sampling in the Frequency Domain
Sampling theorem – Let ga(t) be a band- limite
d signal with CTFT Ga(jΩ) =0 for |Ω|>Ωm
Then ga(t) is uniquely determined by its samples
ga(nT) ,-∞≤n≤∞ if
ΩT≥2Ωm
where ΩT=2π/T
§4.2.1 Effect of Sampling in the Frequency Domain
The condition ΩT≥2Ωm is often referred to a
s the Nyquist condition
The frequency ΩT/2 is usually referred to as t
he folding frequency
§4.2.1 Effect of Sampling in the Frequency Domain
Given ga(nT) , we can recover exactly ga(t) by generating an impulse train
n
ap nTtnTgtg )()()(
and then passing it through an ideal lowpass filter Hr(jΩ) with a gain T and a cutoff frequency Ωc satisfying
m < c < (T - m )
§4.2.1 Effect of Sampling in the Frequency Domain
The highest frequency Ωm contained in ga(t) is usually called the Nyquist frequency
since it determines the minimum sampling
frequency ΩT=2Ωm that must be used to fully recover ga(t) from its sampled version
The frequency 2Ωm is called the Nyquist rate
§4.2.1 Effect of Sampling in the Frequency Domain
Oversampling – The sampling frequency is higher than the Nyquist rate
Undersampling – The sampling frequency is lower than the Nyquist rate
Critical sampling – The sampling frequency is equal to the Nyquist rate
Note: A pure sinusoid may not be recoverable from its critically sampled version
§4.2.1 Effect of Sampling in the Frequency Domain
In digital telephony, a 3.4 kHz signal bandwidth is acceptable for telephone conversation
Here, a sampling rate of 8 kHz, which is greater than twice the signal bandwidth, is used
§4.2.1 Effect of Sampling in the Frequency Domain
In high-quality analog music signal processing, a bandwidth of 20 kHz has been determined to preserve the fidelity
Hence, in compact disc (CD) music
systems, a sampling rate of 44.1 kHz, which is slightly higher than twice the signal bandwidth, is used
§4.2.1 Effect of Sampling in the Frequency Domain
Example - Consider the three continuous- time sinusoidal signals:
ga(t)=cos(6πt)
ga(t)=cos(14πt)
ga(t)=cos(26πt) Their corresponding CTFTs are:
G1(jΩ)=π[δ(Ω-6π)+δ(Ω+6π)]
G2(jΩ)=π[δ(Ω-14π)+δ(Ω+14π)]
G3(jΩ)=π[δ(Ω-26π)+δ(Ω+26π)]
§4.2.1 Effect of Sampling in the Frequency Domain
These three transforms are plotted below
§4.2.1 Effect of Sampling in the Frequency Domain
These continuous-time signals sampled at a rate of T = 0.1 sec, i.e., with a sampling frequency ΩT=20πrad/sec
The sampling process generates the continuous-time impulse trains, g1p(t),
g2p(t), and g3p(t) Their corresponding CTFTs are given by
31,)(10)( k Tp kjGjG
§4.2.1 Effect of Sampling in the Frequency Domain
Plots of the 3 CTFTs are shown below
§4.2.1 Effect of Sampling in the Frequency Domain
These figures also indicate by dotted lines the frequency response of an ideal lowpass filter with a cutoff at Ωc=ΩT/2=10π and a gain T=0.1
The CTFTs of the lowpass filter output are also shown in these three figures
In the case of g1 (t), the sampling rate satisfies the Nyquist condition, hence no aliasing
§4.2.1 Effect of Sampling in the Frequency Domain
Moreover, the reconstructed output is precis
ely the original continuous-time signal
In the other two cases, the sampling rate do
es not satisfy the Nyquist condition, resulting
in aliasing and the filter outputs are all equal
to cos(6πt)
§4.2.1 Effect of Sampling in the Frequency Domain
Note: In the figure below, the impulse appearing at Ω = 6π in the positive frequency passband of the filter results from the aliasing of the impulse in G2(jΩ) at Ω = −14π
Likewise, the impulse appearing at Ω = 6π in the positive frequency passband of the filter results from the aliasing of the impulse in G3(jΩ) at Ω = 26π
§4.2.1 Effect of Sampling in the Frequency Domain
We now derive the relation between the
DTFT of g[n] and the CTFT of gp[t] To this end we compare
n
njj engeG ][)(
n
nTjap enTgjG ][)(
and make use of g[n]=ga[nT] ,-∞<n< ∞
with
§4.2.1 Effect of Sampling in the Frequency Domain
Observation: We have
Tpj jGeG /)()(
Tj
p eGjG )()(
k
Tap kjGT
jG ))((1
)(
From the above observation and
or, equivalently,
§4.2.1 Effect of Sampling in the Frequency Domain
we arrive at the desired result given by
ka
kTa
Tk
Taj
T
kj
TjG
T
jkT
jGT
jkjGT
eG
)2
(1
)(1
)(1
)( /
§4.2.1 Effect of Sampling in the Frequency Domain
The relation derived on the previous slide can be alternately expressed as
)(1
)(
k TaTj jkjG
TeG
Tj
p eGjG )()(
Tpj jGeG /)()(
it follows that G(ejω) is obtained from Gp(jΩ) by applying the mapping Ω=ω/T
or from
From
§4.2.1 Effect of Sampling in the Frequency Domain
Now, the CTFT Gp(jΩ) is a periodic function
of Ω with a period ΩT=2π/T
Because of the mapping, the DTFT G(ejω)
is a periodic function of ω with a period 2π
§4.2.2 Recovery of the Analog Signal
We now derive the expression for the output
ĝa(t) of the ideal lowpass reconstruction filter Hr(jΩ) as a function of the samples g[n]
The impulse response hr (t) of the lowpass reconstruction filter is obtained by taking the inverse DTFT of Hr(jΩ) :
)( jHr
c
cT
,0,
§4.2.2 Recovery of the Analog Signal
Thus, the impulse response is given by
np nTtngtg )(][)(
tt
t
deTdejHth
T
c
tjtjrr
c
c
,2/
)sin(2
)(21)(
The input to the lowpass filter is the impulse train gp (t) :
§4.2.2 Recovery of the Analog Signal
Therefore, the output ĝa(t) of the ideal lowpass filter is given by:
TnTt
TnTtngtg
na /)(
]/)(sin[][)(ˆ
nrpra nTthngtgthtg )(][)()()(^
*
Substituting hr (t) =sin(Ωct)/(ΩTt/2) in the above and assuming for simplicity
c= T/2= /T, we get
§4.2.2 Recovery of the Analog Signal
The ideal bandlimited interpolation process is illustrated below
§4.2.2 Recovery of the Analog Signal
It can be shown that when Ωc= ΩT/ 2 in
2/
)sin()(
t
tth
T
cr
TnTt
TnTtngtg
na /)(
]/)(sin[][)(ˆ
)(][)( rTgrgrTg aa for all integer values of r in the range-∞<r<∞
we observe
hr(0)=1 and hr(nT)=0 for n≠0 As a result, from
§4.2.2 Recovery of the Analog Signal
The relation
)(][)(ˆ rTgrgrTg aa
holds whether or not the condition of the sampling theorem is satisfied
However, ĝa(rT)=ga(rT) for all values of t only if the sampling frequency ΩT satisfies the condition of the sampling theorem
§4.2.3 Implication of the Sampling Process
Consider again the three continuous-time signals: g1(t) =cos(6πt) ,g2(t) =cos(14πt),
and g3(t) =cos(26πt)
The plot of the CTFT G1p (jΩ) of the sampled version g1p(t) of g1(t) is shown below
§4.2.3 Implication of the Sampling Process
From the plot, it is apparent that we can recover any of its frequency-translated versions cos[(20k±6)πt] outside the baseband by passing g1p(t) through an ideal analog bandpass f
ilter with a passband centered at Ω= (20k±6)π
§4.2.3 Implication of the Sampling Process
For example, to recover the signal cos(34πt), it will be necessary to employ a bandpass filter with a frequency response
otherwise,0)34()34(,1.0
)(
jHr
where ∆ is a small number
§4.2.3 Implication of the Sampling Process
Likewise, we can recover the aliased baseband component cos(6πt) from the
sampled version of either g2p(t) or g3p(t) by passing it through an ideal lowpass filter with a frequency response:
otherwise,0)6()6(,1.0
)(
jHr
§4.2.3 Implication of the Sampling Process
There is no aliasing distortion unless the original continuous-time signal also contains the component cos(6πt)
Similarly, from either g2p(t) or g3p(t) we can recover any one of the frequency-translated versions, including the parent continuous-time signal g2 (t) or g3(t) as the case may be, by employing suitable filters
§4.3 Sampling of Bandpass Signals
The conditions developed earlier for the unique representation of a continuous-time signal by the discrete-time signal obtained by uniform sampling assumed that the continuous-time signal is bandlimited in the frequency range from dc to some frequency Ωm
Such a continuous-time signal is commonly referred to as a lowpass signal
§4.3 Sampling of Bandpass Signals There are applications where the continuou
s- time signal is bandlimited to a higher frequency range ΩL≤|Ω|≤ΩH with ΩL> 0
Such a signal is usually referred to as thebandpass signal
To prevent aliasing a bandpass signal can of course be sampled at a rate greater than twice the highest frequency, i.e. by ensuring
ΩT ≥2ΩH
§4.3 Sampling of Bandpass Signals However, due to the bandpass spectrum of
the continuous-time signal, the spectrum of the discrete-time signal obtained by sampling will have spectral gaps with no signal components present in these gaps
Moreover, if ΩH is very large, the sampling
rate also has to be very large which may not be practical in some situations
§4.3 Sampling of Bandpass Signals
A more practical approach is to use under- sampling
Let ΔΩ = ΩH - ΩL define the bandwidth of the bandpass signal
Assume first that the highest frequency ΩH
contained in the signal is an integer multiple of the bandwidth, i.e.,
ΩH =M(ΔΩ )
§4.3 Sampling of Bandpass Signals
We choose the sampling frequency ΩT to satisfy the condition
T = 2() = 2H/M
which is smaller than 2ΩH , the Nyquist rate
Substitute the above expression for ΩT in
kTaTp kjGjG )()( 1
§4.3 Sampling of Bandpass Signals This leads to
k kjjGjG aTp )()( 21
As before,Gp (jΩ) consists of a sum of Ga (jΩ) and replicas of Ga (jΩ) shifted by integer multiples of twice the bandwidth ∆Ω and scaled by 1/T
The amount of shift for each value of k ensures that there will be no overlap between all shifted replicas →no aliasing
§4.3 Sampling of Bandpass Signals
Figure below illustrate the idea behind
-H -L L H0
Gp(j)
-H -L L H0
Gp(j)
§4.3 Sampling of Bandpass Signals As can be seen, ga(t) can be recovered from
gp(t) by passing it through an ideal bandpass filter with a passband given by ΩL≤|Ω|≤ΩH and a gain of T
Note: Any of the replicas in the lower frequency bands can be retained by passing
gp(t) through bandpass filters with passbands ΩL≤ - k(ΔΩ) ≤|Ω|≤ΩH - k(ΔΩ) , 1≤k≤ M-1 providing a translation to lower frequency ranges
§4.4.1 Analog Lowpass Filter Specifications
Typical magnitude response |Ha(jΩ)| of an
analog lowpass filter may be given as indicated below
§4.4.1 Analog Lowpass Filter Specifications
In the passband, defined by 0≤Ω≤Ωp, we require 1-p |Ha(j)| 1+ p , || p
i.e., |Ha(jΩ)| approximates unity within an error of ±δp
In the stopband, defined by Ωs≤Ω<∞, we require |Ha(j)| s , s
i.e., |Ha(jΩ)| approximates zero within an error of δs
§4.4.1 Analog Lowpass Filter Specifications
Ωp – passband edge frequency Ωs – stopband edge frequency δp – peak ripple value in the passband δs – peak ripple value in the stopband Peak passband ripple
dBpp )1(log20 10
dBss )(log20 10 Minimum stopband attenuation
§4.4.1 Analog Lowpass Filter Specifications
Magnitude specifications may alternately be given in a normalized form as indicated below
§4.4.1 Analog Lowpass Filter Specifications
Here, the maximum value of the magnitude in the passband assumed to be untiy
21/1 – Maximum passband deviation,
given by the minimum value of the magnitude in the passband
1/A – Maximum stopband magnitude
§4.4.1 Analog Lowpass Filter Specifications
Two additional parameters are defined –
(1) Transition ratio k = p/s
For a lowpass filter k<1
121
A
k(2) Discrimination parameter
Usually k<<1
§4.4.2 Butterworth Approximation
The magnitude-square response of an N-th order analog lowpass Butterworth filter is given by
Nc
a jH 22
)/(1
1)(
First 2N-1 derivatives of |Ha(jΩ)|2 at Ω=0 are equal to zero
The Butterworth lowpass filter thus is said to have a maximally-flat magnitude at Ω=0
§4.4.2 Butterworth Approximation
Gain in dB is G (Ω)=10 log10|Ha(jΩ)|2 As G(0)=0 and
G (Ωc )= 10 log10(0.5)=−3.0103 -≅ 3dB
Ωc is called the 3-dB cutoff frequency
§4.4.2 Butterworth Approximation
Typical magnitude responses with Ωc=1
0 1 2 30
0.2
0.4
0.6
0.8
1
Mag
nitu
de
Butterworth Filter
N = 2N = 4N = 10
§4.4.2 Butterworth Approximation
Two parameters completely characterizing a Butterworth lowpass filter are Ωc and N
These are determined from the specified bandedges Ωp and Ωs, and minimum
21/1 passband magnitude , and maximum stopband ripple 1/A
§4.4.2 Butterworth Approximation Ωc and N are thus determined from
22
2
11
)/(11)(
N
cppa jH
22
2 1)/(1
1)(A
jH Ncs
sa
)/1(log)/1(log
)/(log]/)1[(log
21
10
110
10
2210
kkA
Nps
Solving the above we get
§4.4.2 Butterworth Approximation Since order N must be an integer, value obt
ained is rounded up to the next highest integer
This value of N is used next to determine Ωc by satisfying either the stopband edge or the passband edge specification exactly
If the stopband edge specification is satisfied, then the passband edge specification is exceeded providing a safety margin
§4.4.2 Butterworth Approximation
Transfer function of an analog Butterworth lowpass filter is given by
N
Nc
NN
Nc
Na
pssdssDCsH
1
1
0)()(
)(
Nep NNjc
1,]2/)12([
Denominator DN(s) is known as the Butterworth polynomial of order N
where
§4.4.2 Butterworth Approximation Example – Determine the lowest order of a
Butterworth lowpass filter with a 1-dB cutoff frequency at 1kHz and a minimum attenuation of 40 dB at 5kHz
Now1
1
1log10
210
401
log10210
Awhich yields A2=10,000
which yields ε2=0.25895and
§4.4.2 Butterworth Approximation Therefore
51334.19611 2
1
A
k
51
p
s
k
2811.3)/1(log
)/1(log
10
110 =k
kN
We choose N=4
Hence
and
§4.4.3 Chebyshev Approximation The magnitude-square response of an N-th
order analog lowpass Type 1 Chebyshev filter is given by
1),coshcosh(
1),coscos()(
1
1
N
NTN
)/(1
1)( 22
2
pNa
TsH
where TN(Ω) is the Chebyshev polynomial of order N:
§4.4.3 Chebyshev Approximation Typical magnitude response plots of the ana
log lowpass Type 1 Chebyshev filter are shown below
0 1 2 30
0.2
0.4
0.6
0.8
1
Mag
nitu
de
Type 1 Chebyshev Filter
N = 2N = 3N = 8
§4.4.3 Chebyshev Approximation
If at Ω=Ωs the magnitude is equal to 1/A, then
2222 1
)/(11)(
ATjH
psNsa
)/1(cosh)/1(cosh
)/(cosh)/1(cosh
11
1
1
21
kkA
Nps
Order N is chosen as the nearest integer greater than or equal to the above value
Solving the above we get
§4.4.3 Chebyshev Approximation The magnitude-square response of an N-th
order analog lowpass Type 2 Chebyshev (also called inverse Chebyshev) filter is given by
22
2
)/(
)/(1
1)(
sN
psNa
T
TjH
where TN(Ω) is the Chebyshev polynomial of order N
§4.4.3 Chebyshev Approximation Typical magnitude response plots of the ana
log lowpass Type 2 Chebyshev filter are shown below
0 1 2 30
0.2
0.4
0.6
0.8
1
Mag
nitu
de
Type 2 Chebyshev Filter
N = 3N = 5N = 7
§4.4.3 Chebyshev Approximation The order N of the Type 2 Chebyshev is det
ermined from given ε, Ωs, and A using
)/1(cosh)/1(cosh
)/(cosh)/1(cosh
11
1
1
21
kkA
Nps
6059.2)/1(cosh
)/1(cosh1
11
k
kN
Example – Determine the lowest order of a Chebyshev lowpass filter with a 1-dB cutoff frequency at 1 kHz and a minimum attenuation of 40 dB at 5 kHz -
§4.4.4 Elliptic Approximation The square-magnitude response of an ellipti
c lowpass filter is given by
)/(1
1)( 22
2
pNa
RjH
where RN(Ω) is a rational function of order N satisfying RN(1/Ω)=1/RN(Ω), with the roots of its numerator lying in the interval 0< Ω<1 and the roots of its denominator lying in the interval 1< Ω<∞
§4.4.4 Elliptic Approximation
For given Ωp , Ωs ,ε, and A, the filter order can be estimated using
)/1(log)/4(log2
10
110
k
N
21' kk
)'1(2'1
0 kk
130
90
500 )(150)(15)(2
where
§4.4.4 Elliptic Approximation
Example - Determine the lowest order of a elliptic lowpass filter with a 1-dB cutoff frequency at 1 kHz and a minimum attenuation of 40 dB at 5 kHzNote: k = 0.2 and 1/k=196.5134
Substituting these values we get
k’ = 0.979796, ρ0 =0 00255135,ρ= 0 0025513525
and hence N = 2.23308 Choose N = 3
§4.4.4 Elliptic Approximation Typical magnitude response plots with Ωp=1 a
re shown below
0 1 2 30
0.2
0.4
0.6
0.8
1
Mag
nitu
de
Elliptic Filter
N = 3N = 4
§4.4.6 Analog Lowpass Filter Design Example – Design an elliptic lowpass filter of lo
west order with a 1-dB cutoff frequency at 1 kHz and a minimum attenuation of 40 dB at 5 kHz
Code fragments used[N, Wn] = ellipord(Wp, Ws, Rp, Rs, ‘s’);[b, a] = ellip(N, Rp, Rs, Wn, ‘s’);with Wp = 2*pi*1000;
Ws = 2*pi*5000; Rp = 1; Rs = 40;
§4.4.6 Analog Lowpass Filter Design Gain plot
§4.5 Design of Analog Highpass Bandpass and Bandstop Filters
Steps involved in the design process: Step 1 – Develop of specifications of a prototype analog lowpass filter HLP(s) from specifications of desired analog filter HD(s)using a frequency transformation
Step 2 – Design the prototype analog lowpass filter
Step 3 – Determine the transfer function HD(s) of desired analog filter by applying the inverse frequency transformation to HLP (s)
§4.5 Design of Analog Highpass Bandpass and Bandstop Filters
Let s denote the Laplace transform variable of prototype analog lowpass filter HLP(s) and ŝ denote the Laplace transform variable of desired analog filter HD(ŝ)
The mapping from s-domain to ŝ-domain is given by the invertible transform s=F(ŝ)
Then
)(ˆ
)ˆ(
1)ˆ()(
)()ˆ(
sFsDLP
sFsLPD
sHsH
sHsH
HLP (s) and is the passband edgep
§4.5.1 Analog Highpass Filter Design
Spectral Transformation:
ss pp
ˆ
ˆ
ˆpp
frequency of HHP (ŝ) On the imaginary axis the transformation is
where Ωp is the passband edge frequency of
§4.5.1 Analog Highpass Filter Design
ˆ
ˆpp
§4.5.1 Analog Highpass Filter Design
Example - Design an analog Butterworth highpass filter with the specifications:
Fp=4 kHz, Fs=1 kHz,αp=0.1 dB,αs=40 dB Choose Ωp=1 Then
41000
4000ˆ
ˆ
ˆ2
ˆ2
s
p
s
ps
F
F
F
F
Analog lowpass filter specifications: Ωp=1, Ωs=4, αp=0.1 dB αs=40 dB
§4.5.1 Analog Highpass Filter Design
Code fragments used[N, Wn] = buttord(1, 4, 0.1, 40, ‘s’);[B, A] = butter(N, Wn, ‘s’);[num, den] = lp2hp(B, A, 2*pi*4000);
Gain plots
§4.5.2 Analog Bandpass Filter Design
Spectral Transformation
w
pB
ˆ
ˆˆ 220
upper passband edge frequencies of desired bandpass filter HBP (ŝ)
HLP (s), and and are the lower and2ˆ
p1ˆ
p
where Ωp is the passband edge frequency of
§4.5.2 Analog Bandpass Filter Design
On the imaginary axis the transformation is
w
pB
ˆ
ˆˆ 220
where is the width of passband and is the passband center frequency of the bandpass filter
passband edge frequency ±Ωp is mapped into and , lower and upper passband edge frequencies
12ˆˆ
ppwB
0
1ˆ
p 2ˆ
p
§4.5.2 Analog Bandpass Filter Design
w
pB
ˆ
ˆˆ 220
§4.5.2 Analog Bandpass Filter Design
If bandedge frequencies do not satisfy the above condition, then one of the frequencies needs to be changed to a new value so that the condition is satisfied
2121ˆˆˆˆ
sspp
Stopband edge frequency ±Ωs is mapped into and , lower and upper stopband edge frequencies
Also,2
ˆs1
ˆs
§4.5.2 Analog Bandpass Filter Design
increase any one of the stopband edges or decrease any one of the passband edges as shown below
Case 1:
to make we can either2121ˆˆˆˆ
sspp 2121
ˆˆˆˆsspp
§4.5.2 Analog Bandpass Filter Design
(1) Decrease to
larger passband and shorter leftmost transition band
1ˆ
p 221ˆ/ˆˆ
pss
(2) Increase to
No change in passband and shorter leftmost transition band
1ˆ
s 221ˆ/ˆˆ
spp
§4.5.2 Analog Bandpass Filter Design
Note: The condition
can also be satisfied by decreasing
which is not acceptable as the passband is reduced from the desired value
212120
ˆˆˆˆsspp
2ˆ
p
Alternately, the condition can be satisfied by increasing which is not acceptable as the upper stopband is reduced from the desired value
2ˆ
s
§4.5.2 Analog Bandpass Filter Design
decrease any one of the stopband edges or increase any one of the passband edges as shown below
Case 2:to make we can either
2121ˆˆˆˆ
sspp
2121ˆˆˆˆ
sspp
§4.5.2 Analog Bandpass Filter Design
(1) Increase to
larger passband and shorter rightmost transition band
2ˆ
p 121ˆ/ˆˆ
pss
(2) Decrease to
No change in passband and shorter rightmost transition band
121ˆ/ˆˆ
spp 2ˆ
s
§4.5.2 Analog Bandpass Filter Design
Note: The condition
can also be satisfied by increasing
which is not acceptable as the passband is reduced from the desired value
1ˆ
p2121
20
ˆˆˆˆsspp
Alternately, the condition can be satisfied by decreasing which is not acceptable as the lower stopband is reduced from the desired value
1ˆ
s
§4.5.2 Analog Bandpass Filter Design
Example – Design an analog elliptic bandpa
ss filter with the specifications:
Now and621 1028ˆˆ pp FF 6
21 1024ˆˆ ss FF
Since we choose2121ˆˆˆˆ
sspp FFFF
571428.3ˆ/ˆˆˆ2211 pssp FFFF
dB22,dB1,kHz8ˆ,kHz3ˆ,kHz7ˆ,kHz4ˆ
2
121
sps
spp
F
FFF
§4.5.2 Analog Bandpass Filter Design
We choose Ωp=1 Hence
4.137/25
9-24s
)(
Analog lowpass filter specifications: Ωp=1, Ω
s=1.4, αp=1dB,αs=22dB
§4.5.2 Analog Bandpass Filter Design
Code fragments used
[N, Wn] = ellipord(1, 1.4, 1, 22, ‘s’);
[B, A] = ellip(N, 1, 22, Wn, ‘s’);
[num, den]= lp2bp(B, A, 2*pi*4.8989795, 2*pi*25/7); Gain plot
§4.5.3 Analog Bandstop Filter Design
Spectral Transformation
where Ωs is the stopband edge frequency of HLP(s), and and are the lower and upper stopband edge frequencies of the desired bandstop filter HBS(ŝ)
s1 s2
20
2s1s2
s ˆˆ
)ˆˆ(ˆ
s
ss
§4.5.3 Analog Bandstop Filter Design
On the imaginary axis the transformation
220
s ˆˆ
ˆ
wB
where is the widrth of stopband and is the stopband center frequency of the bandstop filter
Stopband edge frequency is mapped into and , lower and upper stopband edge frequencies
s1s2ˆˆ wB
s1 s2s
0
§4.5.3 Analog Bandstop Filter Design
Passband edge frequency ±Ωp is mapped into and , lower and upper passband edge frequencies
p1 p2
§4.5.3 Analog Bandstop Filter Design
If bandedge frequencies do not satisfy the above condition, then one of the frequencies needs to be changed to a new value so that the condition is satisfied
212120
ˆˆˆˆˆsspp
Also,
§4.5.3 Analog Bandstop Filter Design
Case 1: To make we can either incr
ease any one of the stopband edges or decrease any one of the passband edges as shown below
2121ˆˆˆˆ
sspp
2121ˆˆˆˆ
sspp
§4.5.3 Analog Bandstop Filter Design
(1) Decrease to
larger high-frequency passband and shorter rightmost transition band
221ˆ/ˆˆ
pss 2ˆ
p
(2) Increase to
No change in passband and shorter rightmost transition band
221ˆ/ˆˆ
spp 2ˆ
s
§4.5.3 Analog Bandstop Filter Design
Note: The condition
can also be satisfied by decreasing
which is not acceptable as the low – frequency passband is reduced from the desired value
Alternately, the condition can be satisfied by increasing which is not acceptable as the stopband is reduced from the desired value
212120
ˆˆˆˆˆsspp
1ˆ
p
1ˆ
s
§4.5.3 Analog Bandstop Filter Design
Case 1: To make we can either decre
ase any one of the stopband edges or increase any one of the passband edges as shown below
2121ˆˆˆˆ
sspp
2121ˆˆˆˆ
sspp
§4.5.3 Analog Bandstop Filter Design
(1) Increase to larger passband and shorter leftmost transition band
121ˆ/ˆˆ
pss 1ˆ
p
(2) Decrease to
No change in passband and shorter leftmost transition band
121ˆ/ˆˆ
spp 1ˆ
s
§4.5.3 Analog Bandstop Filter Design
Note: The condition
can also be satisfied by increasing
which is not acceptable as the high – frequency passband is decreasd from the desired value
Alternately, the condition can be satisfied by decreasing which is not acceptable as the stopband is decreased
212120
ˆˆˆˆˆsspp
2ˆ
p
2ˆ
s