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Asymmetric Full-Duplex with Contiguous Downlink Carrier Aggregation Dani Korpi, Lauri Anttila, and Mikko Valkama Department of Electronics and Communications Engineering, Tampere University of Technology, Finland e-mail: [email protected]ﬁ Abstract—In this paper, a contiguous carrier aggregation scheme for the downlink transmissions in an inband full-duplex cellular network is analyzed. In particular, we consider a scenario where the base station transmits over a wider bandwidth than the mobiles, while both parties are still using the same center frequency. As a result, the mobiles must cancel their own self- interference over a wider bandwidth, when compared to a situation where the uplink and downlink frequency bands are symmetric. Furthermore, due to the inherent RF impairments in the mobile devices, nonlinear modeling of the self-interference is required in the digital domain to fully cancel it over the whole reception bandwidth. The feasibility of the proposed scheme is demonstrated with real-life RF measurements, using two different bandwidths. In both of these cases, it is shown that the SI can be attenuated below the receiver noise ﬂoor over the whole reception bandwidth. I. I NTRODUCTION Several recent works have shown that the concept of simulta- neous transmission and reception on the same center frequency is practically feasible [1]–[4]. The main challenge in imple- menting such inband full-duplex radios is canceling the own transmission from the overall received signal. This so-called self- interference (SI) can be as much as 100–120 dB more powerful than the signal of interest, and hence advanced techniques are needed to solve this issue. As already mentioned, there are various demonstrator imple- mentations, which have been able to tackle the problem of SI, rendering the inband full-duplex radio a feasible concept. For instance, in [1], a full-duplex relay prototype is presented and it is shown to cancel the SI almost perfectly. In fact, the whole principle of inband full-duplex communications is very well suit- able for relay applications, since there the trafﬁc is symmetric in terms of transmission and reception. This means that the inherent symmetry of an inband full-duplex transceiver is well utilized. Also the more generic implementations of inband full-duplex devices assume perfectly symmetric transmission and reception, at least in terms of the bandwidth [3]–[5]. However, when considering a more practical deployment of a full-duplex device, assuming similar data rates for transmission and reception is typically not realistic. Especially, in a cellular network the uplink (UL) data rate requirements are typically much lower than the downlink (DL) data rates [6]. Hence, in order to fully utilize inband full-duplex radios in a cellular network, a method for asymmetric data transfer is needed. In this work, this problem is addressed from the user equipment (UE) point of view, which means that the UL transmission data rate is assumed to be lower than the DL reception data rate. This can be achieved by having a wider bandwidth for the DL signal, and correspondingly a narrower bandwidth for the UL signal. For a high-quality full-duplex transceiver, this would be a trivial change since the SI cancellation procedure would only have to be performed over the transmit signal band, while the frequency bands outside that could be readily used for reception. However, since the focus of this work is on the UE side, the low quality of the RF components must be taken into consideration. In particular, the transmitter power ampliﬁer (PA) will distort the transmit signal, resulting in a signiﬁcant amount of spectral regrowth also outside the actual UL transmission band [5], [7]. This means that part of the SI is leaking to the adjacent frequency bands, calling for cancellation outside the intended transmission bandwidth. In this work, this type of a contiguous DL carrier aggregation scheme is laid out and analyzed, especially in terms of the required SI cancellation processing. Because of the aforemen- tioned nonidealities in the transmitter, simple linear processing will not provide the required levels of cancellation, and thus nonlinear processing is needed to extend the reception band- width. The proposed scheme is then evaluated with real-life RF measurements, which incorporate also a state-of-the-art RF canceller. This means that the obtained results reﬂect the true overall performance of a mobile scale inband full-duplex device with asymmetric transmission and reception bandwidths. The rest of this paper is organized as follows. In Section II, the basic system model is presented, alongside with the full-duplex device architecture. After this, in Section III, the nonlinear mod- eling of the SI waveform is discussed, together with a detailed description of the nonlinear digital cancellation procedure. The RF measurement results are then shown in Section IV. Finally, the conclusions are drawn in Section V. II. SYSTEM MODEL AND DEVICE ARCHITECTURE This work focuses on analyzing a UE, which is assumed to engage in full-duplex communication with a base station (BS). Hence, the scenario is as illustrated in Fig. 1, where the situation is shown for several UEs. In the forthcoming analysis, however, only a single UE is assumed for ease of presentation. As already discussed, in a mobile cell the trafﬁc between the UE and the BS is inherently asymmetric, since more data is typically transferred in the DL than in the UL. This is a fun- damental challenge for an inband full-duplex system, which is usually assumed to transfer data, or at least deploy spectrum, in a symmetric way. In this article, a solution to this issue is proposed in the form of contiguous carrier aggregation. In particular, we 978-1-5090-1749-2/16/$31.00 c 2016 IEEE

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Asymmetric Full-Duplex with ContiguousDownlink Carrier Aggregation

Dani Korpi, Lauri Anttila, and Mikko Valkama

Department of Electronics and Communications Engineering, Tampere University of Technology, Finlande-mail: [email protected]

AbstractIn this paper, a contiguous carrier aggregationscheme for the downlink transmissions in an inband full-duplexcellular network is analyzed. In particular, we consider a scenariowhere the base station transmits over a wider bandwidth thanthe mobiles, while both parties are still using the same centerfrequency. As a result, the mobiles must cancel their own self-interference over a wider bandwidth, when compared to asituation where the uplink and downlink frequency bands aresymmetric. Furthermore, due to the inherent RF impairments inthe mobile devices, nonlinear modeling of the self-interference isrequired in the digital domain to fully cancel it over the wholereception bandwidth. The feasibility of the proposed scheme isdemonstrated with real-life RF measurements, using two differentbandwidths. In both of these cases, it is shown that the SI can beattenuated below the receiver noise floor over the whole receptionbandwidth.

I. INTRODUCTION

Several recent works have shown that the concept of simulta-neous transmission and reception on the same center frequencyis practically feasible [1][4]. The main challenge in imple-menting such inband full-duplex radios is canceling the owntransmission from the overall received signal. This so-called self-interference (SI) can be as much as 100120 dB more powerfulthan the signal of interest, and hence advanced techniques areneeded to solve this issue.

As already mentioned, there are various demonstrator imple-mentations, which have been able to tackle the problem of SI,rendering the inband full-duplex radio a feasible concept. Forinstance, in [1], a full-duplex relay prototype is presented andit is shown to cancel the SI almost perfectly. In fact, the wholeprinciple of inband full-duplex communications is very well suit-able for relay applications, since there the traffic is symmetric interms of transmission and reception. This means that the inherentsymmetry of an inband full-duplex transceiver is well utilized.Also the more generic implementations of inband full-duplexdevices assume perfectly symmetric transmission and reception,at least in terms of the bandwidth [3][5].

However, when considering a more practical deployment of afull-duplex device, assuming similar data rates for transmissionand reception is typically not realistic. Especially, in a cellularnetwork the uplink (UL) data rate requirements are typicallymuch lower than the downlink (DL) data rates [6]. Hence, inorder to fully utilize inband full-duplex radios in a cellularnetwork, a method for asymmetric data transfer is needed. Inthis work, this problem is addressed from the user equipment(UE) point of view, which means that the UL transmission datarate is assumed to be lower than the DL reception data rate. This

can be achieved by having a wider bandwidth for the DL signal,and correspondingly a narrower bandwidth for the UL signal.

For a high-quality full-duplex transceiver, this would be atrivial change since the SI cancellation procedure would onlyhave to be performed over the transmit signal band, while thefrequency bands outside that could be readily used for reception.However, since the focus of this work is on the UE side, the lowquality of the RF components must be taken into consideration.In particular, the transmitter power amplifier (PA) will distortthe transmit signal, resulting in a significant amount of spectralregrowth also outside the actual UL transmission band [5], [7].This means that part of the SI is leaking to the adjacent frequencybands, calling for cancellation outside the intended transmissionbandwidth.

In this work, this type of a contiguous DL carrier aggregationscheme is laid out and analyzed, especially in terms of therequired SI cancellation processing. Because of the aforemen-tioned nonidealities in the transmitter, simple linear processingwill not provide the required levels of cancellation, and thusnonlinear processing is needed to extend the reception band-width. The proposed scheme is then evaluated with real-lifeRF measurements, which incorporate also a state-of-the-art RFcanceller. This means that the obtained results reflect the trueoverall performance of a mobile scale inband full-duplex devicewith asymmetric transmission and reception bandwidths.

The rest of this paper is organized as follows. In Section II, thebasic system model is presented, alongside with the full-duplexdevice architecture. After this, in Section III, the nonlinear mod-eling of the SI waveform is discussed, together with a detaileddescription of the nonlinear digital cancellation procedure. TheRF measurement results are then shown in Section IV. Finally,the conclusions are drawn in Section V.

II. SYSTEM MODEL AND DEVICE ARCHITECTUREThis work focuses on analyzing a UE, which is assumed to

engage in full-duplex communication with a base station (BS).Hence, the scenario is as illustrated in Fig. 1, where the situationis shown for several UEs. In the forthcoming analysis, however,only a single UE is assumed for ease of presentation.

As already discussed, in a mobile cell the traffic between theUE and the BS is inherently asymmetric, since more data istypically transferred in the DL than in the UL. This is a fun-damental challenge for an inband full-duplex system, which isusually assumed to transfer data, or at least deploy spectrum, in asymmetric way. In this article, a solution to this issue is proposedin the form of contiguous carrier aggregation. In particular, we

978-1-5090-1749-2/16/$31.00 c2016 IEEE

FD-BS FD-UEFD-UE

FD-UE

Fig. 1. An illustration of a full-duplex BS communicating with full-duplexUEs in an asymmetric manner.

DL

f

DL DL

UL

3rd order

5th order

7th order

Fig. 2. A frequency domain illustration of the signal received by the UEunder contiguous full-duplex carrier aggregation. The figure includes also thetransmitter-induced nonlinear distortion up to the 7th order.

assume a system where the primary UL and DL carriers arefully overlapping, but such that there are also additional carriersallocated for DL data transfer, which are directly adjacent to theprimary or common UL/DL carrier. Figure 2 illustrates this inthe frequency domain, where the DL bandwidth is three timeshigher than the UL bandwidth.

In an ideal situation, it would be sufficient for the UE tocancel SI only on the common UL/DL carrier, since there are noongoing transmissions in the DL-only carriers. This would resultin an easier SI cancellation task, since the accurate regenerationof the SI signal is more challenging over wider bandwidths.However, as Fig. 2 illustrates, the ongoing transmission over theUL/DL carrier produces SI also to the DL-only carriers due tothe strong nonlinear distortion generated by the transmitter PA.If the SI cancellation is done only on the common UL/DL carrier,the SINR of the DL-only carriers is heavily compromised due tothis spectral regrowth.

Hence, advanced digital SI cancellation techniques are neededto render the proposed contiguous carrier aggregation solutionfeasible. Simple linear digital cancellation procedures are of nouse, since the linear component of the SI is not producing anyinterference at the DL-only carriers. Thus, in order to cancelthe interference, a nonlinear model for the SI is needed, whichis then used to regenerate the signal that is overlapping withthe adjacent carriers. This can be done by using appropriatenonlinear behavioral models for the overall SI, and not limitingthe cancellation processing and the underlying nonlinear basisfunctions to the purely inband part of the SI, which is the case

in the earlier works, e.g., in [5], [7]. Thus, the procedure isotherwise identical to the traditional inband SI cancellation, onlythe bandwidth is wider.

Since this work involves SI cancellation in the mobile device,also the considered architecture for the full-duplex transceivermust be feasible for such a use-case. For this reason, we considera full-duplex device, whose structure is as shown in Fig. 3. Ascan be observed, the full-duplex transceiver is sharing a singleantenna between the transmitter and the receiver, which is typi-cally a necessary feature for a mobile scale device. In addition, athree-tap wideband RF canceller is used to prevent the saturationof the receiver chain, in particular the low-noise amplifier. Thesmall number of taps is made possible by utilizing the advancedcanceller structure elaborated in more details in [8] and [9], sinceminimizing the complexity of the RF canceller is also a crucialaspect for a mobile scale full-duplex transceiver. After the RFcanceller, the remaining SI is cancelled in the digital domainby the developed nonlinear digital canceller. Ideally, after this,the residual SI is well below the receiver noise floor at all DLcarriers, and the full-duplex operation mode does not degradethe final signal-to-interference-plus-noise ratio (SINR) of thereceived signal of interest.

III. SELF-INTERFERENCE SIGNAL MODEL ANDNONLINEAR DIGITAL CANCELLATION

In a mobile scale device, most of the nonlinear distortion inthe transmit signal is typically produced by the PA [7], [10].Thus, in order to cancel the nonlinear SI, a model for the PAmust be incorporated into the digital SI signal model. A typicalchoice is to utilize the so-called parallel Hammerstein (PH)signal model for the SI observed in the digital domain [7]. Sucha signal model has been shown to be accurate in modeling anactual low-cost PA under realistic operating conditions [1], [2].Using the PH model, the observed SI in the digital domain canbe expressed as follows:

y(n) =

Pp=1p odd

M2m=M1

hp(m)p (x(nm)) + z(n), (1)

where P is the nonlinearity order of the model,M1 is the amountof pre-cursor memory, M2 is the amount of post-cursor memory,p () is the pth-order nonlinear basis function, hp(m) containsthe memory coefficients for the pth-order basis function, x(n)is the original digital transmit waveform, and z(n) representsthe noise and possible model mismatch. Since the nonlinear can-celler is modeling only the transmitter PA, which is producingthe majority of the nonlinear distortion, it is sufficient to consideronly the odd order basis functions.

In order to use the above signal model for SI cancellation, thebasis functions must first be generated such that the nonlinearSI also at the neighboring carriers is included, after which thecorresponding memory coefficients must be estimated. Then,it is a straight-forward matter to perform the nonlinear digitalcancellation. In the subsections below, a brief description ofthese issues is provided.

CO

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Wideband RF

cancellation circuit

CO

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LNA IQ Mixer LPF VGA ADC

LPFIQ MixerVGAPA DAC

Digital

cancellation

To

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tecto

r

Transmitter chain

Receiver chain

+

LO

Tra

nsm

it d

ata

Nonlinear

DSPCONTROL

BPF

Fig. 3. A block diagram of the considered mobile scale inband full-duplex transceiver.

A. Generating the Nonlinear Basis FunctionsIn principle, the pth-order basis function is of the form

p (x(n)) = |x(n)|p1 x(n) (2)

when the nonlinear transformation is applied on the nth transmitdata sample. However, this type of a presentation does not takeexplicitly into account the particular sampling frequency of thetransmit signal x(n), which means that there might be somealiasing in the higher-order basis functions. In particular, thepth-order nonlinear transformation has a bandwidth p times ashigh as the original signal, which means that if the samplingfrequency is not sufficiently high in the basis function generationstage, some of the nonlinear transformations will have frequencycontent higher than the Nyquist frequency. Especially, if thereception bandwidth is wide, this aliasing might even overlapwith the signal of interest after the cancellation procedure. Thistype of a situation can be illustrated with the help of Fig. 2,where nonlinearities up to the 7th order are shown. In this case,if the Nyquist frequency in the basis function processing stageis significantly lower than the highest frequency components ofthe 7th-order distortion, some of the nonlinearities will alias onto the DL signal band, resulting in an inaccurate cancellationsignal.

As an obvious solution to this issue, the basis functionsshould be applied to an oversampled transmit signal, after whichthey can be low-pass filtered and decimated to the desiredsampling frequency for SI cancellation. This ensures that nosignificant frequency content is aliasing on to the signal bandof interest when generating the nonlinearly transformed signalwith (2). This is an especially crucial aspect in the consideredsystem, where the signal band of interest is much wider thanthe bandwidth of the actual transmit signal. In such a case, thetransmit signal must be significantly oversampled to ensure thatthe nonlinear transformations within the final cancellation signalare not aliasing on to the reception bandwidth. Denoting thebandwidth of the transmit signal by Bt and the bandwidth ofinterest by Br, the required sampling rate can be expressed as

Fs =PBt +Br

2. (3)

Using this sampling frequency for generating the basis functionsensures that no aliasing occurs within the reception bandwidth.

From (3) it can be observed that the wider the bandwidth ofthe transmitted or received signal is, the more oversampling isrequired. Also increasing the nonlinearity order of the cancellerresults in a higher sampling rate requirement. For instance,setting P = 11 and having an IQ bandwidth of Bt = Br =20 MHz for both the transmit and receive signals, we getFs = 120 MHz, while having a received signal bandwidth ofBr = 60 MHz results in Fs = 140 MHz. However, since thehighest order nonlinearities tend to be relatively weak outsidethe signal bandwidth, in practice a smaller sampling rate mightalso suffice. This is elaborated further in Section IV.

B. LMS-Based Parameter LearningIn order to apply the above signal model for SI cancellation,

the coefficients in hp(m) must be estimated. To ensure a compu-tationally relaxed learning procedure, the widely used least meansquare (LMS) algorithm is utilized for this task. The procedureis similar to the one used in [2], where the LMS-based nonlinearcanceller was successfully used for regular SI cancellation ina more elementary full-duplex device with symmetric UL andDL bandwidths. Now, the estimation and cancellation procedureitself is identical, meaning that the generated basis functions arefirst orthogonalized to ensure efficient learning, after which theLMS algorithm is applied to the orthogonalized basis functions.

Denoting the orthogonalization matrix by S, which is ob-tained in a similar manner as shown in [2], the orthogonalizedbasis functions are defined as

(n) = S(n), (4)

where (n) = [1(x(n)) 3(x(n)) P (x(n))]T , i.e., itcontains all the basis function samples corresponding to the timeindex n. These orthogonalized basis functions are then used inthe actual cancellation process, as follows:

yc(n) = y(n)P

p=1p odd

M2m=M1

hp,ort(m)p (x(nm))

= y(n) hHortu(n), (5)

where p (x(n)) represents the orthogonalized pth-order nonlin-ear basis function from (4), and hp,ort(m) contains the corre-sponding estimates of the SI channel coefficients. The vectors

PXIe-5645RPXIe-5645R

RF cancellerRF canceller

Circulator andCirculator andantennaantenna

Fig. 4. The RF measurement setup used for determining the integratedperformance of the contiguous downlink carrier aggregation solution.

are defined as

hort = [ h1,ort (M1) h3,ort (M1) hP,ort (M1) hP,ort (M2) ]T

u(n) = [ (n+M1)T (n+M11)T (nM2)T ]T .

The learning and adaptation of the coefficients in hort is donewith the LMS algorithm, whose update rule can be written asfollows:

hort hort + yc(n)u(n), (6)

where is a diagonal matrix containing the individual stepsizesfor each orthogonalized nonlinear basis function. Performing thecancellation process in (5) with the latest estimate of the coeffi-cient vector and then updating the coefficients as shown in (6),and repeating this in every iteration, results in a computationallylightweight and highly adaptive SI cancellation procedure.

IV. RF MEASUREMENT RESULTSIn order to evaluate the proposed contiguous carrier aggrega-

tion based asymmetric full-duplex solution, it is evaluated withreal-life RF measurements. The measurement setup is shown inFig. 4, while all the important parameters are listed in Table I.The measurements are carried out using a National InstrumentsPXIe-5645R vector signal transceiver, which is used both as atransmitter and a receiver. The used transmit signal is either a20 MHz or a 40 MHz LTE waveform, centered at 2.46 GHz,and it is fed through a low-cost Texas Instruments CC2595PA, which amplifies the signal by approximately 23 dB. Thisparticular PA is a commercial chip intended to be used in low-cost battery-powered devices, and thereby it produces significantlevels of nonlinear distortion. Utilizing such a low-quality PA en-sures that the measurement results are representative of a mobilescale device. The PA output signal is then divided between theRF canceller and the circulator, the latter of which is connectedto the shared transmit/receive antenna. The SI is leaking to thereceiver both via the circulator and from the antenna reflections.The overall transmit power in the measurements is in the orderof 68 dBm, which is the highest reachable power level with theutilized hardware. In principle, the proposed algorithm worksalso with higher transmit powers, and demonstrating this is apotential future work item.

The total received signal is then routed from the circulatorback to the RF canceller, which performs the analog cancellation

TABLE ITHE ESSENTIAL RF MEASUREMENT PARAMETERS.

Parameter ValueTransmit signal bandwidth (UL) 20 MHz / 40 MHz

Reception bandwidth (DL) 60 MHz / 120 MHzCenter frequency 2.46 GHz

PA gain 23 dBTransmit power 68 dBm

Number of taps in the RF canceller 3Highest nonlinearity order (P ) 11

Number of pre-cursor taps (M1) 15Number of post-cursor taps (M2) 15

utilizing the PA output signal as described in [8] and [9]. Finally,the processed signal is fed to the receiver (NI PXIe-5645R) andcaptured as digital I- and Q-samples, which are post-processedoffline to implement digital baseband cancellation. In all theresults, the highest nonlinearity order of the digital canceller(P ) is set to 11, and the numbers of pre-cursor (M1) and post-cursor taps (M2) are both set to 15. The step-sizes, containedin , are chosen experimentally to provide the best cancella-tion performance. In the forthcoming results, the LMS-basedalgorithm is first allowed to converge towards the steady-statecoefficient values, after which the cancellation performance ismeasured in steady-state. This ensures that the results show thetrue performance of the digital canceller.

In the measurements, two different bandwidth scenarios areused: (i) 20 MHz transmit signal with 60 MHz reception band-width, and (ii) 40 MHz transmit signal with 120 MHz receptionbandwidth. The measurement results from these two scenariosare illustrated in Figs. 5 and 6, respectively. In the figures, theresidual SI after the digital cancellers represents the true powerof the SI signal from which the noise power has already beenexcluded. This is done in practice by repeatedly transmitting thesame signal and averaging the residual SI over these repetitions.This will obviously reduce the noise while having no effect onthe transmit signal dependent SI components. Thanks to thisprocedure, it is possible to determine the residual SI power evenwhen it is below the receiver noise floor.

When investigating the results for the 20/60 MHz case inFig. 5, it can be observed that the nonlinear digital canceller isindeed capable of attenuating the SI below the receiver noisefloor over the whole reception bandwidth. At transmitter pass-band, the performance is slightly worse, as can be expected,but the power of the residual SI is still approximately at thelevel of the receiver noise floor. Another intuitive observationis that the linear digital canceller can attenuate the SI only at theown transmitter passband, since it is only capable of modelingthe linear coupling characteristics. Hence, it cannot attenuatethe SI outside the own transmission bandwidth, and its overallperformance is significantly worse than that of the developednonlinear digital canceller.

Figure 6 shows the corresponding results for the case wherethe transmit signal has a bandwidth of 40 MHz and the receptionis done over a bandwidth of 120 MHz. Again, the nonlineardigital canceller is capable of attenuating the residual SI to thelevel of the receiver noise floor, regardless of the wide reception

Frequency6iMHzAl5s l4s l3s l2s lfs s fs 2s 3s 4s 5s

PS

D6[d

Bm

NMH

z]

lf2s

lfss

l8s

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l4s

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TX6output6i6746dBmARX6input6ilf3756dBmAADC6input6il6f756dBmALinear6canceller6il76736dBmANonlinear6canceller6il86746dBmARX6noise6floor6il83726dBmA

Fig. 5. The signal spectra after the different cancellation stages, when thebandwidth of the transmit signal is 20 MHz and SI cancellation is done overa 60 MHz bandwidth.

bandwidth. Also in this case, the difference to the linear cancelleris over 10 dB at transmitter passband as well as at neighboringcarriers, demonstrating the necessity of nonlinear modeling in amobile scale device. Overall, these findings regarding the totalintegrated SI cancellation performance show that the consideredmethod for asymmetric UL and DL communication is indeedfeasible. As long as the nonlinearity of the PA is properlyconsidered in the digital canceller, both the linear and nonlinearSI can be attenuated over significantly wide bandwidths.

Regarding the oversampling of the transmit signal, the mea-surements indicate that under these conditions a sufficientlyhigh cancellation performance is achieved when the nonlinearbasis functions are generated with a sampling frequency that istwice the reception bandwidth. Hence, for the 60 MHz receptionbandwidth, a sampling frequency of 120 MHz is sufficient, whilethe 120 MHz reception bandwidth requires a sampling frequencyof 240 MHz for the basis function generation. When calculatingthe required sampling rates with (3), it can be deduced thatnow, in both of these cases, the 11th-order basis function ispartially aliasing on to the reception bandwidth. However, sincethe higher-order distortions typically tend to be weaker in power,this does not affect the final residual SI.

V. CONCLUSIONThis paper addressed digital self-interference cancellation

challenges in a downlink carrier aggregation based mobile full-duplex device where the uplink/downlink rates and bandwidthsare asymmetric. The scheme was considered especially in thecontext of a cellular network, where a full-duplex UE is commu-nicating with a full-duplex capable BS such that the downlinksignal bandwidth is wider than the uplink bandwidth. Due tothe inherent RF impairments in the UE, nonlinear modeling ofthe self-interference is required in the digital domain to fullycancel it over the whole reception bandwidth. The feasibilityof the proposed scheme was demonstrated with real-life RFmeasurements using two different bandwidths. In both of thesecases, the SI was attenuated below the noise floor over the wholereception bandwidth.

Frequency6.MHzAl5s l4s l3s l2s lfs s fs 2s 3s 4s 5s

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TX6output6.86dBmARX6input6.lf49f6dBmAADC6input6.l58936dBmALinear6canceller6.l699f6dBmANonlinear6canceller6.l8s956dBmARX6noise6floor6.l8s926dBmA

Fig. 6. The signal spectra after the different cancellation stages, when thebandwidth of the transmit signal is 40 MHz and SI cancellation is done overa 120 MHz bandwidth.

ACKNOWLEDGMENTThe research work leading to these results was funded by the

Academy of Finland (under the project #259915), and the LinzCenter of Mechatronics (LCM) in the framework of the AustrianCOMET-K2 programme. The research was also supported by theInternet of Things program of DIGILE, funded by Tekes.

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[3] M. Duarte, C. Dick, and A. Sabharwal, Experiment-driven characteri-zation of full-duplex wireless systems, IEEE Transactions on WirelessCommunications, vol. 11, no. 12, pp. 42964307, Dec. 2012.

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[5] D. Bharadia, E. McMilin, and S. Katti, Full duplex radios, in Proc.SIGCOMM13, Aug. 2013, pp. 375386.

[6] Nokia Solutions and Networks, TD-LTE frame configuration primer,Nov. 2013, white paper.

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[8] T. Huusari, Y.-S. Choi, P. Liikkanen, D. Korpi, S. Talwar, and M. Valkama,Wideband self-adaptive RF cancellation circuit for full-duplex radio:Operating principle and measurements, in Proc. IEEE 81st VehicularTechnology Conference (VTC Spring), May 2015.

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[10] E. Ahmed, A. M. Eltawil, and A. Sabharwal, Self-interference cancella-tion with nonlinear distortion suppression for full-duplex systems, inProc. 47th Asilomar Conference on Signals, Systems and Computers,Nov. 2013, pp. 11991203.