Paper Title (use style: paper title) - ER Publications€¦ · Web viewThe antenna radiation pattern for the 0 polarized rhombic antenna in both E-plane (𝜑 = 90 ) and H-plane (𝜑

International Journal of Enhanced Research in Science, Technology & EngineeringISSN: 2319-7463, Vol. 4 Issue 7, July-2015

Wideband Dual-Polarized Crossed-Rhombic Antennas for Future 5G Mobile Communications

Base StationOsama M. Haraz 1

1Electrical Engineering Department, Assiut University, 71515 Assiut, Egypt

ABSTRACT

The design and the results of planar wideband dual-polarized crossed-Rhombic antennas are presented for the future fifth generation (5G) mobile communication base station. The proposed antenna is designed using ANSYS HFSS and consists of two perpendicular Rhombic antennas, which are printed on a single substrate, for dual-polarized operation. The two arms of each Rhombic antenna are printed on the same side of the substrate to reduce the cross polarization. The simulated antenna impedance bandwidth is found to be 22 GHz (24-46 GHz) with a fractional bandwidth (FBW) of about 64.7%. It covers the future millimeter-wave 5G bands of 28 GHz and 38 GHz. The antenna exhibits a gain of 2.958 dBi and 3.471 dBi at 28 GHz and 38 GHz, respectively. Within the operating band, the excellent performances of low mutual coupling, low cross polarization, and stable gain are achieved. In order to improve the gain, a suspended metallic layer is added underneath the antenna. The proposed antenna is considered a good candidate for 5G base stations applications.

Keywords: The author shall provide up to 5-6 keywords (in alphabetical order) to help identify the major topics of the paper. For eg; Engineering, magnetization, systems, conferences.

1. INTRODUCTION

The race to find innovative solutions to enable the fifth generation (5G) era has recently begun worldwide. In early 2013, the European Commission (EC) contributed $77 million to develop 5G mobile technology by 2020. The three leading universities are working together to bring the project to completion, the University of Dresden (Germany), the Kings College London (UK), and the University of Surrey (UK). South Korea, on the other hand is quite aggressive to reach this technological achievement, investing $1.5 billion to be launched in 2020 and a pilot network to roll out in 2017. Once developed, the network will permit Gigabit (1 Gb/s) transmissions on compatible portable devices. The 5G network is expected to be fast enough to download an entire movie in one second, or about 1,000 times faster than existing fourth generation (4G) Long Term Evolution (LTE) networks.

Wireless communications are rapidly moving to millimeter-wave (mmWave) frequency bands due to the needs for greater data throughputs, while data centers and computers of the future are evolving to greater reliance on wireless interconnectivity. High frequencies can carry more data, but the drawback is that they generally can be blocked by buildings and lose intensity over longer distances.

The antennas for base stations should cover the future 5G frequency bands. On the other hand, dual-polarized antennas are adopted on a large scale in base stations to increase the channel capacity and reduce the space occupied [1]. Therefore, a wideband dual-polarized antenna covering the 28 GHz and 38 GHz frequency bands for future 5G base stations is necessary.

Recently, different types of antenna structures have been proposed for wideband base stations applications. For example, a bow-tie slot etched on a square patch excited by two different balance-to-unbalance feedings is proposed in [2]. A stacked patch antenna and a square patch antenna with L-probe feeding are introduced in [3], [4], respectively. In [5], researchers developed a patch antenna with two wideband feeding mechanisms, one is meandering probe (M-probe) while the other is pair of twin-L-probes. A 2x1 microstrip stacked patch array with slot-coupling feeding technique is presented in [6]. The author have proposed a dual-polarized dielectric-loaded monopole antenna [7]. A uniplanar polarization diversity monopole-like slot antenna has been reported in [8]. In [9], researchers have developed a dual-polarized planar circular slot antenna. A differentially-driven dual-polarized magneto-electric dipole antenna is presented

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in [10]. Many researchers have proposed dual-polarized crossed-dipole antennas [11]-[23]. Among all proposed antennas, crossed-dipole ones have drawn a great attention to researchers.

The developed crossed-dipole antennas with good impedance matching, high isolation, and low cross polarization are promising for base station applications. The only problem with those designs is the 3D coaxial balun, which increases the antenna complexity and cost. The 3D coaxial baluns are used to transform the unbalanced feeding line to the balanced dipole. One solution have been proposed in [13] to replace the coaxial Balun with a 3D microstrip balun. A planar crossed-dipole antenna printed on a single substrate is proposed in [12]. Furthermore, integrated microstrip Balun is proposed in order to reduce the complexity and cost of planar crossed-dipole antenna [21]. In this article, a dual-polarized crossed-Rhombic antennas with planar microstrip balun is applied to transform the unbalanced microstrip line to the balanced dipole.

In this article, a new planar wideband dual-polarized crossed-Rhombic antenna for future 5G base station applications, printed on a single substrate, is proposed. The two arms of each dipole are printed on the same side of the substrate to achieve symmetrical dipole for low cross polarization. An integrated microstrip Balun is introduced for the unbalance to balance conversion. Its measured overlapped impedance bandwidth of the two ports with return loss S11 < -10 dB is 22 GHz (24-46 GHz) with a fractional bandwidth (FBW) of about 64.7%. Within this frequency band, the mutual coupling is weak and the cross polarization is low. Compared to [21], the proposed antenna is printed on a single substrate for low cost. Compared to [12], the two arms of each dipole are printed on the same side of the substrate for symmetry structure. The proposed antenna features the full planar structure for low cost, the planar microstrip Balun, and the symmetrical dipole for low cross polarization.

2. DUAL-POLARIZED CROSSED-RHOMBIC ANTENNA

A. Design and Configuration

The geometrical configuration of the proposed dual-polarized crossed-Rhombic antenna with is shown in Figure 1. The front and back views of the proposed antenna are shown in Figure 1(a) and Figure 1(b), respectively. It is printed on a 0.068-mm-thick Rogers RT/Duroid 5880 high-frequency laminate with a relative permittivity of ε r = 2.2 and loss tangent (tanδ) = 0.0009. The losses are incorporated in simulation. All copper layers used in proposed antenna structure have thickness of ½ Oz or 17.5 um. Figure 1(c) shows the three-dimensional (3D) isometric view of the proposed antenna. The crossed-Rhombic antenna consists of two perpendicular rhombic antennas, which are printed on different sides of the substrate. The two arms of each rhombic antenna are printed on the same side of the substrate. The proposed antenna is fed with a 50-Ω microstrip feeding line, stepped impedance matching section. With this layout, the proposed dual-polarized antenna can be printed on a single substrate. The two microstrip baluns are integrated into the two dipoles for the full planar structure of the antenna. In addition, the microstrip balun is also beneficial for impedance matching as discussed in the next section. All optimized geometrical parameters for the proposed dual-polarized crossed-Rhombic antenna are summarized in Table 1.

B. Parametric Studies

The simulated reflection coefficient ¿ S11∨¿ at the input port and mutual coupling between the two ports ¿ S21∨¿ with different Lf1, Lf2 and Lf3 are shown in Figures 2-4, respectively. Due to the similar structure of the two dipoles, only the input impedance of port 1 is shown. Figure 2 shows that, as the length Lf1 increases from 0.2 mm to 1.0 mm, the antenna impedance matching will be improved. By further increasing the length Lf1 from 1.0 mm to 1.4 mm, the impedance matching becomes worse. This phenomenon indicates that the stub functions as a distributed capacitance. in addition, varying the length Lf1 has a great effect on the mutual coupling between the two input ports. The optimum length Lf1 is found to be 0.863 mm. Figure 3 shows that, as the length Lf2 increases, the curves moves down which mean that the impedance matching is improved. This phenomenon verifies that this section of microstrip line functions as an impedance transformer. Thus, excellent impedance matching can be achieved by optimizing the microstrip balun. It is worthy to mention here that the mutual coupling is not strongly affected by varying the length Lf2 compared to length Lf1. The optimum length Lf2 is found to be 0.873 mm. finally, Figure 4 presents the effect of varying the reflection coefficient ¿ S11∨¿ and mutual coupling ¿ S21∨¿ with the length Lf3. results show that the length Lf3 has weak effect on both ¿ S11∨¿ and ¿ S21∨¿.

C. Results and Discussions

The proposed antenna structure is simulated and optimized using ANSYS High Frequency Structure Simulator (HFSS) [24]. The results are further verified by simulating the proposed antenna structure in CST Microwave Studio [25].

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The S-parameters of the proposed antenna have been calculated. A comparison between the simulation results of reflection coefficient ¿ S11∨¿ and isolation ¿ S21∨¿ when port 1 is fed and port 2 is matched, obtained from HFSS and CST are given in Figure 5. Figure 6 shows the reflection coefficient ¿ S22∨¿ and isolation ¿ S12∨¿ when port 2 is fed and port 1 is matched. The calculated input impedance of the proposed antenna at port 1 (Z11) and port 2 (Z22) are presented in Figure 7. It can be noticed that the real part of input impedance is varying around 50Ω while the imaginary part is almost 0Ω at the two frequencies of interest, i.e. 28 GHz and 38 GHz.

Results show that the antenna impedance bandwidth of the 0° polarized Rhombic antenna with ¿ S11∨¿ < -10 dB is found to be 28.5 GHz (24–52.5 GHz) from HFSS while from CST the impedance bandwidth is found to be approximately 27.5 GHz (25.1–52.6 GHz). For 90° polarized rhombic antenna, the antenna impedance bandwidth is found to be 28.4 GHz (23–51.4 GHz) from HFSS while from CST is found to be 27.3 GHz (24.5–51.8 GHz). Within this frequency band of interest, the simulated mutual coupling or isolation ¿ S21∨≈∨S12∨¿ between the two ports is better than -33 dB. The antenna is expected to cover the 28/38 GHz bands for the future fifth generation (5G) base stations applications.

The antenna radiation characteristics have been calculated and summarized in Table 2. The gain of the 0° polarized rhombic antenna is found to be 2.958 dBi and 3.471 dBi at 28 GHz and 38 GHz, respectively. For 90° polarized rhombic antenna, the peak gain is found to be 2.838 dBi and 3.523 dBi at 28 GHz and 38 GHz, respectively. Furthermore, the two antennas have good radiation and total efficiencies. The antenna input impedance at 28 GHz at ports 1 and 2 are 53.37+j0.98Ω and 62.9+j1.98Ω, respectively. At 38 GHz, the input impedances at ports 1 and 2 are 63.41-j12.71Ω and 57.94-j15.82Ω, respectively. Results show that the proposed antenna has a good impedance matching at the two frequency band of interests.

The 3D radiation patterns for the 0° and 90° polarized rhombic antennas at 28 GHz and 38 GHz are presented in Figure 8 and Figure 9, respectively. Results show that the proposed antenna exhibits a dipole-like radiation pattern. The antenna radiation pattern for the 0° polarized rhombic antenna in both E-plane (𝜑 = 90°) and H-plane (𝜑 = 0°) remains similar at the two frequency bands of interest, i.e. 28 GHz and 38 GHz. The comparison of 2D radiation patterns obtained from HFSS and CST at 28 GHz, and 38 GHz in both E-plane and H-plane are shown in Figures 10 and 11, respectively. The radiation patterns of the 0° polarized rhombic antenna are almost like that of dipole antenna which consist of a letter eight shape in the E-plane and omnidirectional in the H-plane.

In case of the 90° polarized rhombic antenna, the radiation patterns in both E-plane (φ = 0°) and H-plane (φ = 90°) have been calculated and presented in Figure 12 and 13, respectively.

From the above results, it is noted that the peak gain obtained is only 3-3.5 dBi which is not enough for the future 5G base station applications at 28 GHz and 38 GHz. Therefore, we added a suspended metallic layer underneath the antenna to improve the gain.

For better understanding the proposed antenna operation mechanism, the simulated electric field distributions at 28 GHz and 38 GHz with port 1 and port 2 exciting are illustrated in Figure 14. Simulation results show that the electric field distributions of port 1 and port 2 are perpendicular to each other. This phenomenon results in the low mutual coupling and better isolation between the two ports.

3. GAIN IMPROVEMENT WITH SUSPENDED METALLIC LAYER

A. Design and Configuration

Front and side views of proposed antenna with suspended metallic layer are shown in Figure 15. It consists of the base antenna as explained in Section 2 and a suspended metallic layer is added underneath it. A ROHACELL® foam layer of permittivity ε r = 1.05 is sandwiched between base antenna and metallic layer as a support. The suspended metallic layer has a thickness T, width, length Wg (= Lg), and the height H from the base antenna. The effect of varying all these parameters are studied and optimized in the next section.

B. Parametric Studies

The variation of reflection coefficient ¿ S11∨¿ with different suspended metallic layer height (H = 1 mm, 5, and 9 mm) with a constant thickness of T = 0.2 mm and size Wg = Lg = 35 mm is shown in Figure 16. Table 3 summarized

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the influence of suspended metallic layer dimensions on antenna radiation characteristics such as peak gain and side lobe level (SLL) in both E- and H-planes. It is noted that even though the maximum gain is achieved for a suspended metallic layer height of H = 1 mm as compared to H = 5 mm or 9 mm, the impedance matching is worst as investigated in Table 3. Results show also that changing the suspended metallic layer size from Wg = 20 mm to Wg = 40 mm does not strongly affect the impedance matching of the antenna. It affect only the antenna radiation characteristics. Increasing the size of suspended metallic layer will slightly increase the peak gain and strongly enhance the SLL.

C. Results and Discussions

Figure 17 shows the CST simulated S-parameters curves versus frequency of the rhombic antenna with suspended metallic layer. The achieved impedance bandwidth covers both 28 GHz and 38 GHz frequency bands of interests. The isolation between the two ports is better than -35 dB and -40 dB at 28 GHz and 38 GHz, respectively.

Figure 18 and Figure 19 present the simulated radiation patterns of both 0° and 90° polarized rhombic antenna with suspended metallic layer at 28 GHz, and 38 GHz, respectively. The 2D co- and cross polarization radiation pattern in both E-, and H-planes at 28 GHz and 38 GHz for both 0° and 90° polarized rhombic antenna with suspended metallic layer are shown in Figure 20 and Figure 21, respectively.

Finally, the simulated electric field distributions at 28 and 38 GHz when exciting the two ports have been calculated and presented in Figure 22

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Table 1: Optimized Geometrical Parameters for the Proposed Dual-Polarized Crossed-Rhombic Antenna

Parameter W L W1 L1 W2 L2 RValue (mm) 8.5 8.5 1.566 1.392 1.988 1.842 0.854

Parameter S Wf1 Lf1 Wf2 Lf2 Wf3 Lf3Value (mm) 0.224 0.021 0.863 0.021 0.873 0.063 0.27

(a) Front view (b) Back view

(c) 3D isometric view

Figure 1. Proposed dual-polarized crossed-Rhombic antenna geometry.

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20 25 30 35 40 45 50-50

-40

-30

-20

-10

00

Frequency, GHz

|S11

|, dB

Lf1

= 0.2 mm

Lf1

= 0.6 mm

Lf1 = 1.0 mm

Lf1

= 1.4 mm

20 25 30 35 40 45 50-80

-70

-60

-50

-40

-30

-20

-10

Frequency, GHz

|S21

|, dB

Lf1

= 0.2 mm

Lf1

= 0.6 mm

Lf1

= 1.0 mm

Lf1

= 1.4 mm

(a) Reflection coefficient ¿ S11∨¿ (b) Mutual coupling ¿ S21∨¿

Figure 2. Variation of S-parameters with a balun dimension Lf1.

20 25 30 35 40 45 50-50

-40

-30

-20

-10

0

Frequency, GHz

|S11

|, dB

Lf2

= 0.2 mm

Lf2

= 0.6 mm

Lf2

= 1.0 mm

Lf2

= 1.4 mm

20 25 30 35 40 45 50

-45

-40

-35

-30

-25

Frequency, GHz

|S21

|, dB

Lf2

= 0.2 mm

Lf2

= 0.6 mm

Lf2

= 1.0 mm

Lf2

= 1.4 mm



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20 25 30 35 40 45 50-50

-40

-30

-20

-10

0

Frequency, GHz

|S11

|, dB

Lf3

= 0.1 mm

Lf3

= 0.3 mm

Lf3

= 0.5 mm

Lf3

= 0.7 mm

20 25 30 35 40 45 50

-45

-40

-35

-30

-25

Frequency, GHz

|S21

|, dB

Lf3

= 0.1 mm

Lf3

= 0.3 mm

Lf3

= 0.5 mm

Lf3

= 0.7 mm



20 25 30 35 40 45 5050-70

-60

-50

-40

-30

-20

-10

0

Frequency, GHz

S-pa

ram

eter

s, d

B

S11 (HFSS)

S11 (CST)

S21 (HFSS)

S21 (CST)

Figure 5. Simulated reflection coefficient ¿ S11∨¿ and isolation ¿ S21∨¿ when port 1 is fed and port 2 is matched.

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20 25 30 35 40 45 5050-70

-60

-50

-40

-30

-20

-10

0

Frequency, GHz

S-pa

ram

eter

s, d

B

S22 (HFSS)

S22 (CST)

S12 (HFSS)

S12 (CST)

Figure 6. Simulated reflection coefficient ¿ S22∨¿ and isolation ¿ S12∨¿ when port 2 is fed and port 1 is matched.

20 25 30 35 40 45 50 55-80

-60

-40

-20

0

20

40

60

80

Frequency, GHz

Impe

danc

e Z 11

,

Real part (HFSS) Imaginary part (HFSS) Real part (CST) Imaginary part (CST)

20 25 30 35 40 45 50 55-80

-60

-40

-20

0

20

40

60

80

Frequency, GHz

Impe

danc

e Z 22

,

Real part (HFSS) Imaginary part (HFSS) Real part (CST) Imaginary part (CST)

(a) (b)Figure 7. Simulated input impedance of the proposed antenna (a) Z11 and (b) Z22.

Table 2: Performance Characteristics for the Proposed Dual-Polarized Crossed-Rhombic Antenna

f = 28 GHz f = 38 GHzPort # 1 Port # 2 Port # 1 Port # 2

Gain, dBi 2.958 2.838 3.471 3.523Radiation Efficiency, % 99.99 99.99 99.92 99.94Total Efficiency, % 99.56 99.57 97.79 97.69Input impedance, Ω 53.37+j0.98 62.9+j1.98 63.41-j12.71 57.94-j15.82

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(a) (b)

Figure 8. CST simulated radiation patterns of 0° polarized rhombic antenna at (a) 28 GHz, (b) 38 GHz.

(a) (b)

Figure 9. CST simulated radiation patterns of 90° polarized rhombic antenna at (a) 28 GHz, (b) 38 GHz.

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-40

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-10

0

60

120

30

150

0

180

30

150

60

120

90 90

-180 -135 -90 -45 0 45 90 135 180180-60

-50

-40

-30

-20

-10

0

10

Angle (), degrees

Gai

n m

agni

tude

, dB

HFSS (Co-polarization) HFSS (Cross polarization) CST (Co-polarization) CST (Cross polarization)

-40

-30

-20

-10

0

60

120

30

150

0

180

30

150

60

120

90 90

(a) 28 GHz (b) 38 GHz

Figure 10. 2D radiation pattern in E-plane with co- and cross polarization for the 0° polarized rhombic antenna.

-40

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-10

0

60

120

30

150

0

180

30

150

60

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90 90

-180 -135 -90 -45 0 45 90 135 180180-60

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0

10

Angle (), degrees

Gai

n m

agni

tude

, dB


-40

-30

-20

-10

0

60

120

30

150

0

180

30

150

60

120

90 90

(a) 28 GHz (b) 38 GHz

Figure 11. 2D radiation pattern in H-plane with co- and cross polarization for the 0° polarized rhombic antenna.

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-40

-30

-20

-10

0

60

120

30

150

0

180

30

150

60

120

90 90

-180 -135 -90 -45 0 45 90 135 180180-60

-50

-40

-30

-20

-10

0

10

Angle (), degrees

Gai

n m

agni

tude

, dB


-40

-30

-20

-10

0

60

120

30

150

0

180

30

150

60

120

90 90

(a) 28 GHz (b) 38 GHz

Figure 12. 2D radiation pattern in E-plane with co- and cross polarization for the 90° polarized rhombic antenna.

-40

-30

-20

-10

0

60

120

30

150

0

180

30

150

60

120

90 90

-180 -135 -90 -45 0 45 90 135 180180-60

-50

-40

-30

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-10

0

10

Angle (), degrees

Gai

n m

agni

tude

, dB


-40

-30

-20

-10

0

60

120

30

150

0

180

30

150

60

120

90 90

(a) 28 GHz (b) 38 GHz

Figure 13. 2D radiation pattern in H-plane with co- and cross polarization for the 90° polarized rhombic antenna.

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f = 28 GHz (a) f = 38 GHz

f = 28 GHz (b) f = 38 GHz

Figure 14. Simulated electric field distributions at 28 and 38 GHz when exciting (a) port 1, (b) port 2.

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Figure 15. Proposed dual-polarized crossed-Rhombic antenna with suspended metallic layer, ground plane size Wg × Lg.

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20 25 30 35 40 45 50 55-50

-40

-30

-20

-10

0

Frequency, GHz

|S11

|, dB

H = 1.0 mm, Wg = 35 mm

H = 5.0 mm, Wg = 35 mm

H = 9.0 mm, Wg = 35 mm

Wg = 20 mm, H = 3.3 mm

Wg = 30 mm, H = 3.3 mm

Wg = 40 mm, H = 3.3 mm

Figure 16. Variation of S-parameters with suspended metallic layer dimensions.

Table 3: Performance of the Proposed Dual-Polarized Crossed-Rhombic Antenna with Suspended Metallic Layer

Parametric study case Peak Gain, dBi Side Lobe Level (SLL), dB28 GHz 38 GHz E-plane H-plane

28 GHz 38 GHz 28 GHz 38 GHzWg = 20 mm, H = 3.3 mm 7.30 7.53 -21.2 -24.4 -21.8 -26.6Wg = 30 mm, H = 3.3 mm 7.27 7.93 -22.5 -26.9 -25.6 -32.4Wg = 40 mm, H = 3.3 mm 7.12 8.09 -29.1 -26.3 -34.8 -5.1Wg = 35 mm, H = 1.0 mm 9.52 9.91 -28.5 -29.4 -28.5 -28.1Wg = 35 mm, H = 5.0 mm 7.38 8.37 -19.8 -8.2 -23.6 -30.1Wg = 35 mm, H = 9.0 mm 8.35 8.83 -14.0 -3.9 -3.1 -3.1Wg = 35 mm, H = 3.3 mm 7.28 7.93 -22.5 -26.9 -26.1 -32.4

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20 25 30 35 40 45 50-50

-40

-30

-20

-10

0

Frequency, GHz

S-pa

ram

eter

s, d

B

|S11| |S12| |S21| |S22|

Figure 17. Simulated S-parameters of the rhombic antenna with suspended metallic layer.

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(a) (b)

Figure 18. CST simulated radiation patterns of 0° polarized rhombic antenna with suspended metallic layer at (a) 28 GHz, (b) 38 GHz.

(a) (b)

Figure 19. CST simulated radiation patterns of 90° polarized rhombic antenna with suspended metallic layer at (a) 28 GHz, (b) 38 GHz.

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-40

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-10

0

60

120

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150

0

180

30

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90 90

0 0.2 0.4 0.6 0.8 10

0.2

0.4

0.6

0.8

1

Co-polarization @ 28 GHzCo-polarization @ 38 GHzCross polarization @ 28 GHzCross polarization @ 38 GHz

-40

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0

60

120

30

150

0

180

30

150

60

120

90 90

(a) E-plane (b) H-plane

Figure 20. 2D co- and cross polarization radiation pattern in (a) E-plane, (b) H-plane at 28 GHz and 38 GHz for the 0° polarized rhombic antenna with suspended metallic layer.

-40

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0

60

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0

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30

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90 90

0 0.2 0.4 0.6 0.8 10

0.2

0.4

0.6

0.8

1

Co-polarization @ 28 GHzCo-polarization @ 38 GHzCross polarization @ 28 GHzCross polarization @ 38 GHz

-40

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0

60

120

30

150

0

180

30

150

60

120

90 90

(a) E-plane (b) H-plane

Figure 21. 2D co- and cross polarization radiation pattern in (a) E-plane, (b) H-plane at 28 GHz and 38 GHz for the 90° polarized rhombic antenna with suspended metallic layer.

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f = 28 GHz (a) f = 38 GHz

f = 28 GHz (b) f = 38 GHz

Figure 22. Simulated electric field distributions at 28 and 38 GHz when exciting (a) port 1, (b) port 2.

4. CONCLUSION

A planar dual-polarized crossed-Rhombic antenna for the future fifth generation (5G) base station operations is proposed. Two perpendicular rhombic antennas are printed on a single substrate for dual-polarized operation. The two arms of each rhombic antenna are printed on the same side of the substrate for low cross polarization. The integrated microstrip balun is utilized for the unbalance to balance conversion. A prototype shows that the achieved impedance bandwidth of the two ports with ¿ S11∨≈ ¿S22∨¿ < -10 dB is 22 GHz (24-46 GHz). The gain of the antenna for the single element is found to be almost 3 dBi in the desired frequency band of operation and is further improved to a maximum of 8 dBi with an addition of suspended metallic layer underneath the antenna. Within this frequency band, the mutual coupling of better than -35 dB, the cross polarization of about -30 dB, and the gains of 8 dBi are achieved. The proposed antenna with simple structure and excellent performances is promising for future 5G base stations applications.

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Paper Title (use style: paper title) - ER Publications€¦ · Web viewThe antenna radiation pattern for the 0 polarized rhombic antenna in both E-plane (𝜑 = 90 ) and H-plane (𝜑