Chapter 7

POWER·COMBINING

In Chapter 5, we explored the limitations imposed on the power output of microwave semiconductor devices, and found that they are quite fundamental in nature. While semiconductor devices have the advantage of only needing low voltage power supplies, and are very compact in size and have low weight, they are for these same reasons also limited in their power output. Electron tubes typically use much higher voltages and currents, but are bulky and weigh considerably more than their semiconductor counterparts. There are still ap­plications for which electron tubes are the best choice, even after so many years of development of semiconductor devices. For example, traveling wave tube amplifiers are the prevalent solution for achieving a high power output in communication satellite transmitters, despite the disadvantages of larger weight and volume. Other examples are some radar transmitters (pulsed mag­netrons) and microwave ovens (CW magnetrons). A comparison of the power output of typical electron tubes, and some major semiconductor devices is given in Figure 7.1.

A general method for achieving higher power out from semiconductor de­vices involves combination of the output of several devices, "power combining". These methods may allow semiconductor devices to challenge electron tubes in some further application areas in the future, especially for millimeter wave frequencies, where, as we have seen, the power output falls quite rapidly as the frequency increases. Simply increasing the area of a device runs into the limita­tion presented by a too low device impedance, see Chapter 5. Power combining is accomplished by using appropriate circuits to add the power from several individual devices, which have essentially the optimum power output possible for a single device. In a special case, the individual devices are allowed to radiate into free space, combining their power outputs to a single beam. The latter method is called "spatial power-combining" .

A further distinction between power combining methods is between those which combine the power of oscillators and amplifiers, respectively, illustrated in Figure 7.2. Amplifiers and oscillators, based on a particular device, ideally have approximately the same maximum power output. However, two-terminal devices used as stable amplifiers require additional components (circulators). On the other hand, high power three-terminal devices are almost universally designed as amplifiers. An oscillating power combiner may be employed di­rectly as a source, and is in this sense simpler than an amplifying one. Ampli­fying power combiners may yield higher output, if three-terminal devices are being combined, but also require an original source to amplify. This source may often be a high-quality, low-noise, one, allowing the entire power-combiner

S. Yngvesson, Microwave Semiconductor Devices© Kluwer Academic Publishers 1991

184 Microwave Semiconductor Devices

APPROXIMATE lIMrTS OF MICROWAVE poweR SOURCES

10Mw

t--... ~YSTRON \ ,

'Mw

100kW

~ ~ 10llW w

~

~ lOW

'OOW

~ GYROTAON

\

~, \

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\ H~llX+ \ \ \

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FREQUENCY (GHz)

Figure 7.1. Typical mazimum CW output power for microwave power sources, versus frequency. "Solid state" {i.e. semiconductor} devices have been con­densed into one single curve. Reprinted from STAPRANS, A. {1989}. "Evo­lution of Microwave Power Sources," Microwave Systems News, Vol. 19, June 1989, p. 20, with permission.

Chapter 7 185

(a)

Diodes

Cavity

Output

(b)

Figure 7.2. fllustration of the distinction between amplifying and oscillating power combiners.

186 Microwave Semiconductor Devices

to be low-noise. Similar results may be accomplished for oscillating power­combiners by injection locking (over the locking bandwidth). Amplifying and oscillating power combiners also have different characteristics with respect to interactions between the circuit and the active device: Some types of oscillat­ing combiners operate with substantial mutual coupling, and thereby become "inter-iIJjection-locked" (Stephan, 1986), while amplifiers need to be isolated from each other in order to be stable. There clearly are thus a number of not so obvious trade-offs to consider in choosing between amplifying and oscillat­ing power-combiners. Next, we want to discuss specific implementations of the actual combiners. Often, the following discussion will be applicable to either oscillating or amplifying combiners. When it is not, we shall make a note of this fact.

We can now distinguish the following major techniques of power-combining (Russel, 1979; Chang and Sun, 1983):

1. Chip-level combining

2. Circuit-level combining

3. Spatial (Quasi-Optical) combining

4. Multi-level combining (combinations of 1), 2) and 3))

CHIP-LEVEL POWER-COMBINING

We have shown an example of chip-level power-combining of IMPATT devices in Chapter 5. This type of combining works well for a very smail number of devices (2-4). In the IMPATT case, the total area of the devices still is limited by impedance considerations, and the main advantage of combining the power output from several chips is the lower total thermal resistance, which increases the power output if the thermal power limit dominates (see Chapter 5). Chip-level power combining of IMPATT devices has also been described by Rucker et ai. (1979,1981).

Three-terminal power devices*) are often combined at the chip level. An early such design is described by R. Pucel (1985). Two monolithic MESFET oscillators were designed, employing 100 ohm microstrip transmission lines at the input and output. Microstrip lines of 50 ohm impedance are connected to the two 100 ohm input lines through a simple power splitter, as shown in Figure 7.3, with a corresponding arrangement at the output. The result is an amplifier which is matched to 50 ohms at the input and the output. The approach was extended to a four device combiner by starting at the 25 ohm level. In this case, a simple L-C transformation circuit matches the 25 ohm

*) We will discuss power-combining of three-terminal devices in this chapter although a detailed treatment of such devices is deferred to Chapters 10-12. The present discussion only requires a very general concept of such devices.

Chapter 7 187

Figure 7.3. Chip-level combining of two MESFET monolithic power ampli­fier,. Repoduced from PUCEL, R.A. (1985). "Application, of Monolithic Microwave Integrated Circuit6," in Gallium Ar,enide Technology, David K. Ferry, Ed., Howard W. Sam" Indianapoli" IND, Vol. 1, 2-49.

lines to 50 ohms at the input and output,respectively. The maximum output power nearly doubled for each doubling of the number of devices, and 2 watts was measured for the four-FET combiner. Some extra losses are expected due to the use of matching circuits at higher impedance levels, but the net result clearly was worth the effort in this case. We can also see that the limitations in terms of realizable impedance levels of microstrips make it difficult to extend the method to larger numbers of devices. An improved design may also require isolation resistors which will dissipate further power (see the next section). We may conclude that chip level power combining often is convenient for a small number of devices.

CIRCUIT LEVEL POWER COMBINING

Nonresonant cOlD.bining

A typical method of circuit level power-combining is to use a corporate structure (Figure 7.4). Russel (1979) shows how the combining efficiency de­pends on the number of devices combined, with the loss in each adder stage as

188 Microwave Semiconductor Devices

a parameter. As can be seen from Figure 7.5, the combining efficiency becomes quite low as the number of devices grows, even for as small losses per adder as a few tenths of a dB. The corporate structure still is superior in this re­spect to the chain structure, which consists of a number of directional couplers connected in a chain configuration, with each active device feeding into one of the coupled ports (Russel, 1979). Typical adder elements in the corporate structure may be couplers such as the Wilkinson, Lange, branch line, or cou­pled line types in microstrip, or waveguide hybrid couplers (see (Pozar, 1990) for example). Trade-offs exist for different coupler types: For example, the Wilkinson coupler is simple and has low loss, but it does not provide as good isolation between different active elements as the Lange or branchline couplers. The latter have higher loss, however. Either amplifiers or oscillators may be combined, but if oscillator devices are being employed, one needs to provide injection locking to ensure uniform phase.

In a recent 20 GHz MESFET combiner design, Auricchio, Jr., et al (1989) combined eight MESFET power amplifier modules. The individual 'modules produced 2 watts of power with 25% typical power-added efficiency. The first combining step used a Lange coupler, with a loss of 0.15 dB. Four of these 3.5 watt units were then combined to yield 12 watts, employing a waveguide 4-way combiner with 0.1 dB loss (plus .2 dB for the microstrip-waveguide tran­sition). The final power-added efficiency then decreased to 15.5%, in approx­imate agreement with the predictions from Figure 7.5. The 1 dB bandwidth was 8.5%. It is noteworthy that the 4-way adder had to be designed in waveg­uide - a microstrip version would have been too lossy! This is mostly a problem at very high frequencies, as evidenced by a 5 GHz 4-way MESFET combiner developed by Yanagawa et al. (1990), which had 28% power-added efficiency at an output power of 33 Watts. The adder in this case was designed using a combination of lumped and distributed matching elements. If we go above 20 G Hz, we expect considerably lower efficiencies due to the higher loss of adders for these frequencies. This fact points to the desirability of other approaches for millimeter wave power-combiners, to be discussed later in this chapter.

Nonresonant combiners at millimeter wave frequencies have made use of waveguide hybrid couplers. The basic combiner unit is shown in Figure 7.6. Power incident at port 1 divides equally between the two active devices at ports 2 and 3, and is reflected (amplified) to port 4. If the amplifiers are well balanced, no power will be returned to port 1. The condition for balance is quite critical, and limits the number of sources which may be combined to four. A CW power of 2.5 watts at 61 GHz was generated by a four-diode combiner of this type (Ma et al., 1980), using IMPATTs.

A very broadband type of nonresonant combiner is the traveling-wave type. A 10-30 GHz monolithic version has been described by Tserng and Saunier (1985).

Chapter 7

ADDERS

22

ADDERS 2

ADDERS ADDER

N STAGES 0' P;;~W-ER-C-O-MB-IN-'N-G---~

189

Figure 7.4. A corporate 6tructure for power-combining. Reprinted from R us­SEL, K.J. {1979}. "Microwave Power Combining Technique6," IEEE Tran6. Microw. Theory Tech., MTT-fn, i72, @1979 IEEE.

Figure 7.5. Combining efficiency of corporate power combining 6tructure6, as a function of the number of elements combined. The loss of each adder i" a8.!lumed to be constant, and is given as a parameter. Reprinted from RUSSEL, K.J. {1979}. "Microwave Power Combinin9 Technique"," IEEE TranI. Microw. Theory Tech., MTT-27, i72, @1979 IEEE.

190 Microwave Semiconductor Devices

INPUT OUTPUT

3dB HYBRID COUPLER

x:

Figure 7.6. A hybrid coupler power combiner for two sources. Reprinted from CHANG, K., and SUN, C. (1983). "Millimeter- Wave Power-Combining Techniques," IEEE Trans. Microwave Theory ·Tech., MTT-31, 91, @1983 IEEE.

N-way Nonresonant Combining

The inherent symmetry of a radial line or other circularly symmetric ar­rangements can be exploited in order to design an efficient power-combiner (see (Belohoubeck et al., 1986) for example). Isolation resistors are typically inserted between individual devices and the combining efficiency is ideally in­dependent of the number of devices combined. Goel (1984) used an 8-way microstrip radial combiner (see Figure 7.7) for 8.2 W output power at 17.7-19.1 GHz, with MESFET amplifying active elements.

To summarize, circuit level power combining with nonresonant combiners has the advantages of wide bandwidth, and good efficiency, up to about 20 GHz. A corporate adder network is sufficient if a moderately large number of devices are to be combined «12), while N-way combiners can accommodate larger numbers of devices. Above 20 GHz, hybrid waveguide couplers can be used, but are limited to combining four modules, when used in the balanced mode described above.

Resonant Combining

An early, very successful, power-combining technique was the Kurokawa waveguide combiner (Kurokawa and Magalhaes, 1971), see Figure 7.8. It em­ploys a rectangular waveguide cavity, resonant in the TElon. mode. Coaxial

Chapter 7

5oU11lIAT[ . • (ntO" G.U~ llilC~

OIOl(II1I(116I.l lIClIli

191

Figure 7.7. Ezample 0/ amplifying power-combiner. The top figure shows the connection 0/ the eight power amplifiers, with the power splitter and power combiner, respectively. The lower figure shows the outline 0/ the stripline con­figuration/or the power splitter/combiner. Reprinted from GOEL, J. (198-1). "A K-Band GaAs FET Amplifier with 8.2- W Output Power," IEEE Trans. Microwave Theory Tech., MTT-32, 317, @198-1 IEEE.

192 Microwave Semiconductor Devices

modules containing active devices are positioned along the narrow walls at points for which the microwave magnetic field has maxima. Magnetic coupling is thus established between the cavity and the active modules. The original combiner contained 12 IMPATT oscillating devices, and produced 10.5 watts CWat 9.1 GHz. An amplifying version is also possible, but more cumbersome since waveguide inputs need to be connected to all devices. A later develop­ment due to Harp and Stover (1973) introduced a cylindrical cavity operating in the T Momo mode, see Figure 7.9. Again, coaxial modules are placed around the periphery of the cavity: due to the azimuthal symmetry of the TMomo mode, the modules can be spaced essentially as close as is physically possible « 0.2 inches at X-band), whereas the Kurokawa combiner requires a spacing of >..g/2. While typically the TMo10 mode is used, larger numbers of devices can be combined in cavities resonant in the T Mo2o and T M o4o modes. The output power may be coupled via a coaxial probe through the center of the top of the cavity (this is the maximum point for the E-field). The combining efficiency of resonant cavity combiners in general is independent of the num­ber of active elements combined ('n'), and may actually even increase as more elements are added. This feature is easy to understand, since the diodes all use the same circuit, which has essentially constant losses, independent of n.

Waveguide cavity combiners are operated either as injection-locked or free­running oscillators. They naturally have a fairly narrow bandwidth (a few percent). Conventional mode charts for cylindrical cavities may be used to find dimensions of the cavity in which the desired mode is far from other resonances (see for example (Pozar, 1990). If T MOmo modes with m > 1 are used, it may be necessary to suppress unwanted modes by techniques such as cutting radial slots and filling these with absorber material. Such moding phenomena ultimately limit the number of devices which can be combined in a single cavity.

The design of the basic coaxial module has been described by Chang and Ebert (1980), Allen et al. (1981), and Bialkowski (1986). The load in one end of the module, made from absorber material, ensures that the diode sees a broadband, lossy termination at frequencies outside the cavity resonance. The best stability will result if the load is tapered for a good match. For this case, a larger amount of power will be lost in the load, however, and the efficiency can be increased while still maintaining adequate stability by making the taper of the load less gradual.

A large number of cylindrical cavity power-combiners have been built, most commonly using IMPATT diode active elements. Output powers as high as 1 kW peak (300 W average) have been obtained at X-band (Drubin et al., 1982). Russel (1979) reviews results for combiners of this type with up to 32 diodes. More recently, Tokumitsu et al. (1984), designed a cylindrical cavity combiner for MESFET amplifiers, with 80 watt CW output power at 6 GHz. This resonant combiner was unconventional in the sense that the active elements were completely outside the cavity - cavities were used as

Chapter 7

DIODE

TAPERED TERMINATION

193

Figure 7.8. A Kurokawa-type waveguide power combiner, with siz elements shown. Reprinted from CHANG, K., and SUN, C. (1983). "Millimeter­Wave Power Combining Techniques," IEEE Trans. Microw. Theory Tech., MTT-31, 91, @1983 IEEE.

COAXIAL LINES

MAGNETIC FIELD

INPUT -OUTPUT

ECCOSORB TERMINATING ELEMENT ----lU

COAXIAL LINES

BIAS

COMBINER CAVITY

COAXIAL

ll===::::;;::::.l-t.===::I==1f- ~?NU:lI NG CI RCULATOR --~1.-,-....... ,--___ ~"...... __ ---I

WAVEGUIDE-TO-COAXIAL TRANSITION

Figure 1.9. A circular waveguide type power combiner. Reprinted from CHANG, K., and SUN, C. (1983). "Millimeter-Wave Power Combining Techniques," IEEE Trans. Microw. Theory Tech., MTT-31, 91, @1983 IEEE.

194 Microwave Semiconductor Device.

both dividers and combiners, and were designed with 8 ports with coaxial connectors, to which the amplifiers were connected. Also, a broader bandwidth (10%) was obtained by using dual coupled cavities operating in the TMolO and TMo20 modes, respectively.

Cylindrical cavity combiners have been developed for frequencies up to 30-40 GHz. Adlerstein and Fines (1989) discuss the modeling and design of a 35 GHz IMPATT combiner source, with 38 W peak power at a 30 percent duty cycle. The active elements are modeled, using the large-signal model which we reviewed in Chapter 3. This source consists of two stages, each of which is a circulator-coupled injection-locked oscillator, see Figures 7.10 and 7.11. The first stage combines two diodes in a rectangular cavity, while the second stage employs eight diodes in a TMo20 mode cylindrical cavity. Adlerstein and Fines show how the oscillator efficiency can be predicted, based on S­parameters which can be derived from measurements on the cavity, with short circuits placed at the diode ports. This is useful to know, since in the limit of infinite gain, the power added efficiency approaches the oscillator efficiency. A power-added circuit efficiency of 65-70 percent was deduced for the eight diode combiner. Further improvements are expected, since single-diode tests gave 7 W peak power, while the power added per diode in the combiner was 4.7 W. The DC to RF conversion efficiency was 8-10%.

Resonant power combiners for frequencies above 40 GHz have used rect­angular cavities up to 217 GHz (Chang et al., 1983). At 94 GHz, Thoren and Virostko (1983) achieved 1.89 W peak, with up to 35% duty factor, using a 4-diode combiner in the final stage. This source was mechanically tunable from 90 to 99 GHz. Yen and Chang (1981) used four two-diode combiners in rectangular cavities, in turn combined with hybrid couplers, to achieve a pulse output of 63 watts peak, but with a lower duty cycle « 1%).

SPATIAL (QUASI-OPTICAL) POWER-COMBINING

In a spatial power-combiner, each active module is connected to an an­tenna element, which radiates the power either into a quasi-optical resonator, or directly into free space. A definite phase-relationship must be maintained between the fields radiated from the individual antenna elements. Defined in this manner, we realize that any phased antenna array, in which separate transmitter modules are used behind the phased array elements, is a spatial power combiner. As an example, Figure 7.12 shows a monolithic millimeter wave phased array (Kinzel, 1990). The basic idea is, of course, the same as for the power combiners we have described above. In the case of the phased array, the use of many active modules fits naturally since many antenna el­ements are used, anyway. No substantial further losses need occur, once the power has been coupled from the active module to the antenna element; i.e. the combining efficiency can be essentially 100%. Large arrays of this type have been built as parts of radar systems at centimeter waves for many years.

Chapter 7

Stage 1

Two-Diode Cavity

Stage 2

Eight-Diode Cavity

195

Figure 7.10. fllustration of a two-stage circulator-coupled cavity power com­biner. Reprinted /rom ADLERSTEIN, M.G., and FINES, J. (1989). "A Multi-IMPATT Injection-Locked Oscillator at 35 GHz, " IEEE Trans. Mi­crow. Theory Tech., MTT-37, 571, @1989 IEEE.

Figure 7.11. Photograph of a the hardware for a two-diode cavity power­combiner, described in ADLERSTEIN, M.G., and FINES, J. (1989). "A Multi-IMPATT Injection-Locked Oscillator at 35 GHz," IEEE Trans. Mi­crowave Theory Tech., MTT-37 ,571 @1989 IEEE. The coazial module with impedance tranllformer and matched load can be uen at the center. Courte8Y of Dr. D. Masse', Raytheon Co., Lezington, MA.

196 Microwave Semiconductor Device.

The topic of discussion in this chapter is instead the use of these principles for power-combining of millimeter waves. One should realize that the output power, once radiated from the array, does not have to continue further into "free space", but can equally well be focussed and picked up by for example a waveguide horn, see Figure 7.12b. This makes the array into a self-contained source of the general type which we discussed earlier in this chapter - with the distinction that the division and combining of power occurs in free space. The name "spatial combining" thus usually refers to "free space combining" .*) It is also called "quasi-optical power-combining", where quasi-optical refers to the use of optical components, such as reflectors, lenses, etc., which have maximum dimensions which are maybe tens of wavelengths (as compared to the tradi­tional optical components, with dimensions of 104 to 10& wavelengths). Since spatial power combining arrays have dimensions which are of this order, it is appropriate to use the term "quasi-optical". In this section, we will discuss power-combiners which are termed either "spatial" or "quasi-optical," by their originators.

One type of quasi-optical power combiner is arranged so that the individ­ual antenna elements radiate into a quasi-optical ("Fabry-Perot") cavity, which is in this sense similar to the resonant combiners discussed above. The main distinctions are (1) the size of the cavity, which is larger in the quasi-optical combiner, and (2) the fact that the cavity is of the "open" type. A combiner of this kind is shown in Figure 7.13 (Mink, 1986).

Spatial power combiners generally should be very efficient, and are espe­cially convenient for millimeter waves. The efficiency is predicted to be essen­tially independent of the number of elements combined, whereas any combiner which involves transmission lines will have low efficiency in the millimeter wave range. Some versions of spatial power combiners have a wide bandwidth. We will next illustrate the spatial power combining concept with Bome examples of the different types which have been discussed so far.

A.DlpliCying Phased A.rray CODlbiners

Millimeter wave monolithic phased arrays have been studied rather in­tensely in the last few years. A typical such array has microstrip patch antenna elements, which are either fabricated on a separate substrate, or directly on semi-insulating GaAs (Kinzel, 1990). Modules with active elements are thus either placed below the antenna array substrate, or are completely integrated with the radiators, i.e. on the same substrates. The modules would include phase shifters for scanning of the radiated beam. An array of this type, with fixed phase shifts for each element, could also be used in a focussing mode, and constitute a source of the type shown in Figure 7.12b. An alternative

*) In a sense, the N-way nonresonant combiner is also a "spatial" combiner, since it spreads the active modules in a spatial arrangement. It employs cir­cuit/transmission line elements for this purpose, however, not free space.

Chapter 7

Brick Construct Radialors, MUtes, Coo.of CitaJitry, Thennal "'ilnagemenl

RF & DC Oislribution

§ 3·inch \@J Wafer

• Accommodates large size! number ot MMICs

• Replaceable modules

Very thin

RF & DC Distribution

Thermal Wwgement

r@3.inch \@' Water

Circular polarization straightforward

but... • Finesse circuit processing yields Interconnections between layers

very difficult • Hermetlclty straightforward

but ..•

• Circular polarization more difficult

Lens

a

Power combining

array

Amplifying modules.

b

Lens

Possibly other

quasI-optical components

197

Figure 7.12. (a) Con6truction of monolithic pha6ed arraY6. Reproduced from KINZEL, J. (1990). "Recent Advance6 in Monolithic Millimeter Wave Ar­ray6," IEEE AP-S Intern. Symp. Digest, Vol. IV, 1402, @1990 IEEE. (b) An amplifying quasi-optical power combining array.

198 Microwave Semiconductor Device.

~z x

Figure 1.13. A qua.i-optical Fabry-Perot cavity type power combiner. Reprinted from MINK, J.M. (1986). "Qua6i-Optical Power Combining of Solid-State Millimeter Wave Source., II IEEE 7ran •. Microw. Theory Tech., MTT-3., !J73, @1986 IEEE.

arrangement has been developed by Rascoe et al. (1990), as shown in Figure 7.14. (also, see (Chang et al., 1990». This version uses exponentially tapered ("Vivaldi") endfire antenna elements, which have narrower element beams than microstrip patch radiators. Microstrip patches have to be spaced of the order of 0.5 to 1.0 free space wavelengths apart (depending on the desired angular scan range), in order to avoid large losses due to grating lobes. The narrower element beamwidth of the Vivaldi elements cuts the grating lobe losses down to a negligible level. The designer thus has greater freedom to choose the spac­ing of the antenna elements, as well as the spacing of the active amplifiers and phase shifters. A very important consideration in spatial power combiners of this type is the dissipation of power from the active modules, which is facil­itated by the larger element spacing allowed. A combining efficiency of 62% has been measured in a 32 GHz 21 element prototype array, and it is expected that future improvements will bring this to the 70-80% range. The main losses are due to the connections between power amplifiers and antenna elements, as well as some radiation losses. With an expected output power of 350 to 500 mW from the HEMT power amplifiers, the combined output power will be 5-7 Watts, scannable by ±10° away from a direction perpendicular to the array. A 63 element array is also being planned for possible use as a transmitter source for future deep space missions.

Chapter 7 199

Figure 7.14. Photograph of a Vivaldi antenna array developed for a ~1 ele­ment amplifying power combiner. From CHANG, J., SCHAUBERT, D.H., YNGVESSON, K.S., HUANG, J., JAMNEJAD, V., RASCOE, D., and RI­LEY, A.L. (1990). IEEE AP-S Intern. Symp. Digest, 1150, @1990 IEEE.

Quasi-Optical Resonator Combiners

The concept of a quasi-optical resonator combiner was proposed and ana­lyzed by Mink (1986), following up work by Wandinger and Nabandian (1983). The idea is similar to that of a laser oscillator. An active array consisting of small, driven, source dipoles is assumed to be placed near the flat mirror of a quasi-optical Fabry-Perot resonator (Figure 7.13). One may thus compare the function of the source array to that of an active laser medium, which feeds power into the same kind of resonator. In the laser case, it is well known that most of the power will be radiated into the fundamental resonator mode, for which the field as a function of radius follows a Gaussian function The entire source array could be fabricated as one monolithic semiconductor wafer. Mink showed that for source arrays which were at least of a size = 5 x 5, about 80% of the radiated power would be transferred to the fundamental resonator mode if the sources are uniformly excited. If the source amplitudes are instead given a Gaussian variation, which matches the field variation versus radius in the resonator mode, then one can transfer essentially 100% of the power to the

200 Microwave Semiconductor Devices

Wave guide

~L spherical mirror grooved mirror

Figure 7.15. A Quasi-optical power-combiner consisting of a Fabry-Perot resonator, with a grooved fiat mirror. Reprinted from NAKAYAMA, M., HIEDA, M., TANAKA, T., and MIZUNO, K. (1990). IEEE MTT-S Int. Microw. Symp. Digest, 1209, @1990 IEEE.

fundamental mode, provided that the source spacing is close to its optimum value. As in the laser case, it can then be coupled through the curved mir­ror, which would be made partially transparent. This general concept is the basis for a number of recently investigated spatial combiner designs. A larger number of elements (50-100) can be combined, although some restrictions are likely to exist for combining much larger numbers than, say, 100 elements. The principle should work well into the sub-millimeter wave (THz) range, and an intriguing possibility is to produce levels of power which would be useful for practical applications at frequencies for which no fundamental frequency oscillators presently exist, which are based on semiconductors (harmonic mul­tipliers are used extensively, see Chapter 9). A promising approach would be to combine a monolithic array of RTD devices which are likely to have an upper frequency limit as high as 1 THz (see Chapter 4).

At the time of this writing (mid-1990), preliminary results have been ob­tained by several groups. In an early experiment, Wandinger and Nalbandian (1983) used dielectric rod antenna elements to successfully oscillate two 60 GHz InP Gunn elements in a quasi-optical resonator. Young and Stephan (1987) combined the power of two X-band Gunn sources coupled via planar microstrip patch resonators, placed on the flat mirror ofa Fabry-Perot. Stephan (Stephan

Chapter 7 201

and Itoh, 1990) also successfully oscillated an RTD device in a Fabry-Perot cav­ity. Xue et a1. (1988) have combined two packaged Gunn sources, which were not planar, and were placed close to the center of the cavity instead of close to one mirror.

Recently, larger arrays of devices have been combined, in configurations which are getting closer to the one proposed by Mink. Mizuno et al. (1988; also (Nakayama et aI., 1990» proposed the use of a grooved flat mirror, which allows devices to be mounted in, and biased through, the grooves, see Figure 7.15. (the grooved mirror is constructed from a number of plates, which are electrically insulated to make the biasing possible). With this configuration, 18 Gunn diodes and (separately) 6 MESFETs have been combined so far. Experiments at 50 GHz with this resonator showed that it does indeed have a high Q-value.

Compton and York (see Rutledge et aI., 1990) operated a 4 x 4 array of X-band Gunn diodes coupled to microstrip patch antennas. In its original form, this combiner acts like an oscillating version of a phased array, in which the active elements are injection-locked due to mutual coupling. The addition of a dielectric plate effectively couples the combiner to a low-Q Fabry-Perot, and the locking of the individual oscillators becomes much more stable. In the most complete demonstration to-date of a quasi-optical resonator combiner, a (10 X 10) array of MESFET elements coupled to a wire grid was operated by Rutledge et al. (1990), with a total output power of 0.625 W at about 5 GHz (Figure 7.16). An aluminum mirror at a distance of about a wavelength acts as the second mirror. Radiation patterns show that all devices are locked in phase. This realization, as well as the one of Mizuno et al., are quite close to the original idea for a quasi-optical resonator combiner. As millimeter wave versions are developed, the resonator will likely take on more of the traditional quasi-optical Fabry-Perot form, and the advantages of its high Q-value will no doubt show up as a further narrowing of the FM-spectrum (compare Chapter 6). Also, monolithic active device arrays are already feasible, and can be expected to appear in the next generation of spatial combiners.

Quasi-Optical Grid Multipliers

In an arrangement similar to the quasi-optical grid oscillator (Rutledge et al., 1990), a grid loaded with diodes which have a nonlinear capacitance­versus-voltage characteristic may be used as a quasi-optical multiplier. Har­monic multipliers in general are discussed in Chapter 9, where we shall see that efficiencies of 20-50 percent can be achieved if the diodes have sufficiently high cut-off frequencies, compared with the frequencies of operation. A single diode may be able to handle an input power of a few tens of milliwatts, so that monolithic arrays of diodes with, say, a thousand elements, can produce powers of the order of watts. A prototype monolithic tripler grid has indeed produced output powers at about 100 GHz in this range (Hwu et a1., 1990). The non-

202 Microwave Semiconductor Device.

Figure 7.16. A 100 element wire·grid MESFET power combiner, operating at 5 GHz. Reprinted from RUTLEDGE, D.B., POPOVIC', Z.B., WEIKLE,II, M., POTTER, K.A., COMPTON, R.C., and YORK, R.A. (1990). "QUali­Optical Power-Combining Arrays," IEEE MTT-S Intern. Microw. Symp. Dige.t, 1201, @1990 IEEE.

linear elements were 6000 back-to-back BIN (Barrier-Intrinsic-n+) diodes on a 15 cm2 wafer. These diodes have a more nonlinear CoY-characteristic than, for example, Schottky-barrier diodes. The tripler efficiency measured in this experiment was 8.5%, whereas 25% was predicted based on the diode character­istics. This is therefore a very promising approach for extending the frequency range of millimeter wave sources. A number of other nonlinear elements may also become important in the future.

Summarizing, it is clear that many versions of spatial power-combiners will be investigated in the future, and make available considerably higher powers from semiconductor sources in the millimeter/sub millimeter wave range.

Chapter 7 203

Problems, Chapter 7

1. a) Calculate the diameter of aT MolO-mode cavity (see e.g. Pozar (1990)) for use as a power combiner at 10 GHzj Assume that each diode module has a diameter = 5 mm and center the diode modules on the periphery of the cavity. How many diodes can be used, maximally? b) Choose the highest power diode at 10 GHz from Figure 5.6, and calculate the maximum CW power output from the combiner in a). c) Same question at 44 GHzj (scale the cavity and diode module sizes with the frequency ratio).

2. You want to design a power combiner with a power output of 5 W at 50 GHz, and have available monolithic GaA .. three-terminal devices with 200 mW output each. Compare the following approaches.

a) A corporate combining structure. Sketch a (partial) layout of the de­vices and the adders. The devices are 1 mm wide. Estimate the minimum loss per adder as the loss in the length of micros trip in your layout. Typical loss of microstrip at this frequency is about 0.3 dBI >.,. Assume f. = 12.8 and a reasonable effective f. The final output power is radiated from a waveguide horn. The transition to the horn has 0.8 dB loss and the radiative efficiency of the horn is 80%.

b) A quasi-optical combiner, see Figure 7.12.h. The loss from each ampli­fier output to the radiative element is 0.3 dB, and the radiative efficiency of each element is 70%. Which method is more efficient? How many elements are needed in each case?

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206 Microwave Semiconductor Devicell

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FURTHER READING

CHANG, K. (editor) (1990). "Handbook of Microwave and Optical Compo­nents," Volume 2. John Wiley & Sons, New York. See Chapters 6, 7 and 10.

KUROKAWA, K. (1971). "The Single-Cavity Multiple-Device Oscillator," IEEE Tran". Microw. Theory Tech., MTT-19, 793.

___ , (1973). "Injection-Locking of Microwave Solid-State Oscillators," Proc. IEEE, 61, 1386.