Chapter 9Power electronics and its applications

9.1 Introduction

Power electronics deals with conversion and control of electrical power in the wide range of milliwatts to gigawatts with the help of power semiconductor devices. The applications of power electronics may include dc and ac regulated power supplies, uninterruptible power supply (UPS) systems, electrochemical processes (such as electroplating, electrolysis, anodizing, and metal refining), heating and lighting control, electronic welding, power line static VAR compensators [SVCs, static VAR generator, or static synchronous compensator (STATCOM)], active harmonic filters, high voltage dc (HVDC) systems, photovoltaic (PV) and fuel cell (FC) power conversion, solid state dc and ac circuit breakers, high-frequency heating, and motor drives. Motor drives area may include applications in computers and peripherals, solid state starters for motors, transportation (electric/hybrid electric vehicles (EV/HEV), subway, etc.), home appliances, paper and textile mills, wind generation system, air-conditioning and heat pumps, rolling and cement mills, machine tools and robotics, pumps and compressors, ship propulsion, etc. In addition to applications in energy systems and industrial automation, power electronics is now playing a significant role in global energy conservation that is indirectly helping in the environmental pollution control, i.e., solving the global warming problem.

9.2 Power Semiconductor SwitchesPower semiconductor switches are primarily used to control the flow of electrical energy between the energy source and the load, and to do so with great precision, with extremely fast control times, and with low dissipated power. The application of IC technologies on state-of-the-art power semiconductor devices has resulted in advanced components with low power dissipation, simple drive characteristics, good control dynamics, and switching power extending into the megawatt range.

In electrical energy transfer, electronic devices are generally required to operate in switch mode. This means they should have ideal switch-like characteristics: they appear like a short-circuit passing current with minimal voltage drop across it in the on state; in the other side, they block the flow of current by supporting full supply voltage across it appearing like an open circuit in the off state. They operate in a different mode from power amplifying devices, which allow power transfer according to a linear relationship with an input signal, such as audio amplification. In switch mode operation, an electronic control signal is applied to turn the switch ON, and removed to turn the device OFF. For present devices, the control signal is typically in the 512 V range while the power supply voltage can be in the 20 V8 kV range.

Solid state switch mode devices have been used for controlling power transfer for over 50 years. Demands for the rational use of energy, miniaturization of electronic systems, and electronic power management systems have been the driving force behind the revolutionary development of power semiconductor devices over the last five decades.

The power semiconductor switches cover all applications in the power range from 1 W needed for charging the battery of a mobile phone, up to the Giga Watts range needed for energy transmission lines (HVDC lines). The bipolar devices (e.g., thyristor, integrated gate-commutated thyristor [IGCT]) are a key technology for ultrahigh power systems while the MOS controlled devices (e.g., insulated gate bipolar transistor [IGBT], power MOSFET including smart power systems) are the driving components for medium and low power electronic conversion systems. In the top power end, the switching frequency is below several 100 Hz, the medium power is dominated in the range of 10 kHz, but the system development for lower power is driven by several 100 kHz.

Advances in power electronic systems over the last three to four decades have been marked by five major inventions. Light-triggered thyristors and IGCTs in the top-end power range, IGBTs in the mid and high-end power range, power MOSFET in the low-end power range, and SMART power systems for monolithic system integration, are mainly applied in automotive power. The bipolar transistor and the gate turn-off (GTO) thyristor do not play a significant role in present development.

In the following sections, we will have a comprehensive discussion of a few power semiconductor devices.

9.2.1 Silicon Controlled Rectifier (SCR)

The thyristor, also called a silicon-controlled rectifier (SCR), is basically a four-layer three-junction pnpn device. It has three terminals: anode, cathode, and gate. The device is turned on by applying a short pulse across the gate and cathode. Once the device turns on, the gate loses its control to turn off the device. The turn-off is achieved by applying a reverse voltage across the anode and cathode. The thyristor symbol and its voltampere characteristics are shown in Fig. 9.1. There are basically two classifications of thyristors: converter grade and inverter grade. The difference between a converter-grade and an inverter grade thyristor is the low turn-off time (on the order of a few microseconds) for the latter. The converter grade thyristors are slow type and are used in natural commutation (or phase-controlled) applications.

Fig. 9.1 Thyristor symbol and (b) voltampere characteristicsInverter-grade thyristors are used in forced commutation applications such as DC-DC choppers and DC-AC inverters. The inverter-grade thyristors are turned off by forcing the current to zero using an external commutation circuit. This requires additional commutating components, thus resulting in additional losses in the inverter.

Fig. 9.2 Power thyristor structure and its doping profile abbreviation The basic structure for an N+PNP+ power thyristor is illustrated in Fig. 9.2. The structure is usually constructed by starting with a lightly doped N-type silicon wafer whose resistivity is chosen based upon the blocking voltage rating for the device. The anode P+ region is formed by the diffusion of dopants from the backside of the wafer to a junction depth xJA. The P-base and N+ cathode regions are formed by the diffusion of dopants from the front of the wafer to a depth of xJB and xJK, respectively. Electrodes are formed on the front side of the wafer to contact the cathode and P-base regions, and on the backside of the wafer to contact the anode region. No contact electrode is usually attached to the N-drift (N-base) region.

Thyristors are used in applications at very high power levels which require a large voltage-blocking capability, typically in excess of 3,000 V. To achieve such high-voltage blocking capability in both forward and reverse operating quadrants, the P+ anode/N-drift junction and the P-base/N-drift junction must have a highly graded doping profile. This increases the blocking voltage capability due to voltage supported on the more highly doped side of the junction and allows the utilization of the positive and negative bevels at the edges to suppress premature breakdown at the junction termination. The highly graded junctions can be produced by using gallium and aluminium as the P-type dopants instead of boron.

The thyristor structure contains three PN junctions that are in series as indicated in Fig. 10.2. When a negative bias is applied to the anode terminal of the device, the P+ anode/N-drift junction (J1) and the N+ cathode/P-base junction (J3) become reverse biased while the P-base/N-drift junction (J2) becomes forward biased. Due to high doping concentrations on both sides of the N+ cathode/P-base junction (J3), it is capable of supporting less than 50 V. Consequently, most of negative bias applied to the anode terminal is supported by the P+ anode/N-drift junction (J1). The reverse-blocking voltage capability for the device is determined by the doping concentration and thickness of the N-drift region. Note that an open base bipolar transistor is formed within the thyristor structure between junction J1 and junction J2. Consequently, the breakdown voltage is not determined by the avalanche breakdown voltage but by the open-base transistor breakdown voltage. The width of the N-drift region between these two junctions must be carefully optimized to maximize the blocking voltage capability and minimize the on-state voltage drop.

When a positive bias is applied to the anode terminal of the thyristor, the P+ anode/N-drift junction (J1) and the N+ cathode/P-base junction (J3) become forward biased while the junction (J2) between the P-base region and the N-drift region becomes reverse biased. The applied positive bias is mostly supported across the N-drift region. As in the case of reverse-blocking operation, the blocking voltage capability is determined by open-base transistor breakdown rather than avalanche breakdown.

9.2.2 Power Bipolar Junction Transistors

Power Bipolar Junction Transistors (BJTs) play a vital role in power circuits. Like most other power devices, power transistors are generally constructed using silicon. The use of silicon allows operation of a BJT at higher currents and junction temperatures, which leads to the use of power transistors in AC applications where ranges of up to several hundred kilowatts are essential.

The power transistor is part of a family of three-layer devices. The three layers or terminals of a transistor are the base, the collector, and the emitter. Effectively, the transistor is equivalent to having two pn-diode junctions stacked in opposite directions to each other. The two types of a transistor are termed npn and pnp. The npn-type transistor has a higher current-to-voltage rating than the pnp and is preferred for most power conversion applications. The easiest way to distinguish an npn-type transistor from a pnp-type is by virtue of the schematic or circuit symbol. The pnp type has an arrowhead on the emitter that points toward the base. Fig. 9.3 shows the structure and the symbol of a pnp-type transistor. The npn-type transistor has an arrowhead pointing away from the base. Fig. 9.4 shows the structure and the symbol of an npn-type transistor.

Fig. 9.3 pnp transistor structure (a) and circuit symbol (b).

Fig. 9.4 npn transistor structure (a) and circuit symbol (b).

The Volt-Ampere Characteristics of a BJT: The volt-ampere characteristics of a BJT are shown in Fig. 9.5. Power transistors have exceptional characteristics as an ideal switch and they are primarily used as switches. In this type of application, they make use of the common emitter connection shown in Fig. 9.6. The three regions of operation for a transistor that must be taken into consideration are the cutoff, saturation, and the active region. When the base current (IB) is zero, the collector current (IC) is insignificant and the transistor is driven into the cut-off region. The transistor is now in the OFF state. The collectorbase and baseemitter junctions are reverse-biased in the cut-off region or OFF state, and the transistor behaves as an open switch. The base current (IB) determines the saturation current. This occurs when the base current is sufficient to drive the transistor into saturation. During saturation, both junctions are forward-biased and the transistor acts like a closed switch. The saturation voltage increases with an increase in current and is normally between 0.5 to 2.5 V. The active region of the transistor is mainly used for amplifier applications and should be avoided for switching operation. In the active region, the collectorbase junction is reversed-biased and the baseemitter junction is forward-biased.

Fig 9.5 BJT V-I characteristics.

Fig. 9.6 Biasing of a transistor.

BJT Power Losses: The four types of transistor power losses are the ON-state and OFF-state losses and turn-ON and turn-OFF switching loss. OFF-state transistor losses are much lower than ON-state losses since the leakage current of the device is within a few milliamps. Essentially, when a transistor is in the off state, whatever the value of collectoremitter voltage, there is no collector current. Switching losses depend on switching frequency. The highest possible switching frequency of the transistor is limited by the losses due to the rate of switching. In other words, the higher the switching frequency, the more power loss in the transistor.

BJT Testing: Testing of the state of a transistor can be done with a multimeter. When a transistor is forward-biased, the basecollector and baseemitter regions should have a low resistance. When reverse-biased, the basecollector and baseemitter regions should have a high resistance. When testing the resistance between the collector and the emitter, the resistance reading should result in a much higher than forward bias basecollector and baseemitter resistance. However, faulty power transistors can appear shorted when measuring resistance across the collector and emitter, but still pass both junction tests.

9.2.3 Insulated Gate Bipolar Junction Transistor (IGBT)

The introduction of Power MOSFET was originally regarded as a major threat to the power bipolar transistor. However, initial claims of infinite current gain for the power MOSFETs were diluted by the need to design the gate drive circuit capable of supplying the charging and discharging current of the device input capacitance. This is especially true in high frequency circuits where the power MOSFET is particularly valuable due to its inherently high switching speed. On the other hand, MOSFETs have a higher on state resistance per unit area and consequently higher on state loss. This is particularly true for higher voltage devices (greater than about 500 volts) which restricted the use of MOSFETs to low voltage high frequency circuits (eg. SMPS).

With the discovery that power MOSFETs were not in a strong position to displace the BJT, many researches began to look at the possibility of combining these technologies to achieve a hybrid device which has a high input impedance and a low on state resistance. The obvious first step was to drive an output npn BJT with an input MOSFET connected in the Darlington configuration. However, this approach required the use of a high voltage power MOSFET with considerable current carrying capacity (due to low current gain of the output transistor). Also, since no path for negative base current exists for the output transistor, its turn off time also tends to get somewhat larger. An alternative hybrid approach was investigated at GE Research center where a MOS gate structures was used to trigger the latch up of a four layer thyristor. However, this device was also not a true replacement of a BJT since gate control was lost once the thyristor latched up.

After several such attempts it was concluded that for better results MOSFET and BJT technologies are to be integrated at the cell level. This was achieved by the GE Research Laboratory by the introduction of the device IGT and by the RCA research laboratory with the device COMFET. The IGT device has undergone many improvement cycles to result in the modern Insulated Gate Bipolar Transistor (IGBT). These devices have near ideal characteristics for high voltage (> 100V) medium frequency (< 20 kHZ) applications. This device along with the MOSFET (at low voltage high frequency applications) have the potential to replace the BJT completely.

Constructional Features of an IGBT: Vertical cross section of a n channel IGBT cell is shown in Fig 9.7. Although p channel IGBTs are possible n channel devices are more common.

Fig. 9.7 Vertical cross section of an IGBT.The major difference with the corresponding MOSFET cell structure lies in the addition of a p+ injecting layer. This layer forms a pn junction with the drain layer and injects minority carriers into it. The n type drain layer itself may have two different doping levels. The lightly doped n- region is called the drain drift region. Doping level and width of this layer sets the forward blocking voltage (determined by the reverse break down voltage of J2) of the device. However, it does not affect the on state voltage drop of the device due to conductivity modulation as discussed in connection with the power diode. This construction of the device is called Punch Trough (PT) design. The Non-Punch Through (NPT) construction does not have this added n+ buffer layer. The PT construction does offer lower on state voltage drop compared to the NPT construction particularly for lower voltage rated devices. However, it does so at the cost of lower reverse break down voltage for the device, since the reverse break down voltage of the junction J1 is small. The rest of the construction of the device is very similar to that of a vertical MOSFET including the insulated gate structure and the shorted body (p type) emitter (n+ type) structure. The doping level and physical geometry of the p type body region however, is considerably different from that of a MOSFET in order to defeat the latch up action of a parasitic thyristor embedded in the IGBT structure. A large number of basic cells as shown in Fig 9.7 are grown on a single silicon wafer and connected in parallel to form a complete IGBT device.

Fig. 9.8 Parasitic thyristor in an IGBT. (a) Schematic structure (b) Exact equivalent circuit (c) Approximate equivalent circuit

The IGBT cell has a parasitic p-n-p-n thyristor structure embedded into it as shown in Fig. 9.8 (a). The constituent p-n-p transistor, n-p-n transistor and the driver MOSFET are shown by dotted lines in this figure. Important resistances in the current flow path are also indicated.

Fig. 9.8 (b) shows the exact static equivalent circuit of the IGBT cell structure. The top p-n-p transistor is formed by the p+ injecting layer as the emitter, the n type drain layer as the base and the p type body layer as the collector. The lower n-p-n transistor has the n+ type source, the p type body and the n type drain as the emitter, base and collector respectively. The base of the lower n-p-n transistor is shorted to the emitter by the emitter metallization. However, due to imperfect shorting, the exact equivalent circuit of the IGBT includes the body spreading resistance between the base and the emitter of the lower n-p-n transistor. If the output current is large enough, the voltage drop across this resistance may forward bias the lower n-p-n transistor and initiate the latch up process of the p-n-p-n thyristor structure. Once this structure latches up the gate control of IGBT is lost and the device is destroyed due to excessive power loss.

A major effort in the development of IGBT has been towards prevention of latch up of the parasitic thyristor. This has been achieved by modifying the doping level and physical geometry of the body region. Fig.9.9 shows the circuit symbol of an IGBT.

Fig. 9.9 Circuit symbol of an IGBT

Operating principle of an IGBT: Operating principle of an IGBT can be explained in terms of the schematic cell structure and equivalent circuit of Fig. 9.8 (a) and (c). From the input side the IGBT behaves essentially as a MOSFET. Therefore, when the gate emitter voltage is less then the threshold voltage no inversion layer is formed in the p type body region and the device is in the off state. The forward voltage applied between the collector and the emitter drops almost entirely across the junction J2. Very small leakage current flows through the device under this condition. In terms of the equivalent current of Fig. 9.8 (c), when the gate emitter voltage is lower than the threshold voltage the driving MOSFET of the Darlington configuration remains off and hence the output p-n-p transistor also remains off.

When the gate emitter voltage exceeds the threshold, an inversion layer forms in the p type body region under the gate. This inversion layer (channel) shorts the emitter and the drain drift layer and an electron current flows from the emitter through this channel to the drain drift region. This in turn causes substantial hole injection from the p+ type collector to the drain drift region. A portion of these holes recombine with the electrons arriving at the drain drift region through the channel. The rest of the holes cross the drift region to reach the p type body where they are collected by the source metallization.

From the above discussion it is clear that the n type drain drift region acts as the base of the output p-n-p transistor. The doping level and the thickness of this layer determines the current gain of the p-n-p transistor. This is intentionally kept low so that most of the device current flows through the MOSFET and not the output p-n-p transistor collector. This helps to reduced the voltage drop across the body spreading resistance shown in Fig. 9.8 (b) and eliminate the possibility of static latch up of the IGBT.

The total on state voltage drop across a conducting IGBT has three components. The voltage drop across J1 follows the usual exponential law of a pn junction. The next component of the voltage drop is due to the drain drift region resistance. This component in an IGBT is considerably lower compared to a MOSFET due to strong conductivity modulation by the injected minority carriers from the collector. This is the main reason for reduced voltage drop across an IGBT compared to an equivalent MOSFET. The last component of the voltage drop across an IGBT is due to the channel resistance and its magnitude is equal to that of a comparable MOSFET.

Steady state characteristics of an IGBT: The i-v characteristics of an n channel IGBT is shown in Fig. 9.10 (a). They appear qualitatively similar to those of a logic level BJT except that the controlling parameter is not a base current but the gate-emitter voltage.

Fig. 9.10 Static characteristics of an IGBT (a) Output characteristics; (b) Transfer characteristics

When the gate emitter voltage is below the threshold voltage only a very small leakage current flows though the device while the collector emitter voltage almost equals the supply voltage (point C in Fig. 9.10(a)). The device, under this condition is said to be operating in the cut off region. The maximum forward voltage the device can withstand in this mode (marked VCES in Fig. 9.10 (a)) is determined by the avalanche break down voltage of the body drain p-n junction. Unlike a BJT, however, this break down voltage is independent of the collector current as shown in Fig. 9.10(a). IGBTs of Non-punch through design can block a maximum reverse voltage (VRM) equal to VCES in the cut off mode. However, for Punch Through IGBTs VRM is negligible (only a few tens of volts) due the presence of the heavily doped n+ drain buffer layer.

As the gate emitter voltage increases beyond the threshold voltage the IGBT enters into the active region of operation. In this mode, the collector current ic is determined by the transfer characteristics of the device as shown in Fig. 9.10(b). This characteristic is qualitatively similar to that of a power MOSFET and is reasonably linear over most of the collector current range. The ratio of ic to (VgE vgE(th)) is called the forward transconductance (gfs) of the device and is an important parameter in the gate drive circuit design. The collector emitter voltage, on the other hand, is determined by the external load line ABC as shown in Fig. 9.10(a).

As the gate emitter voltage is increased further ic also increases and for a given load resistance (RL) vCE decreases. At one point vCE becomes less than vgE vgE(th). Under this condition, the driving MOSFET part of the IGBT enters into the ohmic region and drives the output p-n-p transistor to saturation. Under this condition the device is said to be in the saturation mode. In the saturation mode the voltage drop across the IGBT remains almost constant reducing only slightly with increasing vgE.

In power electronic applications an IGBT is operated either in the cut off or in the saturation region of the output characteristics. Since vCE decreases with increasing vgE, it is desirable to use the maximum permissible value of vgE in the ON state of the device. vgE(Max) is limited by the maximum collector current that should be permitted to flow in the IGBT as dictated by the latch-up condition discussed earlier. Limiting VgE also helps to limit the fault current through the device. If a short circuit fault occurs in the load resistance RL , the fault load line is given by CF. Limiting vgE to vgE6 restricts the fault current corresponding to the operating point F. Most IGBTs are designed to with stand this fault current for a few microseconds within which the device must be turned off to prevent destruction of the device.

IGBT does not exibit a BJT-like second break down failure. Since, in an IGBT most of the collector current flows through the drive MOSFET with positive temperature coefficient the effective temperature coefficient of vCE in an IGBT is slightly positive. This helps to prevent second break down failure of the device and also facilitates paralleling of IGBTs.9.2.4 Power MOSFET

Power MOSFETs are marketed by different manufacturers with differences in internal geometry and with different names such as MegaMOS, HEXFET, SIPMOS, and TMOS. They have unique features that make them potentially attractive for switching applications. They are essentially voltage-driven rather than current-driven devices, unlike bipolar transistors.

The gate of a MOSFET is isolated electrically from the source by a layer of silicon oxide. The gate draws only a minute leakage current of the order of nanoamperes. Hence the gate drive circuit is simple and power loss in the gate control circuit is practically negligible. Although in steady state the gate draws virtually no current, this is not so under transient conditions. The gate-to-source and gate-to-drain capacitances have to be charged and discharged appropriately to obtain the desired switching speed, and the drive circuit must have a sufficiently low output impedance to supply the required charging and discharging currents. The circuit symbol of a power MOSFET is shown in Fig. 9.11.

Fig. 9.11 Power MOSFET circuit symbol. (Source: B.K. Bose, Modern Power Electronics: Evaluation, Technology, and Applications, p. 7. 1992 IEEE.)

Power MOSFETs are majority carrier devices, and there is no minority carrier storage time. Hence they have exceptionally fast rise and fall times. They are essentially resistive devices when turned on, while bipolar transistors present a more or less constant VCE(sat) over the normal operating range. Power dissipation in MOSFETs is Id2RDS(on).

At low currents, therefore, a power MOSFET may have a lower conduction loss than a comparable bipolar device, but at higher currents, the conduction loss will exceed that of BJTs. Also, the RDS(on) increases with temperature. An important feature of a power MOSFET is the absence of a secondary breakdown effect, which is present in a bipolar transistor, and as a result, it has an extremely rugged switching performance. In MOSFETs, RDS(on) increases with temperature, and thus the current is automatically diverted away from the hot spot. The drain body junction appears as an anti-parallel diode between source and drain. Thus power MOSFETs will not support voltage in the reverse direction. Although this inverse diode is relatively fast, it is slow by comparison with the MOSFET.

9.2.5 DIAC

A DIAC is a three-layer, low-voltage, low-current semiconductor switch. The DIAC symbol is shown in Fig. 9.12(a). The DIAC structure is shown in Fig. 9.12(b). The DIAC can be switched from the OFF to the ON state for either polarity of applied voltage.

Fig. 9.12 (a) The DIAC symbol; (b) the DIAC structureThe volt-ampere characteristic of a DIAC is shown in Fig. 9.13. When Anode-1 is made more positive than Anode 2, a small leakage current flow until the break-over voltage VBO is reached. Beyond VBO, the DIAC will conduct. When Anode-2 is made more positive relative to Anode-1, a similar phenomenon occurs. The break-over voltages for the DIAC are almost the same in magnitude in either direction. DIACs are commonly used to trigger larger thyristors such as SCRs and TRIACs.

Fig. 9.13 The DIAC characteristics.9.2.6 TRIAC

The TRIAC is a member of the thyristor family. But unlike a thyristor which conducts only in one direction (from anode to cathode) a TRIAC can conduct in both directions. Thus a TRIAC is similar to two back to back (anti parallel) connected thyristors but with only three terminals. As in the case of a thyristor, the conduction of a TRIAC is initiated by injecting a current pulse into the gate terminal. The gate looses control over conduction once the TRIAC is turned on. The TRIAC turns off only when the current through the main terminals become zero. Therefore, a TRIAC can be categorized as a minority carrier, a bidirectional semi-controlled device. They are extensively used in residential lamp dimmers, heater control and for speed control of small single phase series and induction motors.

Fig. 9.14 Circuit symbol and schematic construction of a TRIAC(a) Circuit symbol (b) Schematic constructionFig. 9.14 (a) and (b) show the circuit symbol and schematic cross section of a TRIAC respective. As the TRIAC can conduct in both the directions the terms anode and cathode are not used for TRIACs. The three terminals are marked as MT1 (Main Terminal 1), MT2 (Main Terminal 2) and the gate by G. As shown in Fig 9.14 (b) the gate terminal is near MT1 and is connected to both N3 and P2 regions by metallic contact. Similarly MT1 is connected to N2 and P2 regions while MT2 is connected to N4 and P1 regions.

Since a TRIAC is a bidirectional device and can have its terminals at various combinations of positive and negative voltages, there are four possible electrode potential combinations as given below:

1. MT2 positive with respect to MT1, G positive with respect to MT1 2. MT2 positive with respect to MT1, G negative with respect to MT1 3. MT2 negative with respect to MT1, G negative with respect to MT1 4. MT2 negative with respect to MT1, G positive with respect to MT1 The triggering sensitivity is highest with the combinations 1 and 3 and are generally used. However, for bidirectional control and uniforms gate trigger mode sometimes trigger modes 2 and 3 are used. Trigger mode 4 is usually avoided. Fig 9.15 (a) and (b) explain the conduction mechanism of a TRIAC in trigger modes 1 & 3 respectively.

In trigger mode-1 the gate current flows mainly through the P2 N2 junction like an ordinary thyristor. When the gate current has injected sufficient charge into P2 layer the TRIAC starts conducting through the P1 N1 P2 N2 layers like an ordinary thyristor.

Fig. 9.15 Conduction mechanism of a TRIAC in trigger modes 1 and 3(a) Mode 1 , (b) Mode 3 .In the trigger mode-3 the gate current Ig forward biases the P2 P3 junction and a large number of electrons are introduced in the P2 region by N3. Finally the structure P2 N1 P1 N4 turns on completely.

Steady State Output Characteristics of TRIAC: From a functional point of view a TRIAC is similar to two thyristors connected in anti parallel. Therefore, it is expected that the V-I characteristics of TRIAC in the 1st and 3rd quadrant of the V-I plane will be similar to the forward characteristics of a thyristors. As shown in Fig. 9.16, with no signal to the gate the triac will block both half cycle of the applied ac voltage provided its peak value is lower than the break over voltage (VBO) of the device. However, the turning on of the triac can be controlled by applying the gate trigger pulse at the desired instance. Mode-1 triggering is used in the first quadrant where as Mode-3 triggering is used in the third quadrant. As such, most of the thyristor characteristics apply to the triac (ie, latching and holding current). However, in a triac the two conducting paths (from MT1 to MT2 or from MT1 to MT1) interact with each other in the structure of the triac. Therefore, the voltage, current and frequency ratings of triacs are considerably lower than thyristors. At present triacs with voltage and current ratings of 1200V and 300A (rms) are available. TRIACs also have a larger on state voltage drop compared to a thyristor.

Fig. 9.16 Steady state V I characteristics of a TRIAC9.2.7 Uni-Junction Transistor

The unijunction transistor (UJT) is often described as a voltage-controlled diode. This device is not considered to be an amplifying device. It is, however, classified as a unipolar transistor. The UJT and an N-channel JFET are often confused because of the similarities in their schematic symbols and crystal construction. The operation and function of each device is entirely different.

Fig. 9.17 Unijunction transistor.Fig. 9.17 shows the crystal construction, schematic symbol, and element names of a UJT. Note that the UJT is a three-terminal, single-junction device. The crystal is an N-type bar of silicon with contacts at each end. The end connections are called Base-1 (B1) and Base-2 (B2). A small, heavily doped P region is alloyed to one side of the silicon bar. This serves as the emitter (E). A PN junction is formed by the emitter and the silicon bar.

The arrow of the symbol Points in N, which means that the emitter is P material and the silicon bar is N material. The arrow of the symbol is slanted, which distinguishes it from the N-channel JFET. An inter-base resistance (RBB) exists between B1 and B2. Typically, RBB is between 4 and 10 k . This value can be easily measured with an ohmmeter. The resistance of the silicon bar is represented by RBB. This resistance can be divided into two values. RB1 is between the emitter and B1. RB2 is between the emitter and B2. Normally, RB2 is somewhat less than RB1. The emitter is usually closer to B2 than B1. When a UJT is made operational, the value of RB1 will change with different emitter voltages.

In circuit applications B1 is usually placed at circuit ground or the source voltage negative. The emitter then serves as the input to the device. B2 provides circuit output. A change in EB1 voltage will cause a change RBB. The output current of the device will increase when E B1 turns on. A UJT is normally used as a trigger device. It is used to control some other solid-state device. No amplification is achieved by a UJT.

Fig. 9.18 Characteristic curve for a UJT

A characteristic curve for a typical UJT is shown in Fig. 9.18. The vertical part of this graph shows the emitter voltage as VE. An increase in V causes the curve to rise vertically. The horizontal part of the graph shows the emitter current (IE). Note that the curve has peak voltage (VP) and valley voltage (VV) points. An increase in IE causes VP to rise until it reaches the peak point. A further increase in IE will cause the emitter voltage to drop to the valley point. This condition is called the negative resistance region. A device that has a negative resistance characteristic is capable of regeneration or oscillation.

In operation, when the emitter voltage reaches the peak voltage point, EB1 becomes forward biased. This condition draws holes and electrons to the P-N junction. Holes are injected into the B1 region. This action causes B1 to be more conductive. As a result, the EB1 region become low resistant. A sudden drop in RB1 will cause a corresponding increase in B1-B2 current. This change in current can be used to trigger or turn on other devices. The trigger voltage of a UJT is a predictable value. Knowing this value will permit the device to be used as a control element.

9.2.8 Comparison between IGBT and MOSFET

IGBTMOSFET

Terminals are Gate, Emitter, Collector

High Input impedance

Voltage controlled devices

It will not increases

Designed for higher voltage ratings Terminals are Gate, Source and drain

High Input impedance

Voltage controlled devices

On-state voltage drop and losses increases with rise in temperature

Voltage drop increases with rise in voltage so its voltage rating is limited

9.3 Power Conversion

Power conversion deals with the process of converting electric power from one form to another. The power electronic apparatuses performing the power conversion are called power converters. Because they contain no moving parts, they are often referred to as static

power converters. The power conversion is achieved using power semiconductor devices, which are used as switches. The power devices used are SCRs (Silicon Controlled Rectifiers, or thyristors), TRIACs, power transistors, power MOSFETs, insulated gate bipolar transistors (IGBTs) and MCTs (MOS-controlled thyristors). The power converters are generally classified as:

a. ac-dc converters (phase-controlled converters)

b. direct ac-ac converters (cycloconverters)

c. dc-ac converters (inverters)

d. dc-dc converters (choppers, buck and boost converters)

9.3.1 AC-DC Converters

In power electronics, AC to DC converters are often called as phase-controlled converter. The basic function of a phase-controlled converter is to convert an alternating voltage of variable amplitude and frequency to a variable dc voltage. The power devices used for this application are generally SCRs. The average value of the output voltage is controlled by varying the conduction time of the SCRs. The turn-on of the SCR is achieved by providing a gate pulse when it is forward-biased. The turn-off is achieved by the commutation of current from one device to another at the instant the incoming ac voltage has a higher instantaneous potential than that of the outgoing wave. Thus there is a natural tendency for current to be commutated from the outgoing to the incoming SCR, without the aid of any external commutation circuitry. This commutation process is often referred to as natural commutation.

A single-phase half-wave converter is shown in Fig. 9.19. When the SCR is turned on at an angle , full supply voltage (neglecting the SCR drop) is applied to the load. For a purely resistive load, during the positive half cycle, the output voltage waveform follows the input ac voltage waveform. During the negative half cycle, the SCR is turned off. In the case of inductive load, the energy stored in the inductance causes the current to flow in the load circuit even after the reversal of the supply voltage, as shown in Fig. 9.19(b). If there is no freewheeling diode DF, the load current is discontinuous. A freewheeling diode is connected across the load to turn off the SCR as soon as the input voltage polarity reverses, as shown in Fig. 9.19(c). When the SCR is off, the load current will freewheel through the diode. The power flows from the input to the load only when the SCR is conducting. If there is no freewheeling diode, during the negative portion of the supply voltage, SCR returns the energy stored in the load inductance to the supply. The freewheeling diode improves the input power factor.

Fig. 9.19 Single-phase half-wave converter with freewheeling diode.(a) Circuit diagram; (b) waveform for inductive load with no freewheeling diode;(c) waveform with freewheeling diode.

The controlled full-wave dc output may be obtained by using either a center-tap transformer (Fig. 9.20) or by bridge configuration (Fig. 9.21). The bridge configuration is often used when a transformer is undesirable and the magnitude of the supply voltage properly meets the load voltage requirements. The average output voltage of a single-phase full-wave converter for continuous current conduction is given by:

Vdc = 2(Em/) Cos

Where Em is the peak value of the input voltage and is the firing angle. The output voltage of a single-phase bridge circuit is the same as that shown in Fig. 9.20. Various configurations of the single-phase bridge circuit can be obtained if, instead of four SCRs, two diodes and two SCRs are used, with or without freewheeling diodes.

Fig. 9.20 Single-phase full-wave converter with transformer.

Fig. 9.21 Single-phase bridge converter.

9.3.2 AC- AC convertersAC-AC converters are frequency converters. They produce an ac voltage in which both the frequency and voltage can be varied directly from the ac line voltage, e.g., from a 60- or 50-Hz source.Cycloconverters: Cycloconverters are direct ac-to-ac frequency changers. The term direct conversion means that the energy does not appear in any form other than the ac input or ac output. The output frequency is lower than the input frequency and is generally an integral multiple of the input frequency. A cycloconverter permits energy to be fed back into the utility network without any additional measures. Also, the phase sequence of the output voltage can be easily reversed by the control system. Cycloconverters have found applications in aircraft systems and industrial drives. These Cycloconverters are suitable for synchronous and induction motor control.

Fig. 9.22 Cycloconverter control of an induction motor.

Fig. 9.22 shows a drive using a three-pulse half-wave or 18-thyristor cycloconverter. Each output phase group consists of positive and negative converter components which permit bidirectional current flow. The firing angle of each converter is sinusoidally modulated to generate the variable-frequency, variable-voltage output required for ac machine drive. Speed reversal and regenerative mode operation are easy. The cycloconverter can be operated in blocking or circulating current mode. In blocking mode, the positive or negative converter is enabled, depending on the polarity of the load current. In circulating current mode, the converter components are always enabled to permit circulating current through them. The circulating current reactor between the positive and negative converter prevents short circuits due to ripple voltage. The circulating current mode gives simple control and a higher range of output frequency with lower harmonic distortion.

9.3.3 DC-to-AC Converters

The dc-to-ac converters are generally called inverters. The ac supply is first converted to dc, which is then converted to a variable-voltage and variable-frequency power supply. This generally consists of a three-phase bridge connected to the ac power source, a dc link with a filter, and the three-phase inverter bridge connected to the load. In the case of battery-operated systems, there is no intermediate dc link. Inverters can be classified as Voltage Source Inverters (VSIs) and Current Source Inverters (CSIs). A voltage source inverter is fed by a stiff dc voltage, whereas a current source inverter is fed by a stiff current source. A voltage source can be converted to a current source by connecting a series inductance and then varying the voltage to obtain the desired current. Fig. 9.23 shows a Three-phase thyristor full bridge configuration along with its associated waveforms.

Fig. 9.23 (a) Three-phase thyristor full bridge configuration;(b) Output voltage and current waveforms.

Fig. 9.24 (a) Three-phase converter and voltage source inverter configuration;(b) three-phase square-wave inverter waveforms.

A VSI can also be operated in current-controlled mode, and similarly a CSI can also be operated in the voltage-control mode. The inverters are used in variable frequency ac motor drives, uninterrupted power supplies, induction heating, static VAR compensators, etc.

Voltage Source Inverter: A three-phase voltage source inverter configuration is shown in Fig. 9.24(a). The VSIs are controlled either in square-wave mode or in pulsewidth-modulated (PWM) mode. In square-wave mode, the frequency of the output voltage is controlled within the inverter, the devices being used to switch the output circuit between the plus and minus bus. Each device conducts for 180 degrees, and each of the outputs is displaced 120 degrees to generate a six-step waveform, as shown in Fig. 30.13(b). The amplitude of the output voltage is controlled

by varying the dc link voltage. This is done by varying the firing angle of the thyristors of the three-phase bridge converter at the input. The square-wave-type VSI is not suitable if the dc source is a battery. The six-step output voltage is rich in harmonics and thus needs heavy filtering. In PWM inverters, the output voltage and frequency are controlled within the inverter by varying the width of the output pulses. Hence at the front end, instead of a phase-controlled thyristor converter, a diode bridge rectifier can be used. A very popular method of controlling the voltage and frequency is by sinusoidal pulsewidth modulation. In this method, a high-frequency triangle carrier wave is compared with a three-phase sinusoidal waveform, as shown in Fig. 9.25. The power devices in each phase are switched on at the intersection of sine and triangle waves. The amplitude and frequency of the output voltage are varied, respectively, by varying the amplitude and frequency of the reference sine waves. The ratio of the amplitude of the sine wave to the amplitude of the carrier wave is called the modulation index.

Fig. 9.25 Three-phase sinusoidal PWM inverter waveforms.

The harmonic components in a PWM wave are easily filtered because they are shifted to a higher-frequency region. It is desirable to have a high ratio of carrier frequency to fundamental frequency to reduce the harmonics of lower-frequency components. The most notable ones PWM are selected harmonic elimination, hysteresis controller, and space vector PWM technique.

In inverters, if SCRs are used as power switching devices, an external forced commutation circuit has to be used to turn off the devices. Now, with the availability of IGBTs above 1000-A, 1000-V ratings, they are being used in applications up to 300-kW motor drives. Above this power rating, GTOs are generally used. Power Darlington transistors, which are available up to 800A, 1200 V, could also be used for inverter applications.

Current Source Inverter: Contrary to the voltage source inverter where the voltage of the dc link is imposed on the motor windings, in the current source inverter the current is imposed into the motor. Here the amplitude and phase angle of the motor voltage depend on the load conditions of the motor.

Resonant-Link Inverters: The use of resonant switching techniques can be applied to inverter topologies to reduce the switching losses in the power devices. They also permit high switching frequency operation to reduce the size of the magnetic components in the inverter unit. In the resonant dc-link inverter shown in Fig. 9.26, a resonant circuit is added at the inverter input to convert a fixed dc voltage to a pulsating dc voltage. This resonant circuit enables the devices to be turned on and turned off during the zero voltage interval. Zero voltage or zero current switching is often termed soft switching. Under soft switching, the switching losses in the power devices are almost eliminated. The electromagnetic interference (EMI) problem is less severe because resonant voltage pulses have lower dv/dt compared to those of hard-switched PWM inverters. Also, the machine insulation is less stretched because of lower dv/dt resonant voltage pulses. In Fig. 9.26, all the inverter devices are turned on simultaneously to initiate a resonant cycle. The commutation from one device to another is initiated at the zero dc-link voltage. The inverter output voltage is formed by the integral numbers of quasi-sinusoidal pulses.

The circuit consisting of devices Q, D, and the capacitor C acts as an active clamp to limit the dc voltage to about 1.4 times the diode rectifier voltage Vs. There are several other topologies of resonant link inverters mentioned in the literature. There are also resonant link ac-ac converters based on bidirectional ac switches, as shown in Fig. 9.27. These resonant link converters find applications in ac machine control and uninterrupted power supplies, induction heating, etc. The resonant link inverter technology is still in the development stage for industrial applications.

Fig. 9.26 Resonant dc-link inverter system with active voltage clamping.

Fig. 9.27 Resonant ac-link converter system showing configuration of ac switches.

9.3.4 DC-DC Converters

DC-dc converters are used to convert unregulated dc voltage to regulated or variable dc voltage at the output. They are widely used in switch-mode dc power supplies and in dc motor drive applications. In dc motor control applications, they are called chopper-controlled drives. The input voltage source is usually a battery or derived from an ac power supply using a diode bridge rectifier. These converters are generally either hard-switched PWM types or soft-switched resonant-link types. There are several dc-dc converter topologies, the most

common ones being buck converter, boost converter, and buck-boost converter, shown in Fig. 9.28.

Buck Converter: A buck converter is also called a step-down converter. Its principle of operation is illustrated by referring to Fig. 9.28(a). The IGBT acts as a high-frequency switch. The IGBT is repetitively closed for a time ton and opened for a time toff. During ton, the supply terminals are connected to the load, and power flows from supply to the load. During toff, load current flows through the freewheeling diode D1, and the load voltage is ideally zero. The average output voltage is given by: Vout = D.Vinwhere D is the duty cycle of the switch and is given by D = ton/T, where T is the time for one period. 1/T is the switching frequency of the power device IGBT.

Fig. 9.28 DC-DC converter configurations: (a) buck converter; (b) boost converter; (c) buck-boost converter.

Boost Converter: A boost converter is also called a step-up converter. Its principle of operation is illustrated by referring to Fig. 9.28(b). This converter is used to produce higher voltage at the load than the supply voltage. When the power switch is on, the inductor is connected to the dc source and the energy from the supply is stored in it. When the device is off, the inductor current is forced to flow through the diode and the load. The induced voltage across the inductor is negative. The inductor adds to the source voltage to force the inductor current into the load. The output voltage is given by:

Vout = Vin / (1-D)

Thus for variation of D in the range 0 < D < 1, the load voltage Vout will vary in the range Vin < Vout