951 32. Design, Fabrication and Control of …...951 32. Design, Fabrication and Control of MicroactuatorsDesign, Fabric for Dual-Stage Servo Systems in Magnetic Disk Files This chapter

951

Design, Fabric32. Design, Fabrication and Control of Microactuatorsfor Dual-Stage Servo Systems in Magnetic Disk Files

This chapter discusses the design and fabricationof electrostatic MEMS microactuators and thedesign of dual-stage servo systems in diskdrives. It introduces fundamental requirements ofdisk drive servo systems, along with challengesposed as storage densities increase. It describesthree potential dual-stage configurations andfocuses on actuated slider assemblies usingelectrostatic MEMS microactuators. The authorsdiscuss major electrostatic actuator design issues,such as linear versus rotary motion, electrostaticarray configuration, and differential operation.Capacitive and piezoresistive elements may be usedto sense relative slider position, while integratedgimbals and structural isolation may prove usefulin improving performance. A detailed designexample based on a translational, micromoldedactuator illustrates several of these concepts andis accompanied by theoretical and experimentalresults.

The chapter continues to discuss MEMSmicroactuator fabrication. It describes severalprocesses for obtaining appropriate electrostaticdevices including micromolding, deep-reactiveion etching, and electroplating. The primary goalof these processes is to obtain very high-aspectratio structures, which improve both actuationforce and structural robustness. Other fabricationissues, such as electrical interconnect formation,material selection, and processing cost, are alsoconsidered. Actuated slider fabrication is comparedto that of actuated suspension and actuated headassemblies; this includes instrumentation ofa suspension with strain sensors to aid in vibrationdetection.

The section on controller design reviewsdual-stage servo control design architectures andmethodologies. The considerations for controllerdesign of MEMS microactuator dual-stage servosystems are discussed. The details of controldesigns using a decoupled SISO design methodand the robust multivariable design method ofµ-synthesis are also presented. In the decoupleddesign, a self-tuning control algorithm has been

32.1 Design of the Electrostatic Microactuator 95232.1.1 Disk Drive Structural Requirements . 95232.1.2 Dual-Stage Servo Configurations .... 95332.1.3 Electrostatic Microactuators:

Comb-Drives vs. Parallel-Plates ..... 95432.1.4 Position Sensing .......................... 95632.1.5 Electrostatic Microactuator Designs

for Disk Drives.............................. 958

32.2 Fabrication .......................................... 96232.2.1 Basic Requirements ...................... 96232.2.2 Electrostatic Microactuator

Fabrication Example ..................... 96232.2.3 Electrostatic Microactuator Example

Two ............................................ 96332.2.4 Other Fabrication Processes ........... 96632.2.5 Suspension-Level

Fabrication Processes.................... 96732.2.6 Actuated Head Fabrication ............ 968

32.3 Servo Control Design of MEMSMicroactuator Dual-Stage Servo Systems . 96832.3.1 Introduction

to Disk Drive Servo Control ............. 96932.3.2 Overview of Dual-Stage Servo

Control Design Methodologies ........ 96932.3.3 Track-Following Controller Design

for a MEMS MicroactuatorDual-Stage Servo System ............... 972

32.3.4 Dual-Stage Seek Control Design ..... 978

32.4 Conclusions and Outlook ....................... 979

References .................................................. 980

developed to compensate for the variations inthe microactuator’s resonance mode. In theµ-synthesis design, robust controllers can besynthesized by using additive and parametricuncertainties to characterize the un-modeleddynamics of the VCM and the variations in themicroactuator’s resonance mode. The chapteralso introduces a dual-stage short-span seekcontrol scheme based on decoupled feed forwardreference trajectories generations.

TS0 The running title is too long. Please provide a shortened version for the chapter and/or section title.

Editor’s or typesetter’s annotations (will be removed before the final TEX run)

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32

952 Part E Industrial Applications and Microdevice Reliability

This chapter discusses the design and fabrication ofelectrostatic MEMS microactuators, and the design ofdual-stage servo systems, in disk drives. The focus ofthe chapter is an actuated slider assembly using anelectrostatic MEMS microactuator. We discuss majordesign issues, including linear versus rotary actuation,electrostatic array configuration, and integrated sensingcapability. We describe several fabrication processes forobtaining the necessary devices, such as micromolding,deep-reactive ion etching, and electroplating. Dual-stageservo control design architectures and methodologiesare reviewed as well. We also present in detail track-following controller designs based on a sensitivityfunction decoupling single-input-single-output designmethodology and the robust µ-synthesis design method-ology. Finally we introduce a 2-DOF short span seekcontrol design using dual-stage actuator.

Since the first hard disk drive (HDD) was inventedin the 1950s by IBM, disk drives’ storage density hasbeen following Moore’s law, doubling roughly every18 months. The current storage density is 10 milliontimes larger than that of the first HDD [32.1]. Historic-ally, increases in storage density have been achieved byalmost equal increases in track density, the number oftracks encircling the disk, and bit density, the numberof bits in each track. But, because of superparamag-netism limitations, future areal storage density increasesin HDDs are predicted to be achieved mainly through anincrease in track density [32.2].

Research in the HDD industry is now targeting anareal density of one terabit per square inch. For a pre-dicted bit aspect ratio of 4 : 1, this translates to a linearbit density of 2 M bits per inch (BPI) and a radial trackdensity of 500 K tracks per inch (TPI), which in turn im-plies a track width of 50 nm. A simple rule of thumbfor servo design in HDDs is that three times the statisti-cal standard deviation of the position error between thehead and the center of the data track should be less than1/10 of the track width. Thus, to achieve such a storage

density, nanometer-level precision of the servo systemwill be required.

A disk drive stores data as magnetic patterns, form-ing bits, on one or more disks. The polarity of each bitis detected (read) or set (written) by an electromagneticdevice known as the read/write head. The job of a diskdrive’s servo system is to position the read/write headover the bits to be read or written as they spin by onthe disk. In a conventional disk drive, this is done bysweeping over the disk a long arm consisting of a voicecoil motor (VCM), an E-block, suspensions, and slid-ers, as shown in Figs. 32.1 and 32.2. A read/write headis fabricated on the edge of each slider (one for eachdisk surface). Each slider is supported by a suspensionand flys over the surface of disk on an air-bearing. TheVCM actuates the suspensions and sliders about a pivotin the center of the E-block. We describe this operationin more detail in the following section.

The key to increasing HDD servo precision is toincrease servo control bandwidth. But, the bandwidthof a traditional single-stage servo system as shown inFig. 32.2 is limited by the multiple mechanical resonancemodes of the pivot, the E-block, and the suspension be-tween the VCM and the head. Nonlinear friction ofthe pivot bearing also limits achievable servo preci-sion. Dual-stage actuation, with a second stage actuatorplaced between the VCM and the head, has been pro-posed as a solution that would increase servo bandwidthand precision.

Several different secondary actuation forces and con-figurations have been proposed, each having strengthsand weaknesses given the requirements of HDDs. Thedual-stage configurations can be categorized into threegroups: actuated suspension, actuated slider, and ac-tuated head. Within these, actuation forces includepiezoelectric, electrostatic, and electromagnetic. In thischapter, we discuss design, fabrication and control of anelectrostatic MEMS microactuator (MA) for actuatedslider dual-stage positioning.

32.1 Design of the Electrostatic Microactuator

The servo system of a hard disk drive is the mechatronicdevice that locates and reads data on the disk. In essence,it is a large arm that sweeps across the surface of the disk.At the end of the arm is the read-write head, containingthe magnetic reading and writing elements that transferinformation to and from the disk.

32.1.1 Disk Drive Structural Requirements

This read-write head is contained in a box-like structureknown as a slider. The slider has a contoured lowersurface that acts as an air bearing between the headand the disk. The high-velocity airflow generated by

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TS0 32.1 Design of the Electrostatic Microactuator 953

the spinning disk pushes up on the air-bearing surface,maintaining the slider and read-write head at a constantdistance from the disk, despite unevenness of the disk,permitting reliable data reading and writing.

The arm over an HDD’s disk has three primarystages: the voice-coil motor (VCM), the E-block, andthe suspension. In a conventional disk drive, the VCMperforms all positioning of the head, swinging it backand forth across the disk. The E-block lies between theVCM and the suspension and contains the pivot point.The suspension projects from the E-block over the diskas a thin flexible structure, generally narrowing to a pointat the location of the slider.

For a disk drive servo to operate effectively, it mustmaintain the read-write head at a precise height abovethe disk surface and within a narrow range betweenthe disk tracks arranged in concentric circles aroundthe disk. It must also be able to seek from one trackon the disk to another. Information about the track thatthe head is following is encoded in sectors radiatingfrom center of the disk, allowing the head to identifyits position and distance from the center of the track.To maintain a correct flying height, the suspension mustbe designed with an appropriate stiffness in the verticaldirection to balance an air-bearing force correspondingto the slider design in use in the drive. Meanwhile, thesuspension must be flexible to roll and pitch at the sliderlocation to permit adaptation to unevenness of the disksurface. This is accomplished using a gimbal structure.The suspension as a whole, however, should not bend ortwist during the operation, as this would misdirect thehead away from the track it is following.

As data densities in HDDs increase and track widthsdiminish, single-stage, conventional servo systems be-come less able to position the head precisely. Because theVCM/E-block/suspension assembly is large and massiveas a unit, the speed at which the head can be controlled islimited. Furthermore, the assembly tends to have a lownatural frequency, which can accentuate vibration in thedisk drive and cause off-track errors. At track densitiesapproaching one Terabit per square inch in the future,the vibration induced by airflow in a disk drive alone isenough to force the head off-track.

A solution to these problems is to complementthe VCM with a smaller, secondary actuator to forma dual-stage servo system. The VCM continues to pro-vide rough positioning, while the second stage actuatordoes fine positioning and damps out vibration and otherdisturbances. The smaller second stage actuator cantypically be designed to have a much higher naturalfrequency and less susceptibility to vibration than the

Head Voice coilmotor(VCM)

Data track

Disk Spindlemotor

Fig. 32.1 A schematic diagram of an HDD

Suspension

VCM

Pivot

E-block Head

Fig. 32.2 VCMactuator in anHDD

VCM. Any actuator used in a dual-stage system shouldbe inexpensive to build, require little power to operate,and preserve the stiffness properties described abovenecessary to preserve the flying height.

32.1.2 Dual-Stage Servo Configurations

In the past six years, much research work has been ded-icated to the exploration of suitable secondary actuatorsfor constructing dual-stage servo systems for HDDs.These dual-stage configurations can be categorized intothree groups: “actuated suspension,” “actuated slider,”and “actuated head.”

Actuated SuspensionIn this approach, the suspension is redesigned to accom-modate an active component, typically a piezoelectricmaterial. This piezoelectric material stretches or flexesthe suspension to position the slider and magnetic head.Piezoelectric material is an active actuation elementthat produces a large actuation force but small actua-tion stroke. In the “actuated-suspension” configuration,therefore, the piezoelectric actuators are usually im-plemented in a leverage mechanism that can convertsmall actuation displacements into large head displace-ments. Typically this is done by placing the piezoelectric

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actuators away from the magnetic head (between theE-block and suspension) so that they can have a longleverage arm to gain mechanical amplification andproduce a sufficient magnetic-head motion. The ad-vantage of this approach is that the suspension can befabricated by a conventional suspension-making pro-cess, and its dual-stage servo configuration is effectivein attaining low frequency runout attenuation in thepositioning servo loop. The major drawback of thisapproach is that the system is still susceptible to in-stabilities due to the excitation of suspension resonancemodes. Thus, track-per-inch (TPI) servo performancecan be increased but remains limited when comparedto the next two approaches. This approach, however,is expected to be the first deployed in commercialHDDs [32.3, 4].

Actuated SliderIn this approach, a microactuator is placed between theslider and gimbal to position the slider/magnetic heads.The resulting servo bandwidth can be higher than theprevious approach because the secondary actuation by-passes the mechanical resonances of the suspension.This approach uses existing sliders and microactuatorsthat can be batch fabricated, and thus could be costeffective. But, the size and mass of the microactuatorare significant relative to those of current sliders andmay interfere with the slider flying stability. Currentsuspensions, therefore, need to be redesigned to adoptthis secondary actuator. Suitable driving forces in thisapproach include electrostatic, electromagnetic, andpiezoelectric [32.5–8]. To further reduce the assemblytask of placing the microactuator in between gimbalstructure and slider, some researchers have proposedmicroactuators that are either integrated with the gimbalstructure [32.6] or the slider [32.9].

Actuated HeadIn this approach, the slider is redesigned so that themicroactuators can be placed inside the slider block andactuate the magnetic heads with respect to the rest ofthe slider body. As these microactuators are very small,they only slightly increase the slider weight and arethus capable of working with the current suspension as-sembly. Researchers have successfully demonstrated theintegrated fabrication process for fabricating the elec-trostatic microactuators and magnetic heads within onepiece of ceramic block (slider). The embedded electro-static microactuator has its resonance close to 30 kHzand was able to position the magnetic heads relative tothe rest of the slider body by 0.5 µm [32.10, 11]. Full-

fledged integration of slider, actuator, and read/writehead remains a challenge.

In this chapter, we focus on actuated slider con-figurations, as they involve a great deal of interestingmicroscale engineering. In particular, we will dis-cuss electrostatic actuation, probably the most commonmethod of implementing microactuation in micro-electromechanical devices. Any such microactuator willexhibit certain features:

• a fixed base, which attaches to the suspension,• a movable platform, upon which the slider rests,• springs between the base and platform, flexible inthe direction of desired motion, and stiff in all otherdirections, and• an electrostatic actuation array that generates theforce used to move the platform and slider.

Microactuators must also include a wiring scheme fortransferring signals to and from the slider and often in-corporate a structure for sensing the motion of the sliderrelative to the suspension. Electrostatic microactuatorsto be discussed in this chapter include HexSil and DRIEfabricated actuators from the University of California,Berkeley, and electroplating-formed actuators by IBMand the University of Tokyo.

Electromagnetic or piezoelectric force are alterna-tives to electrostatic actuation in the actuated-sliderconfiguration. Electromagnetic microactuators use fer-romagnetic films to produce force perpendicular to anapplied electric field. This type of actuation has po-tentially low voltage requirements but requires specialfabrication techniques to integrate the magnetic compo-nents into the assembly. A microactuator of this type forhard disk drives is under development at Seagate, withresults as yet unpublished. Piezoelectric microactuatorsuse a piezoelectric material, which expands or contractsin response to applied voltage, to move the slider. Theseactuators have simple fabrication, the patterning a pieceof piezoelectric material to sit between the suspensionand slider, but it is difficult to obtain an adequate range ofmotion. A short stroke from the piezoelectric piece mustbe leveraged into a much larger motion at the read/writehead. A piezoelectric microactuator has been producedby TDK corporation with a 0.5 µm TS

1 stroke length at10 V, with a 10 V bias [32.12].

32.1.3 Electrostatic Microactuators:Comb-Drives vs. Parallel-Plates

Electrostatic microactuators have been studied as thesecondary actuators in HDDs for their relative ease

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TS1 “um” has been changed to “µm”. Please confirm.



of fabrication, particularly in the configurations of“actuated slider” and “actuated head,” since the struc-tural material needs only to be conductive rather thanferromagnetic or piezoelectric. Electrostatic force isgenerated by applying a voltage difference between themoving shuttle and a fixed stator element. Depending onthe designated motion for the shuttle, electrostatic actu-ators are often categorized into two groups: comb-drivesand parallel-plates, as illustrated in Fig. 32.3.

The magnitude of the electrostatic force generatedequals the rate of change of energy that is retained withinthe finger-like structure and varied by shuttle motion.Therefore, the electrostatic force for comb-drive actua-tors, in which the designated shuttle motion moves alongthe x-direction, as shown in Fig. 32.3, equals

Fcomb = ∂E

∂x= εh

2dV 2 , (32.1)

where ε is the permittivity of air, x is the overlap betweentwo adjacent plates, h is plate thickness, and d is the gapbetween two parallel plates. Similarly, the electrostaticforce for parallel-plates actuators is

Fparallel = ∂E

∂y= εxh

2d2 V 2 . (32.2)

As indicated in (32.1) and (32.2), the electrostaticforce for comb-drives actuators does not depend on thedisplacement of the moving shuttle and thus allowsa long stroke while maintaining a constant electro-static force. The electrostatic force for parallel-platesactuators, in contrast, is a nonlinear function of itsshuttle motion (∝ 1/d2), and the maximum stroke islimited by the nominal gap between shuttle and stator.A longer stroke is achieved with a larger gap, at theexpense of lower electrostatic force. For applicationsthat require small stroke but large force output, parallel-plates actuators are preferred since the output forcefrom parallel-plates can be x/d times larger than theforce from comb-drives. The following equation (32.3)is easily derived from (32.1) and (32.2).

Fparallel

Fcomb= x

d. (32.3)

A simplified second-order differential equation isoften utilized to describe the dynamic response of anelectrostatic microactuator

mx(t)+bx(t)+ Kmx(t) = F [V, x(t)] , (32.4)

where m is the mass of the moving shuttle, b is the damp-ing coefficient of the microactuator, Km is the spring

Stator

Shuttle

Parallel-plate motionComb-drive motion

Spring

h

d

xSpring

Shuttle

Stator

y

x

Fig. 32.3 Electrostatic microactuators. Comb-drives versus parallel-plates

constant of the mechanical spring that connects the mov-ing shuttle to an anchor point, and F is the electrostaticforce that can be obtained from (32.1) and (32.2).

Differential DrivesBecause electrostatic force is always attractive, elec-trostatic microactuators need other features to activelycontrol the direction of shuttle motion in servo applica-tions, as opposed to relying on the restoring force froma mechanical spring. For this reason, the differential-drives approach, as shown in Fig. 32.4, is frequentlyadopted in electrostatic microactuator designs. Basedon the differential-drive configuration, the simplifiedsecond-order differential equation (32.4) is rewrittenas,

mx(t)+bx(t)+Kmx(t) = F [Vbias+Vdr, xo−x(t)]−F [Vbias−Vdr, xo+x(t)] ,

(32.5)

where x0 is the nominal position of the moving shuttle. Ifthe differential drive is operated at the bias voltage (Vbias)with a small perturbation voltage (Vdr), the nonlinearforce input in (32.5) can be linearized with a first-order

Isolationplug

Drive-elektrodes

w/ isolation w/o isolation

V3

V2V1 V1 V2

V3

Drive-elektrodes

Shuttle

Shuttle

Fig. 32.4 Differential parallel-plates actuators and electrical-iso-lation features

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approximation.

mx(t)+bx(t)+ (Km − Ke)x(t) = KvVdr

Ke = 2∂F

∂x|Vbias

Kv = 2∂F

∂Vdr|Vbias . (32.6)

Here Ke represents a softening electrostatic spring con-stant, and Kv represents the voltage-to-force gain. Theelectrostatic spring constant acts as a negative springduring the electrostatic microactuator operation, and itsvalue varies with the bias voltage (Vbias). When theelectrostatic spring constant Ke exeeds the spring con-stant Km from the mechanical spring, the microactuatorbecomes unstable; this is often described as pull-in in-stability. As shown in (32.5) and (32.6), the differentialconfiguration cancels the even order harmonics in volt-age and thus linearizes the voltage–force relation tosome extent. Furthermore, in parallel-plates actuators,the differential configuration reduces the nonlinearity inactuation voltage as well as in shuttle displacement.

Electrical IsolationBecause electrostatic actuation requires multiple volt-age levels for actuation force and position sensing (asdiscussed in the following section), electrical isolation isanother challenge for designing an electrostatic micro-actuator. Generally speaking, when multiple voltagelevels are needed in MEMS devices, electrical isola-tion is achieved by breaking up the parts that need tobe on different voltage level and anchoring them sepa-rately to a nonconductive substrate. This approach hasmany drawbacks, not only because it requires a sub-strate in a device but also because structures have to bemechanically separated to be electrically isolated. Theelectrical isolation problem is far more severe in parallel-plates microactuators than comb-drive microactuatorssince parallel-plates actuation generally requires differ-ent voltage levels for stator fingers pulling in oppositedirections.

Figure 32.4 shows an example of how an electrical-isolation feature can be utilized to increase the actuationforce output in a differential parallel-plates micro-actuator design. As shown in the figure, without theproper electrical isolation, drive-electrodes with differ-ent voltage potentials have to be placed in separategroups and result in the same voltage difference acrossdrive-electrodes and shuttle on both sides of each shuttlefinger [32.5, 6]. Since electrostatic forces are always at-tractive, gaps on two sides of the interlaced structurecannot be made equal, otherwise the forces on two

sides of a shuttle finger will be equal and the shut-tle’s movement direction will be uncontrollable. Withsuch electrical-isolation features as the “isolation plug,”shown on the left in Fig. 32.4, gaps of the interlacedstructure can be the same width, since different voltagescan be applied on the two sides of the shuttle fingers. Asshown in Fig. 32.4, the design with integrated electrical-isolation features is more compact than without isolationfeatures. Consequently, more finger structures can fit inthe same amount of space, and the actuation voltage canbe reduced.

32.1.4 Position Sensing

Most proposed HDD dual-stage servo controllers utilizeonly the position of the magnetic head relative to thecenter of data track, known in the industry as the posi-tion error signal, or PES, for closed-loop track followingcontrol. These systems have a single-input-multi-output(SIMO) control architecture. In some instances, how-ever, it is also possible to measure the relative positionerror signal (RPES) of the magnetic head relative tothe VCM. In this case, the control architecture is multi-input-multi-output (MIMO). As shown in [32.13], RPEScan be used in a MIMO controller to damp out the sec-ond stage actuator’s resonance mode and enhance theoverall robustness of the servo system.

Capacitive position sensing and piezoresistive pos-ition sensing are two popular sensing mechanismsamong electrostatic microactuator designs. Each of thesesensing mechanisms is discussed in more detail in thefollowing sections.

Capacitive Position SensingCapacitive position sensing is based on shuttle move-ment causing a capacitance change between the movingshuttle and fixed stators. By measuring the changein capacitance, it is possible to determine the shuttlelocation relative to the fixed stator. The output volt-age (Vo) for both differential drives in Fig. 32.5 equals2δC/CiVs. The capacitance change due to shuttle move-ment (dC/dx) can be derived and the output voltage(Vo) for the comb-drives and parallel-plates can beformulated as a function of shuttle displacement.

Vcomb = 2Cs

Ci

δx

x0Vs ,

Vparallel = 2Cs

Ci

y0δy

y20 − δy2

Vs ,

≈ 2Cs

Ci

δy

y0Vs , (32.7)

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X

Differential comb-drives

Y

Ci

Stator

Shuttle

x0

Stator

Shuttle

Cs–δC

Differential parallel-plates

y0

Cs+δC

Cs–δC

Cs+δCVs

–Vs

Vs

–Vs

Vo

Ci

Vo

Fig. 32.5 Capacitive position sensing. Comb-drives versusparallel-plates motion

where x0 is the nominal overlap for interlaced fingersand y0 is the nominal gap between overlapped fingers,as shown in Fig. 32.5.

As indicated by (32.7), the voltage output for comb-drives sensing structure,Vcomb, is linear with shuttledisplacement. On the other hand, the sensing config-uration that makes use of parallel-plates motion hasbetter sensitivity for detecting shuttle motion since y0is usually smaller than x0. Although the nonlinearity inparallel-plates sensing can be linearized by the differ-ential drive configuration to some extent, in a designexample of 4 µm gap with 1 µm stroke, the linear modelshown in (32.7) still produces 6% deviation from thenonlinear model.

Among electrostatic microactuators that use cap-acitive position sensing, the capacitance variation dueto shuttle motion (dC/dx) is typically at the level of100 fF/µm. In order to obtain 10 nm position sensingresolution, the capacitance sensing circuit must be ableto detect capacitance variation of 1 fF in the presenceof parasitic capacitance and offset/mismatches fromop-amps, which can easily result in an output voltageorders of magnitude larger than the output voltage fromthe designated capacitance variation. In most capacitiveposition sensing, the limiting factor for the sensing res-olution is not the thermal noise but the sensing circuit’sdesign.

Here we introduce two basic capacitance sensingcircuits suitable for high-resolution position sens-ing [32.14]. The concept of the “synchronous scheme,”as shown in Fig. 32.6, is to reduce the impedance ofsense capacitors as well as the offset and 1/f noisesfrom op-amps by applying modulation techniques onthe sense voltage (Vs). The Rdc resistor on the feed-back loop sets the DC voltage level at the input nodesof the “charge integrator.” The effect of the presenceof parasitic capacitance (Cp) is nullified by the virtualground condition from the op-amps. The major draw-back of the synchronous scheme is that the DC-settingresistor (Rdc) has to be large to ensure the proper gainfor the capacitance sensing [32.14], which introducesexcessive thermal noise into the sensing circuit. In addi-tion, a large resistor usually consumes a large die spacein implementation.

A switched-capacitance scheme, as shown at the bot-tom in Fig. 32.6, is one alternative that avoids the use ofDC-setting resistor. The capacitance sensing period isbroken into two phases: reset phase and sense phase.During the reset phase, input/output nodes of capacitors

Ci

Stator

Shuttle

Shuttle

+

_VO

+Vs

Gain

Sense Phase : ΦSN Sensing period (1/fs)

ΦRS

Stator

Shuttle

Shuttle

+_

LPF

+1–1

Gain

I. Synchronous scheme

Reset Phase : ΦRS

II. Switched-capacitance scheme

ΦRS

Charge integrator

Pre-amplifier

ΦSN

ΦSNΦRS ΦSN

Ch

ΦRS ΦSN

Cp

CS

CS

ΦSN

ΦRS–Vs

+Vs

–Vs

Ci

CpCS

CS

Rdc

VO

+Vs

–Vs

Fig. 32.6 Capacitive position sensing circuits. Synchronous schemeversus switched-capacitance scheme

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and input voltage to op-amps are set to ground or ref-erence level to ensure proper DC voltage for the chargeintegrator. During the sense phase, sense voltage ±Vsis applied to the sense capacitors, and the amount ofcharge proportional to the mismatch in the sense cap-acitors is integrated on the capacitor Ci, thus producingan output voltage proportional to the capacitance mis-match from the sense capacitors. This approach replacesthe large DC-setting resistor by capacitors and switchesand results in a much smaller die compared to the syn-chronous scheme. Furthermore, the switching techniqueallows more design flexibility for system integration andperformance improvement because of the ability to al-locate separate phases for various operations. The majordrawback of this design is that it draws noise into sens-ing circuits from switches and sampling capacitor Ch.But, these noises can be compensated by dividing thesense phase into 2 ∼ 3 sub-sense phases [32.14], at theexpense of a complicated circuit design.

Piezoresistive SensingPiezoresistive films have been widely used as strain-sensitive components in a variety of MEMS devices,including pressure sensors and vibration sensors. Gener-ally speaking, piezoresistive sensing techniques requireless complicated sensing circuits and perform better ina severe environment than other sensing techniques.When a piezoresistive film is subjected to stress, thefilm resistivity and dimensions change. The fractionalchange of resistance is proportional to the deformation ofthe piezoresistive film. For a small change of resistance,this relation can be expressed as,

∆R

R= K · ε , (32.8)

where R is resistance of the piezoresistive film, K is itsgage factor and ε is strain. In microactuator designs, thepiezoresistive film is usually applied to the spring struc-ture that connects the moving shuttle to an anchor point.When the shuttle moves, it stretches the spring as well asthe piezoresistive film, consequently, the piezoresistivefilm produces a deformation signal proportional to theshuttle displacements. As a result, piezoresistive sensingis easier to implement than capacitive position sensing,but the sensing resolution is usually less accurate dueto higher thermal noises introduced by the resistance ofthe piezoresistive film.

Another application of piezoresistive sensing, asidefrom measuring relative slider position, is to detect vi-bration in the suspension itself. The idea is to senseairflow-induced vibration of the suspension and feed that

information forward to an actuated slider to damp outmotion at the head. Piezoresistors used for this purposecan be made from metal or semiconductor materials, ar-ranged as a strip or series of strips oriented along thedirection of vibration strain. It is important that thesesensors observe all vibration modes that contribute tooff-track error, so a number of optimization schemesfor locating the sensors have been developed [32.15].One method is to maximize the minimum eigenvalueof the observability matrix of the sensor or sensors.This ensures that all relevant modes are observed. An-other method is to minimize a linear quadratic gaussiancontrol problem over potential sensor locations [32.16].This serves to determine an optimal placement from theperspective of a linear controller.

32.1.5 Electrostatic Microactuator Designsfor Disk Drives

Various electrostatic microactuators have been designedfor the secondary actuation in HDDs. To incorporate anelectrostatic microactuator into a HDD without alteringmuch of current suspension configuration, many designconstrains are imposed. In this section, we will first dis-cuss some design issues and then present one specificdesign example.

Translational Microactuatorsversus Rotary Microactuators

Depending on the motion of the magnetic head actu-ated, microactuator designs are categorized into twogroups: translational actuators and rotary actuators. Ei-ther type can be implemented by comb-drives [32.8,17]or parallel-plates [32.5, 6, 18] actuation.

When employing a translational microactuator ina dual-stage HDD servo, previous research [32.13] hasshown that a force coupling between the suspensionand the translational microactuator exists, consisting oftransmitted actuation force from the VCM and suspen-sion vibration induced by windage. The force couplingfrom the VCM not only complicates the dual-stage servocontroller but also imposes a design constraint on a trans-lational microactuator design, in that the translationalmicroactuator has to provide a large force output to coun-terbalance the coupling force. When the VCM makesa large movement, as in seeking a new data track, themicroactuator may be overpowered. One solution is topull the actuator to one side and lock it momentarilyin place. Even then, the use of the two actuation-stagesmust be carefully coordinated to moderate the influ-ence of the VCM on the microactuator. On the other

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hand, the linear springs in the translational microactuatorcan also aid in damping out motion of the suspen-sion. The portion of suspension vibration induced bywindage mostly consists of high frequency excitation, sothe resulting magnetic head’s position error can be pas-sively attenuated by low resonant frequency translationalmicroactuators.

Generally speaking, rotary actuators are more dif-ficult to design/analyze than translational actuatorsbecause of their nonuniform gap between shuttle andstator. Still, their different operating properties have bothstrengths and weaknesses. Unlike a translational actu-ator, no obvious force coupling is transmitted from thesuspension to a rotary actuator, as the microactuatoris nearly always attached to the end of suspension atthe microactuator’s center of rotation, which acts asa pivot point. With no mechanical coupling, the dual-stage servo system using a rotary microactuator doesnot suffer from the force coupling between VCM andmicroactuator seen in translational designs. But, the ro-tary microactuator has to compensate for the magnetichead’s position error induced by suspension vibrationwithout any passive attenuation of the vibration. Overall,a rotary acuator is likely to behave better than a trans-lational microactuator during track seeking and worseduring track following.

Gimballed Microactuator DesignProper flying height and orientation of a slider andread-write head over a hard disk is maintained by theinteraction of the suspension, the air bearing of theslider, and a gimbal structure. The gimbal structureis located at the tip of the suspension and holds theslider/microactuator in its center coupon. The dynamiccharacteristic requirements of gimbal structures are thatthey be flexible in pitch and roll motion but stiff inin-plane and out-of-plane bending motion. To meet allthese requirements using only one piece of metal isa highly challenging task. For this reason, most com-mercially available suspension/gimbal designs consistof two to three pieces of steel, each with differentthickness.

The goal of a gimballed microactuator design isto integrate seamlessly an actuator and gimbal intoa one-piece structure, as shown in Fig. 32.7, to simplifyboth suspension design and HDD assembly. A full in-tegrated suspension that includes suspension, gimbal,and microactuators in one part has also been pro-posed [32.19]. The dimple structure, existing in mostcurrent gimbal structures, is excluded from the gim-balled microactuator design and electrical interconnects

Translational mircoactuator Rotary mircoactuator

Gimbal torsion bars

Fig. 32.7 Schematics of translational microactuator versus rotarymicroactuator. Courtesy from Lilac Muller

are in situ fabricated on the gimballed microactuator,replacing the flexible cable in current HDD suspensionassembly.

The dimple structure in current suspension assem-blies provides out-of-plane stiffness while preserving thenecessary torsional compliance in the gimbal structure.Without a dimple structure in the suspension assembly,the gimbal itself must provide high out-of-plane bendingstiffness. Otherwise it would unbalance the suspensionpre-load, which is an overbend of the suspension thatbalances the upward air-bearing force on the suspen-sion during operation. The electrical interconnects areimplemented to transmit data between magnetic heads,located at the center coupon of the gimbal structure, andIC circuits located at the end of the suspension. Thein situ fabricated electrical interconnects are inevitablypassed through torsion bars of the gimbal and thus seta design constraint for the minimum width of torsionalbars. Furthermore, both the gimbal structure and micro-actuator should be the same thickness to simplify theMEMS fabrication process.

To summarize the design constrains discussed above,the integrated gimbal structure has to meet perfor-mance requirements with a single, uniform piece ofmaterial that would previously have been achieved bytwo to three metal pieces with different thicknesses,while the minimum width of any torsion bars thatmay be used pre-determined. To solve this problem,Muller [32.6] proposed a T-shaped structure (a beamstructure with overhang surface sheet) for the torsionbars, and Chen [32.19] proposed “double-flexured” tor-sion bars. Additionally, many suspension manufacturershave developed new gimbal structures for their sus-pensions designed specifically for use with MEMSmicroactuators, moving the gimbal location back to thesuspension from the microactuator.

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An Electrostatic Microactuator Design ExampleFigure 32.8 shows a translational microactuator designsuitable for the HDD dual-stage actuation by Horsleyin 1998 [32.5]. The translational electrostatic micro-actuator dimensions are 2.2 mm×2.0 mm×0.045 mmand weight 67 µg. The dimensions of the pico-slideron the top are 1.2 mm×1.0 mm×0.3 mm and weight1.6 mg. This microactuator design does not include“electrical-isolation” features, and thus the electrical iso-lation and electrical-interconnects were fabricated ona separate substrate and subsequently bonded to themicroactuator. This microactuator uses parallel-platesfor actuation force but does not have dedicated position

Pico-slider

Fixedstator

Parallel-platesactuation

Movingshuttle

Fig. 32.8 Pico-slider mounted on a translational microactuator.Courtesy from Horsely 1998

Table 32.1 Parameters of the electrostatic microactuator design by Horsley 1998 [32.5]

Parameters Source∗ Value

Nominal gap D 10 µm

Structure thickness D 45 µm

Rotor mass, m I 44 µg

dC/dx C 68 fF/µm

Actuation voltage, bias voltage D 40 V

Actuation voltage, maximum driving voltage D ±40 V

Voltage-to-force gain Kv I 50 nN/V

Mechanical spring constant Km I 29 N/m

Electrostatic spring constant Ke I 9.6 N/m

Damping coefficient, b I 1.03×10−4 N/(m/s)

Voltage-to-position DC gain M 0.05 µm/V

Resonance frequency, wr M 550 Hz∗: D=Design value, C=Calculation, M=Measurement, I=Inferred from measurements

sensing structures due to fabrication process limitation.Table 32.1 summarizes key parameters of this micro-actuator design.

Based on these parameters, the characteristics of thiselectrostatic microactuator can be estimated by the lineardifferential equation shown in (32.6).

Figure 32.9 shows the schematics of a circuit de-sign by Wongkomet in 1998 [32.14], in which actuationdriving voltage and capacitive position sensing were im-plemented for the electrostatic microactuator designedby Horsley. As mentioned before, the electrostaticmicroactuator design doesn’t have dedicated structurefor a position sensing; as a consequence, the inputnodes for actuation and output nodes for capacitiveposition sensing have to share the same electrodes. Ca-pacitors Cc and Cc0, shown in Fig. 32.9, are carefullydesigned to shield the high voltage presented in ac-tuation circuit from sensing circuit, which is mostlylow voltage, and thus enable driving/sensing circuitintegration.

The driving circuit, shown in the left in Fig. 32.9,demonstrates how to generate the bias voltage (Vbias,±40 V) and drive voltage (Vdr, −40–+40 V) from the0–5 V CMOS compatible circuits. The switches at theoutput of charge pumps were synchronized with theswitching period φRS. During the sense phases φSN1 andφSN2, therefore, the switches are left open and thus novoltage fluctuation is seen by the sensing circuits. This

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Charge pump

Charge pump

Actuationdriving circuit Gain Buf-

fer

Position sensing circuitElectrostaticmicroactuatorDriving circuit

ΦRS

Cs–

Vdr

ΦSN1 ΦSN2

ΦRS ΦRS

ΦSN2,ΦRS ΦSN2

Cs+

Cp

Cp

ΦRS

ΦRS

ΦRS

Vbias+

Vdd+

Vdd–

Vbias–

Vdd+

Vdd+

VS

VS

Ci

Ci

ΦRSΦRS

ΦSN2,ΦRS ΦSN2

Ch

Ch

ΦSN1

ΦSN2

ΦRS

V

Cc

Cc0

Cc

+2.5 ~ –2.5 V

0 V

+5 V+40 V

+40 ~ – 40 V

– 40 V Cp0

Fig. 32.9 A simplifiedschematic including drivingand sensing circuits for theelectrostatic microactuator.Courtesy from Wongkomet1998

arrangement was utilized to reduce feedthrough fromthe driving circuit to position sensing circuit.

The design target for the capacitive position sens-ing circuit was to achieve position sensing resolutionof 10 nm, and this goal was approached by two maintechniques implemented in the circuit: differential sens-ing and Correlated-Double-Sampling (CDS). The mainbenefits of the differential sensing scheme are reducednoise coupling and feedthrough, elimination of even-order harmonics, and improvement of dynamic rangeby doubling the output swing. To adapt this differentialsensing scheme in a differential parallel-plates electro-static microactuator, a bias voltage (Vbias) was appliedto the stator and the drive voltage (Vdr) was applied tothe shuttle. The CDS technique, a modified capacitancesensing technique based on the switched-capacitancescheme, was implemented along with the differentialsensing scheme to compensate for sensing noises includ-ing 1/f, KT/C, switch charge injection and offset fromop-amps [32.14]. The concepts of CDS can be briefly de-scribed as follows: the sense period is broken into threephases: one reset phase, φRS, and two sense phases, φSN1and φSN2. During the φRS sense phase, the voltages forcapacitors and input nodes to op-amps are set to the ref-erence level of the DC-voltage setting for the chargeintegrator, same as for the switched-capacitance schemediscussed in Sect. 32.1.4. During the φSN1 sense phase,a sensing voltage −Vs is applied to the shuttle and resultsin a voltage difference, α(−Vs)+ Verror, across the sam-pling capacitor Ch, where α is the transfer function from

sense voltage to the output voltage at the pre-amplifier,shown as Gain in the plot, and Verror is the voltage at theoutput node of pre-amplifier resulting from noise, leak-age charge, and offset of op-amps. Lastly, during theφSN2 sense phase, the sensing voltage is switched from−Vs to Vs and the switch next to Ch is switched open.This results in a voltage output α(Vs)+ Verror at the out-put node of the pre-amplifier and a voltage, 2α(Vs), atthe input node of the buffer. As a consequence, the volt-

– 20

– 30

– 40

– 50

– 60

– 70

– 80

– 90

–100

–110

Drive voltage/position sending voltage (dB)

102 103 104

Frequency (Hz)

Fig. 32.10 Open-loop and closed-loop frequency response of theprototype microactuator with a pico-slider. The capacitive positionmeasurement (solid line) is compared to the measurements fromLDV (dashed line). Courtesy from Naiuavudhi Wongkomet

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age resulting from sensing error (Verror), which appearsat the output node of pre-amplifier, disappears from thevoltage-input node to the buffer. Switches utilized inthe circuit and their correspondent timing are shown inFig. 32.9. Be aware that for presentation simplicity com-ponents in the pre-amplifier and buffer are not shown indetail in Fig. 32.9.

The frequency response, both open-loop andclosed-loop with a PD controller, of the electrostaticmicroactuator measured by a capacitive position sensingcircuit is shown in Fig. 32.10. The position measure-ments from an LDV are also shown in the same plot

for comparison. The deviation between measurementsfrom the different position sensing devices appears inthe high frequency region at the plot and has beenidentified as the feedthrough from the capacitancesensing circuit. The effect from feedthrough was neg-ligible at low frequency region but becomes significantand results in a deviation in magnitude for the trans-fer function of −80 dB ∼ −90 dB after 2 kHz. Thefeedthrough presented in the capacitance sensing cir-cuits limited the position sensing resolution to the levelof 10 nm.

32.2 Fabrication

While there are several approaches to building electro-static microactuators suitable for hard disk drives, theyall exhibit certain common features from a fabricationstandpoint.

32.2.1 Basic Requirements

As we discussed earlier, nearly all electrostatic micro-actuators rely on a system of interlaced fingers or platesto provide actuation force. As a result, a method forproducing arrays of these fingers or plates with narrowgaps between them is usually the central concern in de-veloping a fabrication process. The resulting structuremust then be strong enough to support both the slider onthe microactuator and the microactuator on the suspen-sion, particularly when loaded by the air-bearing thatsupports the slider above the hard drive’s spinning disk.In addition, the design and fabrication process must in-clude a way to perform electrical interconnection on themicroactuator. This involves transferring signals to andfrom the actuator and slider and isolating the parts of themicroactuator requiring different voltage levels.

In addition, the microfabrication process is subject tocertain basic constraints. The materials used in fabrica-tion must either be thermally and chemically compatiblewith any processing steps that take place after their de-position or must somehow be protected during steps thatwould damage them. This often constrains the choiceof materials, deposition techniques, and processing or-der for many microdevices. Another concern is that thesurface of the structure be planar within photoresist spin-ning capabilities and lithography depth-of-focus limitsif patterning is to be performed. This can be a majorchallenge for disk drive microactuators, which are large

in size and feature high-aspect ratio trenches comparedto other MEMS devices.

32.2.2 Electrostatic MicroactuatorFabrication Example

Section 32.1.5 described the design and operation ofa translational electrostatic microactuator. This sectionexamines the fabrication process by which that micro-actuator was built [32.5]. The process is a variation ona micro-molding process known as HexSil [32.20]. Ina micro-molding process, a mold wafer defines the struc-ture of the microactuator and may be reused many timeslike dies in macro-scale molding; in the HexSil versionof the procedure, the mechanical parts of the micro-actuator are formed in the mold by polysilicon. A secondwafer is used to create metallized, patterned target dies.Upon extraction from the mold, the HexSil structure isbonded to this target substrate, which is patterned todetermine which sections of the HexSil structure areelectrically connected.

Naturally, the mold wafer is the first item to be pro-cessed and is quite simple. It is a negative image of thedesired structure, etched down into the wafer’s surface,as shown in Fig. 32.11a. A deep but very straight etch iscritical for successful fabrication and subsequent oper-ation of the devices. If the trench is too badly bowed or isundercut, the finished devices will be stuck in the moldand difficult to release. On the other hand, a tall devicewill have a larger electrostatic array area, will generatemore force, and be better able to support a slider whilein a disk drive.

Fabrication of the HexSil structure forms the ma-jority of the processing sequence Fig. 32.11b–g. HexSil

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TS0 32.2 Fabrication 963

upside down

Etch trenches for mold

Coat with SiO2

Refill with polysilicon

Pattern polysilicon

Evaporate Cr/Cu seed layer

Electroplate Pads

Remove seed layer, oxide

Bond to target substrate

Lightly doped silicon

Highly doped silicon

Copper

LPCVD oxide

Cr/Cu seed layer

Photoresist

Indium

a)

f)

a)

b)

c)

d)

e)

g)

h)

Fig. 32.11 Fabrication example 1: HexSil TS3

fabrication begins by coating the surface of the moldwafer with a sacrificial silicon oxide. This layer, de-posited by low-pressure chemical vapor deposition(LPCVD), must coat all surfaces of the mold, so thatremoval of the oxide at the end of the process willleave the device completely free of the mold. Thismicroactuator uses a 3 to 4 µm thick oxide layer toensure clearance between the polysilicon structure andthe trench walls during release. The mold is then re-filled completely with LPCVD polysilicon, which willform the desired mechanical structure. Polysilicon ischosen for its conductivity and good conformality in re-filling trenches. This also leaves a planar surface readyfor photolithography to cover bonding locations. Afterthe lithography step, the polysilicon deposited on top ofwafer is removed, except where the bonding points weredefined.

The structure is then prepared for bonding by form-ing a soldering surface. First a chrome/copper seed layeris evaporated on the surface of the wafer. The chromepromotes copper adhesion to the silicon, while the cop-per forms the starting point for electroplating. The seedlayer is covered with photoresist, which is cleared onlyin the locations where contacts are desired. There, elec-troplating will produce a metal film, forming solderingpoints for the device to the target substrate that will formits base. In this case, copper is used as the plating mater-ial, thanks to its high conductivity and solderability. Thedevice is released from the mold by ion milling awaythe thin seed layer, then dissolving the oxide lining witha hydroflouric (HF) acid wet etch. The free standing por-tion of the microactuator is thus ready for bonding to thetarget. The mold, meanwhile, may be used again aftera simple cleaning step.

The target substrate may be prepared in several ways.The target shown above consists simply of indium solderbumps and interconnects on a patterned seed layer; it isknown as a “plating bus” arrangement. The seed layer isused for the same reasons as on the HexSil structure, butin this case the seed layer is patterned immediately, withthe unwanted portions removed by sputter etching. Thisprovides isolation where desired without having to etchthe seed layer after the indium is in place. Photoresist isspun again and patterned to uncover the remaining seedlayer. A layer of indium is then electroplated everywherethe seed layer is visible. Indium acts as the solder whenthe two parts of the microactuator are pressed together,forming a cold weld to the copper at 200–300 MPa thatwas found suitable for hard disk requirements, as shownin Fig. 32.11h.

32.2.3 Electrostatic Microactuator ExampleTwo

The high-aspect ratio microactuator described in thissection is in development at the University of Cal-ifornia, Berkeley, and demonstrates another way tocreate the features required of an actuated slider. Themicroactuator is translational, with parallel-plates actu-ation, and includes two of the design options describedpreviously: integrated isolation plugs, as described inSect. 32.1.3, and a capacative sensing array, as describedin Sect. 32.1.4. A picture of the basic design is shown inFig. 32.12; a central shuttle holding the slider is sup-ported by four folded-flexure springs and driven byparallel-plates arrays.

Basic Process: Silicon-on-Insulator

TS3 In the manuscript we couldn’t see different colors for LPCVD oxide, copper and photoresist. Please show what is what in the

draw.TS3 In the manuscript we couldn’t see different colors for LPCVD oxide, copper and photoresist. Please show what is what in the

draw.


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Shuttle

SpringSpring

Spring SpringSensingcapacitor, Cs

–

Sensingcapacitor, Cs

+

Drivingarray

Drivingarray

Fig. 32.12 Translational high-aspect ratio microactuator

Successful processing of the design is centered aroundthe ability to etch very straight, narrow trenches usingdeep reactive ion etching (DRIE). DRIE is a specialplasma etching sequence that uses polymerization ofthe sidewalls of a trench to keep the walls straighteven at extreme aspect ratios. Deep, narrow trenchesmake possible a microactuator with both closely packedelectrostatic arrays and good mechanical strength. Con-ventional surface micromaching is used to produce themajority of the electrical interconnects.

The basic fabrication process uses a silicon-on-insulator (SOI) wafer to control DRIE trench depth. SOIwafers are very useful in microfabrication as they pro-vide a layer of single-crystal silicon (the device layer)separated from the bulk of the wafer (the handle layer)by a thin layer of buried silicon oxide. This gives a verywell controlled thickness to finished device but is veryexpensive. The microactuators described here are fab-ricated from an SOI wafer with a 100 µm device layer.A variation on the processing sequence and layout elim-inates the need for SOI wafers and will be discussed inthe following section.

The first stage of fabrication is creation of deep iso-lation trenches (see Fig. 32.13a–c). This procedure hasbeen adapted to MEMS from integrated circuit pro-cessing for isolating thick MEMS structures [32.21].The isolation pattern is formed by photolithography andetched by DRIE down to the buried oxide. With trenchesonly 2 µm wide, this corresponds to an aspect ratio of50 : 1. The wafer surface and trenches are then coatedwith LPCVD silicon nitride, which acts as the electric-al insulator. The trench is then refilled in its entirety

DRIE of isolation trenches

Nitride & poly-Si refill

Etch back and nitride cap

Etch contact holes

Metallization

Pattern interconnect

DRIE with thickphotoresist

HF release

Aluminium

Thermal oxide

LPCVD oxide

Spin-on glass

Photoresist



Undoped polysilicon

Silicon nitride

a) e)

b)

c)

d) h)

h)

f)

Fig. 32.13 Basic silicon-on-insulator fabrication process TS3

with LPCVD polysilicon, that a more conformal mater-ial better fills the trench than silicon nitride would alone;polysilicon also has much lower residual stresses thansilicon nitride. The refill leaves a layer of polysiliconon the surface of the substrate, which must be etchedor polished back to the silicon nitride. After etch-back,a second layer of LPCVD nitride completes isolationbetween regions.

Next, electrical interconnects are formed, as shownin Fig. 32.13d–f. Contact holes to the substrate are de-fined by photolithography and etched by reactive ionetching of the silicon nitride. Metal lines may then bepatterned directly by photolithography or formed by lift-off. In a lift-off process, photoresist is deposited andpatterned before the metal. The metal is then depositedvertically, so that the sidewalls of the resist are uncov-ered. Dissolving the resist allows the metal on top to

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float away, leaving behind the interconnect pattern. Inthe figures shown here, the interconnects are formed byevaporated aluminum, eliminating the need for electro-plating and a seed layer. The ability to isolate portionsof the substrate is useful for simplifying the intercon-nect layout, as one interconnect can cross another bypassing underneath it through an isolated portion of thesubstrate.

Finally, structural lithography is performed anda second etch by DRIE is done, as shown in Fig. 32.13g.These trenches will define the shape of the micro-actuator, and are larger, 4 µm wide, to allow sufficientrotor travel. After this etch step, the device layer onthe SOI buried oxide consists of fully defined micro-actuators. When the buried oxide is removed by wetetching, the microactuator is released from the substrateFig. 32.13h and, after removal of the photoresist that pro-tected the device surface from HF during release, readyfor operation. Thanks to the high-aspect ratios of DRIEand the efficient integrated isolation scheme, the micro-actuator can theoretically be operated with a DC biasvoltage of 15 V on the rotor, and a dynamic stator volt-ages below 5 V. In practice, higher voltages are requireddue to the spreading of trench walls over the course ofDRIE. A picture of the electrostatic structure, with therotor fingers on the left and the stator fingers on the right,divided down the center by isolation trenches, is shownin Fig. 32.14.

Variation: Anisotropic Backside ReleaseAn alternative fabrication approach is to build the micro-actuator from a regular silicon wafer (∼ 550 µm thick),but then, since the microactuator shouldn’t be more than100 µm thick to fit in a HDD, etch away the backside ofthe wafer to obtain devices of that size. The main reasonfor using a silicon-only wafer is to reduce the need for

Fig. 32.14 Electrostatic driving array of microactuator fromSOI

expensive SOI wafers, which cost approximately tentimes as much as single-material wafers. Another benefitis the elimination of the need for etch holes to reachthe buried oxide layer, which increase vulnerability ofmicroactuators to particle or moisture contamination.But, the thickness of the devices cannot be controlled asaccurately as with the SOI process, and the processingsequence is more complicated.

Control of backside etching is difficult, making ithard to obtain devices with uniform thickness acrossa wafer. A proposed solution is to use an anisotropicwet etchant during the backside etch step. Anisotropicetchants work very slowly on certain crystal planes ofa single-crystal silicon wafer; a coating on trench side-walls can protect fast-etching planes when the etchantreaches the trenches from the back, causing slow-etchingcrystal planes to begin coming together slowing the etch.The result is a nearly self-stopping etch with a V-shapedprofile that prevents overetching even in the presence ofnonuniformities in etch rate and trench depth across thewafer. The most common anisotropic etchants are potas-sium hydroxide (KOH) and tetramethyl ammoniumhydroxide (TMAH). This release process is referred tohere as an anisotropic backside release (ABR).

The key to a good release by this method is to achieveexcellent protection of the top of the wafer and the deeptrenches from the etchant, and this neccessitates changesin the process flow from that for an SOI wafer. Withouta good protective film, the interconnects on the sur-face of the wafer could be destroyed, or the fast-etchingplanes might be attacked along the trench sidewalls. Ofthe etchants mentioned above, TMAH is preferred, dueto high selectivity to both silicon oxide and nitride and,especially, thermally grown oxide, which can be usedas protective films. But, growing thermal oxide and/ordepositing high-quality nitride are high temperature pro-cesses not compatible with most metals, so the structuraltrenches must be etched and oxidized before forming theelectrical interconnects. As shown in Fig. 32.15c–d, theisolation trenches are formed as before, but the struc-tural trenches are patterned and etched in the very nextstep. A thermal oxide is then grown in the trenches toproduce the protective oxide layer Fig. 32.15e.

To do lithography for the interconnects at this point,then, the surface of the wafer must be planarized. Thisis done by refilling the trenches with spin-on-glass. Thespin-on-glass plugs the trenches nearly to the top andcan be sacrificed along with the thermal oxide at theend of the process. Extra glass on the wafer surface canbe removed by a quick HF dip or chemical mechanic-al polishing. The resulting surface is uneven, as shown

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32.2


Aluminium

Thermal oxide

LPCVD oxide

Spin-on glass

Photoresist



Undoped polysilikon

Silicon nitride

d) DRIE structural trenches

Isolation as before

e) Oxidize trench surface

f) SOG refill, etchback

Metallization

h) Deposit protective coating

i) Anisotropic backside etch

j) Strip protective films

c) g)

Fig. 32.15 Deep trench fabrication process adapted forbackside release TS3

in Fig. 32.15f, but a thick photoresist layer can be ap-plied over the trenches evenly enough to accomplishcontact and metal patterning lithographies. After metal-lization much the same as for the SOI wafer Fig. 32.15g,a final coating of etch resistant material is required toprotect the metal during backside release, as shown inFig. 32.15h–j. After ABR, the backside of the devicewill be shaped by crystal planes, but the thickness ofthe device at the edge of DRIE features will be fixed.While a combination of silicon oxide and silicon nitridehas been effective over a “metallization” of undopedpolysilicon, an effective combination of true metal in-terconnects and protective films has yet to be found andis an ongoing area of research for improving the ABRprocess sequence.

Fig. 32.16 Microactuator with integrated gimbal fabricatedby ABR

A prototype microactuator with integrated gimbalfabricated using ABR is shown in Fig. 32.16. This pro-totype was operated with a 30 V DC bias and ±8 V ACdriving voltage for ±1 µm displacement, while a largerversion (shown in Fig. 32.12 TS

4 ) was operated at 15 Vbias and ±3 V driving. This layout and fabrication con-cept is currently being adjusted to fit a MEMS-readysuspension for more advanced testing.

32.2.4 Other Fabrication Processes

Where the DRIE-based process described above “digs”the microactuator out of a wafer, a procedure developedby IBM “grows” a microactuator with high-aspect ratiotrenches on top of a wafer [32.22].

IBM Electroplated MicroactuatorThis is done by a clever sequence of electroplating stepsand sacrificial depositions. An oxide sacrificial film isfirst patterned then covered with a metal seed layer.When later removed, the sacrificial will have left a gapbetween the microactuator and substrate portions of themicroactuator needed to move freely. A polymer, 40 µmthick, is spun onto the wafer and patterned with a reverseimage of the main structural layer. Trenches are etcheddown to the sacrificial layer by plasma etching thenrefilled by electroplating, much like micro-molding. An-

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32.2

TS4 Is Fig. 32.12 here correct?



other sacrificial polymer, photoresist this time, separatesthe top of the fixed portions of the structure from the plat-form upon which the slider will sit, while convenientlyplanarizing the surface for further lithography. The plat-form is created by two more electroplating steps on topof the parts of the structure that will move freely. Re-moval of the sacrificial layers (photoresist, polymer, andoxide) releases the microactuator for use.

This process has been used to produce both rotaryand linear microactuators and is comparatively advancedfrom a commercial standpoint. The resulting devices arethinner than those described in the previous example, buta similar aspect ratio (20 : 1) can be achieved. This pro-cedure also has the benefits of being a low temperatureprocess, which helps decrease processing cost and iscompatible with the thin film magnetic head manufac-turing process. Moreover it includes an upper structurethat covers the fingers, which improves device reliabilityby shielding the fingers from particles.

Additional fabrication details and dynamic testingresults for the IBM design may be found in refer-ences [32.22] and [32.23]. Certain control results aredescribed in Sect. 32.3.2.

Electroplated Microactuators with Fine GapsFinally, fabrication processes developed at the Univer-sity of Tokyo demonstrate additional techniques forobtaining very small gaps between electrostatic fin-gers [32.24]. Similar to the IBM process, electrostaticfingers are electroplated, in this case by nickel, withina polymer pattern. The gap between closely spaced fin-gers, however, is then formed by a sacrificial metal.Photoresist is used to cover the fingers except for the sur-faces where a small gap is desired; these are electroplatedwith sacrificial copper, making narrow, well-definedgaps possible, in perhaps the most reliable of the pro-cesses described here. Continuing with a second nickelelectroplating step creates the interlaced fingers, whichwill also support the slider; the wafer is polished down tolevel the structure. Finally the copper, photoresist, andseed layer are etched away, leaving the free standingstructure.

In another version of process, photoresist alone sep-arates the stator and rotor fingers. In this case, a thinlayer of photoresist is left where small gaps are de-sired, instead of growing a layer of copper. This methodis dependent on excellent alignment but can eliminatethe need for polishing after the second electroplatingstep.

32.2.5 Suspension-Level FabricationProcesses

The long, highly integrated fabrication processes re-quired by actuated slider microactuators contrast greatlywith the fabrication of actuated suspension micro-actuators.

PZT Actuated SuspensionsIn most cases, actuated suspensions use piezoelectric(PZT) drivers cut or etched as single, homogeneouspieces and attached to the steel suspension. Neverthe-less, microfabrication techniques can still be usefulin suspension processing. For example, some recentsuspensions have used thin film PZT deposited on a sub-strate by sputtering for incorporation in a suspension.This is intended to produce a higher quality PZT filmsand begins to introduce MEMS-style processing tech-niques even into suspension-scale manufacturing [32.25,26].

Integrated Silicon SuspensionAnother area where silicon processing has been sug-gested for use on a suspension is the gimbal region, oreven the entire suspension, as described in [32.19]. Anintegrated silicon suspension would incorporate aspectsof both actuated suspension and actuated slider fabrica-tion. Force is provided by four piezes of bulk PZT cutdown to size, as in other actuated suspensions, but therest of the suspension is fabricated from single-crystalsilicon by bulk micromachining.

The fabrication process for the integrated suspensionis the precursor to the ABR process used to fabricate themicroactuator in Sect. 32.2.3. Layers of LPCVD siliconnitride and polysilicon are deposited on a bare siliconwafers. The polysilicon is doped by ion implantationand covered with a second layer of nitride. The outlineof the suspension and any spaces within it are then litho-graphically patterned. After a reactive-ion nitride etchto reach the substrate, trenches are etched by DRIE.The trenches are then plugged with spin-on-polymer forfurther lithography. First, the top layer of polysiliconis patterned into piezoresisistive strips for sensing vi-bration. Second, a layer of photoresist is patterned forcopper lift-off, to create metal interconnects. Last, theplanarizing polymer is removed, and the entire surfaceis coated with silicon nitride. This final coating protectsthe top surface during an anisotropic backside wet etch,in this case by potassium hydroxide.

Instrumented Suspensions

PartE

32.2


Finally, microfabrication techniques may be useful forinstalling sensors on conventional steel suspensions. Theconcept is to deposit a dielectric and a piezoresistivematerial on the stainless steel sheet that will be formedinto the central piece of the suspension and will patternpiezoresistors directly by lithography and plasma etch-ing. A second lithography may be used to add metallines back to the E-block. Vibration sensors on the sus-pension are very desirable for controlling slider position,and the use of thin-film deposition and photolithographypermits the sensors to be located at the points most ef-fective for detecting vibration. An example of a sensorformed by this method is shown in Fig. 32.17.

32.2.6 Actuated Head Fabrication

Several interesting fabrication processes have been pro-posed for another approach to creating a dual-stage diskdrive servo: the actuated head. In an actuated head,the actuator is built into the slider and moves just theread/write head. This could potentially eliminate manyof the mechanical limits of actuated suspensions or ac-tuated sliders. Actuation techniques are typically similarto those of actuated sliders, with the key complicationbeing the need to integrate an actuator fabrication pro-cess with a slider fabrication process. We discuss waysin which this might be done below.

One fabrication approach to actuating a read/writehead is to enclose an electrostatic driving array insidea slider. The slider is built on a glass or silicon substrate,beginning with the air-bearing surface (ABS), followedby a microactuator, and then surrounded by the remain-der of the slider. The ABS is formed by silicon oxidedeposited over aluminum and tapered photoresist lay-ers that give it a contoured shape. The majority of theABS is then covered with photoresist, while a block inthe center is plated with nickel and etched to form theelectrostatic actuator. The electrostatic microactuator, inturn, is covered while the remainder of the slider bodyis electroplated. The slider may then be bonded to a sus-pension and the ABS surface released from the substrateby etching away the original aluminum and photoresistsurface [32.9].

The sacrificial metal technique described inSect. 32.2.4 is another possibility for building an elec-

Fig. 32.17 Piezoresistive strain sensor on steel substrate

trostatic array onto a slider. It can be used to formsmall array with very narrow gaps, and the process iscompatible with slider and head materials [32.27].

The third actuated-head process makes use of anSOI wafer and silicon based materials. For this structure,a 20 µm device layer is coated with silicon nitride. Thenitride is patterned with contact holes for connectionsto the substrate and used as insulation over the rest ofthe device. A layer of molybdenum is sputter depositedto perform interconnections to both the head and theelectrostatic array. This array is formed by DRIE tothe buried oxide, using a lithographically patterned hardmask of tetraethylorthosilicate. The result is a simple,tiny, actuator that must be bonded to the edge of a sliderafter release from the SOI wafer [32.28].

32.3 Servo Control Design of MEMS Microactuator Dual-Stage ServoSystems

The objective of disk drive servo control is to movethe read/write head to the desired track as quickly as

PartE

32.3

TS0 32.3 Servo Control Design of MEMS Microactuator Dual-Stage Servo Systems 969

possible, referred to as track seek control, and, once on-track, position the head on the center of the track asprecisely as possible, referred to as track-following con-trol, so that data can be read/written quickly and reliably.The implementation of the servo controller relies on theposition error signal (PES), which is obtained by read-ing the position information encoded on the disk’s datatracks.

–

d

GCPESr +

GPxP

n

Fig. 32.18 Disk drive servo control

32.3.1 Introduction to Disk Drive ServoControl

The position error is also called track mis-registration(TMR) in the disk drive industry. Major TMR sourcesin the track-following mode include spindle runout,disk fluttering, bias force, external vibration/shock dis-turbance, arm and suspension vibrations due to airturbulence, PES noise, written-in repeatable runout, andresidual vibration due to seek/settling [32.29]. Thesesources can be categorized as runout, r, input distur-bance, d, and measurement noise, n, by the locationsthey are injected into the control system, as shown inFig. 32.18. In Fig. 32.18, G P(s) and GC(s) representthe disk drive actuator and the controller, respectively;r and xp represent track runout and head position, re-spectively.

From Fig. 32.18, the PES can be written as

PES = S(s)r + S(s)G P(s)d − S(s)n , (32.9)

where S(s) is the closed loop sensitivity function definedby

S(s) = 1

1+ GC(s)G P(s). (32.10)

The higher the bandwidth of the control system, thehigher the attenuation of the sensitivity function S(s)below the bandwidth. Thus, one of the most effectivemethods to reduce PES and increase servo precision isto increase the control system bandwidth.

Traditional disk drive servo systems utilize a singlevoice coil motor (VCM) to move the head. Multiplestructural resonance modes of the E-block arm andthe suspension located between the pivot and the headimpose a major limitation on the achievable controlbandwidth. A dual-stage servo system using a MEMSactuated-slider microactuator can achieve high band-width because the microactuator is located between thesuspension and the slider, thus by-passing the pivot,E-block arm, and suspension resonance modes.

One basic task of dual-stage servo control designis to increase control bandwidth using the second stageactuator. Many design methodologies have been devel-oped to accomplish this objective. In this section, we firstreview major dual-stage servo control design method-ologies, and then we discuss design considerations forcontroller design of a MEMS microactuator dual-stageservo system. As design examples, the details of track-following control designs using a sensitivity functiondecoupling design method and the µ-synthesis designmethod are presented in Sect. 32.3.3. In Sect. 32.3.4,

PartE

32.3


x1C1

er

–

+

G1

xp

C2 G2xr

Fig. 32.19 Master-slave design structure

x1C1

er

–

+

G1

xp

C2 G2xr

+

Fig. 32.20 Decoupled control design structure

x1C1

er

–

+

G1

xp

xrC2 G2

C0

Fig. 32.21 PQ control design structure

x1C1

er

–

+

G1

xp

xrC2 G2

Fig. 32.22 Parallel control design structure

we introduce a short-span seek control scheme usinga dual-stage actuator based on decoupled feed forwardreference trajectories generations.

32.3.2 Overview of Dual-Stage ServoControl Design Methodologies

Various control design architectures and methodologieshave been developed for dual-stage servo control design.They can be largely classified into two categories: thosebased on decoupled or sequential single-input-single-output (SISO) designs, and those based on modernoptimal design methodologies, such as LQG, LQG/LTR,

H∞, and µ-synthesis, in which the dual-stage controllersare obtained simultaneously.

Two constraints must be considered in dual-stageservo control design. First, the contribution from eachactuator must be properly allocated. Usually the firststage actuator, or the coarse actuator, has a large movingrange but a low bandwidth while the second stage actu-ator, or the fine actuator, has a high bandwidth but smallmoving range. Second, any destructive effect, in whichthe two actuators fight each other by moving in oppositedirections, must be avoided.

Classical SISO Design MethodologiesSeveral architectures and design methodologies havebeen proposed to transform the dual-stage control de-sign problem into decoupled or sequential multipleSISO compensators design problems, for example,master-slave design, decoupled design [32.30], PQmethod [32.31], and direct parallel design [32.32].Figure 32.19 to 32.22 shows the block diagrams of dual-stage controller designs using these methods. In thesefigures, G1 and G2 represent the coarse actuator (VCM)and fine actuator (microactuator) respectively; x1 is theposition of the coarse actuator; xr is the position of thefine actuator relative to the coarse actuator; xp is the to-tal position output; r is the reference input (runout), andPES is the position error, e = r − xp.

Master-slave Design. In a traditional master-slavestructure, the absolute position error is fed to the fineactuator, and output of the fine actuator is fed to thecoarse actuator, as shown in Fig. 32.19. The positionerror will be compensated by the high bandwidth fineactuator. The coarse actuator will follow the fine actuatorto prevent its saturation.

Decoupled Design. Figure 32.20 shows the decoupleddesign structure [32.30], which is similar to the master-slave structure. Instead of feeding just the fine actuatoroutput to the coarse actuator, however, the summationof the fine actuator output and the position error is fed,which equals the position error of the coarse actuator.A very nice feature of this structure is that the con-trol system is decoupled into two independent controlloops, and the total sensitivity function is the prod-uct of the sensitivity functions of each of the controlloops [32.13]. Thus the two compensators C1 and C2can be designed independently. Decoupled design isalso called decoupled master-slave design, or sensitiv-ity function decoupling design. In Sect. 32.3.3, we will

PartE

32.3


discuss details of a track-following controller designedusing this method.

Both the master-slave and the decoupled designs usethe relative position of the fine actuator, xr. If a relativeposition sensor is unavailable, xr can be estimated usinga model of the fine actuator.

PQ Design Method. The PQ method is another innova-tive design technique for control design of dual-input,single-output system [32.31]. A block diagram of a dual-stage control design using this method is shown inFig. 32.21.

In PQ design, P is defined by

P = G1

G2, (32.11)

and a dual-stage controller can be designed in two steps.The first involves the design of an auxiliary compen-sator Q for plant P, which is defined by

Q = C1

C2. (32.12)

Q is designed to parameterize the relative contributionof the coarse and fine actuators. The 0 dB crossoverfrequency and phase margin of the open loop trans-fer function PQ are the design parameters in the designof Q. At frequencies below the 0 dB crossover frequencyof PQ, the output is dominated by the coarse actuator,while at frequencies above the 0 dB crossover frequency,the output is dominated by the fine actuator. At the 0 dBcrossover frequency, the contributions from the two actu-ators are equal. A large phase margin of PQ will ensurethat the two actuators will not fight each other whentheir outputs are close in magnitude, thus avoiding anydestructive effects.

The second step in the PQ design methodology isto design a compensator C0 for SISO plant PQ suchthat the bandwidth (crossover frequency), gain margin,phase margin, and error rejection requirements of theoverall control system are satisfied.

Direct Parallel Design. It is also possible to designthe dual-stage controller directly using a parallel struc-ture, as shown in Fig. 32.22, by imposing some designconstraints and by sequential loop closing [32.32].

The two constraints for parallel design in terms ofthe PES open loop transfer functions are [32.32]:

C1(s)G1(s)+C2(s)G2(s) → C2(s)G2(s) , (32.13)

at high frequencies and

|C1(s)G1(s)+C2(s)G2(s)| � |C2(s)G2(s)| ,

(32.14)

at low frequencies. The first constraint implies that openloop frequency response of the dual-stage control systemat high frequencies approximately equals that of the fineactuator control loop. Thus the compensator C2(s) canbe first designed independently as a SISO design prob-lem to satisfy the bandwidth, gain margin, and phasemargin requirements of the dual-stage control system.Compensator C1(s) can then be designed for the SISOplant model with the fine actuator control loop closed,such that the low frequency constraint and overall stabil-ity requirement are satisfied. This model is defined by:

G(s) = G1(s)

1+C2(s)G2(s). (32.15)

A dual-stage controller for a MEMS dual-stage ac-tuator servo system has been designed using this methodand implemented at IBM. An open loop gain crossoverfrequency of 2.39 kHz with gain margin 5.6 dB andphase margin 33◦ was obtained. In experimental testing,a 1-σ TMR of 0.024 µm has been achieved [32.32].

Modern MIMO Design MethodologiesSince the dual-stage actuator servo system is a MIMOsystem, it is natural to utilize modern MIMO opti-mal design methodologies, such as LQG, LQG/LTR,H∞, and µ-synthesis, to design the dual-stage con-troller. Usually MIMO optimal designs are based onthe parallel structure shown in Fig. 32.22, augmentedwith noise/disturbances models and other weightingfunctions to specify the control design performanceobjectives.

Linear quadratic Gaussian (LQG) control combinesa Kalman filter and optimal state feedback controlbased on the separation principle. But, the Kalmanfilter weakens the desirable robustness properties ofthe optimal state feedback control. Linear quadraticGaussian/loop-transfer recovery (LQG/LTR) control re-covers robustness by a Kalman filter redesign process.Dual-stage control designs using LQG and LQG/LTRhave been reported in [32.33–35], and so on.

The LQG design methodology minimizes the H2norm of the control system, while the H∞ designmethodology minimizes the H∞ norm of the controlsystem. µ-synthesis design methodology is based on theH∞ design and accounts for plant model uncertaintiesduring the controller synthesis process with guaranteedrobustness. Dual-stage control designs based on the H∞and µ-synthesis design methodologies have been re-ported in [32.33, 36, 37], and so on. Some details ofa control design using µ-synthesis for the MEMS micro-

PartE

32.3


actuator dual-stage servo system will be presented inSect. 32.3.3

Other advanced control theories also have been ap-plied to dual-stage servo control designs, such as slidingmode control [32.38], neural networks [32.39], and soforth.

32.3.3 Track-Following Controller Designfor a MEMS MicroactuatorDual-Stage Servo System

MEMS microactuators for dual-stage servo applicationare usually designed to have a single flexure reso-nance mode between 1–2 kHz [32.8,40]. This resonancemode is usually very lightly damped and can havea ±15% variations due to the differences in the fabrica-tion processes. Since the uncertain resonance frequencyis relatively low and close to the open loop gain crossoverfrequency of the control system, robustness to thesevariations must be considered in the controller design.

RPES

xVGVxp

GM

τp

τ*p Fig. 32.23 Dual-stage controlplant

RPES

VPES

KRR GR

+

PESr+

GVxV

GM

KV

l/KRSKMxp

– –

Fig. 32.24 Decoupled dual-stage control design block dia-gram

r VPES

GR

PES

RPESKMKVGV

SV SM

ST

– –

Fig. 32.25 The sensitivity block diagram

Controller Design Considerations for MEMSMicroactuator Dual-stage Servo System

Capacitive sensing can be utilized in MEMS micro-actuators to measure the position of the microactuatorrelative to the VCM actuator [32.14, 40]. But, this re-quires additional sensing electronics and wires to andfrom the head gimbal assembly (HGA), which result inadditional fabrication and assembly costs. Thus whetheror not a relative position sensor will be used in MEMSdual-stage servo systems is still an open question. If themicroactuator’s resonance mode is lightly damped, itis susceptible to airflow turbulence and external distur-bances. The relative position signal can be utilized todamp the microactuator’s resonance mode. Dual-stagecontrol can be classified as MIMO or SIMO designs ac-cording to the availability of the relative position signal.

Given the fact that the inertia of the MEMS micro-actuator is very small compared to that of the VCM, theeffect of the motion of the microactuator on the VCM canbe neglected. By feeding the control input of the VCMto the input of the microactuator with a proper gain, wecan cancel the coupling effect of the VCM on the micro-actuator [32.13]. Thus the dual-stage control model canbe decoupled as the sum of the outputs of the VCMand the microactuator, as shown in Fig. 32.23, in whichGV, GM are the VCM and microactuator models respec-tively, and xp is the absolute position of the read/writehead, which equals the absolute position of the VCM,xv, and the position of the microactuator relative to theVCM, RPES.

Decoupled Track-Following Control Designand Self-Tuning Control

Figure 32.24 shows a block diagram used for decoupledcontrol design of MEMS microactuator dual-stage servosystems [32.13].

Sensitivity Function Decoupling Design Methodology.The part enclosed in the dashed box on the upper-rightcorner is the dual-stage plant model shown in Fig. 32.23.In the figure, GV and GM are the VCM and micro-actuator models, respectively; xP is the absolute positionof the read/write head; xv is the absolute position ofthe VCM actuator; and RPES is the position of themicroactuator relative to the VCM; r represents the trackrunout, and PES is the position error signal of the headrelative to the data track.

The decoupling control approach, originally intro-duced in [32.30], utilizes the PES and RPES signals togenerate the position error of the VCM relative to the

PartE

32.3


data track center, labeled as VPES,

VPES = PES + RPES = r − xv , (32.16)

and this signal is fed to the VCM loop compensator.As shown in the block diagram, three compen-

sators need to be designed: the VCM loop compensatorKV; the microactuator PES loop compensator KM;and the microactuator RPES minor loop compensatorKMM = KRR/KRS, which is used to damp the micro-actuator’s flexure resonance mode and place the closedloop poles of the microactuator RPES loop at appro-priate locations. The damped microactuator closed looptransfer function GR, shown in the lower-middle dashedbox, is defined by

GR = GM

KRS + GM KRR, (32.17)

while the total dual-stage open loop transfer functionfrom r to xP, GT, is

GT = KV GV + KM GR + KM GR KV GV . (32.18)

It can be shown that the block diagram in Fig. 32.24is equivalent to the sensitivity block diagram shown inFig. 32.25, and the total closed loop sensitivity functionfrom r to PES equals the product of the VCM and micro-actuator loop sensitivities, SV and SM, respectively:

ST = 1

1+ GT= SV SM , (32.19)

where

SV = 1

1+ KV GV, SM = 1

1+ KM GR. (32.20)

Thus the dual-stage servo control design can be de-coupled into two independent designs: the VCM loopand the microactuator loop design. The VCM loop sen-sitivity can be designed using traditional single-stageservo design methodologies. The microactuator loopsensitivity is designed to expand the bandwidth andincrease the attenuation of low frequency runout anddisturbances.

0 (db)

0 (db)

0 (db)

ωVO ωMR ωVO ωMC

ωVO

ωMR

ωVO

ωMC

AS

Fig. 32.26 Illustration of dual-stage sensitivity ST design

Closed Loop Sensitivity Design by Pole Placement. Fora typical sampled second order system with the discretetime transfer function

G(

q−1)

= q−1 Bo(q−1

)

Ao(q−1

) , (32.21)

where q−1 is one step delay operator; Bo(q−1

)is

the plant zero polynomial, and Ao(q−1

) = 1+a1q−1 +a2q−2 is the plant pole polynomial. The closed loopsensitivity function can be designed by pole placement,solving the following Diophantine equation [32.41]:

Ac

(q−1

)= Ao

(q−1

)S

(q−1

)

+q−1 Bo

(q−1

)R

(q−1

), (32.22)

where Ac(q−1

)is the desired closed loop characteristic

polynomial. The closed loop sensitivity function, GS, is

GS

(q−1

)= Ao

(q−1

)S

(q−1

)

Ac(q−1

) , (32.23)

PartE

32.3


and the discrete time controller C(q−1

)is

C(

q−1)

= R(q−1

)

S(q−1

) . (32.24)

Ac(q−1

)can be chosen such that the required band-

width and system response are satisfied. Usually it ismore intuitive to describe it with its continuous timeequivalent parameters: the damping ratio, ζ , and the nat-ural frequency, ωn . The damping ratio is directly relatedto the phase margin of the open loop transfer function.For a typical design, it can be chosen to be equal toor greater than one, in order to ensure adequate phasemargin. ωn is related to the control system’s bandwidth,which is limited by sampling frequency and time delay,or the high frequency structural dynamics of the system.

Dual-Stage Sensitivity Function Design. The com-pensators in the dual-stage servo system depicted inFig. 32.24 can be designed by a two-step design process,as illustrated in Fig. 32.26.

First, the VCM loop compensator KV is designedto attain a desired VCM closed loop sensitivity SV, asshown in the top part of Fig. 32.26. Its bandwidth, ωVCin Fig. 32.26, is generally limited by the E-block andsuspension resonance modes. The design of this com-pensator can be accomplished using conventional SISOfrequency shaping techniques.

The second step of the design process involves thedesign of the microactuator loop compensators to at-tain additional attenuation SM, as shown in the middlepart of Fig. 32.26. This step is itself accomplished intwo steps. First, the minor RPES loop compensator,KRR/KRS, is determined in order to damp the micro-actuator resonance mode and place the poles of GR, orequivalently ωMR in Fig. 32.26, at desired locations. The

VPES

RPES

KV

PES+

+

– –

KM

KRR

l/KRR

Tuning

GV

GM

GR

r uMR uM

xV

xp

Fig. 32.27 Self-tuning control of the microactuator

compensator can be obtained by solving a Diophantineequation by pole placement as discussed in the previoussubsection. The poles of GR will become the zeros ofthe microactuator loop sensitivity function, SM.

Finally, the PES loop compensator KM is designedto place the poles, or equivalently. ωMC in Fig. 32.26,of the microactuator loop closed loop sensitivity SM.ωMC is limited by the PES sampling frequency andcomputational time delay [32.42].

The total dual-stage sensitivity is shown in the bot-tom part of Fig. 32.26. For a given ωMC, the additionalattenuation AS, provided by the microactuator loop, willbe determined by ωMR. In our proposed procedure, theinitial value of ωMR can be chosen to be the same as ωVC.It is then adjusted so the desired attenuation and phasemargin requirements of the overall dual-stage system aresatisfied. Decreasing ωMR increases the low frequencyattenuation of the closed loop sensitivity function ST butalso generally reduces the phase margin of open looptransfer function GT.

For SIMO design when the RPES is not available,the microactuator model can be used as an open loopobserver to estimate the RPES. The combination of theopen loop observer and the minor loop compensator isequivalent to a notch filter.

Self-Tuning Control to Compensate Variations in theMicroactuator’s Resonance Mode. An adaptive controlscheme can be combined with the decoupled discretetime pole placement design methodology describedabove to compensate variations in the microactuator’sresonance mode by tuning the microactuator RPESinner loop compensator.

A block diagram for the microactuator inner loopself-tuning control is shown in Fig. 32.27. The param-eter adaptation algorithm (PAA) is a direct self-tuningalgorithm, based on the microactuator inner loop poleplacement design.

Consider the microactuator open loop transfer func-tion in (32.21). Since the microactuator’s resonancemode is lightly damped, the zero of the microactuator’sdiscrete time transfer function is very close to 1 (zo ≈ 1).Thus it is possible to factor out the “known” term(1+ zoq−1) from the Diophantine equation (32.22). Theresulting RPES minor-loop closed loop dynamics isgiven by

Ac

(q−1

)y(k) = q−1b0

(1+ zoq−1

)

×[

KRS

(q−1

)uMR(k)+ KRR

(q−1

)y(k)

],

PartE

32.3


where uMR is the control input to the microactuator;y denotes the position of the microactuator relative tothe VCM, i.e. y = RPES. Defining:

S(

q−1)

= b0KRS

(q−1

)= s0 + s1q−1 , (32.25)

R(

q−1)

= b0KRR

(q−1

)= r0 +r1q−1 , (32.26)

the regressor vector, φ(k) and filtered regressor vec-tor, φf(k),

φ(k) = [uMR(k) uMR(k −1) y(k) y(k −1)

],

(32.27)

Ac

(q−1

)φf(k) =

(1+ zoq−1

)φ(k) , (32.28)

and the controller parameter vector θ = [s0 s1 r0 r1]T ,the closed loop RPES dynamics (32.25) can be rewrittenas

y(k) = θT φf(k −1) . (32.29)

From (32.29), the controller parameter vector esti-mate θ(k) = [s0(k) s1(k) r0(k) r1(k)]T can be updatedusing a standard recursive least square algorithm(RLS) [32.41]:

θ(k) = θ(k −1)+ P(k)φf(k)eo(k) , (32.30)

eo(k) = y(k)− θT (k −1)φf(k −1) , (32.31)

P(k) =[

P(k −1)−

P(k −1)φf(k −1)φTf (k −1)P(k −1)

1+φTf (k −1)P(k −1)φf(k −1)

],

(32.32)

K VCM

MARPES

xV

RPES

PES

RPES

PES

r xp

uVCM

dVCM

uVCMuMA

dMA

uMA

nRPES

nPES

WRPES

WNRPES WDMA

WUMA

WDVCMWU1

WUVCM

WNPES

WPES

+

++

+–

δ1

δ1

WRr

Fig. 32.28 µ-synthesis de-sign block diagram

The control law is

S(

k, q−1)

uMR(k)

= So(k)uM(k)− R(

k, q−1)

y(k) , (32.33)

with

S(

k, q−1)

= s0(k)+ s1(k)q−1 , (32.34)

R(

k, q−1)

= r0(k)+ r1(k)q−1 , (32.35)

and uMR(k) being the output of the microactuator fixedouter loop compensator KM, uM(k) being the controlinput to the microactuator.

Dual-Stage Track-Following Control DesignUsing µ-Synthesis

µ, the structured singular value, is a measure of how biga perturbation to a system needs to be to make the systemunstable. By incorporating a fictitious uncertainty block,robust performance in terms of H∞ norm of the closedloop transfer function can be related to the value of µ.µ-synthesis is a robust optimal controller design tech-nique that attempts to minimize µ through an iterativeprocess [32.43].

Design Methodology. For controller design usingµ-synthesis, the model uncertainties are representedusing linear fractional transformations (LFT). Distur-bances and outputs are weighted to characterize thereal plant environment and the performance require-ments. A block diagram used for dual-stage µ-synthesiscontroller design is shown in Fig. 32.28.

PartE

32.3


In the block diagram, two model uncertainties,δ1 and δ2, are considered. δ1 is an additive uncertaintyused to describe the VCM unmodeled dynamics. δ2 rep-resents the parametric uncertainty in the stiffness of themicroactuator.

Disturbance signals accounted for in the model in-clude the track runout r, input disturbances to the VCM,and microactuator, dVCM and dMA, respectively, the PESsensor noise nPES and the RPES sensor noise nRPES.These disturbance signals are generated by passing nor-malized signals r, dVCM, dMA, nPES, and nRPES throughweighting functions Wr , WdVCM , WdMA, Wn PES, andWnRPES, respectively, which can be either constants orfrequency shaping filters. These weights are selectedby the designer so that disturbances are modeled withsufficient fidelity.

The output signals in the synthesis model are headposition error signal, PES, the microactuator relative po-sition signal, RPES, the VCM control input uVCM, andthe microactuator control input uMA. These signals aremultiplied by scaling factors, WPES, WRPES, WuVCM,and WuMA, respectively, to produce the weighted per-formance signals PES, RPES, uVCM, uMA. The scalingfactors are selected to characterize the performancerequirements.

Given a set of input and output weights and plantuncertainties, if the synthesized controller achieves

µ ≤ β , (32.36)

the closed loop transfer function, T , from the normalizeddisturbances

d = [r dVCM dMA nPES nRPES

], (32.37)

to the weighted performance signals

e = [PES RPES uVCM uMA

], (32.38)

will have infinity norm

∥∥T∥∥∞ ≤ β , (32.39)

for perturbations

‖∆‖∞ =∥∥∥∥∥

[δ1 0

0 δ2

]∥∥∥∥∥∞≤ 1

β. (32.40)

The interpretation of H∞ as the RMS gain of thesinusoid to help understand the design is as follows.Assume that each element di of the disturbance input

vector in (32.37) is a sinusoid of the form

di(t) = Di sin(ωi t +ψi) , (32.41)

such that5∑

i=1

D2i ≤ 1 . (32.42)

Then the steady state response of output in (32.38) willalso be a sinusoid of the form

ei(t) = Ei sin(ωi t +φi) , (32.43)

and4∑

i=1

E2i ≤ β , (32.44)

under the perturbations in (32.40).

Weighting Functions for µ-Synthesis Design. Perfor-mance weights are chosen based on limits that the errorsignals should not exceed. The limit on the PES is basedon a rule of thumb commonly used in the disk drive in-dustry which states that the position error of the headshould not exceed 1/10 of a track width during trackfollowing. RPES is limited by the stroke limit of themicroactuator. Thus,

WPES = 10

track width; WRPES = 1

stroke limit.

(32.45)

Weights on control inputs are based on the control inputsaturations.

WUVCM = 1

(iV)max; WUMA = 1

(vM)max,

(32.46)

where (iV)max and (vM)max are the maximum drive cur-rent of the VCM and the maximum drive voltage of themicroactuator, respectively.

The weights for PES noise, nPES, and the dis-turbance to the VCM, wdVCM, were determined byextrapolating the PES decomposition results presentedin [32.29]. The weight of the RPES sensing noise,nRPES, is assumed to be 8 nm [32.14]. The disturbanceweight, wdMA, is assumed to be 0.1 V in the design.

The runout weighting is a low pass filter that cap-tures the combined effects of track runout, mechanicalvibrations, and bias force disturbance to PES,

Wr = 2∗10−8(s +2π ∗10)2

(s +2π ∗1000)2 . (32.47)

PartE

32.3


100 101 102 103 104

100

50

0

– 50

– 100

0

– 50

– 100

– 150

– 200

– 250

– 300

Frequency (rad/s)

Magnitude (dB) Bode diagrams

Phase (deg)

100 101 102 103 104

150

100

50

0

– 50

0

– 50

– 100

– 150

– 200

– 250

– 300

Frequency (rad/s)


Phase (deg)

100 101 102 103 104

50

0

– 50

– 100

– 150

300250200150100

0– 50

– 100

Frequency (rad/s)


Phase (deg)

100 101 102 103 104

50

0

– 50

– 100

– 150

300250200150100

0– 50

– 100

Frequency (rad/s)


Phase (deg)

SIMO

MIMO

MIMO

SIMO

MIMO

SIMO

To: Y(1)

From: U(1)

SIMO

MIMO

a) b)

c) d)

Fig. 32.29 (a) Open loop Bode plot of the decoupled design; (b) open loop Bode plot of the µ-synthesis design; (c) closed loopsensitivity Bode plot of the decoupled design; (d) closed loop sensitivity Bode plot of the µ-synthesis design

The combination of WPES and Wr determines the closedloop sensitivity function from r to PES, S(s), of thedual-stage system.

‖WPES S(s)Wr‖∞ ≤ β . (32.48)

If β = 1, the magnitude of bode plot of S(s) lies belowthat of 1/WPESWr.

For SIMO design, the controller to be synthesizedhas only one input. In the synthesis model, RPES is notfed to the controller K , and RPES noise nr pes and itsweights WnRPES are not used.

Design and Simulation ResultsFigure 32.29 shows the Bode plots of the open looptransfer functions from r to xp p and the closed loopsensitivity functions of the two designs, respectively.Sampling frequency was chosen to be 20 kHz in bothdesigns.

For the decoupled MIMO design, the gain crossoverfrequency, gain margin, phase margin of the open loopsystem are 2,201 Hz, 9.1 dB, and 48.8◦, respectively. For

PartE

32.3


0 0.005 0.01 0.015 0.02 0.025 0.03 0.035 0.04 0.045

422

– 2

0 0.005 0.01 0.015 0.02 0.025 0.03 0.035 0.04 0.045

– 4– 6– 8

– 10

0 0.005 0.01 0.015 0.02 0.025 0.03 0.035 0.04 0.045

1.5

1

0.5

0 0.005 0.01 0.015 0.02 0.025 0.03 0.035 0.04 0.045

10.5

0– 0.5

r0

tuneddesired

r1

s0

s1

Time (s)

Fig. 32.30 Control parameters adaptation response

the decoupled SIMO design, they are 2,381 Hz, 8.7 dB,and 30.6◦, respectively.

For µ-synthesis MIMO design, the gain crossoverfrequency, gain margin, phase margin of the open loopsystem are 2,050 Hz, 9.5 dB, 66.0◦, respectively. Forµ-synthesis SIMO design, they are 2,223 Hz, 2.9 dB,32.0◦, respectively.

The above controller designs have not been im-plemented on a MEMS microactuator dual-stage servosystem. Controllers designed using both methods havebeen tested on a PZT actuated suspension dual-stageservo system, however, validating their effective-ness [32.44].

Figure 32.30 shows the simulation of control param-eters adaptation for the self-tuning controller, when thereal microactuator resonance frequency is 1.2 times itsnominal value. The controller parameters converged totheir desired values. Similar responses were obtainedwhen the real resonance frequency is 0.8 times thenominal value.

32.3.4 Dual-Stage Seek Control Design

Because the inertia of a microactuator is much smallerthan that of the VCM, it can produce a largeracceleration and move faster than the VCM. The mo-tion range of a microactuator is usually limited to

xV

xp

VPESKV

PES+

+

–

–

KM

KRR

l/KRS

GV

GM

+KRF

KVF

xdV

xdR xd

P

–

+

+

RPES

Fig. 32.31 Dual-stage seek control design

Dual-stage one track seek

Time (s)

Position (m)12

10

8

6

4

2

0

– 2

× 10–7

0 0.2 0.4 0.6 0.8 1 1.2 1.4 1.6

RPES

xPxV

Fig. 32.32 Dual-stage short span seek response

a few micrometers, however. Thus the performanceimprovement in seek control made possible by using

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32.3

TS0 32.4 Conclusions and Outlook 979

dual-stage actuation will mainly be in short distanceseeks.

Two degree-of-freedom (DOF) position controlhas been a very popular control technique in shortdistance seeks. A 2-DOF control technique utiliz-ing decoupled feed forward reference trajectories hasbeen developed for short span seek control usinga PZT actuated suspension dual-stage servo sys-tem [32.45]. The same technique can also be applied toseek control of MEMS microactuator dual-stage servosystem.

Figure 32.31 shows a block diagram for dual-stageshort span seek control design using this method. Inthe figure, xd

V and xdR are the desired seek trajecto-

ries of the VCM and the microactuator respectively.KVF and KRF are Zero Phase Error Tracking FeedForward Controllers (ZPETFFC) generated with theVCM model GV and the damped microactuator modelGR [32.46].

To minimize the residual vibration after seek opera-tion, minimum jerk seek trajectories were applied to boththe VCM and the microactuator. These can be generated

by [32.47]

xdV(t)=60 ds

[1

10

(t

TV

)5

− 1

4

(t

TV

)4

+ 1

6

(t

TV

)3]

,

(32.49)

xdR =

60(ds − xd

V(T))

×[

110

( tT

)5− 14

( tT

)4+ 16

( tT

)3]

t ≤ T

ds − xdV(t) t > T

,

(32.50)

where ds is the distance of the head from the targettrack; T is the time when the head reaches the targettrack if dual-stage actuator is used, while TV is the timewhen the head reaches the target track if only VCM is

used. TV and T can be chosen based on the control forcesaturation and seek performance requirements.

Figure 32.32 shows the 1 µm seek responses of thedual-stage actuator for the MIMO design. Overshoot waseliminated, and no obvious residual vibrations occured.The seek time using the dual-stage actuator is about0.25 ms, compared to a seek time of about 0.7 ms if onlythe VCM is used.

32.4 Conclusions and Outlook

In order to increase hard disk drive storage density,high bandwidth dual-stage servo systems are necessaryto suppress disturbances and increase servo precision.Various prototype MEMS microactuators have been de-signed and fabricated to provide a dual-stage actuationsystem. The most common approach for these proto-types is to use electrostatic driving arrays produced byMEMS fabrication methods.

MEMS microactuators provide a potentially highperformance and low cost solution for achieving theservo requirements for extremely high HDD storagedensity. Simulations and experiments show that manyMEMS microactuator features, such as integrated sil-icon gimbals and capacitive relative position sensingarrays, can meet important requirements for operationof HDDs. The use of structural electrical-isolation in themicroactuator design could considerably reduce micro-actuator driving voltages, while processes that avoid theuse of silicon-on-insulator wafers or high-temperatureprocessing steps could reduce manufacturing costs. Highbandwidth dual-stage track-following controllers havebeen designed using both a decoupled SISO design

method and robust µ-synthesis design method, amongothers. In addition, short-span seeking control using a2-DOF control structure with decoupled feed forwardtrajectory generation can greatly reduce the short spanseek time.

But research remains to be done in the areasof system integration, reliability, and performanceenhancement for this technology to be utilized in com-mercial products. Cost and reliability are probably themost important obstacles to commercial implementationof actuated slider dual-stage servo systems at this time.Streamlining of microactuator fabrication processes ordevelopment of a process compatible with that of theslider and head fabrication could reduce manufacturingcosts, which are not quite yet economical. Meanwhile,dynamic behavior and reliability of the microactuatorunder disturbances from airflow, head-disk interaction,and particle presence are being experimentally studiedby industry.

Further in the future, to achieve nanometer servoprecision, third-generation dual-stage servo systems willhave to employ an actuated head approach. Research in

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32.4


this area has just started, with the key problem being howto combine microactuator fabrication with read/writehead fabrication. Another area of research is the useof MEMS technology to incorporate additional sen-sors, such as accelerometers and strain gauge vibrationsensors, to suppress the TMR due to airflow and ex-ternal disturbances excited structural vibrations. Newrobust, adaptive MIMO control architectures and algo-rithms must be developed for such a multisensing andmultiactuation servo system.

Besides the application of MEMS microactuatorsin dual-stage servo control of traditional rotating me-dia magnetic HDDs, recent advances in MEMS andnanotechnology have made possible the developmentof a new class of miniaturized ultra-high density storagedevices that use probe arrays for recording and read-ing data. This new architecture abandons the traditional

rotating media and flying slider paradigm in favor of par-allel x-y scanning of entire probe arrays over a storagemedium. The probe arrays and/or the media are movedwith electrostatic or electromagnetic microactuators thatgenerate x-y in-plane relative motion, as well as z-axis out-of-plane relative motion. One example of thisnew generation of storage devices is IBM’s “millipede,”which utilizes atomic force microscopy (AFM) record-ing technology [32.48]. Data is recorded on a polymermedia by tiny depressions melted by a thermomechan-ical process by the AFM tips. 400 Gb/in.2 bit patternshave been demonstrated using thermal probes.

By any of the paths above, MEMS and nano-technology will become a basic enabling technology forthe development of future data storage devices, offeringthe capabilities of small size, low power consumption,ultra-high densities, low cost, and high performance.

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TS5 Please provide volume number.TS6 Please provide place of proceeding, editors, publisher, place and year of publication.TS7 Please provide place of proceeding, editors, and place and year of publication.TS8 Should “bicroactuator” be “microactuator”?


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951 32. Design, Fabrication and Control of …...951 32. Design, Fabrication and Control of MicroactuatorsDesign, Fabric for Dual-Stage Servo Systems in Magnetic Disk Files This chapter