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Silicon Carbide BJT’s in Boost ApplicationsEfficiency is becoming more and more important as well as size and cost. In boost DC/DC converters,typically used in PV inverters and PFC circuits, increased switching frequency makes a big impact on bothsize and cost. Silicon Carbide (SiC) bipolar junction transistors (BJT’s) offer low-loss high speed switchingcombined with low conduction losses enabling higher switching frequency and maintaining high efficiency.SiC BJT’s combine the best properties from Silicon unipolar and Bipolar technologies in a normally-offdevice. The design and performance of a 1kW boost circuit based on the SiC BJT is presented in this article.Peter Haaf and Martin Domeij, Fairchild Semiconductor Germany and Sweden

SiC BJTs using a vertical NPN structurewere fabricated and assembled in anindustrial standard TO-247 package. Thistype of transistor combines very lowconduction losses with fast and low lossswitching behavior [1]. The high criticalfield strength of SiC gives the possibility tohave low saturation voltages withoutdriving the transistor into hard saturation.Since there is no channel region in a BJTthe VCE(SAT) is determined mainly by thecollector series resistance.It is not necessary to drive the SiC BJT

into hard saturation which gives themexcellent switching properties. There is nocurrent tailing and a minimal storage delay(5 ns) at turn-off. The SiC BJT is also avery robust device with a wide RBSOA,good short circuit capability [2] and canoperate at high temperatures [3]. SiC BJT’shave suffered from bipolar degradation,

making the gain decrease significantly overtime [4]. The BJT’s discussed do not havethis problem (Figure 1).

Base drive circuit and the challengesA SiC bipolar transistor is a current drivendevice. The driver must deliver enoughcurrent to turn the device on and off, plusload and unload the Miller charges quicklyenough. Nevertheless the losses in thedrive circuit should be limited. One targetfor this development was to decrease thedriver losses as much as possible, withoutimpacting the switching speed.To have enough safety margin the base

current should be oversized by a factor of1.5 compared with the calculated basecurrent (Ib = beta • Ic). To simplify the basedrive circuit using a constant base currentis recommended.We developed a driver board with an

adjustable dual supply, Vcc and Vee, a highspeed opto-isolator and an additionalcapacitor C10 in parallel to the baseresistor R2 to boost the base current for ashort moment, just during turn-on andturn-off of the BJT.When the BJT is turned on, Vbe is equal

to 2.9 V with a slightly negativetemperature coefficient. The positivesupply voltage of the driver is divided intoa small voltage drop in the driver, the Vbe

and the voltage drop across the baseresistance R2. The capacitor C10 inparallel to R2 will be charged in thismoment to the voltage across R2. Whenthe device is turned off, this capacitanceprovides a short increase in the drivecurrent to increase the switching speed ofthe BJT.Two big obstacles for fast switching are

the parasitic inductances of the packageand the PCB. To reduce the influence ofthese parasitics the supply voltage for thedriver could be increased, but this wouldlead to significant higher drive lossesduring turn-on. The alternative would betwo separate drivers, one for fastswitching, the other for supplying the basecurrent. As this would make the circuitexpensive and complicated, we decided touse the additional boost capacitor C10 inparallel to the base resistance R2. In anycase, the negative supply for the drivercannot be avoided for fast switching.

Double pulse test setupThe base drive circuit has been optimizedwith the double pulse test setup. Turn-onand turn-off waveforms can be analyzedseparately without the need for a fullthermal design. The test set up is optimized, the

inductor is a 1 mH coreless choke and thecapacitors have been placed in shortest

Figure 1: Gain (hFE) and VCE(SAT) before and after 660 hour DC stress test (open collector, IBE=0.25 A)

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distance to the semiconductors in order toreduce the current loop inductance.By changing the position of the diode,

the same test setup can be used for theboost function. In this case the diode willcharge an additional bank of capacitorsconnected to the load. That means that atleast the primary side circuit and the drivecircuit are identical and the test results canbe compared easily.Some differences from SiC BJTs to

conventional switches are characterized bythe turn-on and turn-off waveforms shownin Figure 4 and 5. The drive circuit issupplied by Vcc = 7V and Vee = -3V. Thered curve M1 shows Vbe, which ismeasured by the differential signalbetween base and emitter lead. Whenturning the BJT on, the signal starts to risefrom -3V to +3V by charging Cbe. At thethreshold of 3 V the current starts tocommutate from the diode to the BJT at adi/dt of around 1000 A/µs.The relatively big Cbe capacitance gives

the BJT stability, as well at slightly noisygate signal. The first drop of Vce is causedby the induced voltage across the loopinductor (Vind = -Lloop • di/dt). As soon as the diode current drops to

zero, the Vce of the BJT starts to drop. Astrong base driver is now needed todischarge the Miller capacitance of theBJT.During turn-off, both the current and

voltage waveform start to change at thesame time. In the first nanoseconds thebase current must charge the Millercapacitance. As soon as Vce reaches thebus voltage, the change in the di/dt of thecurrent drop is visible. Due to the idealfeatures of the SiC material the influenceof the removal of the stored carrier chargeis negligible, this causes just a delay ofabout 5 ns in the begin of the turn-offprocedure.It is often said that TO247 packages

cannot handle the high switching speedsof the SiC devices. In order to doublecheck we performed some tests. As thedriver board is opto-isolated, theconnection to the BJT is rather flexible.The GND-trace of the driver is connectedto two different points of the emitter leadof the BJT.The most significant difference is visible

at highest currents and at maximum di/dt.During turn-on at a di/dt of 1000 A/µs,

Figure 2: Base drive circuit for SiC BJT

Figure 3: Double pulse test setup

Figure 4: Turn-on of the SiC BJT (Eon = 226 µJ,Ch3 magenta...Vce = 600V, Ch4, blue...Ic = 10 A)

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we measured an increase of Eon lossesfrom 117 µJ to 172 µJ and the peakvoltage drop across this lead has beenmeasured at 7 V. A similar effect canpresumably be seen when considering thevoltage drop across the emitter bond wireunder the similar conditions. The goodthing is the relatively large base-emittercapacitance which prevents the devicefrom a parasitic turn-off. The usage ofpackages with separated power-emitterpins and drive-emitter pin will solve thisproblem and will make the device easierto control.

The boost converterThree different boost configurations werereviewed. The first circuit was acontinuous conduction mode (CCM)boost circuit using the 12 A/1200 V SiCBJT and 20 A/1200 V SiC diode. This wastested at the two switching frequencies 20kHz and 40 kHz. The second circuit was aboundary conduction mode (BCM) boostcircuit using the same devices. The ZVSapproach of the BCM boost looks verypromising in comparison to the hardswitched CCM mode. Finally we operatedour high speed IGBT FGL40N120AND incombination with the SiC diode at 20 kHzin CCM mode as a reference.For the CCM operation we used a 10

mH coreless inductor. In order to reducethe drive losses the positive operationvoltage the driver Vcc was reduced to 6 V.The negative driver supply Vee has beenincreased to -6V without any impact onthe drive losses. The waveforms at maxpower at Vout = 812 V are shown inFigures 6 and 7.The big obstacle for BCM operation is

high ripple currents and varyingfrequencies. A typical power range of apractical 2-phase interleaved PFCapplication is about 1 kW down to 90VAC input voltage. Increasing the inputvoltage to 400 VAC, it is easily possibleto increase the power range to 4 kWwithout changing the current level. Bygoing from PFC to boost or from 2-phaseto 4-phase interleaved operation a powerlevel of 8 kW with this topology will beno problem. Figures 8 and 9 show thetest results.The very smooth decrease of the Vce

and the turn-on without any losses isclearly visible. The parasitic capacitance ofthe inductor and the junction capacitanceof the SiC diode limit the dV/dt in Figure 9

LEFT Figure 7: Eoff measurement in CCM (Eoff = 51µJ, M1 red...Vbe, M3 dark blue...Eon/Eoff)

LEFT Figure 5: Turn-off of the SiC BJT (Eoff = 60 µJ, M1, red...Vbe = Vb - Ve, M3, darkblue...Eon, Eoff)

ABOVE Figure 6: Eon measurement in CCM (Eon = 113 µJ, Ch3 magenta...Vce, Ch4 blue...Ic)

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to 5 kV/µs. All transitions look verysmooth and promising, especiallyregarding EMI performance. For theinductor we chose a Kaschke E-core withL= 0.68 µH and multistrand wire, which isoptimized for high di/dt.Finally we compared the very well

known behavior of a fast switching 1200VIGBT to the SiC BJT in CCM mode. Theidentical power circuit operates at 20 kHz,the driver is as well the FAN3122, but forIGBTs it operates between Vcc = 15V andGND. The gate resistor is equal to 4.7 �.The IGBT at a TB=100ns shows about 10xhigher Eoff losses. The Eon losses of the SiCBJT and IGBT are in the same range.The boost measurements have been

performed with a constant Vin = 210 VDC.The output voltage was adjusted bychanging the duty cycle or/and thefrequency in 100 V steps from Vout = 400V to 800 V. The load is a parallel andseries operation of incandescent bulbs of880 W in total. The BCM boost shows best results, the

CCM boost at 20 kHz and 40 kHzmedium and the Si IGBT comes third inthis competition. The higher efficiency forthe 40 kHz CCM circuit versus the 20 kHzCCM is attributed to the lower ripplecurrent seen in the inductor and capacitorat 40 kHz compared with the 20 kHzsolution. Figure 10 shows the measuredefficiency of the variations of the boostconverters. Actually we would have expected in

general a better total efficiency for allmeasurements. Similar to tests seen onvery low RDS(ON) MOSFETs in DC/DCapplications, the package and layouteffects start to dominate as the losses ofthe semiconductors become lesssignificant. Based on the accompanying

LEFT Figure 8: Smooth turn-on of the SiC BJT inZVS (Eon = 0 µJ, Vout = 800 V, Ch3 magenta...Vce,Ch4 blue...Ic)

Figure 9: Turn-off waveform in BCM at Vout = 800 V (Eoff = 54 µJ, M1 red...Vbe, M3 dark blue...Eon/Eoff)

Figure 10: Measured efficiency of the variations of the boost converters

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scope measurements for Eon and Eoff, aloss calculation for the switches has beenperformed (see table 1). The weak pointof the test setup is a symmetric EMI filter,which should prevent noise getting in thepower analyser. A big portion of the totallosses are dissipated in this device. Morethan 80 % of the losses are dissipated inthe passive components.

ConclusionThe SiC BJT offers significant efficiencyadvantages in boost applicationscompared with silicon IGBT technology.Similar to when using other highperformance technologies, the passive

and layout parasitic elements start todominate the losses. This article is derivedfrom a paper given at PEE’s Special PCIM2012 Session [5].

Literature[1] M. Domeij et. al., “2.2 kV SiC BJTs

with low VCESAT and fast switching” tobe published in the Proceedings of theInternational Conference on SiliconCarbide and Related Materials(ICSCRM),2009. [2] Y. Gao et. al., “Comparison of

static and switching characteristics of1200 V 4H-SiC BJT and 1200 V Si-IGBT”, IEEE Transactions on Industry

Applications, Volume 44, No. 3, p. 887,2008 [3] A. Lindgren, M. Domeij, “1200 V 6

A High Temperature SiC BJTs”,Proceedings of HiTEC Conference,2009. [4] M. Domeij et. al., “Degradation-

free 1200 V 6 A SiC BJTs with very lowVCESAT and fast switching”,Proceedings of PCIM Conference, 2009. [5] P. Haaf, M. Domeij, “Silicon

Carbide BJT’s in boost applications”,PEE Special Session “High FrequencySwitching Devices and Technologies forGreen Applications”, PCIM 2012,Nuremberg.

Table 1: Loss calculation of the switches in the application at Vin=210 V, Vout=800 V, Pout=735 W (assumption for conduction losses: SIC BJT Ron=100 m�; IGBT VCE(SAT)=1 V)

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