14A Non-Isolated Flyback Report

7/22/2019 14A Non-Isolated Flyback Report

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7.5W NON-ISOLATED FLYBACK CONVERTER

PROJECT REPORT

SUBMITTED IN PARTIAL FULFILLMENT OF THE REQUIREMENTS FOR THE

AWARD OF THE DEGREE OF

BACHELOR OF TECHNOLOGY

IN

ELECTRICAL AND ELECTRONICS ENGINEERING

BY

K.HARIBABU (07241A0234)

M.SRINIVAS (08245A0203)

J.BHEEMARAY (08245A0206)

Under the guidance of

Ms. U.VIJAYA LAXMI

Assistant Professor

Department of EEE

Department of Electrical and Electronics Engineering

Gokaraju Rangaraju Institute of Engineering and Technology

(Affiliated to Jawaharlal Nehru Technological University)

Hyderabad

2011


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GOKARAJU RANGARAJU INSTITUTE OF ENGINEERING

AND TECHNOLOGY

(Affiliated to Jawaharlal Nehru Technological University)

Hyderabad, Andhra Pradesh

Department of Electrical and Electronics Engineering

CERTIFICATE

This is to certify that the project report entitled

7.5W NON ISOLATEDFLYBACK CONVERTER

is being submitted by

K.HARIBABU (07241A0234)

M.SRINIVAS (08245A0203)

J.BHEEMARAY (08245A0206)

In partial fulfillment for the award of the Degree of Bachelor of Technology in

Electrical and Electronics Engineering from Jawaharlal Nehru Technological University,

Hyderabad during the academic year 2010-2011 is a bonafide record of work carried out by

them under our guidance and supervision.

Head of Department

Prof.P. M. SARMA

Professor

External Examiner Internal Guide

U.VIJAYA LAXMI

Assistant Professor


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ACKNOWLEDGMENT

It would be a great pleasure for us to express our thanks to Dr. S.N. Saxena, Professor and Dean

of placements, Department of Electrical and Electronics Engineering, Gokaraju Rangaraju

Institute of Engineering and Technology, who has supported us through the project work and

thesis.

We express our sincere thanks to Prof.P.M Sarma, head of the department of Electrical and

Electronics Engineering Department Gokaraju Rangaraju Institute of Engineering and

Technology, for his suggestions, motivation and encouragement to work with this project.

We are greatly indebted to our guide Ms.U.Vijaya Laxmi, Assistant professor, Department of

Electrical and Electronics Engineering, Gokaraju Rangaraju Institute of Engineering and

Technology, for her valuable guidance in presenting this project both in theoretical and practical

aspects and rendering us moral support.

We heartly thankful to all staff members in Electrical and Electronics Engineering who helped us

in preparing working model.

K.Haribabu

M.Srinivas

J.Bheemaray


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ABSTRACT

This project covers a closed loop controlled non-isolated flyback converter with

switching element MOSFET is switched ON and OFF using PWM controller UC494 with

switching frequency of 100kHz. The converter is operating from 25V battery (15V to 25V)

providing regulated output power at 15V (0.5A).

The flyback converter is a DC to DC converter in which the energy is received from the

input source when the switch is ON and then pumps energy into the output side during flyback of

the switch. Energy flow from the input side to output side when the switch is turned OFF that is,

when it is made to flyback, hence the name flyback converter. The non-isolated has no

transformer or electrical isolation, allowing it to be smaller and less expensive. Non-isolated

units which make up the majority of POL (point of load) converters, come in assorted board-

friendly compact package styles and are typically used in computing applications such as

memory motherboards. Non-isolated POLs offer a lower price for a given voltage and current

output. Further, they're usually housed in small-size formats like single-in-line packages (SIPs),

dual-in-line packages (DIPs), or surface-mount (SMT) modules. Their more compact packaging

yields significant real estate savings on the pc board and allows them to be placed close to their

loads. In turn, their close load placement reduces I2R copper losses on the board and improves

transient response.

This project consist of a combination of startup circuit, maximum pulse width circuit,

controller circuit, power circuit of power converter, feedback circuit. This circuits are used to

control of switching pulse width using feedback circuit and controller by PWM technique.

Advantages of this project areMost flexible low cost, low-power topology, with constant

Multiple DC voltage. Disadvantages of this project are switching losses are more, and cannot be

designed for high power rating due to isolation problems.

Applications of flyback converter are Low power switch-mode power supplies(cell

phone charger), High voltage generation(xenon flash lamps, lasers), High voltage supply for the

CRT in TVs, monitors, Low cost multiple-output power supplies(main pc supply less than

250W).


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ABBREVIATIONS

RT Timing Resistor

CT Timing Capacitor

N1 Number of Primary Turns

N2 Number of Secondary Turns

V1 Primary voltage of Inductor

V2 Secondary voltage of Inductor

PWM Pulse Width Modulation

TON On Time of The Switch

TOFF Off Time of The Switch

T Time Period

f Frequency

S Switch

VREF Reference voltage

MOSFET Metal oxide semi conductor field effect transistor

D Diode

PCB Printed circuit board

V0 Output voltage

TP Test point

V Volts

A Amperes

MMF Magneto Motive Force


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CONTENTS

Acknowledgement i

Abstract ii

Abbreviations iii

Contents iv

List of Figures viii

List of Figures ix

1. INTRODUCTION1.1. Switch mode conversion

1.2. Types of converters

1.2.1. Non-Isolated converter

1.2.2. Isolated converter

1

2

3

3

4

2. FLYBACK CONVERTERS

2.1. Fly-Back Converter

2.1.1. Basic Topology of Flyback converter

2.1.2. Principle of Operation of Flyback converter

2.2. Flyback circuit Waveforms

2.3. Practical Flyback converter

5

5

5

6

1112

3. PROJECT BLOCK DIAGRAM REPRESENTATION

3.1. Block diagram of project Circuit

(Non-Isolated flyback Converter)

14

14

4. START-UP AND MAXIMUM PULSE WIDTH CIRCUIT4.1. Start-up circuit

4.1.1 Operation of start-up circuit

4.2. Maximum pulse width circuit

1515

16

16


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5. CONTROLLER CIRCUIT

5.1 Pulse width modulation Technique

5.2 UC494c Controller

5.2.1. Pin configuration of UC494C controller

5.2.2. Features of UC494C controller IC

5.3 Controller circuit

5.3.1. Operation of controller circuit

17

17

18

18

19

19

20

6. Feedback circuit

6.1 Closed loop control

6.2 Compensator(LEAD/LAG)

21

21

22

7. POWER CIRCUIT

7.1 Main components of power circuit

7.2 MOSFET

7.2.1. MOSFET as a switch

7.2.2. MOSFET characteristics curve

7.3 Couple Inductor

7.4 Voltage regulator(LM7815)7.4.1. Features of LM7815

7.5 Diode(MUR110)

7.6 Circuit diagram of non-Isolated Flyback converter

Power circuit

7.6.1. Operation of Power circuit

24

24

24

25

26

30

3435

35

36

36

8. SIMULATION OF NON-ISOLATED FLYBACK CONVERTER

CIRCUIT IN MATLAB SOFTWARE

8.1. Simulation circuit description

8.1.1. Simulation circuit diagram

8.1.2. Simulation waveforms

8.1.3. Tabulation of input voltage and output voltage

37

37

37

38

39


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9. NON-ISOLATED FLYBACK CONVERTER HARDWARE KIT

9.1. Hardware kit PCB layout

9.2. PCB with components soldered

9.3. Practical output waveforms for 15Volts input supply

9.4. Practical output waveforms for 25Volts input supply

9.5. Tabulation of input voltage and output voltage With

voltage regulator(LM7815) on load side

40

40

41

42

44

47

10. APPLICATIONS AND LIMITATION

10.1 Applications of Non-Isolated Flyback converter

10.2 Limitations of Non-Isolated Flyback converter

48

48

48

11: CONCLUSION

11.1 Conclusion

11.2 Results

11.3 Difficulties encountered during the Project

50

50

50

50

12. SCOPE FOR FUTURE WORK

12.1 Scope for future work

51

51

References 52

Appendix

A: List of Project Components

B: Data sheet of UC494C

C: Data sheet of MOSFET (IRFZ44)

D: Datasheet of Voltage regulator(LM7815)

53

53

54

58

60


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LIST OF FIGURES

Fig.2.1 Flyback converter 5

Fig.2.2(a) Current path during Mode-1 of circuit operation 7

Fig.2.2(b) Equivalent circuit in Mode-1 7





Fig.2.5 Flyback circuit waveforms under continuous magnetic flux 11

Fig.2.6 Fly-back circuit waveforms under discontinuous magnetic

Flux

11

Fig. 2.7 Practical Fly Back Converter 12

Fig.3.1 Block diagram representation of Project Circuit 14

Fig.4.1 Startup circuit 15

Fig .4.2 Maximum pulse width circuit 16

Fig.5.1 Pulse width modulation technique 17


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Fig.5.2 Pin configuration of UC494 controller 18

Fig 5.3 Controller circuit 19

Fig.5.4 Connections of amplifier 20

Fig.6.1 Closed loop block diagram 21


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Fig.6.2 Feedback Circuit 23

Fig.7.1 Power MOSFET (a) Schematic , (b)Transfer characteristics,

(c)Device symbol 25

Fig.7.2 Symbol of MOSFETs 26

Fig.7.3 MOSFET characteristics curves 27

Fig.7.4 Cut-off region equivalent diagram 27

Fig.7.5 Saturation region equivalent diagram 28

Fig.7.6 MOSFET as a Switch 29

Fig.7.7 Couple inductors 33

Fig.7.8 Voltage regulator LM7815 35

Fig.7.9 Power circuit 36

Fig. 8.1 Simulation circuit of flyback converter 37

Fig. 8.2 Gate pulses 38

Fig. 8.3 Output voltage(16.6V) 38

Fig. 8.4 Output current(0.43A) 38

Fig.9.1 PCB layout of hardware kit 40

Fig.9.2 Hardware kit PCB with components 41

Fig.9.3 Ramp voltage (3V) for 15V input voltage 42


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Fig.9.4 Gate voltage(12V) for 15V input voltage 42

Fig.9.5 Output voltage(15.2V) for 15V input voltage 43

Fig. 9.7 Secondary Inductor Voltage for 15V input voltage 44

Fig. 9.8 Ramp voltage(3V) for 25V input voltage 44

Fig. 9.9 Gate voltage(20V) for 25V input voltage 45

Fig.9.10 Output voltage(15.2V) for 25V input 45

Fig. 9.11Primary Inductor voltage for 25V input voltage 46

Fig. 9.12Secondary Inductor voltage for 25V input voltage 46


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LIST OF TABLES

Table 8.1 Tabulation of input voltage and output voltage without voltage

regulator on the load side

39

Table 9.1 Tabulation of input voltage and output voltage 47


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CHAPTER 1

INTRODUCTION

Power electronic converters are a family of electrical circuits which convert electricalenergy from one level of voltage/current/frequency to another using semiconductor-based

electronic switches. The essential characteristic of these types of circuits is that the switches are

operated only in one of two states - either fully ON or fully OFF - unlike other types of electrical

circuits where the control elements are operated in a (near) linear active region. As the power

electronics industry has developed, various families of power electronic converters have evolved,

often linked by power level, switching devices and topological origins. The process of switching

the electronic devices in a power electronic converter from one state to another is called

modulation and the development of optimum strategies to implement this process has been the

subject of intensive international research efforts for at least 30 years. Each family of power

converters has preferred modulation strategies associated with it that aim to optimize the circuit

operation for the target criteria most appropriate for that family. Parameters such as switching

frequency, distortion, losses, harmonic generation and speed of response are typical of the issues

which must be considered when developing modulation strategies for a particular family of

converters.

DC to DC converters are important in portable electronic devices such as cellular phones

and laptop computers, which are supplied with power from batteriesprimarily. Such electronic

devices often contain several sub-circuits, each with its own voltage level requirement different

from that supplied by the battery or an external supply (sometimes higher or lower than the

supply voltage). Additionally, the battery voltage declines as its stored power is drained.

Switched DC to DC converters offer a method to increase voltage from a partially lowered

battery voltage thereby saving space instead of using multiple batteries to accomplish the same

thing.

Most DC to DC converters also regulate the output voltage. Some exceptions include

high-efficiency LED power sources, which are a kind of DC to DC converter that regulates the

current through the LEDs and simple charge pumps which double or triple the input voltage.


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Linear regulators can only output at lower voltages from the input. They are very

inefficient when the voltage drop is large and the current is high as they dissipate heat equal to

the product of the output current and the voltage drop; consequently they are not normally used

for large-drop high-current applications.

The inefficiency wastes power and requires higher-rated and consequently more

expensive and larger components. The heat dissipated by high-power supplies is a problem in

itself as it must be removed from the circuitry to prevent unacceptable temperature rises.

They are practical if the current is low, the power dissipated being small, although it may

still be a large fraction of the total power consumed. They are often used as part of a simple

regulated power supply for higher currents: a transformer generates a voltage which when

rectified, is a little higher than that needed to bias the linear regulator. The linear regulator dropsthe excess voltage, reducing hum-generating ripple current and providing a constant output

voltage independent of normal fluctuations of the unregulated input voltage from the transformer

or bridge rectifier circuit and of the load current.

Linear regulators are inexpensive, reliable if good heat sinking is used and much simpler

than switching regulators. As part of a power supply they may require a transformer, which is

larger for a given power level than that required by a switch-mode power supply. Linear

regulators can provide a very low-noise output voltage and are very suitable for powering noise-

sensitive low-power analog and radio frequency circuits. A popular design approach is to use an

LDO, Low Drop-out Regulator, that provides a local "point of load" DC supply to a low power

circuit.

1.1 SWITCHED-MODE CONVERSION

Electronic switch-mode DC to DC converters convert one DC voltage level to another, by

storing the input energy temporarily and then releasing that energy to the output at a different

voltage. The storage may be in either magnetic field storage components (inductors,

transformers) or electric field storage components (capacitors). This conversion method is more

power efficient (often 75% to 98%) than linear voltage regulation (which dissipates unwanted

power as heat). This efficiency is beneficial to increasing the running time of battery operated

devices. The efficiency has increased since the late 1980s due to the use of power FETs, which


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are able to switch at high frequency more efficiently than power bipolar transistors, which incur

more switching losses and require a more complicated drive circuit. Another important

innovation in DC-DC converters is the use of synchronous rectification replacing the flywheel

diode with a power FET with low "On" resistance, thereby reducing switching losses.

Most DC to DC converters are designed to move power in only one direction, from the

input to the output. However, all switching regulator topologies can be made bi-directional by

replacing all diodes with independently controlled active rectification. A bi-directional converter

can move power in either direction, which is useful in applications requiring regenerative

braking. Drawbacks of switching converters include complexity, electronic noise (EMI / RFI)

and to some extent cost, although this has come down with advances in chip design.

DC to DC converters are now available as integrated circuits needing minimal additionalcomponents. DC to DC converters are also available as a complete hybrid circuit component,

ready for use within an electronic assembly.

In these DC to DC converters, energy is periodically stored into and released from a

magnetic field in an inductor or a transformer, typically in the range from 300 kHz to 10 MHz.

By adjusting the duty cycle of the charging voltage (i.e., the ratio of on/off time), the amount of

power transferred can be controlled. Usually, this is applied to control the output voltage, though

it could be applied to control the input current, the output current, or maintain a constant power.

Transformer-based converters may provide isolation between the input and the output. In

general, the term "DC to DC converter" refers to one of these switching converters. These

circuits are the heart of a switched-mode power supply. Many topologies exist.

1.2 TYPES OF CONVERTERS

1.2.1 Non-Isolated Converters: Also called Point of Load converters, these step up or step

down voltage by a low ratio. These have ICs specifically meant for the purpose and a DC path

between its output and input. The four main types of non isolated converters are: Buck, Boost,

Buck-Boost and Cuk converters. While the Buck steps down the voltage, Boost steps it up.

Buck-Boost and Cuk are able to step up as well as step down the voltage.


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Types of Non-Isolated converters

1. Buck Converter (Step-Down Converter)

2. B C ( C)

3. BB C

4. C C

1.2.2 Isolated Converters: These converters are characterized by the presence of an

electrical barrier between the input and output. The barrier is provided by a high frequency

transformer, which can withstand a few hundred volts to several thousand volts. The output of an

isolated converter can be positive or negative and are useful in medical applications. These

devices are available in different types and configurations. The two basic types are flyback and

forward. Both these use the energy stored in the inductor's magnetic field for their operation.

Types of Isolated converters

1. F

2. F

1.2.2a Flyback converter: In this type of power supply converter, a transformer is used to store

energy, rather than a single inductor. It has two discrete phases for energy storage and output

delivery. The magnetic flux of the transformer core never reverses in polarity; hence, to avoid

the resultant magnetic saturation the core must be large enough for the given power level. These

are used in lower power applications, such as Cathode ray tubes and Geiger counter tubes which

draw lesser current.

1.2.2b Forward converter: The transformer transfers the energy between the input and the

output in a single step. This power supply converter can step-up or step-down voltage or offer a

combination of the two. For multiple outputs, all one needs to do is manipulate the turns on the

secondary winding. Applications include car amplifiers, where low battery voltage is stepped up

to obtain higher output for the amplifiers.


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CHAPTER 2

FLYBACK CONVERTER

2.1 FLYBACK CONVERTER

Fly-back converter is the most commonly used SMPS circuit for low output power

applications where the output voltage needs to be isolated from the input main supply. The

output power of flyback type SMPS circuits may vary from few watts to less than 100 watts. The

overall circuit topology of this converter is considerably simpler than other SMPS circuits. Input

to the circuit is generally unregulated dc voltage obtained by rectifying the utility ac voltage

followed by a simple capacitor filter. The circuit can offer single or multiple isolated output

voltages and can operate over wide range of input voltage variation. In respect of energy-

efficiency, fly-back power supplies are inferior to many other SMPS circuits but its simple

topology and low cost makes it popular in low output power range.

The commonly used fly-back converter requires a single controllable switch like, MOSFET and

the usual switching frequency is in the range of 100 kHz. A two-switch topology exists that

offers better energy efficiency and less voltage stress across the switches but costs more and the

circuit complexity also increases slightly. The present lesson is limited to the study of fly-back

circuit of single switch topology.2.1.1 Basic Topology of Flyback Converter: Fig.2.1 shows the basic topology of a

fly-back circuit. Input to the circuit may be unregulated dc voltage derived from the utility ac

supply after rectification and some filtering.

Fig.2.1 Flyback converter


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The ripple in dc voltage waveform is generally of low frequency and the overall ripple

voltage waveform repeats at twice the ac mains frequency. Since the SMPS circuit is operated at

much higher frequency (in the range of 100 kHz) the input voltage, in spite of being unregulated,

may be considered to have a constant magnitude during any high frequency cycle. A fast

switching device (S) like a MOSFET, is used with fast dynamic control over switch duty ratio(ratio of ON time to switching time-period) to maintain the desired output voltage.

The transformer in Fig.2.1, is used for voltage isolation as well as for better matching

between input and output voltage and current requirements. Primary and secondary windings of

the transformer are wound to have good coupling so that they are linked nearly by same

magnetic flux. That will be shown in the next section the primary and secondary windings of the

fly-back transformer dont carry current simultaneously and in this sense fly-back transformer

works differently from a normal transformer. In a normal transformer, under load primary and

secondary windings conduct simultaneously such that the ampere turns of primary winding is

nearly balanced by the opposing ampere-turns of the secondary winding (the small difference in

ampere-turns is required to establish flux in the non-ideal core). Since primary and secondary

windings of the fly-back transformer dont conduct simultaneously they are more like two

magnetically coupled inductors and it may be more appropriate to call the fly-back transformer

as inductor-transformer. Accordingly the magnetic circuit design of a fly-back transformer is

done like that for an inductor. The details of the inductor-transformer design are dealt with

separately in some later lesson. The output section of the fly-back transformer, which consists of

voltage rectification and filtering, is considerably simpler than in most other switched mode

power supply circuits. As can be seen from the circuit (Fig.2.1), the secondary winding voltage is

rectified and filtered using just a diode and a capacitor. Voltage across this filter capacitor is the

SMPS output voltage.

2.1.2 Principle of Operation

During its operation fly-back converter assumes different circuit configurations. Each of

these circuit configurations have been referred here as modes of circuit operation. The completeoperation of the power supply circuit is explained with the help of functionally equivalent

circuits in these different modes.

As may be seen from the circuit diagram of Fig.2.1, when switch S is on, the primary

winding of the transformer gets connected to the input supply with its dotted end connected to

the positive side. At this time the diode D connected in series with the secondary winding gets


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reverse biased due to the induced voltage in the secondary (dotted end potential being higher).

Thus with the turning on of switch S, primary winding is able to carry current but current in the

secondary winding is blocked due to the reverse biased diode. The flux established in the

transformer core and linking the windings is entirely due to the primary winding current. This

mode of circuit has been described here as Mode-1 of circuit operation.

Fig. 2.2(a) shows the current carrying part of the circuit and Fig. 2.2(b) shows the circuit

that is functionally equivalent to the fly-back circuit during mode-1. In the equivalent circuit

shown, the conducting switch or diode is taken as a shorted switch and the device that is not

conducting is taken as an open switch. This representation of switch is in line with our

assumption where the switches and diodes are assumed to have ideal nature, having zero voltage

drop during conduction and zero leakage current during off state.

In Mode-1 the input supply voltage appears across the primary winding inductance and

the primary current rises linearly. The following mathematical relation gives an expression for

current rise through the primary winding:

Edc= Lpri - - - -(2.1)

where Edc is the input dc voltage, Lpri is inductance of the primary winding and ipri is the

instantaneous current through primary winding.

At the end of switch-conduction (i.e., end of Mode-1), the energy stored in the magnetic

field of the fly back inductor-transformer is equal to LpI2

p/2 , where Ipdenotes the magnitude of

primary current at the end of conduction period. Even though the secondary winding does not

conduct during this mode, the load connected to the output capacitor gets uninterrupted current

Fig.2.2(a): Current path during

Mode-1 of circuit o eration

Fig.2.2(b): Equivalent circuit in

Mode-1


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due to the previously stored charge on the capacitor. During mode-1, assuming a large capacitor

the secondary winding voltage remains almost constant and equals to Vsec=EdcN2/N1. During

mode-1, dotted end of secondary winding remains at higher potential than the other end. Under

this condition voltage stress across the diode connected to secondary winding (which is now

reverse biased) is the sum of the induced voltage in secondary and the output voltage(Vdoide=Vo+ EdcN2/N1).

Mode-2 of circuit operation starts when switch S is turned off after conducting for some

time. The primary winding current path is broken and according to laws of magnetic induction,

the voltage polarities across the windings reverse. Reversal of voltage polarities makes the diode

in the secondary circuit forward biased.

Fig. 2.3(a) shows the current path during mode-2 of circuit operation while Fig. 2.3(b) shows the

functional equivalent of the circuit during this mode.

In mode-2, though primary winding current is interrupted due to turning off of the switch

S, the secondary winding immediately starts conducting such that the net MMF produced by

the windings do not change abruptly (MMF is magneto motive force that is responsible for flux

production in the core. MMF in this case, is the algebraic sum of the ampere-turns of the two

windings. Current entering the dotted ends of the windings may be assumed to produce positive

MMF and accordingly current entering the opposite end will produce negative MMF).

Continuity of MMF, in magnitude and direction is automatically ensured as sudden change in

MMF is not supported by a practical circuit for reasons briefly given below.

MMF is proportional to the flux produced and flux in turn, decides the energy stored in

the magnetic field (energy per unit volume being equal to B2/2 , B being flux per unit area and

is the permeability of the medium). Sudden change in flux will mean sudden change in the

Fig.2.3(a): Current path during

Mode-2 of circuit operation

Fig.2.3(b): Equivalent circuit in Mode-2


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magnetic field energy and this in turn will mean infinite magnitude of instantaneous power,

something that a practical system cannot support.

For the idealized circuit considered here, the secondary winding current abruptly rises

from zero to IpN1/N2as soon as the switch S turns off, N1and N2 denote the number of turns in

the primary and secondary windings respectively. The diode connected in the secondary circuitas shown in Fig.2.1 allows only the current that enters through the dotted end. It can be seen that

the magnitude and current direction in the secondary winding is such that the MMF produced by

the two windings does not have any abrupt change. The secondary winding current charges the

output capacitor. The + marked end of the capacitor will have positive voltage. The output

capacitor is usually sufficiently large such that its voltage doesnt change appreciably in a single

switching cycle but over a period of several cycles the capacitor voltage builds up to its steady

state value.

The steady-state magnitude of output capacitor voltage depends on various factors, like

input dc supply, fly-back transformer parameters, switching frequency, switch duty ratio and the

load at the output. Capacitor voltage magnitude will stabilize if during each switching cycle, the

energy output by the secondary winding equals the energy delivered to the load.

As can be seen from the steady state waveforms of Fig.2.4(a) and Fig.2.4(b), the

secondary winding current decays linearly as it flows against the constant output voltage (VO).

The linear decay of the secondary current can be expressed as follows:

Lsec = -Vo - - - -(2.2)

Where Lsecand isec are secondary winding inductance and current respectively. Vo is the

stabilized magnitude of output voltage.

Under steady-state and under the assumption of zero on-state voltage drop across diode,

the secondary winding voltage during this mode equals Vo and the primary winding voltage is

VON1/N2 (dotted ends of both windings being at lower potential). Under this condition, voltage

stress across switch S is the sum total of the induced emf in the primary winding and the dc

supply voltage (Vswitch

= Edc+VoN1/N

2).

In secondary winding while charging the output capacitor (and feeding the load)

transferring energy from the magnetic field of the flyback transformer to the power supply output

is in electrical form. If the off period of the switch is kept large, the secondary current gets


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2.2 FLYBACK CIRCUIT WAVEFORMS

Fig.2.5 Flyback circuit waveforms under continuous magnetic flux

Fig.2.6 Flyback circuit waveforms under discontinuous magnetic flux


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2.3 PRACTICAL FLY-BACK CONVERTER

The flyback converter discussed in the previous sections neglects some of the practical

aspects of the circuit. The simplified and idealized circuit considered above essentially conveys

the basic idea behind the converter. However a practical converter will have device voltage drops

and losses. The coupling between the primary and secondary windings will not be ideal. The loss

part of the circuit is to be kept in mind while designing for rated power. The designed input

power (Pin

) should be equal to Po/, where P

ois the required output power and is the efficiency

of the circuit. A typical value for may be taken close to 0.6 for first design iteration. Similarly

one needs to counter the effects of the non-ideal coupling between the windings. Due to the non-

ideal coupling between the primary and secondary windings when the primary side switch is

turned-off some energy is trapped in the leakage inductance of the winding. The flux associated

with the primary winding leakage inductance will not link the secondary winding and hence the

energy associated with the leakage flux needs to be dissipated in an external circuit (known as

snubber). Unless this energy finds a path, there will be a large voltage spike across the windings

which may destroy the circuit.

Fig. 2.7 Practical Fly Back Converter

Fig.2.7 shows a practical fly-back converter. The snubber circuit consists of a fast

recovery diode in series with a parallel combination of a snubber capacitor and a resistor. The

leakage-inductance current of the primary winding finds a low impedance path through the

snubber diode to the snubber capacitor. It can be seen that the diode end of the snubber capacitor


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will be at higher potential. To check the excessive voltage build up across the snubber capacitor

a resistor is put across it. Under steady state this resistor is meant to dissipate the leakage flux

energy. The power lost in the snubber circuit reduces the overall efficiency of the fly-back type

SMPS circuit. A typical value for efficiency of a fly-back circuit is around 65% to 75%. In order

that snubber capacitor does not take away any portion of energy stored in the mutual flux of thewindings, the minimum steady state snubber capacitor voltage should be greater than the

reflected secondary voltage on the primary side. This can be achieved by proper choice of the

snubber-resistor and by keeping the RC time constant of the snubber circuit significantly higher

than the switching time period. Since the snubber capacitor voltage is kept higher than the

reflected secondary voltage, the worst-case switch voltage stress will be the sum of input voltage

and the peak magnitude of the snubber capacitor voltage. The circuit in Fig.2.7 also shows, in

block diagram a Pulse Width Modulation (PWM) control circuit to control the duty ratio of the

switch. In practical fly-back circuits, for closed loop output voltage regulation one needs to feed

output voltage magnitude to the PWM controller. In order to maintain ohmic isolation between

the output voltage and the input switching circuit the output voltage signal needs to be isolated

before feeding back. A popular way of feeding the isolated voltage information is to use a

tertiary winding. The tertiary winding voltage is rectified in a way similar to the rectification

done for the secondary winding. The rectified tertiary voltage will be nearly proportional to the

secondary voltage multiplied by the turns-ratio between the windings. The rectified tertiary

winding voltage also doubles up as control power supply for the PWM controller. For initial

powering up of the circuit the control power is drawn directly from the input supply through a

resistor (shown as RS

in Fig.2.6) connected between the input supply and the capacitor of the

tertiary circuit rectifier. The resistor Rs is of high magnitude and causes only small continuous

power loss.

In case, multiple isolated output voltages are required the fly-back transformer will need

to have multiple secondary windings. Each of these secondary winding voltages are rectified and

filtered separately. Each rectifier and filter circuit uses the simple diode and capacitor as shownearlier for a single secondary winding. In the practical circuit shown above, where a tertiary

winding is used for voltage feedback it may not be possible to compensate exactly for the

secondary winding resistance drop as the tertiary winding is unaware of the actual load supplied

by the secondary winding.


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CHAPTER 3

PROJECT BLOCK DIAGRAM REPRESENTATION

The entire Circuit of 7.5W Non-isolated flyback converter is combination of following circuits.

C

F

3.1 BLOCK DIAGRAM REPRESENTATION OF PROJECT

The following figure represents the block diagram of 7.5W Non-isolated flyback

converter.

Fig.3.1 Block diagram representation of Project Circuit


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CHAPTER 4

START-UP CIRCUIT AND MAXIMUM PULSE

WIDTH CIRCUIT

4.1 START-UP CIRCUIT

The startup circuit is a scheme for high power DC/DC converters to minimize the effect

of in-rush current during start-up. A single pulse width modulation controller (PWM) is possible

for the present invention for not only start-up but also normal boost modes. A primary circuit can

have a clamping switch or at least two choke diodes. The choke diode can include push-pull

and L-type configurations. A resistor can be used to dissipate energy clamped from the voltage

spike. A startup circuit can be used to eliminate the in-rush current experienced during start-up.

Since the present invention eliminates the need to match characteristics of multiple controllers, it

significantly reduces the cost associated with implementing this type of technology.

A system to DC-DC converter, the system comprising a primary circuit comprising at

least one bridge leg component; a secondary circuit comprising at least two bridge leg

components, a transformer coupling the primary circuit and the secondary circuit; the primary

circuit further comprising a clamping switch; and a start-up circuit comprising a high frequencyrectifier diode, a high frequency capacitor electrically coupled across the high frequency rectifier

diode and an output capacitor electrically coupled across an output of the secondary circuit,

whereby at least one bridge leg component of the primary circuit is protected from in-rush

current in a Start-up mode.

Fig.4.1 Start-up circuit


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4.1.1 Operation:

The start-up circuit consist of power supply, diode, capacitor and voltage divider (R1and

R2). The supply voltage is obtained from a battery or DC regulated power supply from (0V to

30V).The voltage is smoothened by capacitor and given to voltage divider. The voltage divider

divides the voltage into 50:1 ratio and the TP3 is connected in between this two resistors which

is connected to the Dead time control pin of the controller. The Diode blocks the reverse voltage

to the supply.

4.2 MAXIMUM PULSE WIDTH CIRCUIT

The maximum pulse width circuit consist of clamper which clamps amplifier output

voltage that is compensation pin by connecting the base of Q1 to compensation pin.

Fig .4.2 Maximum pulse width circuit

The amplifier output (Compensation pin) is compared with the internal ramp to generate

the duty ratio. The amplifier output requires to be clamped below the peak of the ramp in order

that the maximum duty ratio is well below 3 V, which is the peak of the ramp. For this purpose,

the amplifier output is provided with a clamp circuit consisting of Q1 (2N2222), R6 (10k ), R7

(10k) and Q2 (2N2907). The clamp level is obtained from a biased diode network consisting of

D2, D3, D4 and D5 (1N4148).


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CHAPTER 5

CONTROLLER CIRCUIT

5.1 PULSE WIDTH MODULATION TECHNIQUE

From the derivations for the boost, buck and inverter (flyback) it can be seen that

changing the duty cycle controls the steady-state output with respect to the input voltage. This is

a key concept governing all inductor-based switching circuits. The most common control method

shown in figure 5.1 is pulse-width modulation (PWM). This method takes a sample of the output

voltage and subtracts this from a reference voltage to establish a small error signal (VERROR).

This error signal is compared to an oscillator ramp signal. The comparator outputs a digital

output (PWM) that operates the power switch. When the circuit output voltage changes, VERROR

also changes and thus causes the comparator threshold to change. Consequently, the output pulse

width (PWM) also changes. This duty cycle change that moves the output voltage to reduce to

error signal to zero, thus completing the control loop.

Fig.5.1 Pulse Width Modulation technique


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5.2 UC494C CONTROLLER:

The controller circuit consist of UC494 Advanced Regulating Pulse Width Modulator.

This entire series of PWM modulators each provide a complete pulse width modulation systemin a single monolithic integrated circuit. These devices include a 5V reference accurate to 1%,

two independent amplifiers usable for both voltage and current sensing, an externally

synchronisable oscillator with its linear ramp generator and two uncommitted transistor output

switches. These two outputs may be operated either in parallel for single ended operation or

alternating for push-pull applications with an externally controlled dead-band. These units are

internally protected against double pulsing of a single output or from extraneous output signals

when the input supply voltage is below minimum. . The UC494A is packaged in a 16-pin DIP.

The UC494A is specified for operation over the full military temperature range of -55C to

+125C, while the UC494C is designed for industrial applications from 0C to +70C.

5.2.1 Pin Configuration of UC494 Controller

Fig.5.2 Pin configuration of UC494 controller

5.2.2 Features

1. D 40, 200A

2. 1% A 5


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3. D E A

4. Wide Range, Variable Deadtime

5.

6. H

7. D

8.

5.3 CONTROLLER CIRCUIT

Fig 5.3 Controller circuit

5.3.1 Operation:

The controller circuit is shown in figure 5.2. The Vcc pin of the controller is connected

to the D1cathode(15V to 25V). The most common control method, shown in adjacent figure is


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pulse-width modulation. This method takes a sample of the output voltage and subtracts this

from a reference voltage to establish a small error signal (VERROR). This error signal is compared

to an oscillator ramp signal.

The ramp voltage is generated by Ct(pin 5) and Rt (pin 6) of 3V using oscillator circuit

with switching frequency of 100kHz. The switching frequency is given by 1.11/RtCt. The

controller has two internal amplifiers a and b. The amplifier outputs are wired such that the

higher of the two outputs will prevail (wired OR). The amplifier b is not used and hence it is

biased (non-inverting input to ground and inverting input to 2.5 V) such that its output is low .

Fig.5.4 Connections of amplifier

The amplifier a non-inverting pin is given to reference voltage of 2.5V and inverting pin

connected to feedback loop with PI controller at TP6. The output feedback is compensated by

using lead/lag compensator and given to compensation pin. The output of amplifier with

compensation and the ramp voltage signal is given to comparator which compares both voltage

signal and generates the output pulses. The controller has two uncommitted transistor output

switches. These two outputs may be operated either in parallel for single ended operation or

alternating for push-pull applications with an externally controlled dead-band. The output control

pin is connected to the ground to work the output in single ended or parallel operation. The

emitter pins of the two transistors are tied together and connected to ground and the collector

pins are tied together and given to the gate of MOSFET .


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CHAPTER 6

FEEDBACK CIRCUIT

6.1 CLOSED LOOP CONTROL

This is also often termed as Automatic Control, Process Control, Feedback Control etc.

Here the controller objective is to provide such inputs to the plant such that the output y(t)

follows the input r(t) as closely as possible, in value and over time. The structure of the common

control loop with its constituent elements, namely the Controller, the Actuator, the Sensor and

the Process itself is shown. In addition the signals that exist at various points of the system are

also marked. These include the command (alternatively termed the set point or the reference

signal), the exogenous inputs (disturbances, noise).

The difficulties in achieving the performance objective is mainly due to the unavoidable

disturbances due to load variation and other external factors, as well as sensor noise, the

complexity, possible instability, uncertainty and variability in the plant dynamics, as well as

limitations in actuator capabilities.

Fig.6.1 Closed loop block diagram

Most industrial control loop command signals are piecewise constant signals that indicate

desirable levels of process variables, such as temperature, pressure, flow, level etc., which ensure

the quality of the product in Continuous Processes. In some cases, such as in case of motion

control for machining, the command signal may be continuously varying according to the

dimensions of the product. Therefore, here deviation of the output from the command signal


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results in degradation of product quality. It is for this reason that the choice of the feedback

signals, that of the controller algorithm (such as, P, PI or PID), the choice of the control loop

structure (normal feedback loop, cascade loop or feedforward) as well as choice of the controller

gains is extremely important for industrial machines and processes. Typically the control

configurations are well known for a given class of process however, the choice of controllergains have to be made from time to time, since the plant operating characteristics changes with

time. This is generally called controller tuning.

6.2 COMPENSATOR(LEAD/LAG)

Generally the purpose of the Lead-Lag compensator is to create a controller which has

an overall magnitude of approximately 1. The lead-lag compensator is largely used for phase

compensation rather than magnitude. A pole is an integrator above the frequency of the pole. A

zero is a derivative above the frequency of the zero.

Adding a pole to the system changes the phase by -90 deg and adding a zero changes the

phase by +90 deg. So if the system needs +90 deg added to the phase in a particular frequency

band then you can add a zero at a low frequency and follow that zero with a pole at a higher

frequency.

Lead and lag control are used to add or reduce phase between 2 frequencies. Typically

these frequencies are centered around the open loop crossover frequency. A lead filter typically

has unity gain (0 dB) are low frequencies while the lag provides a non unity gain at low

frequencies.

Where X is the input to the compensator, Y is the output, s is the complex Laplace transform

variable, z is the zero frequency and p is the pole frequency. The pole and zero are both typically

negative. In a lead compensator, the pole is left of the zero in the complex plane | z | < | p | ,

while in a lag compensator | z | > | p | . A lead-lag compensator consists of a lead compensator

cascaded with a lag compensator.


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The overall transfer function can be written as

Typically | p1| > | z1| > | z2 | > | p2 | , where z1and p1are the zero and pole of the lead

compensator and z2and p2are the zero and pole of the lag compensator. The lead compensator

provides phase lead at high frequencies. This shifts the poles to the left, which enhances the

responsiveness and stability of the system. The lag compensator provides phase lag at low

frequencies which reduces the steady state error.

Fig.6.2 Feedback Circuit

The dc gain from duty ratio to output voltage consists of modulator gain and converter

gain. The modulator gain is the reciprocal of the ramp peak in the modulator. In UC494, it is

1/3.5.

This gain varies from 72 to 66. The overall gain is therefore 20.47 to 18.87 for the

converter. The lower gain is at higher voltage. The closed loop control used is a PI controller

with lead/lag compensator. The PI corner frequency [1=(R16C8)] is chosen at 3030 rad/sec. The

lead/lag compensator frequencies [1=(R15C7)] are chosen as 2220 rad/sec and

[1=(C7R3R4R15)]22000 rad/sec .


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CHAPTER 7

POWER CIRCUIT

7.1 MAIN COMPONENTS OF POWER CIRCUIT:

FE (F44)

F ( )

D(110)

(7815C)

7.2 MOSFET

The power MOSFET is commonly presented and regarded as a voltage driven device and

as such there is a natural expectation that it can be driven from any pulse source, irrespective of

that sources energy or current capability. This assumption is partly justified if the system in

question only pulses or switches the MOSFET at a low frequency or in pure DC circuits, where

the transistor may only be used in a toggled state. However, for typical switching frequencies

from several kHz upwards attention must be paid to the gate drive requirements to ensure

efficient and saturated switching of the MOSFET. This must be considered as the gate-source

(g-s) circuit is to a first approximation, essentially a CR network; comprising the g-s capacitance

and the resistance of the metallic/silicon interconnects. To this network must be added the

effective resistance or source impedance of the gate driver circuitry and for true assessments

consideration of the drain-gate (d-g) capacitance and the Miller effect. Due to this network, the

g-s voltage follows an exponential curve as the C elements charge and so either sufficient time

must be given to allow this voltage to reach its target value (thus limiting the operating

frequency and increasing the time spent in the linear region thereby producing high switching

losses), or the R element must be minimised. The MOSFETs are of two types N-channel

MOSFET and P-channel MOSFET. The N-channel MOSFET is made ON using positive gate

voltage and N-channel MOSFET is made ON with negative gate voltage. The N-channel

MOSFETs are most commonly used in the power electronics field.


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7.2.1 MOSFET as a Switch

The N-channel, Enhancement-mode MOSFET operates using a positive input voltage and

has an extremely high input resistance (almost infinite) making it possible to interface with

nearly any logic gate or driver capable of producing a positive output. Also, due to this very high

input (Gate) resistance we can parallel together many different MOSFETs until we achieve the

current handling limit required.

Fig.7.1 Power MOSFET (a) Schematic (b)Transfer characteristics (c)Device symbol

While connecting together various MOSFETs may enable us to switch high currents or

high voltage loads, doing so becomes expensive and impractical in both components and circuit

board space. To overcome this problem Power Field Effect Transistors or Power FET's were

developed.


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P-channel in enhancement and depletion mode

N-channel in enhancement and depletion mode

Fig.7.2 Symbol of MOSFETs

We now know that there are two main differences between field effect transistors,

depletion-mode only for JFET's and both enhancement-mode and depletion-mode for MOSFETs.

In this project we will look at using the Enhancement-mode MOSFET as a Switch as these

transistors require a positive gate voltage to turn "ON" and a zero voltage to turn "OFF" making

them easily understood as switches and also easy to interface with logic gates.

The operation of the enhancement-mode MOSFET can best be described using its I-V

characteristics curves shown below. When the input voltage, ( VIN ) to the gate of the transistor is

zero, the MOSFET conducts virtually no current and the output voltage, ( VOUT) is equal to the

supply voltage VDD. So the MOSFET is "fully-OFF" and in its "cut-off" region.

7.2.2 MOSFET Characteristics Curves

The minimum ON-state gate voltage required to ensure that the MOSFET remains fully-

ON when carrying the selected drain current can be determined from the V-I transfer curves

below. When VINis high or equal to VDD, the MOSFET Q-point moves to point A along the load

line. The drain current ID increases to its maximum value due to a reduction in the channel

resistance. ID becomes a constant value independent of VDD and is dependent only on VGS.

Therefore, the transistor behaves like a closed switch but the channel ON-resistance does not

reduce fully to zero due to its RDS(on)value, but gets very small.


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Fig.7.3 MOSFET characteristics curves

Likewise, when VINis LOW or reduced to zero, the MOSFET Q-point moves from point

A to point B along the load line. The channel resistance is very high so the transistor acts like an

open circuit and no current flows through the channel. So if the gate voltage of the MOSFET

toggles between two values, HIGH and LOW the MOSFET will behave as a "single-pole single-

throw" (SPST) solid state switch and this action is defined as

7.2.2.1. Cut-off region

Here the operating conditions of the transistor are zero input gate voltage ( V IN ), zero

drain current IDand output voltage VDS= VDDTherefore the MOSFET is switched "Fully-OFF".

Cut-off Characteristics

Fig.7.4 Cut-Off region equivalent diagram


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The input and Gate are grounded (0V)

Gate-source voltage less than threshold voltage VGS< VTH

MOSFET is "fully-OFF" (Cut-off region)

No Drain current flows ( ID= 0 )

VOUT= VDS= VDD= "1"

MOSFET operates as an "open switch"

Then we can define the "Cut-Off region" or "OFF mode" of a MOSFET switch as being, gate

voltage, VGS< VTHand ID= 0. For a P-channel MOSFET, the gate potential must be negative.

7.2.2.2. Saturation region

Here the transistor will be biased so that the maximum amount of gate voltage is applied

to the device which results in the channel resistance RDS(on) being as small as possible with

maximum drain current flowing through the MOSFET switch. Therefore the MOSFET is

switched "Fully-ON".

Saturation Characteristics

Fig.7.5 Saturation region equivalent diagram

The input and Gate are connected to VDD

Gate-source voltage is much greater than threshold voltage VGS> VTH

MOSFET is "fully-ON" (saturation region)

Max Drain current flows ( ID= VDD/ RL)

VDS= 0V (ideal saturation)

Min channel resistance RDS(on)< 0.1


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= D= 0.2 (D.D)

MOSFET operates as a "closed switch"

Then we can define the "Saturation region" or "ON mode" of a MOSFET switch as gate-source

voltage, VGS> VTH and ID= Maximum. For a P-channel MOSFET, the gate potential must be

positive.

By applying a suitable drive voltage to the gate of an FET, the resistance of the drain-

source channel, RDS(on) can be varied from an "OFF-resistance" of many hundreds of k's,

effectively an open circuit, to an "ON-resistance" of less than 1, effectively a short circuit. We

can also drive the MOSFET to turn "ON" faster or slower, or pass high or low currents. This

ability to turn the power MOSFET "ON" and "OFF" allows the device to be used as a very

efficient switch with switching speeds much faster than standard bipolar junction transistors.

An example of using the MOSFET as a switch

Fig.7.6 MOSFET as a Switch

In this circuit arrangement an Enhancement-mode N-channel MOSFET is being used to

switch a simple lamp "ON" and "OFF" (could also be an LED). The gate input voltage VGS is

taken to an appropriate positive voltage level to turn the device and therefore the lamp either

fully "ON", ( VGS= +ve ) or at a zero voltage level that turns the device fully "OFF", ( VGS= 0 ).


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If the resistive load of the lamp was to be replaced by an inductive load such as a coil,

solenoid or relay a "flywheel diode" would be required in parallel with the load to protect the

MOSFET from any self generated back-emf.

Above figure7.6 shows a very simple circuit for switching a resistive load such as a lamp

or LED. But when using power MOSFETs to switch either inductive or capacitive loads some

form of protection is required to prevent the MOSFET device from becoming damaged. Driving

an inductive load has the opposite effect from driving a capacitive load. For example, a capacitor

without an electrical charge is a short circuit resulting in a high "inrush" of current and when we

remove the voltage from an inductive load we have a large reverse voltage build up as the

magnetic field collapses, resulting in an induced back-emf in the windings of the inductor.

For the power MOSFET to operate as an analogue switching device it needs to beswitched between its "Cut-off Region" where VGS= 0 and its "Saturation Region" were VGS(on)=

+ve. The power dissipated in the MOSFET ( PD) depends upon the current flowing through the

channel IDat saturation and also the "ON-resistance" of the channel given as R DS(on).

In this project we are using IRFZ44, the main features of this power MOSFET has

dynamic dv/dt rating, 175o operating temperature, fast switching, ease paralleling, simple drive

requirements. The power MOSFET has the current carrying capacity of 1A and block about

minimum 55V.

7.3 COUPLE INDUCTOR

The coupled inductors are of crucial importance as models in many practical applications,

like in electrical transformers, motors and generators. In most cases, such devices are modeled

`from the electrical point of view' without the necessity (or possibility) of modeling detailed

structure of the magnetic field inside the device. Mutual inductance occurs when the change in

current in one inductor induces a voltage in another nearby inductor. It is important as the

mechanism by which transformers work, but it can also cause unwanted coupling between

conductors in a circuit. The mutual inductance M, is also a measure of the coupling between two

inductors. The mutual inductance by circuit i on circuit j is given by the double integral

Neumann formula,


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The mutual inductance also has the relationship,

where

M21 is the mutual inductance and the subscript specifies the relationship of the voltage

induced in coil 2 due to the current in coil 1.

N1is the number of turns in coil 1,

N2is the number of turns in coil 2 and

P21is the permeance of the space occupied by the flux.

The mutual inductance also has a relationship with the coupling coefficient. The coupling

coefficient is always between 1 and 0, and is a convenient way to specify the relationship

between a certain orientation of inductor with arbitrary inductance:

where

k is the coupling coefficient and 0 k 1,

L1is the inductance of the first coil and

L2is the inductance of the second coil.

Once the mutual inductance M, is determined from this factor it can be used to predict the

behavior of a circuit:

where

V1is the voltage across the inductor of interest,

L1is the inductance of the inductor of interest,

dI1/dt is the derivative with respect to time of the current through the inductor of interest,


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dI2/dt is the derivative with respect to time of the current through the inductor that is

coupled to the first inductor and

M is the mutual inductance.

The minus sign arises because of the sense the current I2has been defined in the diagram.

With both currents defined going into the dots the sign of M will be positive. When one inductor

is closely coupled to another inductor through mutual inductance, such as in a transformer, the

voltages, currents and number of turns can be related in the following way,

where

Vsis the voltage across the secondary inductor,Vpis the voltage across the primary inductor (the one connected to a power source),

Nsis the number of turns in the secondary inductor and

Npis the number of turns in the primary inductor.

Conversely the current:

where

Isis the current through the secondary inductor,

Ipis the current through the primary inductor (the one connected to a power source),

Nsis the number of turns in the secondary inductor and

Npis the number of turns in the primary inductor.

Note that the power through one inductor is the same as the power through the other. Also note

that these equations don't work if both transformers are forced (with power sources).

When either side of the transformer is a tuned circuit, the amount of mutual inductance

between the two windings determines the shape of the frequency response curve. Although no

boundaries are defined this is often referred to as loose-, critical- and over-coupling. When two


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tuned circuits are loosely coupled through mutual inductance, the bandwidth will be narrow. As

the amount of mutual inductance increases, the bandwidth continues to grow. When the mutual

inductance is increased beyond a critical point the peak in the response curve begins to drop and

the center frequency will be attenuated more strongly than its direct sidebands. This is known as

overcoupling.

( ) Pair of interacting coils, ( ) coupled electrical inductors.

Fig.7.7 Couple inductors

The couple inductor is used as a flyback transformer in this project. Flyback transformer utilizes

the "flyback" action of an inductor or flyback transformer to convert the input voltage and

current to the desired output voltage and current. Modern flyback transformer and circuit design

now permit use in excess of 300 watts of power, but most applications are less than 50 watts. By

definition a transformer directly couples energy from one winding to another winding.

A flyback transformer does not act as a true transformer. A flyback transformer first

stores energy received from the input power supply (charging portion of a cycle) and then

transfers energy (discharge portion of a cycle) to the output, usually a storage capacitor with a

load connected across its terminals. An application in which a complete discharge is followed by


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a short period of inactivity (known as idle time) is defined to be operating in a discontinuous

mode. An application in which a partial discharge is followed by charging is defined to be

operating in the continuous mode.

this project the main inductor of primary 33 turns and secondary 48 turns is used the

rated current is 1.11A.The ripple current is chosen as 0.22A with maximum ON time of 6 sec,at input voltage of 15V, this gives an inductor value of approximately 400 H with turns ratio of

0.691.

7.4 VOLTAGE REGULATOR (LM7815)

A voltage regulator is an electrical regulator designed to automatically maintain a

constant voltage level. A voltage regulator may be a simple "feed-forward" design or may

include negative feedback control loops. It may use an electromechanical mechanism, or

electronic components. Depending on the design, it may be used to regulate one or more AC or

DC voltages.

Electronic voltage regulators are found in devices such as computer power supplies

where they stabilize the DC voltages used by the processor and other elements. In automobile

alternators and central power station generator plants, voltage regulators control the output of the

plant. In an electric power distribution system, voltage regulators may be installed at a substation

or along distribution lines so that all customers receive steady voltage independent of how much

power is drawn from the line. The linear regulator is the basic building block of nearly every

power supply used in electronics. The IC linear regulator is so easy to use and so inexpensive

that it is usually one of the cheapest components in an electronic assembly.


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Fig.7.8 Voltage regulator LM7815

The voltage regulator LM7815 is used in this project and connected to the output offlyback converter to maintain the constant voltage of 15V for variation of output voltage

with in limit for a input voltage of 15V to 25V. Maximum ratings of LM7815C, 35V input

voltage, power dissipation internally limited, operating junction temperature range is 0 to

+150oc.

7.4.1 Features of LM7815

1A.

4% .

7.5 DIODE (MUR110)

The diode MUR110 is used in the power circuit. The diode carries 0.5A average current

and blocks about 20V and suitable for 100kHz switching. The reverse recovery time has to be

better than 50ns. Therefore MUR110 is selected.


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7.6 CIRCUIT DIAGRAM OF POWER CIRCUIT OF NON-

ISOLATED FLYBACK CONVERTER

Fig.7.9 Power circuit

7.6.1 Operation of Circuit

The above figure7.9 is the circuit diagram of 7.5W Non-isolated flyback converter power

circuit. Operation of power circuit of 7.5W non-isolated fly back converter is same as that the

operation of fly back converter. When supply voltage is given to the converter and gate pulses

are applied to the gate of MOSFET it starts conducting. The gate pulses are obtained from the ICUC494 controller which is a PWM controller.

But due to reverse polarities of transformer, the diode in the secondary circuit will get

reverse biased and does not conduct and therefore the capacitor C4which is already charged in

previous stage will get discharge and maintains the supply to the load.

When MOSFET is in off position, the energy stored in the inductor of the transformer

will make the diode in the secondary circuit forward bias, charges the capacitor C4 and alsosupplies current to load. By this way it maintains the load. The voltage across the capacitor

C5(17V) is more than what we require(15V). In order to get constant desired voltage we are

using LM7815C which gives the constant output voltage of 15V.


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CHAPTER 8

SIMULATION OF FLYBACK CONVERTER USINGMATLAB

8.1 SIMULATION CIRCUIT DESCRIPTION

The flyback converter circuit is simulated in MATLAB software. The switching pulses to

the MOSFET is given from the comparator. The ramp signal and error voltage is given as input

to the comparator inputs. The error voltage is obtained by subtracting the output voltage and

reference voltage. This error voltage, ramp signal and comparator from a pulse width

modulation technique. As the error voltage is generated from output voltage and reference

voltage it act as a closed loop flyback converter. No voltage regulator is used in the power

circuit, therefore the output voltage is not constant on the output side.

8.1.1 Simulation Circuit


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Fig. 8.1 Simulation circuit of flyback converter

8.1.2 Output Wave Forms

Fig. 8.2 Gate Pulses


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Fig. 8.3 Output

Voltage(16.6V)

Fig. 8.4 Output Current(0.43A)

8.1.3 Tabulation of Input Voltage and Output Voltage Without Voltage

Regulator on the Load Side

Table 8.1 The output voltage for variation of input voltage

INPUT VOLTAGE

(VOLTS)

OUPUT VOLTAGE

(VOLTS)

15 11.5

16 12.4

17 12.6

18 13.24

19 13.81

20 14.37

21 14.9

22 15.5


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The above table shows the variation of output voltage for different input voltages. The

output voltage is not constant as the voltage regulator is not connected on the load side of the

power circuit.

23 16

24 16.63

25 17.2


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CHAPTER 9

NON-ISOLATED FLYBACK CONVERTER

HARDWARE KIT

9.1 HARDWARE KIT PCB LAYOUT


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9.2 HARDWARE KIT

Fig.9.2 Hardware kit PCB with components


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9.3 PRACTICAL OUTPUT WAVEFORMS FOR INPUT

VOLTAGE OF 15VOLTS

Fig.9.3 Ramp Voltage (3V)

Fig.9.4 Gate Voltage(12V)

Scale

X-axis 1unit- 10sec

Y-axis 1unit- 2V

Scale

X-axis 1unit- 20sec

Y-axis 1unit- 10V


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Fig.9.5 Output Voltage(15.2V)

Fig.9.6 Primary Inductor Voltage

Scale

X-axis 1unit- 10sec

Y-axis 1unit- 10V

Scale

X-axis 1unit- 20sec

Y-axis 1unit- 20V


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Fig. 9.7 Secondary Inductor Voltage

9.4 PRACTICAL OUTPUT WAVEFORMS FOR INPUT

VOLTAGE OF 25VOLTS

Scale

X-axis 1unit- 20sec

Y-axis 1unit- 20V

Scale

X-axis 1unit- 10sec

Y-axis 1unit- 2V


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Fig. 9.8 Ramp Voltage(3V)

Fig. 9.9 Gate Voltage(20V)

Fig.9.10 Output Voltage(15.2V)

Scale

X-axis 1unit- 20sec

Y-axis 1unit- 10V

Scale

X-axis 1unit- 10sec

Y-axis 1unit- 10V


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Fig. 9.11 Primary Inductor Voltage

Scale

X-axis 1unit- 20sec

Y-axis 1unit- 20V

Scale

X-axis 1unit- 20sec

Y-axis 1unit- 20V


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Fig. 9.12 Secondary Inductor Voltage

9.5 TABULATION OF INPUT VOLTAGE AND OUTPUT VOLTAGE

Table 9.1 The output voltage for variation of input voltage

The above table shows the variation of output voltage for different input voltages. The

output voltage is as the voltage regulator LM7815 is connected on the load side of the power

circuit.

INPUT VOLTAGE

(VOLTS)

OUPUT VOLTAGE

(VOLTS)

15 15.2

16 15.2

17 15.2

18 15.2

19 15.2

20 15.2

21 15.2

22 15.2

23 15.2

24 15.2

25 15.2


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CHAPTER 10

APPLICATIONS AND LIMITATIONS

10.1 APPLICATIONS:

, ( ) ,

.

( , C)

(.., C


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The voltage feedback loop requires a lower bandwidth due to a zero in the response

of the converter.

The current feedback loop used in current mode control needs slope compensation in

cases where the duty cycleis above 50%.

The power switches are now turning on with positive current flow.


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CHAPTER 11

CONCLUSION

11.1. CONCLUSION:

, .

H .

AAB 8.1 ,

FE .

C494C

11.2. RESULTS:

F (H ) 15.2,

15 25.

.

EAD/AG

.

A FE

.

11.3DIFFICULTIES ENCOUNTERED DURING THE PROJECT The simulation difficulties occurred due to non availability of IC PWM controller

UC494C in the MATLAB software.

.


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CHAPTER 12

SCOPE FOR FUTURE WORK

12.1 SCOPE FOR FUTURE WORK:

F F 250 H

, A

.

Resonant switching techniques reduce the switching losses to practically zero; the

switching frequency then may be increased to hundreds of kHz to achieve higher power

densities. Such converters in general are classified as Soft switching converters. In these

converters, the switching transitions occur with zero loss.

With the Active clamp circuit, the transistor turn-off voltage spike is clamped, the

transformer leakage energy is recycled, and zero-voltage-switching (ZVS) of the

MOSFET switches becomes a possibility.

FE ( ) GB,

.

Designing for high power ratings the isolation should be provided for control circuit.


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REFERENCES

BOOKS

1.Keng Chih Wu Pulse Width Modulated DC/DC Converters

2.. H. , E, 2 ., H, E C,

.

3.. , . . , . . , E:

C, A D, 2 ., & ,

, 1995

4.B, (1999), H, GH

5.A. ., D, 2 ., GH,

, 1998

WEBSITES

Flyback Converter http://en.wikipedia.org/wiki/Flyback_converter

DC-DC Converter http://en.wikipedia.org/wiki/DC-to-DC_converter

SMPS http;//en.nptel.iitm.ac.in/courses/.../L21(DP)(PE)

%20((EE)NPTEL).pdf


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Data sheets of components www.alldatasheet.com

SMPS, DC-DC converter


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APPENDIX A

COMPONENTS LIST WITH THEIR RATINGS


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APPENDIX B

DATA SHEET OF UC494C


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APPENDIX C

DATA SHEET OF MOSFET (IRFZ44)


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APPENDIX D

DATA SHEET OF LM7815


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Documents

14A Non-Isolated Flyback Report