Modeling and
estimation for
control
E. Witrant
Control-oriented modeling and
system identification
Outline
Emmanuel [email protected]
April 22, 2014.
Modeling and
estimation for
control
E. Witrant
Course goal
To teach systematic methods for building mathematical models
of dynamical systems based on physical principles and
measured data.
Main objectives:
• build mathematical models of technical systems from first
principles
• use the most powerful tools for modeling and simulation
• construct mathematical models from measured data
Modeling and
estimation for
control
E. Witrant
Class overview
1 Introduction to modeling
Physical Modeling
2 Principles of physical modeling
3 Some basic relationships in physics
4 Bond graphs
Simulation
5 Computer-aided modeling
6 Modeling and simulation in Scilab
System Identification
7 Experiment design for system identification
8 Non-parametric identification
9 Parameter estimation in linear models
10 System identification principles and model validation
11 Nonlinear black-box identification
Towards process supervision
12 Recursive estimation methods
Modeling and
estimation for
control
E. Witrant
Grading policy
Based on homeworks, each due on the next day at the
beginning of the first class.
Modeling and
estimation for
control
E. Witrant
Material
• Lecture notes from 2E1282 Modeling of Dynamical
Systems, Automatic Control, School of Electrical
Engineering, KTH, Sweden.
• L. Ljung and T. Glad, Modeling of Dynamic Systems,
Prentice Hall Information and System Sciences Series,
1994.
• S. Campbell, J-P. Chancelier and R. Nikoukhah, Modeling
and Simulation in Scilab/Scicos, Springer, 2005.
• S. Stramigioli, Modeling and IPC Control of Interactive
Mechanical Systems: A Coordinate-free Approach,
Springer, LNCIS 266, 2001.
• L. Ljung, System Identification: Theory for the User, 2nd
Edition, Information and System Sciences, (Upper Saddle
River, NJ: PTR Prentice Hall), 1999.
Modeling and
estimation for
control
E. Witrant
Class website
• Go to:
http://www.gipsa-lab.grenoble-inp.fr/˜e.witrant/classes/14_Modeling_Lessons_KUL.pdf
• or Google “witrant” then go to “Teaching”, “International”
and “Control-oriented modeling and system identification”
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Modeling and estimation forcontrol
Lecture 1: Introduction to modeling
Emmanuel [email protected]
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Systems and Models
Systems and experiments
• System: object or collection of objects we want to study.• Experiment: investigate the system properties / verify
theoretical results, BUT• too expensive, i.e. one day operation on Tore Supra;• too dangerous, i.e. nuclear plant;• system does not exist, i.e. wings in airplane design.
⇒ Need for models
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
What is a model?
• Tool to answer questions about the process withoutexperiment / action-reaction.
• Different classes:1 Mental models: intuition and experience (i.e. car driving,
industrial process in operator’s mind);2 Verbal models: behavior in different conditions described
by words (i.e. If . . . then . . . );3 Physical models: try to imitate the system (i.e. house
esthetic or boat hydrodynamics);4 Mathematical models: relationship between observed
quantities described as mathematical relationships (i.e.most law in nature).Generally described by differential algebraic equations:
x(t) = f(x(t), u(t), d(t))
0 = g(x(t), u(t), d(t))
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Models and simulation
• models→ used to calculate or decide how the systemwould have reacted (analytically);
• Simulation: numerical experiment = inexpensive and safeway to experiment with the system;
• simulation value depends completely on the model quality.
How to build models?
• Two sources of knowledge:• collected experience: laws of nature, generations of
scientists, literature;• from the system itself: observation.
• Two areas of knowledge:• domain of expertise: understanding the application and
mastering the relevant facts→ mathematical model;• knowledge engineer: practice in a usable and explicit
model→ knowledge-based model.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
• Two different principles for model construction:• physical modeling: break the system into subsystems
described by laws of nature or generally recognizedrelationships;
• identification: observation to fit the model properties tothose of the system (often used as a complement).
How to verify models?
• Need for confidence in the results and prediction, obtainedby verifying or validating the model: model vs. system.
• Domain of validity: qualitative statements (most verbalmodels), quantitative predictions. Limited for all models.
⇒ Hazardous to model outside the validated area.
⇒ Models and simulations can never replace observationsand experiments - but they constitute an important anduseful complement.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Different types of mathematical models
• Deterministic - Stochastic: exact relationships vs.stochastic variables/processes;
• Static - Dynamic: direct, instantaneous link (algebraicrelationships) vs. depend also on earlier applied signals(differential/difference equations);
• Continuous - Discrete time: differential equation vs.sampled signal;
• Distributed - Lumped: events dispersed over the space(distributed parameter model→ partial differentialequation PDE) vs. finite number of changing variables(ordinary diff. eqn. ODE);
• Change oriented - Discrete event driven: continuouschanges (Newtonian paradigm) vs. (random) event-basedinfluences (i.e. manufacture, buffer. . . )
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Examples of Models
Networked control of the inverted pendulum [Springer’07]
• Objective: test control laws for control over networks.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
• Physical model and abstraction:
m2
m1
u(t)
l ol c
θ(t)
z(t)
• Mathematical model• from physics:
[
m1 m1l0m1l0 J + m1z2
] [
zθ
]
+
[
0 −m1zθ2m1zθ 0
] [
zθ
]
+
[
−m1 sin θ− (m1l0 + m2lc) sin θ −m1z cos θ
]
g =
[
10
]
u,
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
• input/ouput representation (x [z, z, θ, θ]⊤):
x1 = x2,
x2 =u
m1− l0x4 + x1x2
4 + g sin (x3),
x3 = x4,
x4 =1
J0(x1) −m1l20[g (m2lc sin (x3) + m1x1 cos (x3))
−m1 (l0x4 + 2x2) x1x4 + −l0 u] ,J0(x1) = J + m1x2
1 ,
y = x1, x2
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
• Fluid-flow model for the network [Misra et al. 2000, Hollotand Chait 2001]: TCP with proportional active queuemanagement (AQM) set the window size W and queuelength q variations as
dWi(t)dt
=1
Ri(t)−
Wi(t)2
Wi(t − Ri(t))Ri(t − Ri(t))
pi(t),
dq(t)dt
= −Cr +N
∑
i=1
Wi(t)Ri(t)
, q(t0) = q0,
where Ri(t) q(t)Cr
+ Tpi is the round trip time, Cr the link
capacity, pi(t) = Kpq(t − Ri(t)) the packet discard functionand Tpi the constant propagation delay. The averagetime-delay is τi =
12 Ri(t)
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
E.g. network with 2 TCP flows:
dW1,2(t)dt
=1
R1,2(t)−
W1,2(t)2
W1,2(t − R1,2(t))R1,2(t − R1,2(t))
p1,2(t)
dq(t)dt
= −300 +2
∑
i=1
Wi(t)Ri(t)
, q(0) = 5
τ(t) = R1(t)/2
Behavior of the network internal states.
q(t)W1(t)
W2(t)
time (s)
q(t)
and
W1,
2(t)
(pac
kets
)
Average queue length and windows sizes
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
• Compare different control laws: in simulation
0 0.2 0.4 0.6 0.8 1-0.5
0
0.5
1
1.5
2
2.5
3
3.5
4
0 0.2 0.4 0.6 0.8 1-15
-10
-5
0
5
10
15
predictor with a variable horizon
predictor with a fixed horizon
state feedback
buffer strategy
θ(t)(r
ad)
time (s)
Simulation with measurement noise
predictor with a variable horizon
predictor with a fixed horizon
state feedback
buffer strategy
θ(t)(r
ad)
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
On the inverted pendulum experiment:
(a) Predictive control with fixed horizon. (b) Predictive control with time-varyinghorizon.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Thermonuclear Fusion with ToreSupra tokamak
Physical model
∂ψ
∂t= η∥(x, t)
[
1µ0a2
∂2ψ
∂x2+
1µ0a2x
∂ψ
∂x
+R0jbs(x, t) + R0jni(x, t)]
jφ(x, t) = −1
µ0R0a2x∂
∂x
[
x∂ψ
∂x
]
Mathematical model [PPCF 2007]
Abstraction
Experimental results
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Identification of temperature profiles [CDC 2011]
• Parameter-dependant first-order dynamics:
τth(t) = eϑt0 Iϑt1p Bϑt2
φ0nϑt3
e Pϑt4tot
dWdt
= Ptot −1τth
W , W(0) = Ptot(0)τth(0)
Te0(t) = AW
→ “free” parameters ϑi determined from a sufficiently richset of experimental data.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
• Model validation:
Central temperature (keV) and power inputs (MW)
|η(x, t) − η(x, t)|
x
time (s)
Te0(t)
Te0(t)ITERL-96P(th)Plh Picrf
Comparison of the model with a shot not included in thedatabase.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Conclusion: all models are approximate!
• A model captures only some aspects of a system:• Important to know which aspects are modelled and which
are not;• Make sure that model is valid for intended purpose;• “If the map does not agree with reality, trust reality”.
• All-encompasing models often a bad idea:• Large and complex hard to gain insight;• Cumbersome and slow to manipulate.
• Good models are simple, yet capture the essentials!
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Models for Systems and Signals
Types of models
• System models (differential / difference equations) andsignal models (external signals / disturbances).
• Block diagram models: logical decomposition of thefunctions and mutual influences (interactions, informationflows), not unique. Related to verbal models.
• Simulation models: related to program languages.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Input, output and disturbance signals
• Constants (system or design parameters) vs. variables orsignals;
• Outputs: signals whose behavior is our primary interest,typically denoted by y1(t), y2(t), . . . , yp(t).
• External signals: signals and variables that influence othervariables in the system but are not influenced by thesystem:• input or control signal: we can use it to influence the
system u1(t), u2(t), . . . , um(t);• disturbances: we cannot influence or choose
w1(t), w2(t), . . . , wr(t).
• Internal variables: other model variables.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Differential equations
• Either directly relate inputs u to outputs y:
g(y(n)(t), y(n−1)(t), . . . , y(t), u(m)(t), u(m−1)(t), . . . , u(t)) = 0
where y(k )(t) = dk y(t)/dtk and g(·) is an arbitrary,vector-valued, nonlinear function.
• or introduce a number of internal variables related by firstorder DE
x(t) = f(x(t), u(t))
with x, f and u are vector-valued, nonlinear functions, i.e.
x1(t) = f1(x1(t), . . . , xn(t), u1(t), . . . , um(t))
x2(t) = f2(x1(t), . . . , xn(t), u1(t), . . . , um(t))...
xn(t) = fn(x1(t), . . . , xn(t), u1(t), . . . , um(t))
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
The outputs are then calculated from xi(t) and ui(t) from:
y(t) = h(x(t), u(t))
• Corresponding discrete time equations:
x(t + 1) = f(x(t), u(t))
y(t) = h(x(t), u(t))
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
The concept of state and state-space modelsDefinitions:
• State at t0: with this information and u(t), t ≥ t0, we cancompute y(t).
• State: information that has to be stored and updatedduring the simulation in order to calculate the output.
• State-space model (continuous time):
x(t) = f(x(t), u(t))
y(t) = h(x(t), u(t))
u(t): input, an m-dimensional column vectory(t): output, a p-dimensional column vectorx(t): state, an n-dimensional column vector
→ nth order model, unique solution if f(x, u) continuouslydifferentiable, u(t) piecewise continuous and x(t0) = x0
exists.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
• State-space model (discrete time:)
x(tk+1) = f(x(tk ), u(tk )), k = 0, 1, 2, . . .
y(tk ) = h(x(tk ), u(tk ))
where u(tk ) ∈ Rm, y(tk ) ∈ Rp, x(tk ) ∈ Rn.→ nth order model, unique solution if the initial valuex(t0) = x0 exists.
Linear models:
• if f(x , u) and h(x, u) are linear functions of x and u:
f(x , u) = Ax + Bu
h(x , u) = Cx + Du
with A : n × n, B : n ×m, C : p × n and D : p ×m.
• if the matrices are independent of time, the system islinear and time-invariant.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Stationary solutions, static relationships and linearizationStationary points: Given a system
x(t) = f(x(t), u(t))
y(t) = h(x(t), u(t))
a solution (x0, u0) such that 0 = f(x0, u0) is called a stationarypoint (singular point or equilibrium).At a stationary point, the system is at rest: x(0) = x0, u(t) = u0
for t ≥ 0⇒ x(t) = x0 for all t ≥ 0.Stability: suppose that x(t0) = x0 gives a stationary solution,what happens for x(t0) = x1? The system is
• asymptotically stable if any solution x(t) close enough tox0 converges to x0 as t → ∞;
• globally asymptotically stable if all solutions x(t) withu(t) = u0 converge to x0 as t → ∞.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Static relationships:
• for asymptotically stable stationary point (x0, u0), theoutput converges to y0 = h(x0, u0). Since x0 dependsimplicitly on u0,
y0 = h(x(u0), u0) = g(u0)
Here, g(u0) describes the stationary relation between u0
and y0.• Consider a small change in the input level from u0 to
u1 = u0 + δu0, the stationary output will be
y1 = g(u1) = g(u0 + δu0) ≈ g(u0) + g′(u0)δu0 = y0 + g′(u0)δu0.
Here g′(u0) : p ×m describes how the stationary outputvaries locally with the input→ static gain.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Linearization:
• system behavior in the neighborhood of a stationarysolution (x0, u0);
• consider small deviations ∆x(t) = x(t) − x0,∆u(t) = u(t) − u0 and ∆y(t) = y(t) − y0, then
∆x = A∆x + B∆u
∆y = C∆x + D∆u
where A , B , C and D are partial derivative matrices off(x(t), u(t)) and h(x(t), u(t)), i.e.
A =
∂f1∂x1
(x0, u0) . . .∂f1∂xn
(x0, u0)
......
∂fn∂x1
(x0, u0) . . .∂fn∂xn
(x0, u0)
;
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
• important and useful tool but• only for local properties;• quantitative accuracy difficult to estimate→ complement
with simulations of the original nonlinear system.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
ExampleFrom lecture notes by K.J. Åstrom, LTH
Model of bicycle dynamics:
d2θ
dt2=
mglJp
sin θ +mlV2
0 cos θ
bJp
(
tan β+a
V0 cos2 β
dβdt
)
where θ is the vertical tilt and β is front wheel angle (control).⇒ Hard to gain insight from nonlinear model. . .
Linearized dynamics (around θ = β = β = 0):
d2θ
dt2=
mglJp
θ +mlV2
0
bJp
(
β+aV0
dβdt
)
has transfer function
G(s) =mlV2
0
bJp×
1 + aV0
s
s2 −mglJp
.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Gain proportional to V20 :
• more control authority at high speeds.
Unstable pole at√
mglJp≈
√
g/l:
• slower when l is large;
• easier to ride a full size bike than a childrens bike.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Homework 1Consider the inverted pendulum dynamics:
x1 = x2,
x2 =u
m1− l0x4 + x1x2
4 + g sin (x3),
x3 = x4,
x4 =1
J0(x1) −m1l20[g (m2lc sin (x3) + m1x1 cos (x3))
−m1 (l0x4 + 2x2) x1x4 + −l0 u] ,J0(x1) = J + m1x2
1 ,
y =
[
x1
x2
]
where
Parameter name Value Meaningm1 0.213 kg Mass of the horizontal rod.m2 1.785 kg Mass of the vertical rod.l0 0.33 m Length of the vertical rod.lc −0.029 m Vertical rod c.g. position.g 9.807 m
s2 Gravity acceleration.J 0.055 Nm2 Nominal momentum of inertia.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
Analyze the system dynamics by
• linearizing the proposed model;
• writing the transfer function;
• interpreting the resulting equations.
Modeling andestimation for
control
E. Witrant
Systems andModelsWhat is a model?
How to build models?
How to verify models?
Mathematical models
ExamplesInverted pendulum
Tore Supra
Conclusions
Models forsystems andsignalsDifferential equations
State-space models
Stationary, stabilityand linearization
Homework
References
1 L. Ljung and T. Glad, Modeling of Dynamic Systems, PrenticeHall Information and System Sciences Series, 1994.
2 V. Misra, W.-B. Gong, and D. Towsley, Fluid-based analysis of anetwork of AQM routers supporting TCP flows with anapplication to RED, SIGCOMM 2000.
3 C. Hollot and Y. Chait, Nonlinear stability analysis for a class ofTCP/AQM networks, CDC 2001.
4 EW, D. Georges, C. Canudas de Wit and M. Alamir, On the useof State Predictors in Networked Control Systems, LNCSSpringer, 2007.
5 EW, E. Joffrin, S. Bremond, G. Giruzzi, D. Mazon, O. Barana etP. Moreau, A control-oriented model of the current profile inTokamak plasma, IOP PPCF 2007.
6 EW and S. Bremond, Shape Identification for DistributedParameter Systems and Temperature Profiles in Tokamaks,CDC 2011.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Modeling and Estimation for ControlPhysical Modeling
Lecture 2: Principles of physical modeling
Emmanuel [email protected]
April 22, 2014.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
The Three Phases of Modeling
“Successful modeling is based as much on a good feeling for
the problem and common sense as on the formal aspects that
can be taught”
1. Structuring the problem
• divide the system into subsystems, determine causes and
effects, important variables and interactions;
• intended use of the model?
• results in block diagram or similar description;
• needs understanding and intuition;
• where complexity and degree of approximation are
determined.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
2. Setting up the Basic Equations
• “fill in” the blocks using the laws of nature and basic
physical equations;
• introduce approximations and idealizations to avoid too
complicated expressions;
• lack of basic equations→ new hypotheses and innovative
thinking.
3. Forming the State-Space Models
• formal step aiming at suitable organization of the
equations/relationships;
• provides a suitable model for analysis and simulation;
• computer algebra can be helpful;
• for simulation: state-space models for subsystems along
with interconnections.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Example: the Mining Ventilation Problem
Objectives:
• propose a new automation strategy to minimize the fans
energy consumption, based on distributed sensing
capabilities: wireless sensor network;
• investigate design issues and the influence of sensors
location;
• find the optimal control strategy that satisfies safety
constraints.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Outline
1 The Phases of Modeling
2 1: Structuring the problem
3 2. Setting up the Basic Equations
4 3. Forming the State-Space Models
5 Simplified models
6 Conclusions
7 Homework
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Phase 1: Structuring the problem
Ask the good questions:
• What signals are of interest (outputs)?
• Which quantities are important to describe what happens
in the system?
• Of these quantities, which are exogenous and which
should be regarded as internal variables?
• What quantities are approximately time invariant and
should be regarded as constants?
• What variables affect certain other variables?
• Which relationships are static and which are dynamic?
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
General tips:
• often need experimental results to assist these steps (i.e.
time constants and influences);
• the intended use determines the complexity;
• use model to get insights, and insights to correct the
model;
• work with several models in parallel, that can have different
complexity and be used to answer different questions;
• for complex systems, first divide the system into
subsystems, and the subsystems into blocs.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Example: for the mining ventilation problem
• Inputs to the system:
• ρ: air density in vertical shaft;
• P: air pressure in vertical shaft;
• ∆H: variation of pressure produced by the fan;
• mj,in: incoming pollutant mass rate due to the engines;
• mj,chem: mass variation due to chemical reactions between
components;
• h: time-varying number of hops in WSN.
• Outputs from the system:
• cj(z, t) pollutants (COx or NOx ) volume concentration
profiles, where z ∈ [0; hroom] is the height in the extraction
room;
• uavg is the average velocity of the fluid in the tarpaulin tube;
• mj pollutant mass in the room;
• τwsn delay due to the distributed measurements and
wireless transmission between the extraction room and the
fan.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
• Division into subsystems:
• fan / tarpaulin tube / extraction room / wireless sensor
network.
• Corresponding block diagram:
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Phase 2: Setting up the Basic
Equations
Main principles:
• formulate quantitative I/O relationships;
• use knowledge of mechanics, physics, economics, . . .
• well-established laws, experimental curves (data sheets)
or crude approximations;
• very problem dependent!
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Two groups of relationships:
1 Conservation laws: relate quantities of the same kind, i.e.
• Pin − Pout = stored energy / unit time;
• inflow rate - outflow rate = stored volume / t;
• input mass flow rate - output mass flow rate = stored mass
/ t;
• Kirchhoff’s laws.
2 Constitutive relationships: relate quantities of differentkinds (i.e. voltage - current, level - outflow, pressure drop -flow)
• material, component or bloc in the system;
• static relationships;
• relate physical to engineering relationships;
• always approximate.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
How to proceed?
• write down the conservation laws for the block/subsystem;
• use suitable constitutive relationships to express the
conservation laws in the model variables. Calculate the
dimensions as a check.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Mining ventilation example (i.e. extraction room):
• Conservation law - conservation of mass for chemical
species j:
mj(t) = mj,in(t) − mj,out(t) − mj,chem(t)
• Constitutive relationship - relate the mass to concentration
profile:
mj(t) = Sroom
∫ hroom
0
cj(z, t)dz
= Sroom
[∫ hdoor
0
cj(z, t)dz + αj(t)∆h
]
,
and hypothesis on the shape (e.g. sigmoid):
cj(z, t) =αj(t)
1 + e−βj(t)(z−γj(t)).
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Phase 3: Forming the
State-Space Models
Straightforward recipe:
1 choose a set of state variables (memory of what has
happened, i.e. storage variables);
2 express the time derivative of each state as a function of
states and inputs;
3 express the outputs as functions of the state and inputs.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Examples of stored quantities:
• position of a mass / tank level (stored potential energy);
• velocity of a mass (stored kinetic energy);
• charge of capacitor (stored electrical field energy);
• current through inductor (stored magnetic energy);
• temperature (stored thermal energy);
• internal variables from step 2.
Make separate models
for the subsystems and diagram interconnections→ modularity
and error modeling diagnostic.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Extraction room model:
• shape parameters α, β and γ chosen as the state:
x(t) = [α, β, γ]T ;
• time derivative from mass conservation:
Ej
αj(t)
βj(t)γj(t)
= mj,in(t) − Bj ufan(t − τtarp) − Djk , with
Ej Sroom
Vint
......
...∂Cj,i
∂αj
∂Cj,i
∂βj
∂Cj,i
∂γj
......
...
+
∆h
0
0
T
Bj 1
hdoor
Vint
...
Cj,i
...
× Starp ν, Djk = Sroom
Vint
...
ηjk ,i Cj,i Ck ,i
...
+ ηjkαjαk∆h
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Number of state variables:
• sufficient if derivatives described by state and inputs;
• harder to determine unnecessary states;
• linear models→ rank of matrices;
• when used in simulation, the only disadvantage is related
to unnessary computations.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Simplified models
Even if a relatively good level of precision can be achieved, the
model has to be manageable for our purpose.
Model simplification:
• reduced number of variables;
• easily computable;
• linear rather than nonlinear;
• tradeoff between complexity and accuracy;
• balance between the approximations;
• three kinds:
1 small effects are neglected - approximate relationships are
used;
2 separation of time constants;
3 aggregation of state variables.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Small effects are neglected - approximate relationships
are used:
• i.e. compressibility, friction, air drag→ amplitude of the
resonance effects / energy losses?
• based on physical intuition and insights together with
practice;
• depends on the desired accuracy;
• linear vs. nonlinear: make experiments and tabulate the
results.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Separation of time constants:
• May have different orders of magnitude, i.e. for Tokamaks:Alfven time (MHD instabilities) 10−6 s
density diffusion time 0.1 − 1 s
heat diffusion time 0.1s-1s (3.4 s for ITER)
resistive diffusion time few seconds (100 − 3000 s for ITER)
• Advices:
• concentrate on phenomena whose time constants match
the intended use;
• approximate subsystems that have considerably faster
dynamics with static relationships;
• variables of subsystems whose dynamics are appreciably
slower are approximated as constants.
• Two important advantages:
1 reduce model order by ignoring very fast and very slow
dynamics;
2 by giving the model time constants that are on the same
order of magnitude (i.e. τmax/τmin ≤ 10 − 100), we get
simpler simulations (avoid stiffness!).
• When different time-scales are desired, consider the use
of different models.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Aggregation of state variables:
To merge several similar variables into one state variable: often
average or total value.
• i.e. infinite number of points in the extraction room→ 3
shape parameters, trace gas transport in firns;
• hierarchy of models with different amount of aggregation,
i.e. economics: investments / private and government /
each sector of economy / thousand state variables;
• partial differential equations (PDE) reduced to ordinary
differential equations (ODE) by difference approximation of
spatial variables.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Example: Heat conduction in a rod
• input: power in the heat source P;
• output: temperature at the other endpoint T ;
• heat equation:∂
∂tx(z, t) = a
∂2
∂z2x(z, t)
where x(z, t) is the temperature at time t at the distance z
from the left end point and a is the heat conductivity
coefficient of the metal;
• hypothesis: no losses to the environment;
• at the end points: a∂
∂zx(0, t) = P(t), x(L , t) = T(t)
• requires to know the whole function x(z, t1), 0 ≤ z ≤ L , to
determine T(t), t ≥ t1,→ infinite dimensional system.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Example: Heat conduction in a rod (2)
Aggregation of state variables: approximate for simulation
• divide the rode (x(z, t), 0 ≤ z ≤ L/3, aggregated into x1(t)etc.) and assume homogeneous temperature in each part
• conservation of energy for part 1:
d
dt(heat stored in part 1) = (power in) − (power out to part 2)
d
dt(C · x1(t)) = P − K(x1(t) − x2(t))
C: heat capacity of each part, K : heat transfer
• similarly:
d
dt(C · x2(t)) = K(x1(t) − x2(t)) − K(x2(t) − x3(t))
d
dt(C · x3(t)) = K(x2(t) − x3(t))
T(t) = x3(t)
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
• Rearrange the equations to obtain the linear state-space
model:
x(t) =K
C
−1 1 0
1 −2 1
0 1 −1
x +1
C
1
0
0
P
y(t) = (0 0 1) x(t)
• Conclusions: essentially the same as using finite
difference approximation on the spacial derivative
(homework), a finer division would give a more accurate
model.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Example: solving the air continuity in polar firns and ice
coresFrom poromechanics, firn =
system composed of the ice
lattice, gas connected to the
surface (open pores) and gas
trapped in bubbles (closed pores).Air transport is driven by:
∂[ρice(1 − ǫ)]
∂t+ ∇[ρice(1 − ǫ)~v ] = 0
∂[ρogasf ]
∂t+ ∇[ρo
gas f(~v + ~wgas)] = −~ro→c
∂[ρcgas(ǫ − f)]
∂t+ ∇[ρc
gas(ǫ − f)~v] = ~ro→c
with appropriate boundary and
initial conditions.Scheme adapted from [Sowers
et al.’92, Lourantou’08].
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
I.e. CH4 transport at NEEM (Greenland)
⇒ Unique archive of the recent (50-100 years) anthropogenic
impact. Can go much further (i.e. > 800 000 years) in ice.
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
From distributed to lumped dynamics
• Consider a quantity q transported in 1D by a flux u = qv
with a source term s (t ∈ [0, T ], z ∈ [0, zf ]):
∂q
∂t+∂
∂z[q v(z, t)] = s(z, t), with
q(0, t) = 0
q(x , 0) = q0(x)
where s(z, t) , 0 for z < z1 < zf and s = 0 for z1 < z < zf .
• Approximate ∂[qv]/∂z, i.e. on uniform mesh:
• backward difference: (uz)i =ui−ui−1
∆z+ ∆z
2(uzz)i
• central difference: (uz)i =ui+1−ui−1
2∆zi− ∆z2
6(uzzz)i
• other second order:
(uz)i =ui+1+3ui−5ui−1+ui−2
4∆zi+ ∆z2
12(uzzz)i −
∆z3
8(uzzzz)i
• third order: (uz)i =2ui+1+3ui−6ui−1+ui−2
6∆zi− ∆z3
12(uzzzz)i
• Provides the computable lumped model: dq/dt = Aq + s
• The choice of the discretization scheme directly affects the
definition of A and its eigenvalues distribution: need to
check stability and precision!
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
E.g. stability: eigenvalues of A for CH4 at NEEM with
dt ≈ 1 week
-2 -1.5 -1 -0.5 0 0.5 1 1.5 2-3
-2
-1
0
1
2
3
Real part of eigenvalues
Imag
inar
y pa
rt o
f eig
enva
lues
0 20 40 60 80 100 120 140 160 180-3
-2.5
-2
-1.5
-1
-0.5
0
0.5
Depth (m)
Rea
l par
t of e
igen
valu
es
FOUCentralFOU + centralFOU + 2nd orderFOU + 3rd order
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
E.g. eig(A) for CH4 at NEEM with dt ≈ 1 week, zoom
-15 -10 -5 0 5
x 10-3
-2
-1
0
1
2
Real part of eigenvalues
Imag
inar
y pa
rt o
f eig
enva
lues
0 20 40 60 80 100 120 140 160 180-3
-2.5
-2
-1.5
-1
-0.5
0
0.5
1
Depth (m)
Rea
l par
t of e
igen
valu
es
FOUCentralFOU + centralFOU + 2nd orderFOU + 3rd order
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Conclusions
→ Guidelines to structure the general approach for modeling
• The clarity of the model and its usage directly depends on
its initial philosophy
• Prevent the temptation to avoid the documentation of
“obvious steps”
• Forecasting the use of experimental knowledge and
sub-model validation strategies during the modeling
phases is essential
Principles of
physical
modeling
E. Witrant
The Phases of
Modeling
1: Structuring
the problem
2. Setting up
the Basic
Equations
3. Forming the
State-Space
Models
Simplified
models
Firn example
Conclusions
Homework
Homework 2Use finite differences to solve the heat conduction
a∂2
∂z2x(z, t) =
∂
∂tx(z, t), T(t) = x(L , t), P(t) = a
∂
∂zx(z, t)|z=0.
1 define the discretized state
X(t) [x1(t) . . . xi(t) . . . xN(t)]T as a spatial discretization
of x(z, t);
2 use the central difference approximation ∂2u∂z2 ≈
ui+1−2ui+ui−1
∆z2
to express dxi(t)/dt as a function of xi+1, xi and xi−1, for
i = 1 . . .N;3 introduce the boundary conditions
• with ∂u∂z(0, t) ≈ u1−u0
∆zto express x0 as a function of x1 and
P, then substitute in dx1/dt ;• with ∂u
∂z(L , t) ≈ uN+1−uN
∆zto express xN+1 as a function of xN ,
then substitute in dxN/dt (suppose that there is no heat
loss: ∂x(L , t)/∂z = 0);
4 write the discretized dynamics in the state-space form;
5 for N = 3 compare with the results obtained in class.
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Modeling and Estimation for ControlPhysical Modeling
Lecture 3: Some Basic Relationships in Physics
Emmanuel [email protected]
April 21, 2014.
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Introduction
• Most common relationships within a number of areas in
physics.
• More general relationships become visible.
⇒ General modeling strategy.
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Outline
1 Physic fundamentals
2 Electrical Circuits
3 Mechanical Translation
4 Mechanical Rotation
5 Flow Systems
6 Thermal System
7 Thermal Systems
8 Conclusions
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Physic fundamentals
Mass conservationFor a closed system: ∆M = 0.
Energy conservation
For an isolated system: ∆E = 0.
1st law of thermodynamics
Heat (Q) and Work (W ) are equivalent and can be exchanged.
∆E = ∆U +∆Ecin +∆Epot = Q + W .
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Electrical Circuits
Fundamental quantities:
voltage u (volt) and current i (ampere).
Components:Nature Relationship (law) Energy
Inductor
(L henry)i(t) =
1
L
∫ t
0
u(s)ds, u(t) = Ldi(t)
dt
T(t) = 12Li2(t)
(magnetic field E storage, J)
Capacitor
(C farad)u(t) =
1
C
∫ t
0
i(s)ds, i(t) = Cdu(t)
dt
T(t) = 12Cu2(t)
(electric field E storage)
Resistor
(R ohm)u(t) = Ri(t)
Nonlinear
resistanceu(t) = h1(t)i(t), i(t) = h2(t)u(t)
P(t) = u(t) · i(t)(loss, in watts, 1 W = 1 J/s)
Ideal
rectifierh2(t) =
x, x > 0
0, x ≤ 0
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Interconnections (Kirkhhoff’s laws):
∑
k
ik (t) ≡ 0 (nodes),∑
k
uk (t) ≡ 0 (loops).
Ideal transformer:transform voltage and current s.t. their product is constant:
u1 · i1 = u2 · i2, u1 = αu2, i1 =1
αi2
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Mechanical Translation
Fundamental quantities:
force F (newton) and velocity v (m/s), 3-D vectors (suppose
constant mass m = 0).
Components:Nature Relationship (law) Energy
Newton’s
force lawv(t) =
1
m
∫ t
0
F(s)ds, F(t) = mdv(t)
dt
T(t) = 12mv2(t)
(kinetic E storage)
Elastic bodies
(k N/m)F(t) = k
∫ t
0
v(s)ds, v(t) =1
k
dF(t)
dt
T(t) = 12k
F2(t)(elastic E storage)
FrictionF(t) = h(v(t))
Air drag
Dampers
h(x) = cx2sgn(x)h(x) = γx
P(t) = F(t) · v(t)(lost as heat)
Dry
frictionh(x) =
+µ if x > 0
F0 if x = 0
−µ if x < 0
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Interconnections:
∑
k
Fk (t) ≡ 0 (body at rest)
v1(t) = v2(t) = . . . = vn(t) (interconnection point)
Ideal transformer:force amplification thanks to levers:
F1 · v1 = F2 · v2
F1 = αF2
v1 =1
αv2
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Example: active seismic isolation control [2]
Mass - spring - damper approximation:
m4x4(t) = γ4(x3 − x4) + k4(x3 − x4)mi xi(t) = [γi(xi−1 − xi) + ki(xi−1 − xi)]
+[γi+1(xi+1 − xi)+ki+1(xi+1 − xi)], i = 2, 3
m1x1(t) = [γ1(x0 − x1) + k1(x0 − x1)]+[γ2(x2 − x1) + k2(x2 − x1)]+u(t)
m1x0(t) = Fearth(t)y(t) = [x0 + x1 x2 − x1]
T
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Mechanical Rotation
Fundamental quantities:
torque M [N ·m] and angular velocity ω [rad/s].
Components:Nature Relationship (law) Energy
Inertia
J [Nm/s2]ω(t) =
1
J
∫ t
0
M(s)ds, M(t) = Jdω(t)
dt
T(t) = 12Jω2(t)
(rotational E storage)
Torsional
stiffness kM(t) = k
∫ t
0
ω(s)ds, ω(t) =1
k
dM(t)
dt
T(t) = 12k
M2(t)(torsional E storage)
Rotational
frictionM(t) = h(ω(t)) P(t) = M(t) · ω(t)
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Interconnections:
∑
k
Mk (t) ≡ 0 (body at rest).
Ideal transformer:a pair of gears transforms torque and angular velocity as:
M1 · ω1 = M2 · ω2
M1 = αM2
ω1 =1
αω2
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Example: printer belt pulley [3]
Spring tension: T1 = k(rθ − rθp) = k(rθ − y)Spring tension: T2 = k(y − rθ)
Newton: T1 − T2 = md2y
dt2
Motor torque (resistance, L = 0: Mm = Km i =Km
Rv2
drives belts + disturb.: Mm = M + Md
T drives shaft to pulleys: M = Jd2θ
dt2+ h
dθ
dt+ r(T1 − T2)
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Flow Systems
Fundamental quantities:
for incompressible fluids, pressure p [N/m2] and flow Q [m3/s].
Fluid in a tube:
Pressure gradient ∇p force p · A
mass ρ · l · A flow Q = v · A
inertance [kg/m4] Lf = ρ · l/A
Constitutive relationships (Newton: sum of forces = mass ×
accel.):
Q(t) =1
Lf
∫ t
0
∇p(s)ds, ∇p(t) = Lf
dQ(t)
dt
T(t) = 12Lf Q
2(t)(kinetic E storage)
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Fluid in a tube (2):
Pressure drop from Darcy-Weisbach’s equation for a circular
pipe:∂P
∂x= f
(
L
D
) (
V2
2g
)
Friction factor for laminar flow (Re < 2300): f = 64Re
. For
turbulent flow, empirical formula or Moody Diagram:
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Flow in a tank:
• Volume V =∫
Qdt , h = V/A , and fluid capacitance
Cf A/ρg [m4s2/kg].
• Constitutive relationships:Bottom pres.
p = ρ · h · gp(t) =
1
Cf
∫ t
0
Q(s)dsT(t) = 1
2Cf p
2(t)(potential E storage)
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Flow through a section reduction:
• Pressure p, Flow resistance Rf , constant H .
• Constitutive relationships:
Pressure drop
Darcy’s law
area change
∇p(t) = h(Q(t))∇p(t) = Rf Q(t)
∇p(t) = H ·Q2(t) · sign(Q(t))
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Interconnections:
∑
k
Qk (t) ≡ 0 (flows at a junction),∑
k
pk ≡ 0 (in a loop)
Ideal transformer:
p1 · Q1 = p2 · Q2, p1 = αp2, Q1 =1
αQ2.
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Thermal System
Fundamental quantities:
Temperature T [K ], Entropy S [J/kg · K ],and heat flow rate q [W ].
3 ways to transfer heat:
• Conduction: Contact between 2 solids at different
temperatures
• Convection: Propagation of heat through a fluid (gas or
liquid)
• Radiation: 3rd principle of thermodynamics : P = ǫSσT4
(T > 0⇒ qrad > 0)
Thermal energy of a body or Fluid: Etherm = M · Cp · T
Heat transported in a Flow: q = m · h (h=enthalpy)
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Heat Conduction
Body heating:
Fourier’s law of conduction in 1D
k ·∂T2
∂x2= ρ · Cp ·
∂T
∂t
q(t) = M · Cp ·∂T
∂t,
where k [W/m · K ] is thermal conductivity of the body,
ρ [kg/m3] and M [kg] are the density and the mass of the body,
and Cp [W/(kg · K)] is the specific Heat of the body.
Interconnections:∑
k
qk (t) ≡ 0 (at one point).
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Heat Convection
Forced convection between a flowing fluid in a pipe:
h · Sw · (Tw(t) − T(t)) = −Mw · Cp,w ·dTw(t)
dt= q(t)
where T [K ] is the fluid temperature,
h [W/m2 · K ] is the heat transfert coefficient,
and Tw [K ], Mw [kg], Sw [m2] Cp,w [J/kg · K ] are the
temperature, mass, surface and specific heat of the pipe.
Interconnections:
∑
k
qk (t) ≡ 0 (at one point).
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Convective Heat Transfer
coefficient
Correlation for Forced internal turbulent Flow:Dittus-Boelter correlation (1930) with 10000 < Re < 120000.
h =k
DNu
where k is thermal conductivity of the bulk fluid, D is the
Hydraulic diameter and Nu is the Nusselt number.
Nu = 0.023 · Re0.8· Prn
with Re = ρ·v ·Dµ
is the Reynolds Number and Pr is the Prandtl
Number. n = 0.4 for heating (wall hotter than the bulk fluid) and
n = 0.33 for cooling (wall cooler than the bulk fluid). Precision
is ±15%
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Thermal Systems: summary
• Conduction in 0D:Thermal capacity
C [J/(K · s)]T(t) =
1
C
∫ t
0
q(s)ds, q(t) = CdT(t)
dt
• Interconnections:
q(t) = W∆T(t) (heat transfer between 2 bodies)∑
k
qk (t) ≡ 0 (at one point).
where W = hSw [J/(K · s)].
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Conclusions
Obvious similarities among the basic equations for different
systems!
Some physical analogies:System Effort Flow Eff. storage Flow stor. Static relation
Electrical Voltage Current Inductor Capacitor Resistor
Mechanical:
- Translational Force Velocity Body (mass) Spring Friction
- Rotational Torque Angular V. Axis (inertia) Torsion s. Friction
Hydraulic Pressure Flow Tube Tank Section
Thermal Temperature Heat flow rate - Heater Heat transfer
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
Characteristics:
1 Effort variable e;
2 Flow variable f ;
3 Effort storage: f = α−1 ·∫
e;
4 Flow storage: e = β−1 ·∫
f ;
5 Power dissipation: P = e · f ;
6 Energy storage via I.: T = 12α
f2;
7 Energy storage via C.: T = 12β
e2;
8 Sum of flows equal to zero:∑
fi = 0;
9 Sum of efforts (with signs) equal to zero:∑
ei = 0;
10 Transformation of variables: e1f1 = e2f2.
• Note: analogies may be complete or not (i.e. thermal).
⇒ Create systematic, application-independent modeling from
these analogies (next lesson).
Principles of
physical
modeling
E. Witrant
Physic
fundamentals
Electrical
Circuits
Mechanical
Translation
Mechanical
Rotation
Flow Systems
Thermal
System
Heat Conduction
Heat Convection
Thermal
Systems
Conclusions
References
1 L. Ljung and T. Glad, Modeling of Dynamic Systems,
Prentice Hall Information and System Sciences Series,
1994.
2 Noriaki Itagaki and Hidekazu Nishimura, “Gain-Scheduled
Vibration Control in Consideration of Actuator Saturation”,
Proc. of the IEEE Conference on Control Applications, pp
474- 479, vol.1, Taipei, Taiwan, Sept. 2-4, 2004.
3 R.C. Dorf and R.H. Bishop, Modern Control Systems, 9th
Ed., Prentice Hall, 2001.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Modeling and Estimation for ControlPhysical Modeling
Lecture 4: Bond Graphs
Emmanuel WITRANT
April 22, 2014
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Basic Concepts behind Bond
Graphs [S. Stramigioli’01]
• Mathematical modeling: mathematical relations, generally
without constraints or physical interpretation.
• Physical modeling: physical concepts and restrict to keep
some physical laws.
Bond-graph
• satisfy 1st principle of thermodynamics: energy
conservation
• self-dual graphs where:
vertices = ideal physical concepts (storage or transformation of
energy)
edges - power bonds - = lossless transfer of energy (i.e. water
pipes, energy from one part to the other in the system)
⇒ excellent tool for describing power-consistent networks of
physical systems.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Causality
• Block diagrams: exchange of information takes place
through arrows, variable x going from A to B = causal
exchange of information
but often physically artificial and not justified, i.e. resistor
• Bond graphs: causality not considered in the modeling
phase, only necessary for simulation.
Energy
• one of the most important concepts in physics
• dynamics is the direct consequence of energy exchange
• lumped physical models: system = network
interconnection of basic elements which can store,
dissipate or transform energy
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Outline
1 Physical Domains and Power Conjugate Variables
2 The Physical Model Structure and Bond Graphs
3 Energy Storage and Physical State
4 Free Energy Dissipation
5 Ideal Transformations and Gyrations
6 Ideal Sources
7 Kirchhoff’s Laws, Junctions and the Network Structure
8 Bond Graph Modeling of Electrical Networks
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Physical Domains and Power
Conjugate Variables
Physical domains:
• Discriminate depending on the kind of energy that a
certain part of the system can store, i.e. kinetic energy of
a stone thrown in the air→ translational mechanical -
potential energy of a capacitor→ electrical domain.
• Most important primal domains:
• mechanical = mechanical potential & mechanical kinetic;
• electromagnetic = electrical & magnetical;
• hydraulic = hydraulic potential & hydraulic kinetic;
• thermic: only one without dual sub-domains, related to the
irreversible transformation of energy to the thermal
domain.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Power conjugate variables:
• Similarity among domains (cf. Lesson 3), i.e. oscillator
• In each primal domain: two special variables, power
conjugate variables, whose product is dimensionally equal
to power
• Efforts and flows:
Domain Effort Flow
Mechanical Translation force F velocity v
Mechanical Rotation torque τ angular velocity ω
Electro-magnetic voltage v current i
Hydraulic pressure p flow rate Q
Thermic temperature T entropy flow E
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
The Physical Model Structure and
Bond Graphs
Energetic ports:
• physical modeling→ atomic elements like the storage,
dissipation, or transformation of energy;
• external variables = set of flows and dual vectors;
• effort-flow pairs = energetic ports since their dual product
represents the energy flow through this imaginary port.
Bond graphs as a graphical language:
1 extremely easy to draw;
2 mechanical to translate into block diagram or differential
equations;
3 a few rules and it is impossible to make the common “sign
mistakes” of block diagrams.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Energetic bonds:
• edges in the graph, represent the flow of energy (e.g.
water pipes);
• notations: effort value above or left, flow under or right;
• rules:
A B A B
1 each bond represents both an effort e and a dual flow f ;
2 the half arrow gives the direction of positive power P = eT f
(energy flows);
3 effort direction can be, if necessary, specified by the causal
stroke & dual flow goes ALWAYS in the opposite direction
(if not an element could set P independently of destination
→ extract infinite energy).
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Network structure:
• if 2 subsystems A and B , both the effort and flow MUST
be the same: interconnection constraint that specifies how
A and B interact;
• more generally, interconnections and interactions are
described by a set of bonds and junctions that generalize
Kirchhoff’s laws.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Energy Storage and Physical
State
Identical structure for physical lumped models
• Integral form characterized by:
1 an input u(t), always and only either effort or flow;
2 an output y(t), either flow or effort;
3 a physical state x(t);4 an energy function E(x).
• State-space equations: x(t) = u(t), y(t) =∂E(x(t))
∂x• Change in stored energy:
E =dE
dt=∂E(x)
∂x
Tdx
dt= yT u = Psupplied
→ half arrow power bonds always directed towards
storage elements (E > 0)!
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
e.g. Capacitor:
x
E
∂∂
∫ q v
u x y
i
flow effortstate
Ce
f
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Bond graphs representations
• Depending whether u is an effort or a flow in the integralform, two dual elements:
• C element: has flow input u and dual effort output y;
• I element: has effort input u and dual flow output y.
• Causal representations:
generalized displacement
q(t) = q(t0) +∫ t
t0f(s)ds
differential form→ γ−1(e)
generalized potential energy E(q)
co-energy
E∗(e)⇒ γ−1(e) =∂E∗(e)
∂e
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
generalized momenta
p(t) = p(t0) +∫ t
t0e(s)ds
differential form→ γ−1(f)
generalized kinetic energy E(p)
co-energy
E∗(f)⇒ γ−1(f) =∂E∗(f)
∂f
• Multidimensional I indicated by I and multidimensional C =
C.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Mechanical domain
C - Spring:
• input u = velocity v, generalized displacement∫
v = x,
stored potential energy E(x) = 12kx2, effort
y =∂E
∂x= kx = F (elastic force);
• holds for ANY properly defined energy function, which is
the ONLY information characterizing an ideal storage of
energy;
• e.g. nonlinear spring: E(x) = 12kx2 + 1
4kx4⇒
y = F =∂E
∂x= kx + kx3;
• linear spring, co-energy E∗(F) =1
2
F2
k,
x = γ−1(F) =∂E∗(F)
∂F=
F
k.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
I - Effort as input, kinetic mechanical domain:
• input u = force F ,∫
F = p = mv (momenta) by Newton’s
law (holds if m(t))⇒ proper physical state for kinetic E
storage: momentum p;
• E(p) =1
2
p2
m, y = v = γ(p) =
∂E
∂p=
p
m;
• kinetic co-energy E∗(v) =1
2mv2,
p = γ−1(v) =∂E∗(v)
∂v= mv.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Electrical domain:
• proper physical states: charge q and flux φ, NOT i and v;
C - Storage in electrostatic domain:
• u = i, physical state∫
i = q (generalized displacement),
stored potential energy E(q) =1
2
q2
C(co-energy
E∗(v) =1
2Cv2), effort y =
∂E
∂q=
q
C= v;
• e.g. nonlinear capacitor: E(q) =1
2
q2
C+
1
4
q4
C⇒
y = v =q
C+
q3
C.
• using co-energy, q = γ−1(v) =∂E∗(v)
∂v= Cv.
I - Ideal inductor:
• u = v,∫
v = φ, E(φ) =1
2
φ2
L, where L induction
constant, y = i =φ
L.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Energy storage:
• Generalized states:
Domain Gen. momentum (∫
e) Gen. displacement (∫
f )
Mech. Translational momentum p displacement x
Mech. Rotational ang. momentum m ang. displacement θ
Electromagnetic flux linkage φ charge q
Hydraulic pressure mom. Pp volume V
Thermic NON EXISTENT entropy E
• Storage elements:
1 what are the real physical states?
2 energy function provides for the equation;
3 argument→ what physical ideal element it represents;
4 the only ideal physical elements to which a state is
associated are energy storage;
5 in bond graphs, the power bond connected to a storage
element must always be directed toward the element.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Duality
• 2 storage / physical domain but thermal (generalized
potential and kinetic energy storage) = dual;
• one major concept in physics: oscillations if interconnected
dual elements, e.g. spring-mass or capacitor-inductor;
• thermal domain does NOT have both = irreversibility of
energy transformation due to a lack of “symmetry”.
Extra supporting states
• states without physical energy;
• e.g. position of a mass translating by itself: physical state
p, position x =∫
v = p/m but if the measurement is x and
not v:(
p
x
)
=
(
0
p/m
)
+
(
u
0
)
, y = x
⇒ total state (p, x)T , physical state p, supporting state x
needed for analysis without associated physical energy.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Free Energy Dissipation
Principle:
• irreversible transformation, e.g. mechanical or electrical→
thermal;
• dissipation of energy is transformation (1st principle of
thermodynamics);
• dissipation of free-energy (math.: Legendre transformation
of energy with respect to entropy), e.g. ideal electrical
resistors or mechanical dampers;
• ideal dissipator characterized by a purely statical
(no-states) effort/flow relation: e = Z(f) (Impedance form)
or f = Y(e) (Admittance form) for which Z(f)f < 0 or
eY(e) < 0 (energy flowing toward the element)
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Electrical domain
• Ohm’s law: u = Ri and i = u/R;
• causally invertible;
• r : constant R of a linear element (r = R).
Mechanical domain
• viscous damping coefficient b: F = bv and v = F/b,
r = b.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Ideal Transformations and
Gyrations
Electrical domain
• elements with two power ports = two power bonds;
• ideal, power continuous, two port elements: power flowing
from one port (input bond) ≡ one flowing out from other
port (output bond)⇒ cannot store energy inside.
• e.g. ideal transformer:
• input and output bonds with positive power flow in and out;
• external variables: (ein, fin) = power flowing in from input
port and (eout , fout ) = power flowing out from other port;
• power continuity: Pin = eTin
fin = eTout fout = Pout
• linear relation between one of the external variable on one
port to one of the external variables on the other port;
• flow-flow→ ideal transformers, flow-effort→ ideal gyrators
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Ideal Transformers
• relation: linear between flows and dependent linear
between efforts;
• characterizing equation: fout = nfin where n: linear
constant characterizing the transformer
• power constraint: ein = neout ⇔ eout =1nein
⇒ if 2 ports belong to same domain and n < 1, ein < eout
but fin > fout .
• e.g. gear-boxes: ein = torque applied on pedal axis and fin= angular velocity around the pedals, (eout , fout ) on the
back wheel;
• n relates the efforts in one way and also the flows in the
other way;
• if n variable: modulated TF (extra arrow).
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Ideal Gyrators
• linear constant between effort of output port and flow of
input port: eout = nfin;
• power constraint: ein = nfout ⇔ fout =1nein;
• e.g. gyrative effect of a DC motor (electrical power flows in
and mechanical power flows out): out torque τ = Ki,
power continuity→ u = Kω (e.m.f.):
Electrical
domain
(
i
u
)
→
←
(
τ
ω
)
Rotational
domain
• if n variable: modulated gyrator.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Multi-bonds
• characteristic constant→ matrix, if variable→ modulated
transformer or gyrator;
• Transformers:
• TF, MTF;
• f2 = Nf1 ⇒ e1 = NT e2 (using eT1
f1 = eT2
f2);
• Gyrators:
• GY, MGY, SGY;
• e2 = Nf1 ⇒ e1 = NT f2;
• e = Sf with S = −ST =
[
0 −NT
N 0
]
• if N = identity matrix: symplectic gyrator SGY (algebraic
relationship, can be used to dualize C into I).
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Ideal Sources
• Supply energy: ideal flow source and ideal effort source.
• Only elements from which the power bond direction goes
out: Psource = eT f .
• Supply a certain effort or flow independently of the value
of their dual flow and effort.
• e.g. ideal voltage and current source in the electrical
domain
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Kirchhoff’s Laws, Junctions and
the Network Structure
• How we place the bricks with respect to each other
determines the energy flows and dynamics
• Generalization of Kirchhoff’s laws, network structure→
constraints between efforts and flows
• Two basic BG structures: 1 junctions = flow junctions and
0 junctions = effort junctions
• Any number of attached bonds
• Power continuous (in = out)
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
1-junctions:
• flow junction: all connected bonds are constrained to have
the same flow values;
• causality: only one bond sets the in flow and all other
bonds use it (strokes constraint);
• equations:
fi1 = · · · = fim = fo1 = · · · = fon (flow equation),m∑
k=1
eik =n∑
k=1
eok (effort equation);
• mesh Kirchhoff ’s law in electrical networks: same current
and the algebraic potential sum = 0;
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Electrical example:
• same current→
flow junction
• all bonds point to R, C and I and
source bond point out→ all
signs are automatically correct;
• I (integral causality) “sets” the
junction current (mesh) and
other elements have this current
as input and voltages as outputs;
• complete dynamics describedby:
• effort equation:
Vs = Vr + Vc + Vl
• I element: φ = Vl and i = φ/L• q element: q = i and
Vc = q/C
• R element: Vr = Ri
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
0-junctions:
• effort junction: all connected bonds constrained to have
same efforts;
• causality: only one bond sets ein and all other bonds use
it;
• equations:
ei1 = · · · = eim = eo1 = · · · = eon (effort equation),m∑
k=1
fik =n∑
k=1
fok (flow equation);
• current Kirchhoff ’s law: in a node, algebraic current sum =
0.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Effort difference:
• need the difference of two efforts to specify power
consistent interconnection with other elements;
• all flows are the same and
m∑
k=1
eik =n∑
k=1
eok ⇒ e1 = e2 + e3 ⇔ e3 = e1 − e2.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Flow difference:
• need the difference of two flows to specify power
consistent interconnection with other elements;
• all efforts are the same and
m∑
k=1
fik =n∑
k=1
fok ⇒ f1 = f2 + f3 ⇔ f3 = f1 − f2.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Bond Graph Modeling of
Electrical Networks
Algorithm:
1 for each node draw a 0-junction which corresponds to the
node potential;
2 for each bipole connected between two nodes, use effort
difference where a bipole is attached and connect the
ideal element to the 0-junction representing the difference.
3 choose a reference (v = 0) and attach an effort source
equal to zero to the corresponding 0-junction.
4 simplify:
• eliminate any junction with only 2 attached bonds and have
the same continuing direction (one in and one out);
• fuse 1 and 0-junctions that are connected through a
single-bond;
• eliminate all junctions after the 0 reference source that do
not add any additional constraint.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Bond Graph Modeling of
Mechanical Systems
Algorithm:
1 for each moving mass draw a 1-junction = mass velocity;
2 add an additional 1-junction for inertial reference with an
attached Sf = 0;
3 for each inertia attach a corresponding I element to the
one junction corresponding to its velocity;
4 for each damper or spring: flow difference for ∆v attach to
the 1-junction;
5 simplify the graph by:
• eliminating all junctions with only two bonds in the same
continuing direction;
• fuse 1 and 0-junctions connected through a single-bond;
• eliminate all the junctions after the reference source which
do not add any additional constraints.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Examples
DC motor example
• 6 interconnected lumps:
• 2 storage elements with corresponding physical states
(φ, p): ideal inductor L and rotational inertia I → 2 states
and order 2 model;
• 2 dissipative elements: the resistor R and the friction b;
• 1 gyration effect K ;
• an ideal voltage source u.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
• Elements equations:
• storage elements and physical states:
Inertia
p = τI
ω =∂EI
∂p=∂
∂p
(
1
2Ip2
)
=p
I
Inductor
φ = ul
i =∂EL
∂φ=∂
∂φ
(
1
2Lφ2
)
=φ
L
• dissipation (linear): ur = Ri and τb = bω (dissipating
torque);
• gyration equations: τ = Ki and um = Kω
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
• Network interconnection:
• use previous algorithms to describe the electrical and
mechanical parts;
• introduce the gyrator to connect the two domains→
inter-domain element;(a) Preliminary diagram drawing:
• 0-junctions of electrical to indicate the connection points of
the bipoles;
• mechanical: 1-junctions = angular rotation of the wheel and
reference inertial frame (source);
• gyrator = relation from flow i to effort τ⇒ 1 to 0 junction;
• torque applied between the wheel and ground.
• simplifications:
(b) eliminate the two zero sources and attached junctions;
(c) eliminate any junction with only two bonds attached to it;
(d) mix all the possible directly communicating junctions of the
same type.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Intuitively:
• electrical part = series connection source, resistor,
inductor and electrical gyrator side→ 1-junction;
• mechanical part: only the velocity w is present, the motor
applies a torque to the wheel, but part of it is “stolen” by
the dissipating element.
• final equations⇒ LTI state-space form:
p = τI = τ − τb = Ki − bω =K
Lφ −
b
Ip,
φ = ul = −um − ur + u = −K
Ip −
R
Lφ+ u
d
dt
(
p
φ
)
=
(
−b/I K/L
−K/I −R/L
)
︸ ︷︷ ︸
A
(
p
φ
)
+
(
0
1
)
︸︷︷︸
B
u
y ω = (1/I 0)︸ ︷︷ ︸
C
(
p
φ
)
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Multidimensional example
• two point masses connected by an elastic translational
spring and a damper;
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
• bond graph:
• note: all bonds attached to 1-junction have the same flows
and all attached to 0-junction the same effort;
• “:: E(q)” = energy function, q = energy variable (p1, p2) for
I and position diff. ∆x for elastic;
• ideal source→ constant force = gravitation for each mass;
• “: b” for dissipative element indicates Fr = b(v2 − vl).
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Conclusions
Bond graphs:
• Provide a systematic approach to multiphysics modeling
• Based on the fundamental laws of energy conservation
• Fundamental theory = port-Hamiltonian systems
• Used in industry with dedicated numerical solvers (e.g.
20-Sim)
• Needs practice!
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Homework 3
Draw the bond graph model of the printer belt pulley problem
introduced in Lesson 3 and check that you obtain the same
equations.
Bond Graphs
E. Witrant
Power
Conjugate
Variables
Structure and
Bond Graphs
Storage and
state
Energy
Dissipation
Transformations
and Gyrations
Ideal sources
Junctions
Electrical
Networks
Mechanical
Systems
Examples
Conclusions
Reference
1 S. Stramigioli, Modeling and IPC Control of Interactive
Mechanical Systems: A Coordinate-free Approach,
Springer, LNCIS 266, 2001.
2 L. Ljung and T. Glad, Modeling of Dynamic Systems,
Prentice Hall Information and System Sciences Series,
1994.
Computer-
aided
modeling
E. Witrant
Computer
Algebra
Analytical
Solutions
Algebraic
Modeling
Automatic
Bond Graphs
Translation
Numerical
Methods
Conclusions
Homework
Modeling and Estimation for ControlSimulation
Lecture 5: Computer-aided modeling
Emmanuel WITRANT
April 22, 2014
Computer-
aided
modeling
E. Witrant
Computer
Algebra
Analytical
Solutions
Algebraic
Modeling
Automatic
Bond Graphs
Translation
Numerical
Methods
Conclusions
Homework
Introduction
x = f(x , u)
y = h(x , u)
• Can contain complex calculations.
• Computer assistance?
• Computer algebra.
• 2 systematic ways to state-space: algebraic and bond
graphs.
• Numerical limitations.
Computer-
aided
modeling
E. Witrant
Computer
Algebra
Analytical
Solutions
Algebraic
Modeling
Automatic
Bond Graphs
Translation
Numerical
Methods
Conclusions
Homework
Outline
1 Computer Algebra
2 Analytical Solutions
3 Algebraic Modeling
4 An Automatic Translation of Bond Graphs to Equations
5 Numerical Methods - a short glance
Computer-
aided
modeling
E. Witrant
Computer
Algebra
Analytical
Solutions
Algebraic
Modeling
Automatic
Bond Graphs
Translation
Numerical
Methods
Conclusions
Homework
Computer Algebra
• Methods for manipulating mathematical formulas (,
numerical calculations).
• Numerous softwares: Macsyma, Maple, Reduce, Axiom,
Mathematica...
• Examples of capabilities:
• Algebraic expressions: (x + y)2 = x2 + 2xy + y2
• Factorizations: x3 − y3 = (x − y)(x2 + xy + y2)• Symbolic differentiation
∂
∂z(x2z + sin yz + a tan z) = x2 + y cos yz +
a
1 + z2
• Symbolic integration
∫
√
1 + x2dx =1
2(arc sinhx + x
√
x2 + 1)
Computer-
aided
modeling
E. Witrant
Computer
Algebra
Analytical
Solutions
Algebraic
Modeling
Automatic
Bond Graphs
Translation
Numerical
Methods
Conclusions
Homework
Analytical Solutions
• May have partial interesting results, i.e.
x1 = f1(x1, x2)
x2 = f2(x1, x2)
solution algorithm generates F(x1, x2) = C if possible,
continue from this to
x1 = φ1(t)
x2 = φ2(t).
F is called the integral of the system, geometrically = path
in x1 − x2 plane, but do not have velocity information.
Computer-
aided
modeling
E. Witrant
Computer
Algebra
Analytical
Solutions
Algebraic
Modeling
Automatic
Bond Graphs
Translation
Numerical
Methods
Conclusions
Homework
Example: the pendulum
θ = ω
ω = −g
lsin θ
has integral 12ω2 − g
lcos θ = C which represents the energy
(kinetic + potential) of the system.
Figure: Pendulum trajectories in θ − ω plane
Computer-
aided
modeling
E. Witrant
Computer
Algebra
Analytical
Solutions
Algebraic
Modeling
Automatic
Bond Graphs
Translation
Numerical
Methods
Conclusions
Homework
Algebraic Modeling→ Transform the equations into a convenient form.
Introduction of state variables for higher-order differential
equations:
• Consider
F(y , y , . . . , yn−1, yn; u) = 0,
• introduce the variables
x1 = y , x2 = y, . . . , xn = yn−1,
• we get
x1 = x2, , x2 = x3, . . . , xn−1 = xn
F(x1, x2, . . . , xn, xn; u) = 0
→ state-space description provided xn can be solved for the
last equation.
Computer-
aided
modeling
E. Witrant
Computer
Algebra
Analytical
Solutions
Algebraic
Modeling
Automatic
Bond Graphs
Translation
Numerical
Methods
Conclusions
Homework
Example
• Let
y(3)2 − y2y4 − 1 = 0.
• With x1 = y, x2 = y, x3 = y , we get
x1 = x2
x2 = x3
x23 − x2
2 x41 − 1 = 0
• The last equation can be solved for x3 and gives
x1 = x2
x2 = x3
x3 = ±√
x22x4
1+ 1
Note: 2 cases if we don’t know the sign of y(3) = x3 from
physical context.
Computer-
aided
modeling
E. Witrant
Computer
Algebra
Analytical
Solutions
Algebraic
Modeling
Automatic
Bond Graphs
Translation
Numerical
Methods
Conclusions
Homework
Systems of higher-order differential equations:
• two higher-order differential equations in 2 variables
F(y, y , . . . , yn−1, yn; v , v, . . . , vm−1; u) = 0
G(y , y , . . . , yn−1; v , v , . . . , vm−1, vm; u) = 0
• introduce the variables
x1 = y , x2 = y, . . . , xn = yn−1,
xn+1 = v , xn+2 = v , . . . , xn+m = vm−1,
• we get
x1 = x2, x2 = x3, . . . , xn−1 = xn
F(x1, x2, . . . , xn , xn; xn+1, . . . , xn+m ; u) = 0
xn+1 = xn+2, . . . , xn+m−1 = xn+m
G(x1, x2, . . . , xn; xn+1, . . . , xn+m, xn+m; u) = 0
⇒ state-space description if xn and xn+m can be solved in F
and G.
Computer-
aided
modeling
E. Witrant
Computer
Algebra
Analytical
Solutions
Algebraic
Modeling
Automatic
Bond Graphs
Translation
Numerical
Methods
Conclusions
Homework
• Example:
y + v + yv = 0 (1)
y2
2+
v2
2− 1 = 0 (2)
Problem: highest order derivatives in same equation
Computer-
aided
modeling
E. Witrant
Computer
Algebra
Analytical
Solutions
Algebraic
Modeling
Automatic
Bond Graphs
Translation
Numerical
Methods
Conclusions
Homework
• Solution:
• differentiate (2) twice gives (3);
• (1)×v-(3) =(4);
• (4)×v2 & vv eliminated with (3) gives (5);
• eliminate v thanks to (2)→ eq. in y only.
• Can be generalized to an arbitrary number of equations
provided all equations are polynomial in the variables and
their derivatives.
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An Automatic Translation of Bond
Graphs to Equations
From a simple example:
• Introduce the state x = αf2 for I: x = e2;
• imagine a list of equations with ei and fi computed from v
and x, e1 = v first and f1 = f2 last (or f1 = f3);
e1 = v
...
f1 = f2
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1) from I element: f2 = x/α, dual
e2 = x2 = e1 − e3 (junction output)→second to last so that e1 and e3 are
calculated before:
e1 = v
f2 =1
αx
...
x = e2 = e1 − e3
f1 = f2
2) What variables are defined by first
2 equation? Junction→ flows and R:
e1 = v
f2 =1
αx
f3 = f2
e3 = βf3
x = e2 = e1 − e3
f1 = f2
⇒ starting from v and x, all variables evaluated in proper order.
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• successive substitutions gives a compact state-space
description:
x = e1 − e3 = e1 − βf3 = e1 − βf2 = e1 −β
αx = v −
β
αx
→ choose 2 lists, forward and backward, instead of one.
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Algorithms for Equation Sorting
1 Choose a source and write its input in forward list and the
equation of its dual in backward list.
2 From adjacent bonds, if some variable is defined in terms
of already calculated variables, write its equation in the
forward list and the equation of the other bond variable in
the backward list, as far as possible.
3 Repeat 1 and 2 until all sources have been treated.
4 Choose an I element and write the equation fi =1αi
xi in
forward list and xi = ei = . . . in backward list.
5 Do the analogy of step 2.
6 Repeat 4 and 5 until all I elements have been processed.
7 Do the analogy of steps 4, 5, and 6 for all C elements
(ei =1βi
xi to forward list and xi = fi backward list.
8 Reverse the order of the backward list and put it after the
forward list.
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Example: DC motor
• State variables:
x1 =
∫ t
v2dτ = L1i2,
∫ t
M2dτ = Jω2
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• Create the list:
Step Forward list Backward list
1 v1 = v i1 = i22 i2 = 1
L1x1 x1 = v2 = v1 − v3 − v4
2 i3 = i2 v3 = R1i32 i4 = i2 v4 = kω1
2 M1 = ki4 ω1 = ω2
4 ω2 = 1Jx2 x2 = M2 = M1 −M3
5 ω3 = ω2 M3 = φ(ω3)
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• Reverse backward list after forward list:
v1 = v
i2 =1
L1
x1
i3 = i2
i4 = i2
M1 = ki4
ω2 =1
Jx2
ω3 = ω2
M3 = φ(ω3)
x2 = M2 = M1 −M3
ω1 = ω2
v4 = kω1
v3 = R1i3
x1 = v2 = v1 − v3 − v4
i1 = i2
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• Eliminating all variables that are not states gives:
x1 = v −R1
L1
x1 −k
Jx2
x2 =k
L1
x1 − φ(x2/J)
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Numerical MethodsPhysical model→ state-space equations→ scaling (same
order of magnitude to avoid numerical problems)→ impact of
discretization in simulation.
Basis of Numerical Methods:
• Consider the state-space model
x = f(x(t), u(t))
where x ∈ Rn. If fixed input u(t) = u(t), u is a time
variation and
x = f(t , x(t))
x(0) = x0
we want an approximation of x at 0 < t1 < t2 < · · · < tf →x1, x2, x3, . . . approximate x(t1), x(t2), x(t3), . . .
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• Simplest algorithm: difference ratio = Euler’s method:
xn+1 − xn
h≈ x(tn) = f(tn, xn), where h = tn+1 − tn
⇒ xn+1 = xn + h · f(tn , xn)
more generally
xn+1 = G(t , xn−k+1, xn−k+2, . . . , xn, xn+1)
where k is the number of utilized previous steps→ k-step
method. If xn+1 not in G: explicit method (i.e. Euler),
otherwise implicit.
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Firn example: gas in open pores (1)
Impact of the convection term discretization on the trace gases
mixing ratio at NEEM (EU hole)
0 10 20 30 40 50 60 70
310
320
330
340
350
360
370
380
Depth (m)
CO2 firn − NEEM EU
CO
2 (
ppm
)
Figure: For 100 (‘ · · · ’), 200 (‘- - -’) and 395 (‘—’) depth levels
(∆z ≈ 0.8, 0.4 and 0.2 m, respectively): Lax-Wendroff (blue,
reference), central (red) and first order upwind (green).
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Firn example: gas in open pores (2)
Impact of time discretization on the trace gases mixing ratio at
NEEM (EU hole, ∆z = 0.2 m and a zoom on specific region)
0 10 20 30 40 50 60375
380
385
390
Depth (m)
CO2 firn − NEEM EU
CO
2 (
ppm
)
Figure: Explicit with a sampling time ts=15 minutes (red), implicit
(blue) with ts = 1 day (‘—’), 1 week (‘– – –’) and 1 month (‘- - -’), and
implicit-explicit (green) with ts = 1 week (‘—’) and 1 month (‘– – –’).
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Firn example: gas in open pores (3)
Averaged simulation time per gas associated with the proposed
time-discretization schemes for NEEM EU (1800 to 2008, full
close-off depth at 78.8 m, 12 gases, left) and South Pole 1995 (1500
to 1995, full close-off depth at 123 m), obtained on a PC laptop
equipped with the processor i5 540 m (2.53 Ghz, 3 Mo):
Method ts ∆z a Simulation time a
Implicit 1 day 0.2 m 4.02 / 22.25 s
Implicit 1 week 0.2 m 0.63 / 3.91 s
Implicit 1 month 0.2 m 0.26 / 1.48 s
Explicit 15 min 0.2 m 5.09 / 29.45 min
Explicit 30 min 0.4 / 0.61 m 24.39 s / 1.34 min
Explicit 1 h 0.8 / 1.23 m 7.19 s / 12.13 s
Imp-explicit b 1 week 0.2 m 0.63 s / 3.77 s
Imp-explicit b 1 month 0.2 m 0.27 s / 1.48 sa : NEEM EU / South Pole; b : Crank-Nicholson.
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• Accuracy determined by the global error
En = x(tn) − xn
but hard to compute→ one-step (provided exact previous
steps), local error
en = x(tn) − zn, zn = G(t , x(tn−k ), x(tn−k+1), . . . , zn)
i.e. for Euler (xn+1 ≈ xn + h · f(tn, xn))
en+1 = x(tn+1) − zn+1 = x(tn+1) − x(tn) − h · f(tn, x(tn))
=h2
2x(ζ), for tn < ζ < tn+1
Note (Taylor):
x(tn+1) = x(tn) + h · f(tn , x(tn)) +h2
2· f ′(tn, x(tn)) + O(3)
→ local error proportional to h2 and global error
proportional to h (number of steps proportional to h−1).
If local error O(hk+1), k is the order of accuracy.
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• Stability is also crucial. i.e.
x = λx , λ ∈ Cx(0) = 1
with Euler: xn+1 = xn + hλxn = (1 + hλ)xn has solution
xn = (1 + hλ)n .
It implies that
xn → 0 if |1 + hλ| < 1
|xn | → ∞ if |1 + hλ| > 1
stable if Re [λ] < 0 AND |1 + hλ| < 1 (h small enough)
→ the stability of the DE does not necessarily coincides
with the one of the numerical scheme!
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The Runge-Kutta Methods:
Consider the integral form
x(tn+1) = x(tn) +
∫ tn+1
tn
f(τ, x(τ))dτ
with central approximation
xn+1 = xn + h · f(tn +h
2, x(tn +
h
2))
and (Euler) x(tn +h2) ≈ xn +
h2f(tn, xn). Consequently, we have
the simplest Runge-Kutta algorithm
k1 = f(tn, xn),
k2 = f(tn +h
2, xn +
h
2k1),
xn+1 = xn + hk2.
Local error x(tn+1) − xn+1 = O(h3)→ 1 order of magnitude
more accurate than Euler.
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• General form:
k1 = f(tn , xn),
k2 = f(tn + c2h, xn + ha21k1,
k3 = f(tn + c3h, xn + h(a31k1 + a32k2)),
...
ks = f(tn + csh, xn + h(as1k1 + · · ·+ as,s−1ks−1)),
xn+1 = xn + h(b1k1 + · · ·+ bsks),
where s, ci, bi and aij chosen to obtain the desired order
of accuracy p, calculation complexity or other criterion→family of Runge-Kutta methods.
• A classic method sets s = p = 4 with
c2 = c3 =1
2, c4 = 1, a21 = a32 =
1
2, a43 = 1,
b1 = b4 =1
6, b2 = b3 =
2
6, (others = 0)
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Adams’ Methods:
• Family of multistep methods
xn = xn−1 +k
∑
j=0
βj fn−j , fi = f(ti , xi)
where βj chosen such that the order of accuracy is as high
as possible. If β0 = 0: explicit form (accuracy k + 1),
Adams-Bashforth, while β0 , 0: implicit form (accuracy k ),
Adams-Moulton.
• Simplest explicit forms:
k = 1 : xn = xn−1 + fn−1h
k = 2 : xn = xn−1 + (3fn−1 − fn−2)h
2
k = 3 : xn = xn−1 + (23fn−1 − 16fn−2 + 5fn−3)h
12
k = 4 : xn = xn−1 + (55fn−1 − 59fn−2 + 37fn−3 − 9fn−4)h
24
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• Simplest implicit forms:
k = 1 : xn = xn−1 + fnh
k = 2 : xn = xn−1 + (fn + fn−1)h/2
k = 3 : xn = xn−1 + (5fn + 8fn−1 − fn−2)h/12
k = 4 : xn = xn−1 + (9fn + 19fn−1 − 5fn−2 + fn−3)h/24
• Why more complicated implicit methods?
(a) Adams-Bashforth (explicit),
k = 1 (–), k = 2 (−−) and
k = 3 (· · · ).
(b) Adams-Moulton (implicit),
k = 2 (–) and k = 3 (−−).
⇒ Larger stability regions. Note: ր k ց stability.
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Variable Step Length:
• Fixed steps often inefficient→ large steps when slow
changes & small steps when rapid changes.
• Automatic adjustment based on local error approximation,
i.e. assume a local error
x(tn+1) − xn+1 = Chp+1 + O(hp+2)
where C depends on the solution (unknown). If 2 steps of
length h, we have approximately (errors are added)
x(tn+2) − xn+2 = 2Chp+1 + O(hp+2) (1)
x value computed for a step of length 2h from tn to tn+2:
x(tn+2) − x = C(2h)p+1 + O(hp+2) (2)
(2) − (1) : xn+2 − x = 2Chp+1(2p − 1) + O(hp+2) (3)
C from (3) in (1) : x(tn+2) − xn+2 =xn+2 − x
2p − 1+ O(hp+2)
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Previous result:
x(tn+2) − xn+2 =xn+2 − x
2p − 1+ O(hp+2)
Assume O(hp+2) negligible→ known estimate of the error.
• The estimate can be used in several ways, in general:
ց h if error > tolerance,
ր h if error≪ tolerance.
Ideally, a given accuracy is obtained with minimum
computational load.
• Crucial issue for embedded control and large-scale plants.
Most of the time, use existing softwares/libraries.
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Stiff differential equations:
• Both fast and slow components and large difference
between the time constants, i.e.
x =
(
−10001 −10000
1 0
)
x
x(0) =
(
2
−1.0001
)
has solution
x1 = e−t + e−10000t
x2 = −e−t − 0.0001e−10000t .
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• Problem: in simulation, start with very small step to follow
the fast term (i.e. e−10000t ), which soon goes to zero:
solution only characterized by slow term. BUTր h implies
stability problems (i.e. −10 000 · h within stability region).
⇒ use methods that are always stable: compromise with
accuracy (implicit in general).
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Comments about Choice of Methods:
• Runge-Kutta most effective for low complexity
(computational work) while Adams better for high
complexity;
• methods for stiff problems - may be - ineffective for nonstiff
problems;
• problem dependent.
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Conclusions
• First step:
• from the physical model (high order or bond graphs), write
the system in a state-space form
• investigate the behavior of the continuous dynamics, e.g.
nonlinearities, time-delays, time constants of the linearized
dynamics . . .
• Second step:
• discretize the dynamics to get computable difference
equations
• check on the impact of the discretization step
• advanced methods with predictor/corrector schemes
• From experience:
• hand-made discretizations are often more tractable
• when doing “equivalent” model transformations, they are
more equivalent in the discretized framework
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Homework 4
a. Write the state-space description for:
• Example 1:
y + v2 + y = 0
y2 + v + vy = 0
• Example 2:
y + v3 + v2 + y = 0
y2 + v + vy = 0
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b. Numerical methods
Consider the differential equa-
tion
y′′(t) − 10π2y(t) = 0
y(0) = 1, y(0) = −√
10π
1 Write this equation in
state-space form.
2 Compute the eigenvalues.
3 Explain the difference
between exact and
numerical difference
expressed in Table 8.6.3.
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References
1 L. Ljung and T. Glad, Modeling of Dynamic Systems,
Prentice Hall Information and System Sciences Series,
1994.
2 W.E. Boyce and R.C. Di Prima, Elementary Differential
Equations and Boundary Value Problems, 6th edition,
John Wiley & Sons, Inc., 1997.
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Simulating ODEs
advanced call
more options
Implicit diff eq
Linear systems
Boundary
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Simulating BVPs
COLNEW
optimal control
Difference
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Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Modeling and Estimation for ControlSimulation
Lecture 6: Simulation with Scilab
Emmanuel WITRANT
April 22, 2014
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Ordinary
differential
equations
Simulating ODEs
advanced call
more options
Implicit diff eq
Linear systems
Boundary
value problems
Simulating BVPs
COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Models for simulation
• Choose the appropriate simulation tool/function depending
on the class of model
• I.e. Scilab provides a wide array of tools for different
models.
• Can use abbreviated commands and defaults parameters.
• Important to know appropriate tools, how the algorithms
are set up and how to face difficulties.
Simulation toolsThree forms:
1 primary tools used by knowledgeable users on challenging
problems;
2 simplified version easier to use and for simpler problems;
3 special cases occurring in specific areas of science and
engineering.
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Ordinary
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Simulating ODEs
advanced call
more options
Implicit diff eq
Linear systems
Boundary
value problems
Simulating BVPs
COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Outline
1 Ordinary differential equations
2 Boundary value problems
3 Difference equations
4 Differential algebraic equations
5 Hybrid systems
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Implicit diff eq
Linear systems
Boundary
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Simulating BVPs
COLNEW
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Difference
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Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Ordinary differential equations
(ODEs)
y = f(t , y), y(t0) = y0
where y, f vector valued, and t ∈ R.
• Higher order models can always be transformed into 1st
order and directly simulated in Scilab, except Boundary
value problems.
• Unique solution if f and ∂f/∂y continuous.
• The most continuous derivatives of f(t , y) exist, the more
derivatives y has. In simulation, accuracy obtained from
error estimates that are based on derivatives.
• Controlled differential equation (DE):
y = f(t , y , u(t))
y has only one more derivative than u→ may create
problems for piecewise continuous inputs.
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Implicit diff eq
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Difference
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Simulation
Differential
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Simulation
Example
Hybrid systems
Simulation
Conclusions
Simulating ODEs: simplest call
y=ode(y0,t0,t,f)
• t0, y0, f(t , y)→ default method and error tolerance, adjust
step size;
• many more solutions than needed: specify also final time
vector t;
• returns y= [y(t0), y(t1), . . . , y(tn)];
• online function definition, i.e. f(t , y) = −y + sin(t)
function ydot = f(t,y)
ydot=-y+sin(t)
endfunction
• interface to ode solvers like ODEPACK.
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COLNEW
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Difference
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Simulation
Differential
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Simulation
Example
Hybrid systems
Simulation
Conclusions
Simulating ODEs: advanced call
[y,w,iw]=ode([type],y0,t0,t [,rtol [,atol]],f [,jac]
... [w,iw])
• ”type”: lsoda (default, automatically selects between
nonstiff predictor-corrector Adams and stiff backward
difference formula BDF), adams, stiff (BDF), rk
(adaptive Runge-Kutta of order 4), rkf (RK 45, highly
accurate), fix (simpler rkf), root (lsodar with root
finding), discrete.
• ”rtol, atol”: real constants or vectors, set absolute and
relative tolerance on y: ǫy(i) = rtol(i) ∗ |y(i)|+ atol(i),computational time vs. accuracy.
• ”jac”: external, analytic Jacobian (for BDF and implicit)
J=jac(t,y).
• ”w,iw”: real vectors for storing information returned by
integration routine.
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Implicit diff eq
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Simulation
Differential
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Simulation
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Hybrid systems
Simulation
Conclusions
Simulating ODEs: more options
odeoptions[itask,tcrit,h0,hmax,hmin,jactyp,mxstep,
maxordn,maxords,ixpr,ml,mu]
• sets computation strategy, critical time, step size and
bounds, how nonlinear equations are solved, number of
steps, max. nonstiff and stiff order, half-bandwidths of
banded Jacobian.
• computational time and accuracy can vary greatly with the
method.
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Ordinary
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advanced call
more options
Implicit diff eq
Linear systems
Boundary
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Simulating BVPs
COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Simulating ODEs: Implicit differential equations
• A(t , y)y = g(t , y), y(t0) = y0. If A not invertible ∀(t , y) of
interest→ implicit DAE, if invertible→ linearly implicit DE
or index-zero DAE.
• Better to consider directly than inverting A (more efficient
and reliable integration).
y=impl([type],y0,ydot0,t0,t [,atol, [rtol]],res,adda
... [,jac])
→ requires also y(t0) and to compute the residuals
(g(t , y) − A(t , y)y) as: r=res(t,y,ydot)
Computer-
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E. Witrant
Ordinary
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Simulating ODEs
advanced call
more options
Implicit diff eq
Linear systems
Boundary
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Simulating BVPs
COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Simulating ODEs: Linear systems
• number of specialized functions for
x = Ax + Bu, x(0) = x0,
y = Cx + Du.
• [sl]=syslin(dom,A,B,C [,D [,x0] ]) defines a
continuous or discrete (dom) state-space system, system
values recovered using [A,B,C,D]=abcd(s1);
• [y [,x]]=csim(u,t,sl,[x0])→ simulation (time
response) of linear system.
Simulating ODEs: Root finding
• to simulate a DE up to the time something happens;
• y,rd[,w,iw]=ode(root”,y0,t0,t [,rtol [,atol]],f [,jac],ng,g
[,w,iw])” integrate ODE f until g(t , y) = 0;
• iteratively reduces the last step to find surface crossing.
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E. Witrant
Ordinary
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Simulating ODEs
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Implicit diff eq
Linear systems
Boundary
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Simulating BVPs
COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Boundary value problems (BVPs)
• DE with information given at 2 or more times:
y = f(t , y), t0 ≤ t ≤ tf ,
0 = B(y(t0), y(tf)).
If y is n-dimensional→ n boundaries.
• More complicated than initial value problems (cf.
Optimization class), where local algorithm move from one
point to the next.
• BVP: need more global algorithm with full t interval→
much larger system of equations.
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Ordinary
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Implicit diff eq
Linear systems
Boundary
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COLNEW
optimal control
Difference
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Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Simulating BVPs: Numerous methods
1 shooting methods: take given IC then guess the rest and
adjust by integrating the full interval→ easy to program
but not reliable on long intervals and stiff problems;
2 multiple shooting: breaks time interval into subinterval and
shoot over these;
3 discretize the DE and solve the large discrete system, i.e.
Euler with step h on y = f(t , y), t0 ≤ t ≤ tf ,
0 = B(y(t0), y(tf )) gives:
yi+1 − yi − f(t0 + ih, yi) = 0, i = 0, . . . ,N − 1,
B(y0, yN) = 0.
usually with more complicated methods than Euler but
large system of (nonlinear) DE→ BVP solver has to deal
with numerical problems and need Jacobian-like
information.
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E. Witrant
Ordinary
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Simulating ODEs
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Implicit diff eq
Linear systems
Boundary
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COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Simulating BVPs: COLNEW
Scilab uses Fortran COLNEW code in bvode, which assumes
that the BVP is of the form
dmi ui
dxmi= fi
(
x , u(x),du
dx, . . . ,
dmi−1u
dxmi−1
)
, 1 ≤ i ≤ nc ,
gi
(
ζj , u(ζj), . . . ,dm∗u
dxm∗
)
= 0, j = 1, . . . ,m∗,
where ζj are x where BC hold and aL ≤ x ≤ aR .
Let m∗ = m1 + m2 + · · ·+ mnc, z(u) =
[
u, dudx, . . . , dm∗u
dxm∗
]
, then
dmi ui
dxmi= fi (x, z(u(x))) , 1 ≤ i ≤ nc , aL ≤ x ≤ aR
gi (ζj , z(u(ζj))) = 0, j = 1, . . . ,m∗,
bvode starts with initial mesh, solve NL system and iteratively
refines the mesh.
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Ordinary
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Simulating ODEs
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Implicit diff eq
Linear systems
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COLNEW
optimal control
Difference
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Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Simulating BVPs: COLNEW implementation
[z]=bvode(points,ncomp,m,aleft,aright,zeta,ipar,ltol,
...tol,fixpnt,...fsub1,dfsub1,gsub1,dgsub1,guess1)
• solution z evaluated at the given points for ncomp≤ 20
DE;
• we have to provide bounds (aleft,aright) for u, BCs and
numerical properties of the model.
Computer-
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E. Witrant
Ordinary
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Simulating ODEs
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Implicit diff eq
Linear systems
Boundary
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optimal control
Difference
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Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Simulating BVPs: Example - optimal controlNecessary conditions: consider the NL controlled system
y = y2 + v, J(y, u) =
∫ 10
0
10v2 + y2 dt
Find v : y(0) = 2→ y(10) = −1, while min J. NC found fromHamiltonian and give the BV DAE
y = y2 + v,
λ = −2y − 2λy,
0 = 20v + λ,y(0) = 2, y(10) = −1.
⇒ BVP :
y = y2 − λ/20,
λ = −2y − 2λy,
y(0) = 2, y(10) = −1.
Ready to be solved by bvode, which gives:
Computer-
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Ordinary
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Implicit diff eq
Linear systems
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optimal control
Difference
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Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Difference equations
• Discrete-time values or values changing only at discrete
times, for discrete processes or because of isolated
observations.
• Integer variable k and sequence y(k) that solves
y(k + 1) = f(k , y(k)), y(k0) = y0,
or with time sequence tk , k ≥ k0:
z(tk+1) = g(tk , z(tk )), z(tk0) = z0.
If evenly spaced events tk+1 − tk = h = cst :
v(k + 1) = g(w(k), v(k)), v(k0) = v0,
w(k + 1) = w(k) + h, w(k0) = tk0
Computer-
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Ordinary
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Simulating ODEs
advanced call
more options
Implicit diff eq
Linear systems
Boundary
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COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Difference equations (2)
• Solution existence simpler than DE: y(k) computed
recursively from y(k0) as long as (k , y(k)) ∈ Df .
• Note: uniqueness theorem for DE (if 2 solutions start at
the same time but with different y0 and if continuity of f , fyholds, then they never intersect) not true for difference
equations.
• Can always be written as 1st order difference equations.
Computer-
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E. Witrant
Ordinary
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Simulating ODEs
advanced call
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Implicit diff eq
Linear systems
Boundary
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COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Simulating difference equations
1 Easier because no choice about time step and no error
from derivatives approximations→ only function
evaluation and roundoff errors.
2 First order y(k + 1) = f(k , y(k)), y(k0) = y0, evaluated
by y=ode(discrete”,y0,k0,kvect,f)” where kvect =
evaluation times.
3 Linear systems
x(k + 1) = Ax(k) + Bu(k), x(0) = x0,
y(k) = Cx(k) + Du(k),
• [X]=ltitr(A,B,U,[x0]) or
[xf,X]=ltitr(A,B,U,[x0]);
• If given by a transfer function
[y]=rtitr(Num,Den,u [,up,yp]) where [,up,yp] are
past values, if any;
• Time response obtained using
[y [,x]]=flts(u,sl [,x0]).
Computer-
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E. Witrant
Ordinary
differential
equations
Simulating ODEs
advanced call
more options
Implicit diff eq
Linear systems
Boundary
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Simulating BVPs
COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Differential algebraic equations
(DAEs)
• Most physical models are differential + algebraic (DAEs):
F(t , y , y) = 0
→ rewrite as ODE or simpler DAE, or simulate the DAE
directly.
• Theory much more complex than ODEs: ∃ solutions only
for certain IC, called consistent IC, i.e.
y1 = y1 − cos(y2) + t ,
0 = y31 + y2 + et ,
requires y1(t0)3 + y2(t0) + et0 = 0.
Computer-
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E. Witrant
Ordinary
differential
equations
Simulating ODEs
advanced call
more options
Implicit diff eq
Linear systems
Boundary
value problems
Simulating BVPs
COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Differential algebraic equations (2)
• Structure→ index definition (≥ 0, 0 for ODE). Index-oneDAE in Scilab: F(t , y , y) = 0 with Fy , Fy is an index-onematrix pencil along solutions and Fy has constant rank:
1 implicit semiexplicit:
F1(t , y1, y2, y1) = 0
F2(t , y1, y2) = 0
where ∂F1/∂y1 and ∂F2/∂y2 nonsingular, y1 is the
differential variable and y2 the algebraic one;
2 semiexplicit:
y1 = F1(t , y1, y2)
0 = F2(t , y1, y2)
with ∂F2/∂y2 nonsingular.
Computer-
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E. Witrant
Ordinary
differential
equations
Simulating ODEs
advanced call
more options
Implicit diff eq
Linear systems
Boundary
value problems
Simulating BVPs
COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Simulating DAEs
• Need information on both y(t0) and y(t0) to uniquely
determine the solution and start integration, i.e.
tan(y) = −y + g(t)→ family of DE
y = tan−1(−y + g) + nπ. Sometimes approximate value of
y(t0) or none at all.
• Scilab uses backward differentiation formulas (BDF), i.e.
backward Euler on F(t , y , y) = 0 gives
F
(
tn+1, yn+1,yn+1 − yn
h
)
= 0
→ given yn, iterative resolution using the Jacobian w.r.t.
yn+1: Fy +1hFy ′ .
• based on DASSL code (for nonlinear fully implicit
index-one DAEs):
[r [,hd]]=dassl(x0,t0,t [,atol,[rtol]],res [,jac]
where x0 is y0 [ydot0], res returns the residue
r=g(t,y,ydot) and info sets computation properties.
Computer-
aided
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E. Witrant
Ordinary
differential
equations
Simulating ODEs
advanced call
more options
Implicit diff eq
Linear systems
Boundary
value problems
Simulating BVPs
COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Example tan(y) = −y + 10t cos(3t), y(0) = 0
y(0) = nπ is a consistent IC→ compare y(0) = 0, π, 2π
implicit ODEs and fully implicit DAEs can have multiple nearby
roots→ integrators must ensure no jump on another solution
when making a step (conservatism in the step size choice).
DAEs and root-finding:
[r,nn,[,hd]]=dasrt(x0,t0,t [,atol,[rtol]]
...,res [,jac],ng, surf [,info] [,hd]): just add
intersection surface.
Computer-
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E. Witrant
Ordinary
differential
equations
Simulating ODEs
advanced call
more options
Implicit diff eq
Linear systems
Boundary
value problems
Simulating BVPs
COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Hybrid systems
• Mixture of continuous- and discrete-time events.
• When an event (discrete variable change) occurs: change
in DE, state dimension, IC (initialization problem). . .
• Interfere with error control of integrators.
• Handled in Scilab and more particularly Scicos.
Computer-
aided
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E. Witrant
Ordinary
differential
equations
Simulating ODEs
advanced call
more options
Implicit diff eq
Linear systems
Boundary
value problems
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COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Simulating Hybrid systems
• Continuous variable yc and discrete variable yd (piecewise
constant on [tk , tk+1[):
yc(t) = f0(t , yc(t), yd(t)), t ∈ [tk , tk+1[
yd(tk+1) = f1(t , yc(tk+1), yd(tk )) at t = tk+1
i.e. sampled data system (u is a control function):
yc(t) = f0(t , yc(t), u(t)), t ∈ [tk , tk+1[,
u(tk+1) = f1(t , yc(tk+1), u(tk )) at t = tk+1.
• yt=odedc(y0,nd,stdel,t0,t,f), where
y0=[y0c;y0d],
stdel=[h, delta] with delta=delay/h,
yp=f(t,yc,yd,flag).
Computer-
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E. Witrant
Ordinary
differential
equations
Simulating ODEs
advanced call
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Implicit diff eq
Linear systems
Boundary
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COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
Conclusions
• Implementing the equations needs some dedicated
thinking
• Need to understand the expected results prior to
computation
• Trade-off:
• computation time vs. precision
• mathematical simplicity vs. code efficiency
• Particularly challenging for real-time modeling
• A code aims to be transmitted to other people: the
structure and comments have to be clear!
Computer-
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E. Witrant
Ordinary
differential
equations
Simulating ODEs
advanced call
more options
Implicit diff eq
Linear systems
Boundary
value problems
Simulating BVPs
COLNEW
optimal control
Difference
equations
Simulation
Differential
algebraic
equations
Simulation
Example
Hybrid systems
Simulation
Conclusions
References
1 S. Campbell, J-P. Chancelier and R. Nikoukhah, Modeling
and Simulation in Scilab/Scicos, Springer, 2005.
2 Scilab website: http://www.scilab.org.
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Modeling and Estimation for ControlSystem Identification
Lecture 7: Signals for System Identification
Emmanuel WITRANT
April 23, 2014
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Basics of System Identification
System identification = use of data in modeling
• Include experimental data in modeling work.
• Used to find constants or complete the model.
• Based on system variables: inputs, outputs and possibly
disturbances.
→ understand how the system works, describe partial
systems and compute values of the constants.
• Three different ways to use identification for modeling:
1 make simple experiments to facilitate problem structuration
(phase 1);
2 describe I/O relationships independently of physical
insights (often linear);
3 use data to determine unknown parameters in physical
models: tailor-made models.
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Estimate system from measurements of u(t) and y(t)
Many issues:
• choice of sampling frequency, input signal (experiment
conditions);
• what class of models, how to model disturbances?
• estimating model parameters from sampled, finite and
noisy data.
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Outline
1 From Continuous Dynamics to Sampled Signals
2 Disturbance Modeling
3 Signal Spectra
4 Choice of Sampling Interval and Presampling Filters
⇒ Introduction to signal analysis and processing
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
From Continuous Dynamics to
Sampled Signals
Continuous-time signals and systemsContinuous-time signal y(t)
Fourier transform Y(ω) =∫ ∞
−∞y(t)e−iωt dt
Laplace transform Y(s) =∫ ∞
−∞y(t)e−st dt
Linear system y(t) = g ∗ u(t)Y(ω) = G(ω)U(ω)Y(s) = G(s)U(s)
Derivation operator p × u(t) = u(t) works as s-variable, but in
time domain.
Example (0 IC) y(t) = 0.5u(t) + u(t)y(t) = (0.5p + 1)u(t)
Y(s) = (0.5s + 1)U(s)
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Discrete-time signals and systemsDiscrete-time signal y(kh)
Fourier transform Y (h)(ω) = h∑∞
k=−∞ y(kh)e−iωkh
z-transform Y(z) =∑∞
k=−∞ y(kh)z−k
Linear system y(kh) = g ∗ u(kh)
Y (h)(ω) = Gd(eiωh)U(h)(ω)
Y(z) = Gd(z)U(z)Shift operator q × u(kh) = u(kh + h) works as z-variable, but in
time-domain.
Example (0 IC) y(kh) = 0.5u(kh) + u(kh − h)y(kh) = (0.5 + q−1)u(kh)Y(z) = (0.5 + z−1)U(z)
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Sampled systems
Continuous-time linear system
x(t) = Ax(t) + Bu(t)
y(t) = Cx(t) + Du(t)
⇒ G(s) = C(sI − A)−1B + D.
Assume that we sample the inputs and outputs of the system
Relation between sampled inputs u[k ] and outputs y[k ]?
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Sampled systems (2)
Systems with piecewise constant input:
• Exact relation possible if u(t) is constant over each
sampling interval.
• Solving the system equations over one sampling interval
gives
x[k + 1] = Adx[k ] + Bdu[k ]
y[k ] = Cx[k ] + Du[k ]
Gd(z) = C(zI − Ad)−1Bd + D
where Ad = eAh and Bd =∫ h
0eAsBds.
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Sampled systems (3)
Example: sampling of scalar system
• Continuous-time dynamics
x(t) = ax(t) + bu(t)
• Assuming that the input u(t) is constant over a sampling
interval
x[k + 1] = adx[k ] + bdu[k ]
where ad = eah and bd =∫ h
0easbds = b
a(eah − 1).
• Note: continuous-time poles in s = a, discrete-time poles
in z = eah .
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Sampled systems (4)
Frequency-domain analysis of sampling
• Transfer function of sampled system
Gd(z) = C(zI − Ad)−1Bd + D
produces same output as G(s) at sampling intervals.
• However, frequency responses are not the same! One has
|G(iω) − Gd(eiωh)| ≤ ωh
∫ ∞
0
|g(τ)|dτ
where g(τ) is the impulse response for G(s).
⇒ Good match at low frequencies (ω < 0.1ωs)⇒ choose
sampling frequency > 10× system bandwidth.
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Sampling of general systems
• For more general systems,
• nonlinear dynamics, or
• linear systems where input is not piecewise constant
conversion from continuous-time to discrete-time is not
trivial.
• Simple approach: approximate time-derivative with finite
difference:
p ≈1 − q−1
hEuler backward
p ≈q − 1
hEuler forward
p ≈2
h×
q − 1
q + 1Tustin’s approximation
(typical for linear systems)
. . .
• I.e. write x(tk ) = x(tk − 1) +∫ tk
tk−1f(τ) dτ and find the
previous transformations using different integral
approximations
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Numerical approximations of the integral [F. Haugen’05]
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Disturbance Modeling
• Discrete-time set-up:⇒ Estimate Gd from
measurements y[k ] and u[k ].The effect of disturbances is
crucial, need for a disturbance
model!• Basic observations:
• disturbances are different from time to time
• some characteristics (e.g., frequency content) persist
• Can be captured by describing disturbances as filtered
white noise v(k) = H(q)e(k)
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Discrete-time stochastic processes
• Discrete-time stochastic process: an infinite sequence
v(k , θ) whose values depend on a random variable θ
• To each fixed value θ∗ of θ, the sequence v(k , θ∗)depends only on k and is called a realization of the
stochastic process
• For a discrete-time stochastic process v[k ], we define its
Expected or mean value mv(k) = Eθv[k ]Auto-correlation function Rv(k , l) = Eθv[k + l]v[k ]
and say that v[k ] is
stationary if mv and Rv are independent of k
ergodic if mv and Rv can be computed from
a single realization
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Some background
• define the real random variable e the possible outcomes
of unpredictable experiment;
• define fe(x) the probability density function:
P(a ≤ e < b) =
∫ b
a
fe(x)dx
• the expectation is
Ee =
∫
R
xfe(x)dx or (discrete) Ee =∑
xiP[X = xi ]
• the covariance matrix is
Cov(e, y) = E[(e − E(e))(y − E(y))T ] = E(ey) − E(e)E(y)
=∑
i,j
(ei − E(e))(yj − E(y))P[e = ei , y = yj ] (discrete)
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Some background (2)
White noise:
• A stochastic process e[k ] is called white noise
if me = 0 and
Re(k , l) =
σ2 if l = 0
0 otherwise
Unpredictable sequence!
Signals and auto-correlation function (ACF)
• Different realizations may look very different.
• Still, qualitative properties captured as:
- slowly varying ACF↔
slowly varying process;
- quickly varying ACF↔
quickly varying process.
• Close to white noise if R(l)→ 0 rapidly as |l| grows.
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Some background (3)
Properties of the auto-correlation function [Wikipedia]
• Symmetry: ACF is even (Rf (−l) = Rf (l) if f ∈ R) or
Hermitian (conjugate transpose, Rf (−l) = R∗f(l) if f ∈ C)
• Peak at the origin (|Rf(l)| ≤ Rf (0)) and the ACF of a
periodic function is periodic with the same period (dirac at
0 if white noise)
•∑
uncorrelated functions (0 cross-correlation ∀l) =∑
ACF of each function
• Estimate: for discrete process X1,X2, . . . ,Xn with known
mean µ and variance σ:
R(l) ≈1
(n − l)σ2
n−l∑
t=1
(Xt − µ)(Xt+k − µ), ∀l < n ∈ N+
• unbiased if true µ and σ
• biased estimate if sample mean and variance are used
• can split the data set to separate the µ and σ estimates
from the ACF estimate
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Signal Spectra
A common framework for deterministic and stochastic
signals
• Signals typically described as stochastic processes with
deterministic components (det. inputs vs. stoch.
disturbances).
• For a linear system with additive disturbance e(t)(sequence of independent random variables with
me(t) = 0 and variances σ2)
y(t) = G(q)u(t) + H(q)e(t)
we have that
Ey(t) = G(q)u(t)
so y(t) is not a stationary process.
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Quasi-Stationary Signals (QSS)
Definition: s(t) is QSS if
1 Es(t) = ms(t), |ms(t)| ≤ C, ∀t (bounded mean)
2 Es(t)s(r) = Rs(t , r), |Rs(t , r)| ≤ C, and the following limit
exists
limN→∞
1
N
N∑
t=1
Rs(t , t − τ) = Rs(τ), ∀τ (bounded autocor.)
where E is with respect to the stochastic components of s(t).
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Quasi-Stationary Signals (2)
• If s(t) deterministic then s(t) is a bounded sequence
such that
Rs(τ) = limN→∞
1
N
N∑
t=1
s(t)s(t − τ)
exists (E has no effect).
• If s(t) stationary, the bounds are trivially satisfied and
Rs(τ) do not depend on t .
• Two signals s(t) and w(t) are jointly quasi-stationary if
both QSS and if the cross-covariance
Rsw(τ) = Es(t)w(t−τ), with Ef(t) limN→∞
1
N
N∑
t=1
Ef(t), exists.
• Uncorrelated signals if Rsw(τ) ≡ 0.
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Definition of Spectra
• Power spectrum of s(t) (freq. content of stoch. process,
always real):
φs(ω) =∞∑
τ=−∞
Rs(τ)e−iτω
e.g. for white noise φs(ω) = σ2: same power at all
frequencies.
• Cross-spectrum between w(t) and s(t) (measures how
two processes co-vary, in general complex):
φsw(ω) =∞∑
τ=−∞
Rsw(τ)e−iτω
ℜ(φsw)→ cospectrum, ℑ(φsw)→ quadrature spectrum,
arg(φsw)→ phase spectrum, |φsw | → amplitude spectrum.
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Definition of Spectra (2)
• From the definition of inverse Fourier transform:
Es2(t) = Rs(0) =1
2π
∫ π
−π
φs(ω)dω
• Example (stationary stochastic process): for the process
v(t) = H(q)e(t), the spectrum is φv(ω) = σ2|H(eiω)|2.
• Example (spectrum of a mixed det. and stoch. signal): for
the signal
s(t) = u(t) + v(t),
where u(t) deterministic and v(t) stationary stochastic
process, the spectrum is φs(ω) = φu(ω) + φv(ω).
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Transformation of spectrum by linear systems
• Theorem: Let w(t) QSS with spectrum φw(ω), G(q)stable and s(t) = G(q)w(t). Then s(t) is also QSS and
φs(ω) = |G(e iω)|2φw(ω)
φsw(ω) = G(e iω)φw(ω)
• Corollary: Let y(t) given by
y(t) = G(q)u(t) + H(q)e(t)
where u(t) QSS, deterministic with spectrum φu(ω), and
e(t) white noise with variance σ2.Let G and H be stable filters, then y(t) is QSS and
φy(ω) = |G(e iω)|2φu(ω) + σ2|H(e iω)|2
φyu(ω) = G(e iω)φu(ω)
⇒ We can use filtered white noise to model the character of
disturbances!
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Spectral factorization
• The previous theorem describes spectrum as real-valued
rational functions of eiω from transfer functions G(q) and
H(q).In practice: given a spectrum φv(ω), can we find H(q) s.t.
v(t) = H(q)e(t) has this spectrum and e(t) is white
noise? Exact conditions in [Wiener 1949] & [Rozanov
1967]
• Spectral factorization: suppose that φv(ω) > 0 is a rational
function of cos(ω) (or eiω), then there exists a monic
rational function (leading coef. = 1) of z, H(z), without
poles or zeros on or outside the unit circle, such that:
φv(ω) = σ2|H(eiω)|2
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Spectral factorization (SF): ARMA process example
If a stationary process v(t) has a rational spectrum φv(ω), wecan represent it as v(t) = H(q)e(t), where
H(q) =C(q)
A(q)=
1 + c1q−1 + · · ·+ cncq−nc
1 + a1q−1 + · · ·+ anaq−na.
We may write the ARMA model:
v(t) + a1v(t − 1) + · · ·+ anav(t − na)
= e(t) + c1e(t − 1) + · · ·+ cnce(t − nc)
• if nc = 0, autoregressive (AR) model:
v(t) + a1v(t − 1) + · · ·+ anav(t − na) = e(t),
• if na = 0, moving average (MA) model:
v(t) = e(t) + c1e(t − 1) + · · ·+ cnce(t − nc).
⇒ SF provides a representation of disturbances in the standard
form v = H(q)e from information about its spectrum only.
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Filtering and spectrum
• Consider the general set-up with u(k) and e(k)uncorrelated:
φy(ω) = |G(eiω)|2φu(ω) + φe(ω)
φyu(ω) = G(eiω)φu(ω)
• Note:
• power spectrum additive if signals are uncorrelated
• cross correlation can be used to get rid of disturbances
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Choice of Sampling Interval and
Presampling Filters
Sampling is inherent to computer-based data-acquisition
systems→ select (equidistant) sampling instances to minimize
information losses.
Aliasing
Suppose s(t) with sampling interval T: sk = s(kT),k = 1, 2, . . ., sampling frequency ωs = 2π/T and Nyquist
(folding) frequency ωN = ωs/2.
If sinusoid with |ω| > ωN, ∃ ω, −ωN ≤ ω ≤ ωN, such that
cosωkT = cos ωkT
sinωkT = sin ωkT
k = 0, 1, . . .
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Aliasing (2)
⇒ Alias phenomenon: part of the signal with ω > ωN
interpreted as lower frequency↔ spectrum of sampled signal
is a superposition (folding) of original spectrum:
φ(T)s (ω) =
∞∑
r=−∞
φcs(ω+ rωs)
where φcs and φ
(T)s correspond to continuous-time and sampled
signals.
To avoid aliasing: choose ωs so that φcs(ω) is zero outside
(−ωs/2, ωs/2). This implies φ(T)s (ω) = φc
s(ω).
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Antialiasing presampling filters
• We loose signals above ωN , do not let folding effect distort
the interesting part of spectrum below ωN → presampling
filters κ(p): sF(t) = κ(p)s(t) ⇒ φcsF(ω) = |κ(iω)|2φc
s(ω)
• Ideally, κ(iω) s.t.
|κ(iω)| = 1, |ω| ≤ ωN
|κ(iω)| = 0, |ω| > ωN
which means that sFk= sF(kT) would have spectrum
φ(T)sF
(ω) = φcs(ω), −ωN ≤ ω < ωN
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
Antialiasing presampling filters (2)
⇒ Sampled spectrum without alias thanks to antialiasing
filter, which should always be applied before sampling if
significant energy above ωN .
• Example - Continuous-time signal: square wave plus
high-frequency sinusoidal
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Noise-reduction effect of antialiasing (AA) filters
• Typically, signal = useful part m(t) + disturbances v(t)(more broadband, e.g. noise): choose ωs such that most
of the useful spectrum below ωN. AA filters cuts away HF.
• Suppose s(t) = m(t) + v(t) and sampled, prefiltered
signal sFk= mF
k+ vF
k, sF
k= sF(kT). Noise variance:
E(vFk )
2 =
∫ ωN
−ωN
φ(T)vF
(ω)dω =∞∑
r=−∞
∫ ωN
−ωN
φcvF(ω+ rωs)dω
→ noise effects from HF folded into region [−ωN , ωN] andintroduce noise power. Eliminating HF noise by AA filter,variance of vF
kis thus reduced by
∑
r,0
∫ ωN
−ωN
φcv(ω+ rωs)dω =
∫
|ω|>ωN
φcv(ω)dω
compared to no presampling filter.
⇒ ց noise if spectrum with energy above ωN.
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Conclusions
• First step to modeling and identification = data acquisition
• Implies computer-based processing and sampled signal
• Models including both deterministic and stochastic
components
• Characterize the spectrum for analysis and processing
• Prepare experimental signal prior to the identification
phase
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
HomeworkSpectrum of a sinusoid function:
u(t) = A cos(ω0t)
1 Show that u(t) is a quasi-stationary signal by computing
the bound Ru(τ).
2 Show that the power spectrum φu(ω) is composed of two
Dirak δ functions.
Hint - you may wish to use the identities:
cos θ+ cos φ = 2 cos
(
θ+ φ
2
)
cos
(
θ − φ
2
)
cos(ω0τ) =1
2
(
eiω0τ + e−iω0τ)
δ(x) =1
2π
∞∑
n=−∞
einx
Signals for
System
Identification
E. Witrant
Basics of
System
Identification
Sampled
Signals
Discrete-time
Sampled systems
Sampling
Disturbance
Modeling
Discrete-time
stochastic processes
Some background
Signal Spectra
Quasi-Stationary
Signals
Definition of Spectra
Transformation by
linear systems
Spectral factorization
Filtering and spectrum
Sampling
Interval and
Filters
Aliasing
Antialiasing filters
Noise-reductioneffect
Conclusions
Homework
References
1 L. Ljung, System Identification: Theory for the User, 2nd
Edition, Information and System Sciences, (Upper Saddle
River, NJ: PTR Prentice Hall), 1999.
2 L. Ljung and T. Glad, Modeling of Dynamic Systems,
Prentice Hall Information and System Sciences Series,
1994.
3 Finn Haugen, Discrete-time signals and systems,
TechTeach, 2005. http://techteach.no/publications/
discretetime_signals_systems/discrete.pdf
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Modeling and Estimation for ControlSystem Identification
Lecture 8: Non-parametric Identification
Emmanuel WITRANT
April 23, 2014
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Class goal
Linear time-invariant model
• described by transfer functions or impulse responses
• determine such functions directly, without restricting the
set of models
• non-parametric: do not explicitly employ finite-dimensional
parameter vector in the search
• focus on determining G(q) from input to output
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Outline
1 Transient-response and correlation analysis
2 Frequency-response analysis
3 Fourier analysis
4 Spectral analysis
5 Estimating the disturbance spectrum
6 Conclusions
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Time-domain methods
Impulse-response analysis
Consider the system:
(input u(t), output y(t) and
additive disturbance v(t))
y(t) =∞∑
k=1
g(k)u(t − k) + v(t)
= G0(q)u(t) + v(t)
• pulse u(t) =
α, t = 0
0, t , 0→ y(t) = αg0(t) + v(t),
by def of G0 and impulse response g0(t)• if v(t) small, then the estimate is g(t) = y(t)/α and error
ǫ(t) = v(t)/α from experiment with pulse input.
• Drawbacks:
• most physical processes do not allow pulse inputs s.t. ǫ(t)negligible
• nonlinear effects may be emphasized
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Step-response analysis
Similarly,
u(t) =
α, t ≥ 0
0, t < 0
• y(t) = αt∑
k=1
g0(k) + v(t)
• g(t) =y(t) − y(t − 1)
αand ǫ(t) =
v(t) − v(t − 1)
α• results in
⊲ large errors in most practical application
⊲ sufficient accuracy for control variables, i.e. time delay,
static gain, dominant time-constants
⊲ simple regulators tuning (Ziegler-Nichols rule, 1942)
⊲ graphical parameter determination (Rake, 1980)
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Correlation analysis
Consider again:
y(t) =∞∑
k=1
g0(k)u(t − k) + v(t)
• If u is QSS with Eu(t)u(t − τ) = Ru(τ) and
Eu(t)v(t − τ) = 0 (OL) then
Ey(t)u(t − τ) = Ryu(τ) =∞∑
k=1
g0(k)Ru(k − τ)
• If u is a white noise s.t. Ru(τ) = αδτ0 then
g0(τ) = Ryu(τ)/α
⊲ An estimate of the impulse response is obtained from an
estimate of Ryu(τ)
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Example: N measurements
RNyu(τ) =
1
N
N∑
t=τ
y(t)u(t − τ)
if u , white noise,
• estimate RNu (τ) =
1
N
N∑
t=τ
u(t)u(t − τ)
• solve RNyu(τ) =
1
N
N∑
k=1
g(k)RNu (k − τ) for g(k)
• if possible, set u such that RNu and RN
yu are easy to solve
(typically done by commercial solvers).
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Frequency-response analysis
Sine-wave testing
• physically, G(z) is such that G(eiω) describes what
happened to a sinusoid
• if u(t) = α cosωt , t = 0, 1, 2, . . . then
y(t) = α|G0(eiω)| cos(ωt + φ) + v(t) + transient
where φ = arg G0(eiω)
⊲ G0(eiω) determined as:
• from u(t), get the amplitude and phase shift of y(t)• deduce the estimate GN(e
iω)• repeat for frequencies within the interesting band
• known as frequency analysis
• drawback: |G0(eiω)| and φ difficult to determine accurately
when v(t) is important
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Frequency analysis by the correlation method
• since y(t) of known freq., correlate it out from noise
• define sums
IC(N) 1
N
N∑
t=1
y(t) cosωt and IS(N) 1
N
N∑
t=1
y(t) sinωt
• based on previous y(t) (ignore transients and cos(a + b))
IC(N) =α
2|G0(e
iω)| cos(φ) + α|G0(eiω)|1
2
1
N
N∑
t=1
cos(2ωt + φ)
︸ ︷︷ ︸
→0 as N→∞
+1
N
N∑
t=1
v(t) cos(ωt)
︸ ︷︷ ︸
→0 as N→∞ if v(t) DN contain ω
• if v(t) is a stat. stoch. process s.t.∑∞
0 τ|Rv(τ)| < ∞ then
the 3rd term variance decays like 1/N
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
• similarly,
IS(N) = −α2|G0(e
iω)| sin(φ) + α|G0(eiω)|1
2
1
N
N∑
t=1
sin(2ωt + φ)
+1
N
N∑
t=1
v(t) sin(ωt) ≈ −α2|G0(e
iω)| sin(φ)
• and we get the estimates
|GN(eiω)| =
√
I2C(N) + I2
S(N)
α/2, φN = arg GN(e
iω) = − arctanIS(N)
IC(N)
• repeat over the freq. of interest (commercial soft.)
(+) Bode plot easily obtained and focus on spec. freq.
(-) many industrial processes DN admit sin inputs & long
experimentation
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Relationship to Fourier analysis
Consider the discrete Fourier transform
YN(ω) =1√N
∑Nt=1 y(t)e−iωt and IC & IS , which gives
IC(N) − i IS(N) =1√
NYN(ω)
• from the periodogram (signal power at frequency ω) of
u(t) = α cosωt , UN(ω) =√
Nα/2 if ω = 2πr/N for some
r ∈ N• then GN(e
iω) =√
NYN(ω)Nα/2
=YN(ω)UN(ω)
• ω is precisely the input frequency
• provides a most reasonable estimate.
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Commercial software example
In practice, you may use Matlab Identification toolboxr to
• import the data in a GUI
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
• pre-process it (remove mean, pre-filter, separate
estimation from validation, etc.)
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
• analyse the signals
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
• get multiple models of desired order and compare the
outputs
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Fourier analysis
Empirical transfer-function estimate
Extend previous estimate to multifrequency inputs
ˆGN(e
iω) =YN(ω)
UN(ω)with (Y/U)N(ω) =
1√
N
N∑
t=1
(y/u)(t)e−iωt
ˆGN = ETFE, since no other assumption than linearity
• original data of 2N numbers y(t), u(t), t = 1 . . .Ncondensed into N numbers (essential points/2)
ReˆGN(e
2πik/N), ImˆGN(e
2πik/N), k = 0, 1, . . . ,N
2− 1
→ modest model reduction• approx. solves the convolution (using Fourier techniques)
y(t) =N∑
k=1
g0(k )u(t − k ), t = 1, 2, . . . ,N
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Properties of the ETFE
If the input is periodic:
• the ETFE is defined only for a fixed number of frequencies
• at these frequencies the ETFE is unbiased and its
variance decays like 1/N
If the input is a realization of a stochastic process:
• the periodogram |UN(ω)|2 is an erratic function of ω, which
fluctuates around φu(ω)
• the ETFE is an asymptotically unbiased estimate of the TF
at increasingly (with N) many frequencies
• the ETFE variance do notց as N ր and is given as the
noise to signal ratio at the considered freq.
• the estimates at different frequencies are uncorrelated
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Conclusions on ETFE
• increasingly good quality for periodic signals but no
improvement otherwise as N ր• very crude estimate in most practical cases
• due to uncorrelated information per estimated parameter
⇒ relate the system behavior at one frequency to another
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Spectral analysis
Smoothing the ETFE
Assumption: the true transfer function G0(eiω) is a smooth
function of ω. Consequences:
• G0(eiω) supposed constant over
2πk1
N= ω0 −∆ω < ω < ω0 +∆ω =
2πk2
N
• the best way (min. var.) to estimate this cst is a weightedaverage of G0(e
iω) on the previous freq., eachmeasurement weighted by its inverse variance:
• for large N, we can use Riemann sums and introduce the
weights Wγ(ζ) =
1, |ζ | < ∆ω0, |ζ | > ∆ω
• after some cooking and simplifications,
GN(eiω0) =
∫ π
−πWγ(ζ − ω0)|UN(ζ)|2 ˆGN(eiζ)dζ
∫ π
−πWγ(ζ − ω0)|UN(ζ)|2dζ
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Connection with the Blackman-Turkey procedure
Noticing that as N → ∞∫ π
−πWγ(ζ − ω0)|UN(ζ)|2dζ →
∫ π
−πWγ(ζ − ω0)φu(ζ)dζ
supposing∫ π
−πWγ(ζ)dζ = 1 then
φNu (ω0) =
∫ π
−πWγ(ζ − ω0)|UN(ζ)|2dζ
φNyu(ω0) =
∫ π
−πWγ(ζ − ω0)YN(ζ)UN(ζ)dζ
GN(eiω0) =
φNyu(ω0)
φNu (ω0)
→ ratio of cross spectrum by input spectrum (smoothed
periodograms proposed by B&T )
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Weighting function Wγ(ζ): the frequency window
• “Wide” fw→ weight many different frequencies, small
variance of GN(eiω0) but far from ω0 = bias
• γ (∼ width−1) = trade-off between bias and variance
• Width and amplitude:M(γ)
∫ π
−π ζ2Wγ(ζ)dζ and W(γ) 2π
∫ π
−πW2γ (ζ)dζ
• Typical windows for spectral analysis:
2πWγ(ω) M(γ) W(γ)
Bartlett1
γ
(
sin γω/2
sinω/2
)22.78
γ0.67γ
Parzen4(2 + cosω)
γ3
(
sin γω/4
sinω/2
)412
γ20.54γ
Hamming1
2Dγ(ω) +
1
4Dγ(ω − π/γ) +
1
4Dγ(ω+ π/γ),
π2
2γ20.75γ
where Dγ(ω) sin(γ+ 1/2)ω
sinω/2
• good approx. for γ ≥ 5, as γր M(γ)ց and W(γ)ր
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
• Example: γ = 5 vs. γ = 10
-3 -2 -1 0 1 2 3
0
0.2
0.4
0.6
0.8
1
1.2
1.4
1.6
ω (rad/s)
Wγ(ω
)
Windows for γ = 5
BartlettParzenHamming
-3 -2 -1 0 1 2 3
0
0.2
0.4
0.6
0.8
1
1.2
1.4
1.6
ω (rad/s)
Wγ(ω
)
Windows for γ = 10
BartlettParzenHamming
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Asymptotic properties of the smoothed estimate
• The estimates ReGN(eiω) and ImGN(e
iω) are
asymptotically uncorrelated and of known variance
• GN(eiω) at , freq. are asymptotically uncorrelated
• γ that min. the mean square estimate (MSE) is
γopt =
(
4M2|R(ω)|2φu(ω)
Wφv(ω)
)1/5
· N1/5
→ frequency window more narrow when more data
available, and leads to MSE ∼ C · N−4/5
• typically, start with γ = N/20 and compute GN(eiω) for
various values of γ,ր γց biasր variance (more details)
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Example
• Consider the system
y(t) − 1.5y(t − 1) + 0.7y(t − 2) = u(t − 1) + 0.5u(t − 2) + e(t)
where e(t) is a white noise with variance 1 and u(t) a
pseudo-random binary signal (PRBS), over 1000 samples.
% Construct the polynomial
m0=poly2th([1 -1.5 0.7],[0 1 0.5]);
% Generate pseudorandom, binary signal
u=idinput(1000,’prbs’);
% Normally distributed random numbers
e=randn(1000,1);
% Simulate and plot the output
y=idsim([u e],m0);
z=[y u]; idplot(z,[101:200])
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
• we get the inputs and ouputs
0 20 40 60 80 100-20
-10
0
10
Time
y1
0 20 40 60 80 100-1
-0.5
0
0.5
1
u1
Time
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
• The ETFE and smoothing thanks to Hamming window
(γ = 10, 50, 200) are obtained as
% Compute the ETFE
ghh=etfe(z);[om,ghha]=getff(ghh);
% Performs spectral analysis
g10=spa(z,10);[om,g10a]=getff(g10);
g50=spa(z,50);[om,g50a]=getff(g50);
g200=spa(z,200);[om,g200a]=getff(g200);
g0=th2ff(m0);[om,g0a]=getff(g0);
bodeplot(g0,ghh,g10,g50,g200,’a’);
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
• we get the ETFE and estimates
10-1
100
10-1
100
101
Frequency (rad/s)
Am
plitu
de
true systemETFE
γ=10
γ=50
γ=200
⇒ γ = 50 seems a good choice
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Estimating the disturbance
spectrum
y(t) = G0(q)u(t) + v(t)
Estimating spectra
• Ideally, φv(ω) given as (if v(t) measurable):
φNv (ω) =
∫ π
−πWγ(ζ − ω)|VN(ζ)|2dζ
• Bias: E φNv (ω) − φv(ω) =
12M(γ)φ′′v (ω) + O(C1(γ))
︸ ︷︷ ︸
γ→∞
+O(√
1/N)︸ ︷︷ ︸
N→∞
• Variance : Var φNv (ω) =
W(γ)
Nφ2
v(ω) + O(1/N)︸ ︷︷ ︸
N→∞
• Estimates at , freq. are uncorrelated
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
The residual spectrum
• v(t) not measurable→ given the estimate GN
v(t) = y(t) − GN(q)u(t)
gives
φNv (ω) =
∫ π
−πWγ(ζ − ω)|YN(ζ) − GN(e
iζ)UN(ζ)|2dζ
• After simplifications: φNv (ω) = φ
Ny (ω) −
|φNyu(ω)|2
φNu (ω)
• Asymptotically uncorrelated with GN
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Coherency spectrum
• Defined as
κNyu(ω)
√√
|φNyu(ω)|2
φNy (ω)φ
Nu (ω)
→ φNv (ω) = φ
Ny (ω)[1 − (κNyu(ω))
2]
• κyu(ω) is the coherency spectrum, i.e. freq. dependent
corr. btw I/O
• if 1 at a given ω, perfect corr. ↔ no noise.
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Conclusions
Nonparametric identification
• direct estimate of transient or frequency response
• valuable initially to provide the model structure (relations
between variables, static relations, dominant
time-constants . . . )
• spectral analysis for frequency fonctions, Fourier = special
case (wide lag window)
• essential user influence = γ: trade-off between frequency
resolution vs. variability
• reasonable γ gives dominant frequency properties
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
Homework
download the User’s guide at
http://www.mathworks.com/access/helpdesk/help/pdf_doc/ident/ident.pdf
System Identification ToolboxTM 7
Suppose that you have some data set with inputs u ∈ R1×Nt
and outputs y ∈ RNy×Nt for which you wish to identify a
deterministic model:
1. Which kind of models can you test with the SysID toolbox?
2. Which functionalities are available for identification and
model analysis?
Nonparametric
identification
E. Witrant
Time-domain
methods
Impulse-response
Step-response
Correlation
Frequency-
response
Sine-wave testing
Correlation method
Relationship to Fourier
Fourier
ETFE definition
ETFE properties
Spectral
Smoothing the ETFE
Blackman-Turkey
procedure
Frequency window
Asymptotic properties
Disturbance
spectrum
Residual spectrum
Coherency spectrum
Conclusions
Homework
References
• L. Ljung, System Identification: Theory for the User, 2nd
Edition, Information and System Sciences, (Upper Saddle
River, NJ: PTR Prentice Hall), 1999.
• L. Ljung and T. Glad, Modeling of Dynamic Systems,
Prentice Hall Information and System Sciences Series,
1994.
• O. Hinton, Digital Signal Processing, EEE305 class
material, Chapter 6 - Describing Random Sequences,
http://www.staff.ncl.ac.uk/oliver.hinton/eee305/Chapter6.pdf .
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
System Identification
Lecture 9: Parameter Estimation in Linear Models
Emmanuel WITRANT
April 24, 2014
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Class goal
Today, you should be able to
• distinguish between common model structures used in
identification
• estimate model parameters using the prediction-error
method
• calculate the optimal parameters for ARX models using
least-squares
• estimate bias and variance of estimates from model and
input signal properties
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
System identification
Many issues:
Les. 7 choice of sampling frequency, input signal (experiment
conditions), pre-filtering;
Les. 8 non parametric models, from finite and noisy data, how to
model disturbances?
Today what class of models? estimating model parameters from
processed data.
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
System identification via parameter estimation
Need to fix model structure before trying to estimate
parameters
• system vs. disturbance model
• model order (degrees of transfer function polynomials)
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Outline
1 Linear models
2 Basic principle of parameter estimation
3 Minimizing prediction errors
4 Linear regressions and least squares
5 Properties of prediction error minimization estimates
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Linear models
Model structuresMany model structures commonly used (BJ includes all others
as special cases)
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Transfer function parameterizations
The transfer functions G(q) and H(q) in the linear model
y[k ] = G(q; θ)u[k ] + H(q; θ)e[k ]
will be parameterized as (i.e. BJ)
G(q; θ) q−nkb0 + b1q−1 + · · ·+ bnb
q−nb
1 + f1q−1 + · · ·+ fnfq−nf
H(q; θ) 1 + c1q−1 + · · ·+ cnc
q−nc
1 + d1q−1 + · · ·+ dndq−nd
where the parameter vector θ contains the coefficients bk ,
fk , ck , dk .
Note: nk determines dead-time, nb , nf , nc , nd order of transfer
function polynomials.
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Model order selection from physical insight
Physical insights often help to determine the right model order:
y[k ] = q−nkb0 + b1q−1 + · · ·+ bnb
q−nb
1 + f1q−1 + · · ·+ fnfq−nf
u[k ] + H(q; θ)e[k ]
If system sampled with first-order hold (input pw. cst, 1 − q−1),
• nf equals the number of poles of continuous-time system
• if system has no delay and no direct term, then nb = nf ,
nk = 1
• if system has no delay but direct term, then nb = nf + 1,
nk = 0
• if continuous system has time delay τ, then nk = [τ/h] + 1
Note: nb does not depend on number of continuous-time zeros!
i.e. compare Euler vs. Tustin discretization
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Basic principle of parameter
estimation
• For given parameters θ, the model predicts that the
system output should be y[t ; θ]
• Determine θ so that y[t ; θ] matches observed output y[t ]“as closely as possible”
• To solve the parameter estimation problem, note that:
1 y[t ; θ] depends on the disturbance model
2 “as closely as possible” needs a mathematical formulation
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
One step-ahead prediction
Consider LTI y(t) = G(q)u(t) + H(q)e(t) and undisturbed
output y∗ = G∗u∗. Suppose that H(q) is monic (h(0) = 1, i.e.
1 + cq−1 for moving average), the disturbance is
v(t) = H(q)e(t) =∞∑
k=0
h(k)e(t − k) = e(t) +∞∑
k=1
h(k)e(t − k)
︸ ︷︷ ︸
m(t−1), known at t − 1
Since e(t) white noise (0 mean), conditional expectation (mean
value of distribution)
v(t |t − 1) = m(t − 1) = (H(q) − 1)e(t) = (1 − H−1(q))v(t)
⇒ y(t |t − 1) = G(q)u(t) + v(t |t − 1)
= G(q)u(t) + (1 − H−1(q))(y(t) − G(q)u(t))
=[
1 − H−1(q)]
y(t) + H−1(q)G(q)u(t)
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Parameter estimation methodsConsider the particular model structureM parameterized using
θ ∈ DM ⊂ Rd: M∗ = M(θ)|θ ∈ DM
• each model can predict future inputs:
M(θ) : y(t |θ) = Wy(q, θ)y(t) + Wu(q, θ)u(t)
i.e. one step-ahead prediction of
y(t) = G(q, θ)u(t) + H(q, θ)e(t) :
Wy(q, θ) = [1 − H−1(q, θ)], Wu(q, θ) = H−1(q, θ)G(q, θ)(multiply by H−1 to make e white noise),
• or nonlinear filterM(θ) : y(t |θ) = g(t ,Z t−1; θ) where
ZN [y(1), u(1), . . . , y(N), u(N)] contains the past
information.
⇒ Determine the map ZN → θN ∈ DM = parameter
estimation method
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Evaluating the candidate models
Given a specific modelM(θ∗), we want to evaluate the
prediction error
ǫ(t , θ∗) = y(t) − y(t |θ∗)
computed for t = 1, 2, . . . ,N when ZN is known.
• “Good model” = small ǫ when applied to observed data,
• “good” prediction performance multiply defined, guiding
principle:
Based on Z t we can compute the prediction error ǫ(t , θ).At time t = N, select θN s.t. ǫ(t , θN), t = 1, 2, . . . ,N,
become as small as possible.
• How to qualify “small”:
1 scalar-valued norm or criterion fct measuring size of ǫ;
2 ǫ(t , θN) uncorrelated with given data (“projections” are 0).
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Minimizing prediction errors
1. Get y(t |θ∗) from the model to compute
ǫ(t , θ∗) = y(t) − y(t |θ∗)
2. Filter ǫ ∈ RN with stable linear L(t):ǫF(t , θ) = L(q)ǫ(t , θ), 1 ≤ t ≤ N
3. Use the norm (l(·) > 0 scalar-valued)
VN(θ,ZN) =
1
N
N∑
t=1
l(ǫF(t , θ))
4. Estimate θN by minimization
θN = θN(ZN) = arg min
θ∈DMVN(θ,Z
N)
⇒ Prediction-error estimation methods (PEM), defined
depending on l(·) and prefilter L(q).
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Choice of LExtra freedom for non-momentary properties of ǫ
• same as filtering I/O data prior to identification
• L acts on HF disturbances or slow drift terms, as
frequency weighting
• note that the filtered error is
ǫF(t , θ) = L(q)ǫ(t , θ) =[
L−1(q)H(q, θ)]−1
[y(t) − G(q, θ)]
⇒ filtering is same as changing the noise model to
HL(q, θ) = L−1(q)H(q, θ)
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Choice of l
• quadratic norm l(ǫ) is first candidate
• other choices for robustness constraints
• may be parameterized as l(ǫ, θ), independently of model
parametrization
θ =
[
θ′
α
]
: l(ǫ(t , θ), θ) = l(ǫ(t , θ′), α)
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Multivariable systems
Quadratic criterion:
l(ǫ) =1
2ǫTΛ−1ǫ
with weight Λ ≥ 0 ∈ Rp×p
• Define, instead of l, the p × p matrix
QN(θ,ZN) =
1
N
N∑
t=1
ǫ(t , θ)ǫT (t , θ)
• and the scalar-valued function
VN(θ,ZN) = h(QN(θ,Z
N))
with h(Q) = 12tr(QΛ−1).
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Linear regressions and least
squares
Linear regressions
Employ predictor architecture (linear in theta)
y(t |θ) = φT (t)θ + µ(t)
where φ is the regression vector, i.e. for ARX
y(t) + a1y(t − 1) + . . .+ anay(t − na)
= b1u(t − 1) + . . .+ bnbu(t − nb) + e(t),
⇒ φ(t) = [−y(t − 1) − y(t − 2) . . . − y(t − na)
u(t − 1) . . . u(t − nb)]T
and µ(t) a known data-dependent vector (take µ(t) = 0 in the
following).
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Least-squares criterion
The prediction error becomes ǫ(t , θ) = y(t) − φT(t)θ and the
criterion function (with L(q) = 1 and l(ǫ) = 12ǫ2)
VN(θ,ZN) =
1
N
N∑
t=1
1
2
[
y(t) − φT(t)θ]2
∴ least-squares criterion for linear regression. Can be
minimized analytically (1st order condition) with
θLSN = arg min VN(θ,Z
N) =
1
N
N∑
t=1
φ(t)φT (t)
−1
︸ ︷︷ ︸
R(N)−1∈Rd×d
1
N
N∑
t=1
φ(t)y(t)
︸ ︷︷ ︸
f(N)∈Rd
the least-squares estimate (LSE). [Exercise: proove this result]
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Example: parameter estimation in ARX models
Estimate the model parameters a and b in the ARX model
y(k) = ay(k − 1) + bu(k − 1) + e(k)
from y(k), u(k) for k = 0, . . . ,N.
⇒ find θLSN
!
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Solution
• θ = [a b]T and φ(t) = [y(t − 1) u(t − 1)]T
• The optimization problem is solved with
R(N) =1
N
N∑
t=1
[
y2(t − 1) y(t − 1)u(t − 1)y(t − 1)u(t − 1) u2(t − 1)
]
and
f(N) =1
N
N∑
t=1
[
y(t − 1)y(t)u(t − 1)y(t)
]
• Note: estimate computed using covariances of u(t), y(t)(cf. correlation analysis).
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Properties of the LSE
Consider the observed data y(t) = φT(t)θ0 + v0(t), θ0 being
the true value:
limN→∞
θLSN − θ0 = lim
N→∞R(N)−1 1
N
N∑
t=1
φ(t)v0(t) = (R∗)−1f∗
with R∗ = Eφ(t)φT (t), f∗ = Eφ(t)v0(t), v0 & φ QSS. Then
θLSN→ θ0 if
• R∗ non-singular (co-variance exists, decaying as 1/N)
• f∗ = 0, satisfied if
1 v0(t) a sequence of independent random variables with
zero mean (i.e. white noise): v0(t)∀t − 1
2 u(t)∀v0(t)& na = 0 (i.e. ARX)→ φ(t) depends on u(t)only.
[Exercise: proove this result]
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Multivariable caseWhen y(t) ∈ Rp
VN(θ,ZN) =
1
N
N∑
t=1
1
2
[
y(t) − φT (t)θ]T
Λ−1[
y(t) − φT(t)θ]
gives the estimate
θLSN = arg min VN(θ,Z
N)
=
1
N
N∑
t=1
φ(t)Λ−1φT (t)
−1
1
N
N∑
t=1
φ(t)Λ−1y(t)
Key issue: proper choice of the relative weight Λ!
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
LS for state-space
Consider the LTI
x(t + 1) = Ax(t) + Bu(t) + w(t)
y(t) = Cx(t) + Du(t) + v(t)
Set
Y(t) =
[
x(t + 1)y(t)
]
, Θ =
[
A B
C D
]
, Φ(t) =
[
x(t)u(t)
]
, E(t) =
[
w(t)v(t)
]
Then Y(t) = ΘΦ(t) + E(t) where E(t) from sampled sum of
squared residuals (provides cov. mat. for Kalman filter).
Problem: get x(t). Essentially obtained as x(t) = LYr where
Yr is a r-steps ahead predictor (cf. basic subspace algorithm).
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Parameter estimation in general model structures
More complicated when predictor is not linear in parameters. In
general, we need to minimize VN(θ) using iterative numerical
method, e.g.,
θi+1 = θi − µiMiV ′N(θi)
[Exercise: analyze the convergence of V ]
Example: Newtons method uses (pseudo-Hessian)
Mi =(
V ′′N (θi))−1
or(
V ′′N (θi) + α
)−1
while Gauss-Newton approximate Mi using first-order
derivatives.
⇒ locally optimal, but not necessarily globally optimal.
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Properties of prediction error
minimization estimates
What can we say about models estimated using prediction
error minimization?
Model errors have two components:
1 Bias errors: arise if model is unable to capture true system
2 Variance errors: influence of stochastic disturbances
Two properties of general prediction error methods:
1 Convergence: what happens with θN as N grows?
2 Accuracy: what can we say about size of θN − θ0 as N ր?
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Convergence
• If disturbances acting on system are stochastic, then so is
prediction error ǫ(t)
• Under quite general conditions (even if ǫ(t) are not
independent)
limN→∞
1
N
N∑
t=1
ǫ2(t |θ) = Eǫ2(t |θ)
and
θN → θ∗ = arg minθ
Eǫ2(t |θ) as N →∞
⇒ Even if model cannot reflect reality, estimate will minimize
prediction error variance! ↔ Robustness property.
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Example
Assume that you try to estimate the parameter b in the model
y[k ] = bu[k − 1] + e[k ]
while the true system is given by
y[k ] = u[k − 1] + u[k − 2] + w[k ]
where u, e, w are white noise sequences, independent of
each other.
[Exercise: What will the estimate (computed using the
prediction error method) converge to?]
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
SolutionThe PEM will find the parameters that minimize the variance
Eǫ2(k) = E(y[k ] − y[k ])2
= E(u[k − 1] + u[k − 2] + w[k ] − bu[k − 1] − e[k ])2
= E((1 − b)u[k − 1] + u[k − 2])2+ σ2w + σ2
e
= (1 − b)2σ2u + σ2
u + σ2w + σ2
e
minimized by b = 1→ asymptotic estimate.
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Convergence (2): frequency analysis
Consider the one-step ahead predictor and true system
y(t) = [1 − H−1∗ (q, θ)]y(t) + H−1
∗ (q, θ)G(q, θ)u(t)
y(t) = G0(q)u(t) + w(t)
⇒ ǫ(t , θ) = H−1∗ (q)[y(t) − G(q, θ)u(t)]
= H−1∗ (q)[G0(q) − G(q, θ)]u(t) + H−1
∗ w(t)
Looking at the spectrum and with Parseval’s identity
θ∗ = limN→∞
θN = arg minθ
∫ π
−π
|G0(eiω) −G(e iω, θ)|2
︸ ︷︷ ︸
made as small as possible
φu(ω)
|H∗(e iω)|2︸ ︷︷ ︸
weighting function
dω
• good fit where φu(ω) contains much energy, or H∗(eiω)
contains little energy
• can focus model accuracy to “important” frequency range
by proper choice of u
• θ∗ can be computed using the ETFE as G0
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Example
Output error method using low- and high-frequency inputs
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Estimation error varianceSupposing that ∃ θ0 s.t.
y(t) − y(t |θ0) = ǫ(t |θ0) = e(t) = white noise with var λ
the estimation error variance is
E(θN − θ0)(θN − θ0)T ≈ 1
NλR−1, where R = Eψ(t |θ0)ψ(t |θ0)
T
and ψ(t |θ) ddθ
y(t |θ) (prediction gradient wrt θ). Then:
• the error varianceր with noise intensity andց with N
• the prediction quality is proportional to the sensitivity of y
with respect to θ (componentwise)
• considering that ψ computed by min. algo., use
R ≈1
N
N∑
t=1
ψ(t |θN)ψ(t |θN)T , λ ≈
1
N
N∑
t=1
ǫ2(t |θN)
• θN converges to a normal distribution with mean θ0 and
variance 1NλR−1
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Error variance (2):frequency domain characterization
The variance of the frequency response of the estimate
Var
G(eiω; θ) ≈n
N
Φw(ω)
Φu(ω)
• increases with number of model parameters n
• decreases with N & signal-to-noise ratio
• input frequency content influences model accuracy
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Identifiability
• Determines if the chosen parameters can be determined
from the data, uniquely.
• A specific parametrization is identifiable at θ∗ if
y(t |θ∗) ≡ y(t |θ) implies θ = θ∗
• May not hold if
• two , θ give identical I/O model properties
• we get , models for , θ but the predictions are the same
due to input deficiencies
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
Conclusions
• Model structure from physical insights
• Seek (next step) model prediction using measurement
history
• Minimize prediction error with proper weights (filters)
• i.e. least squares: regressor & disturbance architecture⇒
optimization using signal covariances
• Evaluate convergence & variance as performance criteria,
check identifiability
Parameter
estimation in
linear models
E. Witrant
Linear models
Model structures
TF parameterizations
From physical insights
Parameter
estimation
Prediction
Estimation methods
Evaluating models
Minimizing
pred error
Choice of L
Choice of l
Multivariable systems
Linear reg. &
LS
LS criterion
Properties of the LSE
Multivariable LS
LS for state-space
General models
PEM
properties
Convergence
Variance
Identifiability
Conclusions
References
• L. Ljung, System Identification: Theory for the User, 2nd
Edition, Information and System Sciences, (Upper Saddle
River, NJ: PTR Prentice Hall), 1999.
• L. Ljung and T. Glad, Modeling of Dynamic Systems,
Prentice Hall Information and System Sciences Series,
1994.
• Lecture notes from 2E1282 Modeling of Dynamical
Systems, Automatic Control, School of Electrical
Engineering, KTH, Sweden.
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
System Identification
Lecture 10: Experiment Design and Model
Validation
Emmanuel WITRANT
April 24, 2014
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Class goal
Today, you should be able to
• use system identification as a systematic model-building
tool
• do a careful experiment design/data collection to enable
good model estimation
• select the appropriate model structure and model order
• validate that the estimated model is able to reproduce the
observed data
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
System identification: an iterative
procedure
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Outline
1 Experiments and data collection
2 Informative experiments
3 Input design for open-loop experiments
4 Identification in closed-loop
5 Choice of the model structure
6 Model validation
7 Residual analysis
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Experiments and data collection
A two-stage approach.
1 Preliminary experiments:
• step/impulse response tests to get basic understanding of
system dynamics
• linearity, stationary gains, time delays, time constants,
sampling interval
2 Data collection for model estimation:
• carefully designed experiment to enable good model fit
• operating point, input signal type, number of data points to
collect, etc.
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Preliminary experiments: step response
Useful for obtaining qualitative information about system:
• indicates dead-times, static gain, time constants and
resonances
• aids sampling time selection (rule-of-thumb: 4-10 samples
per rise time)
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Tests for verifying linearity
For linear systems, response is independent of operating point,
• test linearity by a sequence of step response tests for
different operating points
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Tests for detecting friction
Friction can be detected by using small step increases in input
Input moves every two or three steps.
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Designing experiment for model estimation
Input signal should excite all relevant frequencies
• estimated model accurate in frequency ranges where
input has much energy
• good choice is often a binary sequence with random hold
times (e.g., PRBS)
Trade-off in selection of signal amplitude
• large amplitude gives high signal-to-noise ratio, low
parameter variance
• most systems are nonlinear for large input amplitudes
Many pitfalls if estimating a model of a system under
closed-loop control!
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Informative experiments
• The data set Z∞ is “informative enough” with respect to
model setM∗ if it allows for discremination between 2,
models in the set.
• Transferred to “informative enough” experiment if it
generates appropriate data set.
• Applicable to all models likely to be used.
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Open-loop experiments
Consider the set of SISO linear models
M∗ = G(q, θ),H(q, θ)|θ ∈ DM
with the true model
y(t) = G0(q)u(t) + H0(q)e0(t)
If the data are not informative with respect toM∗ & θ1 , θ2,
then
|∆G(eiω)|2Φu(ω) ≡ 0,
where ∆G(q) G(q, θ1) − G(q, θ2):
⇒ crucial condition on the open-loop input spectrum Φu(ω)
• if it implies that ∆G(eiω) ≡ 0 for two equal models, then
the data is sufficiently informative with respect toM∗
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Persistence of excitation
Def. A QSS u(t) with spectrum Φu(ω) is said persistently
exciting of order n if, ∀Mn(q) = m1q−1 + . . .+ mnq−n
|Mn(eiω)|2Φu(ω) ≡ 0→ Mn(e
iω) ≡ 0
Lem. In terms of covariance function Ru(τ), it means that if
Rn
Ru(0) . . . Ru(n − 1)...
. . ....
Ru(n − 1) . . . Ru(0)
then u(t) persistently exciting⇔ Rn nonsingular.
Lec.PE If the underlying system is y[t ] = θTφ[t ] + v[t ] then θ that
makes the model y[t ] = θφ[t ] best fit measured u[t ] and
y[t ] are given by
θ = (φTNφN
︸︷︷︸
Rn
)−1φTNyN
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Informative open-loop experiments
Consider a setM∗ st.
G(q, θ) =q−nk (b1 + b2q−1 + . . .+ bnb
q−nb+1)
1 + f1q−1 + . . .+ fnfq−nf
then an OL experiment with an input that is persistently exciting
of order n = nb + nf is sufficiently informative with respect to
M∗.Cor. an OL experiment is informative if the input is persistently
exciting.
• the order of excitation = nb of identified parameters
• e.g. Φu(ω) , 0 at n points (n sinusoids)
Rq: immediate multivariable counterpart
⇒ The input should include many distinct frequencies: still a
large degree of freedom!
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Input design for open-loop
experiments
Three basic facts:
• asymptotic properties of the estimate (bias & variance)
depend only on input spectrum, not the waveform
• limited input amplitude: u ≤ u ≤ u
• periodic inputs may have some advantages
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
The crest factor
• cov. matrix typically inversely proportional to input power
⇒ have as much power as possible
• physical bounds u, u→ desired waveform property
defined as crest factor; for zero-mean signal:
C2r =
maxt u2(t)
limN→∞1N
∑Nt=1 u2(t)
• good waveform = small crest factor
• theoretical lower bound is 1 = binary, symmetric signals
u(t) = ±u
• specific caution: do not allow validation against
nonlinearities
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Common input signals
Achieve desired input spectrum with smallest crest factor:
typically antagonist properties.
• Filtered Gaussian white noise (WN): any spectrum with
appropriate filter, use off-line non-causal filters (e.g.
Kaiser & Reed, 1977) to eliminate the transients
(theoretically unbounded)
• Random binary signals (RBS): generate with a filtered
zero-mean Gaussian noise and take the sign. Cr = 1,
problem: filter change spectrum
• Pseudo-Random Binary Signal (PRBS): periodic,deterministic signal with white noise properties.Advantages with respect to RBS:
• cov. matrix can be analytically inverted
• secured second order properties when whole periods
• not straightforward to generate uncorrelated PRBS
• work with integer number of periods to have full PRBS
advantages→ limited by experimental length
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Common input signals (2)
• Low-pass filtering by increasing the clock period: to get
more low-frequency, filter PRBS (no B) and take P
samples over one period:
u(t) =1
P(e(t) + . . .+ e(t − P + 1))
• Multi-sines: sum of sinusoids
u(t) =∑d
k=1 ak cos(ωk t + φk )
• Chirp signals or swept sinusoids: sin. with freq. that
changes continuously over certain band Ω : ω1 ≤ ω ≤ ω2
and time period 0 ≤ t ≤ M
u(t) = A cos(
ω1t + (ω2 − ω1)t2/(2M)
)
instantaneous frequency (d/dt): ωi = ω1 +tM(ω2 − ω1).
Good control over excited freq. and same crest as sin. but
induces freq. outside Ω.
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Periodic inputs
Some guidelines:
• generate PRBS over one full period, M = 2n − 1 and
repeat it
• for multi-sine of period M, choose ωk from DFT-grid
(density functional theory) ωl = 2πl/M, l = 0, 1, . . . ,M − 1
• for chirp of period M, choose ω1,2 = 2πk1,2/M
Advantages and drawbacks:
• period M → M distinct frequencies in spectrum, persistent
excitation of (at most) order M
• when K periods of length M (N = KM), average outputs
over the periods and select one to work with (ց data to
handle, signal to noise ration improved by K )
• allows noise estimation: removing transients, differences
in output responses over , periods attributed to noise
• when model estimated in Fourier transformed data, no
leakage when forming FT
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Example: input consisting of five sinusoids
u = idinput([100 1 20],’sine’,
[],[],[5 10 1]);
% u = idinput(N,type,band,levels)
% [u,freqs] = idinput(N,’sine’,
% band,levels,sinedata)
% N = [P nu M] gives a periodic
% input with nu channels,
% each of length M*P and
% periodic with period P.
% sinedata = [No_of_Sinusoids,
% No_of_Trials, Grid_Skip]
u = iddata([],u,1,’per’,100);
u2 = u.u.ˆ2;
u2 = iddata([],u2,1,’per’,100);
0 200 400 600 800 1000 1200 1400 1600 1800 2000-1
-0.5
0
0.5
1
20 periods of u and u2
0 10 20 30 40 50 60 70 80 90 100-1
-0.5
0
0.5
1
1 period of u and u2
Sample
0 0.05 0.1 0.15 0.2 0.25 0.3 0.35 0.4 0.45 0.510
-30
10-20
10-10
100
1010
Frequency (1/s)
Power Power spectrum for u and u
2
u
u2
Spectrum of u vs. u2: frequency splitting (the square having
spectral support at other frequencies) reveals the nonlinearity
involved.
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Identification in closed-loop
Identification under output feedback necessary if unstable
plant, or controlled for safety/production, or inherent feedback
mechanisms.
Basic good news: prediction error method provides good
estimate regardless of CL if
• the data is informative
• the model sets contains the true system
Some fallacies:
• CL experiment may be non-informative even if persistent
input, associated with too simple regulators
• direct spectral analysis gives erroneous results
• corr. analysis gives biased estimate, since
Eu(t)v(t − τ) , 0
• OEM do not give consistent G when the additive noise not
white
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Example: proportional feedback
Consider the first-order model and feedback
y(t) + ay(t − 1) = bu(t − 1) + e(t), u(t) = −fy(t)
then
y(t) + (a + bf)y(t − 1) = e(t)
⇒ all models a = a + γf , b = b − γ where γ is an arbitrary
scalar give the same I/O description: even if u(t) is persistently
exciting, the experimental condition is not informative enough.
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Some guidelines
• The CL experiment is informative⇔ reference r(t) is
persistently exciting in
y(t) = G0(q)u(t) + H0(q)e(t)
u(t) = r(t) − Fy(q)y(t)
• Non linear, time-varying or complex (high-order) regulators
yield informative enough experiments in general
• A switch between regulators, e.g.
u(t) = −F1(q)y(t) and u(t) = −F2(q)y(t),
s.t. F1(eiω) , F2(e
iω); ∀ω
achieves informative experiments
• Feedback allows to inject more input in certain freq ranges
without increasing output power.
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Choice of the model structure
1 Start with non-parametric estimates (correlation analysis,spectral estimation)
• give information about model order and important
frequency regions
2 Prefilter I/O data to emphasize important frequency ranges
3 Begin with ARX models
4 Select model orders via
• cross-validation (simulate & compare with new data)
• Akaike’s Information Criterion, i.e., pick the model that
minimizes(
1 + 2d
N
) N∑
t=1
ǫ[t ; θ]2
where d = nb estimated parameters in the model
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Model validation
Parameter estimation→ “best model” in chosen structure, but
“good enough”?
• sufficient agreement with observed data
• appropriate for intended purpose
• closeness to the “true system”
Example: G(s) =1
(s + 1)(s + a)has O- & CL responses for
a = −0.01, 0, 0.01
Insufficient for OL prediction, good enough for CL control!
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Validation
• with respect to purpose: regulator design, prediction or
simulation→ test on specific problem, may be limited to
do exhaustively (cost, safety)
• feasibility of physical parameters: estimated values and
variance compared with prior knowledge. can also check
sensitivity for identifiability
• consistency of I/O behavior:
• Bode’s diagrams for , models & spectral analysis
• by simulation for NL models
• with respect to data: verify that observations behaveaccording to modeling assumptions
1 Compare model simulation/prediction with real data
2 Compare estimated models frequency response and
spectral analysis estimate
3 Perform statistical tests on prediction errors
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Example: Bode plot for CL control
Different low-frequency behavior, similar responses around
cross-over frequency
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Model reduction
• Original model unnecessarily complex if I/O properties not
much affected by model reduction
• Conserve spectrum/eigenvalues
• Numerical issues associated with matrix conditioning (e.g.
plasma in optimization class)
Parameter confidence interval
• Compare estimate with corresponding estimated standard
deviation
• If 0∈ confidence interval, the corresponding parameter
may be removed
• Usually interesting if related to a physical property (model
order or time-delay)
• If standard dev. are all large, information matrix close to
singular and typically too large order
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Simulation and prediction
• Split data into two parts; one for estimation and one for
validation.
• Apply input signal in validation data set to estimated model
• Compare simulated output with output stored in validation
data set.
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Residual analysis
• Analyze the data not reproduced by model = residual
ǫ(t) = ǫ(t , θN) = y(t) − y(t |θN)
• e.g. if we fit the parameters of the model
y(t) = G(q, θ)u(t) + H(q, θ)e(t)
to data, the residuals
ǫ(t) = H(q, θ)−1 [y(t) − G(q, θ)u(t)]
represent a disturbance that explains mismatch between
model and observed data.
• If the model is correct, the residuals should be:
⋄ white, and
⋄ uncorrelated with u
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Statistical model validation
• Pragmatic viewpoint: basic statistics from
S1 = maxt|ǫ(t)|, S2
2 =1
N
N∑
t=1
ǫ2(t)
likely to hold for future data = invariance assumption (ǫ do
not depend on something likely to change or on a
particular input in ZN)
⇒ Study covariance
RNǫu(τ) =
1
N
N∑
t=1
ǫ(t)u(t − τ), RNǫ (τ) =
1
N
N∑
t=1
ǫ(t)ǫ(t − τ)
⋄ RNǫu(τ): if small, S1,2 likely to be relevant for other inputs,
otherwise, remaining traces of y(t) not inM⋄ RN
ǫ (τ): if not small for τ , 0, part of ǫ(t) could have been
predicted⇒ y(t) could be better predicted
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Whiteness test
• Suppose that ǫ is a white noise with zero mean and
variance λ, then
N
λ2
M∑
τ=1
(
RNǫ (τ)
)2=
N(
RNǫ (τ)
)2
M∑
τ=1
(
RNǫ (τ)
)2 ζN,M
should be asymptotically χ2(M)-distributed (independency
test), e.g. if ζN,M < χ2α(M), the α level of χ2(M)
• Simplified rule: autocorrelation function√
NRNǫ (τ) lies
within a 95% confidence region around zero→ large
components indicate unmodelled dynamics
• Similarly, independency if√
NRNǫu(τ) within 95%
confidence region around zero:
⋄ large components indicate unmodelled dynamics
⋄ RNǫu(τ) nonzero for τ < 0 (non-causality) indicates the
presence of feedback
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
Conclusions
System identification: an iterative procedure in several steps
• Experiment design
⋄ preliminary experiments detect basic system behavior
⋄ carefully designed experiment enable good model
estimation (choice of sampling interval, anti-alias filters,
input signal)
• Examination and prefiltering of data
⋄ remove outliers and trends
• Model structure selection
• Model validation
⋄ cross-validation and residual tests
Experiment
design and
model
validation
E. Witrant
Experiments
and data
collection
Preliminary
experiments
Designing for
estimation
Informative
experiments
Open-loop
experiments
Persistence of
excitation
Input design
Crest factor
Common input
Periodic inputs
CL Identif
Guidelines
Model
structure
Model
validation
Residual
analysis
Statistical validation
References
• L. Ljung, System Identification: Theory for the User, 2nd
Edition, Information and System Sciences, (Upper Saddle
River, NJ: PTR Prentice Hall), 1999.
• L. Ljung and T. Glad, Modeling of Dynamic Systems,
Prentice Hall Information and System Sciences Series,
1994.
• Lecture notes from 2E1282 Modeling of Dynamical
Systems, Automatic Control, School of Electrical
Engineering, KTH, Sweden.
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
System Identification
Lecture 11: Nonlinear Black-box Identification
Emmanuel WITRANT
April 24, 2014
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Motivation
Linear systems limited when considering:
• Physical models
• large parameter variability
• complex systems
Today’s concerns:
• generic classes of models
• black box: neural networks and AI
• parameter estimation for NL models: back on nonlinear
programming
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Outline
1 Nonlinear State-space Models
2 Nonlinear Black-box Models
3 Parameters estimation with Gauss-Newton stochastic gradient
4 Temperature profile identification in tokamak plasmas
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Nonlinear State-space Models
• General model set:
x(t + 1) = f(t , x(t), u(t),w(t); θ)
y(t) = h(t , x(t), u(t), v(t); θ)
Nonlinear prediction→ no finite-dimensional solution
except specific cases: approximations
• Predictor obtained from simulation model (noise-free)
x(t + 1, θ) = f(t , x(t , θ), u(t), 0; θ)⇔d
dtx(t , θ) = f(·)
y(t |θ) = h(t , x(t , θ), u(t), 0; θ)
• Include known physical parts of the model, but unmodeled
dynamics that can still have a strong impact on the system
→ black-box components.
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Nonlinear Black-box Models:
Basic Principles
Model = mapping from past data Z t−1 to the space of output
y(t |θ) = g(Z t−1, θ)
→ seek parameterizations (parameters θ) of g that are flexible
and cover “all kinds of reasonable behavior” ≡ nonlinear
black-box model structure.
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
A structure for the general mapping: Regressors
Express g as a concatenation of two mappings:
• φ(t) = φ(Z t−1): takes past observation into regression
vector φ (components = regressors), or φ(t) = φ(Z t−1, θ);
• g(φ(t), θ): maps φ into space of outputs.
Two partial problems:
1 How to choose φ(t) from past I/O? Typically, using only
measured quantities, i.e. NFIR (Nonlinear Finite Impulse
Response) and NARX.
2 How to choose the nonlinear mapping g(φ, θ) from
regressor to output space?
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Basic features of function expansions and basis functions
• Focus on g(φ(t), θ) : Rd → Rp, φ ∈ Rd , y ∈ Rp.
• Parametrized function as family of function expansions
g(φ, θ) =n∑
k=1
αk gk (φ), θ = [α1 . . . αn]T
gk referred as basis functions, provides a unified
framework for most NL black-box model structures.
• How to choose gk ? Typically
• all gk formed from one “mother basis function” κ(x);• κ(x) depends on a scalar variable x;
• gk are dilated (scaled) and translated versions of κ, i.e. if
d = 1 (scalar case)
gk (φ) = gk (φ, βk , γk ) = κ(βk (φ − γk ))
where βk = dilatation and γk = translation.
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Scalar examples
• Fourier series: κ(x) = cos(x), g are Fourier series
expansion, with βk as frequencies and γk as phases.
• Piece-wise continuous functions: κ as unit interval
indicator function
κ(x) =
1 for 0 ≤ x < 1
0 else
and γk = k∆, βk = 1/∆, αk = f(k∆): give a PW constant
approximation ∀ f over intervals of length ∆. Similar
version with Gaussian bell κ(x) = 1√2π
e−x2/2.
• PW continuous functions - variant -: κ as unit step function
κ(x) =
0 for x < 0
1 for x > 0
Similar result with sigmoid function κ(x) = 11+e−x
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Classification of single-variable basis functions
• local basis functions, with significant variations in local
environment (i.e. presented PW continuous functions);
• global basis functions, with significant variations over the
whole real axis (i.e. Fourier, Voltera).
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Construction of multi-variable basis functions
(φ ∈ Rd , d > 1)
1 Tensor product. Product of the single-variable function,
applied to each component of φ:
gk (φ) = gk (φ, βk , γk ) =d∏
j=1
κ(βj
k(φj − γj
k))
2 Radial construction. Value depend only on φ’s distance
from a given center point
gk (φ) = gk (φ, βk , γk ) = κ(
||φ − γk ||βk
)
where || · ||βkis any chosen norm, i.e. quadratic:
||φ||2βk
= φTβkφ with βk > 0 matrix.
3 Ridge construction. Value depend only on φ’s distance
from a given hyperplane (cst ∀φ in hyperplane)
gk (φ) = gk (φ, βk , γk ) = κ(βTk (φ − γk ))
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Approximation issues
• For any of the described choices, the resulting model
becomes
g(φ, θ) =n∑
k=1
αkκ(βk (φ − γk ))
• Fully determined by κ(x) and the basis functions
expansion on a vector φ.
• Parametrization in terms of θ characterized by three
parameters: coordinates α, scale or dilatation β, location
γ. Note: linear regression for fixed scale and location.
• Accuracy [Juditsky et al., 1995]: for almost any choice of
κ(x) (except polynomial), we can approximate any
“reasonable” function g0(φ) (true system) arbitrarily well
with n large enough.
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Approximation issues (2)
Efficiency [Barron 1993]:
1 if β and γ allowed to depend on the function g0 then n
much less than if βk , γk fixed a priori;
2 for local, radial approach, necessary n to achieve a degree
of approximation d of s times differentiable function:
n ∼ 1
δ(d/s), δ ≪ 1
→ increases exponentially with the number of regressors
= curse of dimensionality.
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Networks for nonlinear black-box structuresBasis function expansions often referred to as networks.
• Multi-layer networks:
Instead of taking a linear combination of regressors, treat
as new regressors and introduce another “layer” of basis
functions forming a second expansion, e.g. two-hidden
layers network
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Networks for nonlinear black-box structures (2)
• Recurrent networks. When some regressors at t are
outputs from previous time instants φk (t) = g(φ(t − k), θ).
Estimation aspects
Asymptotic properties and basic algorithms are the same as
the other model structures!
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Parameters estimation with
Gauss-Newton stochastic
gradient algorithm
⇒ A possible solution to determine the optimal parameters of
each layer.
Problem description
Consider no system outputs y ∈ Rnm×no , with nm measurements
for each output, and a model output y ∈ Rnm×no .
Objective: determine the optimal set of model parameters θ
which minimizes the quadratic cost function
J(θ) 1
nm
nm∑
i=1
||y(i) − y(θ, i)||22
Output error variance is minimized for θ∗ = arg minθ J(θ).
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Stochastic descent algorithm
Based on the sensitivity of y(θ, i) with respect to θ
S(θ, i) ∂y
∂θ=
[
∂y
∂θ1, . . . ,
∂y
∂θnv
]T
,
the gradient of the cost function writes as
∇θJ(θ) = −2
nm
nm∑
i=1
S(θ, i)(y(i) − y(θ, i))
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Stochastic descent algorithm (2)
θ∗ obtained by moving along the steepest slope −∇θJ(θ) with a
step η, which as to ensure that
θl+1 = θl − ηl∇θJ(θl)
converges to θ∗, where l algorithm iteration index. ηl chosen
according to Gauss-Newton’s method as
ηl (ΨθJ(θl) + υI)−1,
where υ > 0 is a constant introduced to ensure strict
positiveness and ΨθJ(θl) is the pseudo-Hessian, obtained
using Gauss-Newton approximation
ΨθJ(θl) =
2
nm
nm∑
i=1
S(θl , i)S(θl , i)T
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Stochastic descent algorithm (3)
Consider dynamical systems modeled as (t ∈ [0,T ])
dxm
dt= fm(xm(t), u(t), θ), xm(t0) = xm0
y(t) = gm(xm(t), u(t), θ)
xm is the model state and fm(·) ∈ C1, then
S(θ, t) =∂gm
∂xm
∂xm
∂θ+∂gm
∂θ
where the state sensibility∂xm
∂θobtained by solving the ODE
d
dt
[
∂xm
∂θ
]
=∂fm
∂xm
∂xm
∂θ+∂fm
∂θ
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Assumptions
• ni independent system inputs u =
u1, . . . , uni
∈ Rnm×ni ,
available during the optimal parameter search process.
• The set y , u corresponds to historic data and J is the
data variance.
• The set of nm measurements is large enough and well
chosen (sufficiently rich input) to be considered as
generators of persistent excitation to ensure that the
resulting model represents the physical phenomenon
accurately within the bounds of u.
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
For black-box modelsConsider the nonlinear black-box structure
g(φ, θ) =n∑
k=1
αkκ(βk (φ − γk ))
To find the gradient ∇θJ(θ) we just need to compute
∂
∂α[ακ(β(φ − γ))] = κ(β(φ − γ))
∂
∂β[ακ(β(φ − γ))] = α
∂
∂β[κ(β(φ − γ))]φ
∂
∂γ[ακ(β(φ − γ))] = −α
∂
∂γ[κ(β(φ − γ))]
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Example: sigmoid functions family
κj =1
1 + e−βj(x−γj)
The sensibility function is set with
∂v
∂αj=
1
1 + e−βj(x−γj),∂v
∂βj=αje−βj(x−γj)(x − γj)
(1 + e−βj(x−γj))2,
∂v
∂γj
= −αje−βj(x−γj)βj
(1 + e−βj(x−γj))2.
Notes:
• any continuous function can be arbitrarily well
approximated using a superposition of sigmoid functions
[Cybenko, 1989]
• nonlinear function⇒ nonlinear optimization problem
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Temperature profile identification
in tokamak plasmas⇒ Parameter dependant identification of nonlinear distributed
systems
• Grey-box modeling,
• 3-hidden layers approach: spatial distribution, steady-state
and transient behaviour,
• Stochastic descent method with direct differentiation.
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Identification method: TS Temperature profile (L-mode)
Physical model:3
2
∂nT
∂t= ∇ (nχ∇T) + ST
• Input: normalized profile v(x , t) =Te(x ,t)Te0(t)
1. v(x , t) = α
1+e−β(x−γ), ⇒ θf = α, β, γ
2.
αlh = eϑsα0 Iϑsα1p B
ϑsα2
φ0Nϑsα3
∥
(
1 +Picrf
Ptot
)ϑsα4
βlh = −eϑsβ0 Iϑsβ1
p Bϑsβ2
φ0nϑsβ3
e Nϑsβ4
∥⇒ θs = ϑsα i, ϑsβ i , ϑsγ i
γlh = eϑsγ0 Iϑsγ1
p Bϑsγ2
φ0Nϑsγ3
∥
(
1 +Picrf
Ptot
)ϑsγ4(
1 +Picrf
Ptot
)ϑsγ5
3.
τth(t) = eϑt0 Iϑt1p B
ϑt2
φ0nϑt3e P
ϑt4
tot
dW
dt= Ptot −
1
τthW , W(0) = Ptot(0)τth(0)
Te0(t) = AW ⇒ θt = ϑt ,i
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Identification method (2)
v(x, tk )
v(θf (tk ))
θf(uss , θs)
uss(tkss)
f(w, ut , θt , θs)
ut(tkt)
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Results (19 shots - 9500 measurements)
0 50 100 150 200 250 300 350 400
10-4
10-3
0 50 100 150 200 250 300 350 40010
-2
10-1
0 50 100 150 200 250 300 350 4000
2
4
6
8
First layer estimation
Second layer estimation
Third layer estimation
nx∑
1
||v(x
i,t k)−
v(x
i,t k)||
2n
x∑
1
||v(x
i,t k)−
v(u
ss,t
k)||
2
Te
0&
Te
0(k
eV
)
Time (s)
meas.
identif.
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Results (2)
Test case:
0
0.2
0.4
0.6
0.8
1
5 10 15 20 25 30 35 40 45
0 5 10 15 20 25 30 35 40 450
2
4
6
8
0
0.05
0.1
0.15
0.2
Central temperature (keV) and power inputs (MW)
|η(x , t) − η(x , t)|
x
time (s)
Te0(t)
Te0(t)ITERL-96P(th)Plh Picrf
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Conclusions
• Development similar to linear models
• Predictor→ nonlinear function of past observations
• Unstructured black-box models much more demanding
• Clearly identify nonlinearities prior to identification:
semi-physical models give the regressor
• Define sub-models that can be analyzed independently
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
Homework 5
1 Comment and write down the equations corresponding to
the algorithm “fitTe sig.m” below.
2 How should the script be modified to use Gaussian fitting
curves?
function [P,Jf] = fitTe_sig(TE,xval,P)
% TE = input temperature profile
% xval = location of the measurement
% P = Initial Conditions on design parameters
[np,nm] = size(TE); % number of profiles/measurements
xval = xval’; y = TE’;
ni = 1000; % number of iterations
nv = 3; % number of design parameters
nu = .1*eye(nv); % conditioning parameter
J = zeros(ni,1); % cost function
for j = 1:ni % for each iteration
GJ = zeros(nv,1); % Gradient J
DP2(:,j) = P; % design parameters evolution recording
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
y_est = zeros(nm,1); % model output (estimated system output)
sigmoid = P(1)./(1+exp(-P(2).*(xval-P(3)))); % source terms
y_est = y_est + sigmoid;
dsigmoid_e = sigmoid.ˆ2./P(1).*exp(-P(2).*(xval-P(3)));
S = [sigmoid./P(1) (xval-P(3)).*dsigmoid_e -P(2).*dsigmoid_e];
% Sensitivity function dy/dK
PJ = S’*S; % pseudo-hessian \psi
for i=1:np % for each profile
error = y(:,i)-y_est;
% difference between the reference (0 in our case) and model
for k = 1:nv % for each design parameter
GJt(k) = error’*S(:,k);
end
GJ = GJ + GJt’;
J(j) = J(j) + (error’*error)ˆ2/np;
end
GJ = -2/np.*GJ;
PJ = 2.*PJ;
alpha = inv(PJ + nu);
P = P - alpha*GJ; % this is the veriation law for K
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
if J(j) < 1*1e-4
% if the cost function is sufficiently small we are happy and
% get out!
TE_est = y_est’;
break
end
end
Jf = J(j);
Nonlinear
Black-box
Identification
E. Witrant
Nonlinear
State-space
Models
Nonlinear
Black-box
Models
Regressors
Function expansions
and basis functions
Multi-variable basis
functions
Approximation issues
Networks
Parameters
estimation with
Gauss-Newton
Stochastic descent
Assumptions
For black-box models
Identification in
tokamaks
Identification method
Results
Conclusions
References
1 L. Ljung, “System Identification: Theory for the User”, 2nd
Edition, Information and System Sciences, (Upper Saddle
River, NJ: PTR Prentice Hall), 1999.
2 E. Witrant and S. Bremond, ”Shape Identification for
Distributed Parameter Systems and Temperature Profiles
in Tokamaks”, Proc. of 50th IEEE Conference on Decision
and Control, Orlando, USA, December 12-15, 2011.
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
System Identification
Lecture 12: Recursive Estimation Methods
Emmanuel WITRANT
April 24, 2014
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Motivation
On-line model when the system is in operation to take decision
about the system, i.e.
• Which input should be applied next?
• How to tune the filter parameters?
• What are the best predictions of next
outputs?
• Has a failure occurred? Of what type?
⇒ Adaptive methods (control, filtering, signal processing and
prediction).
Recursive estimation methods
• completed in one sampling interval to keep up with
information flow;
• carry their estimate of parameter variance;
• also competitive for off-line situations.
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Overview
• General mapping of data set to parameter space
θt = F(t ,Z t) may involve an unknown large amount of
calculation for F .
• Recursive algorithm format:
• X(t): information state
X(t) = H(t ,X(t − 1), y(t), u(t))
θt = h(X(t))
• H & h: explicit expressions involving limited calculations
⇒ θt evaluated during a sampling interval
• small information content in latest measurements pair (γt
and µt ):
θt = θt−1 + γt Qθ(X(t), y(t), u(t))
X(t) = X(t − 1) + µtQX(X(t), y(t), u(t))
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Outline
1 The Recursive Least-Squares Algorithm
2 The Recursive IV Method
3 Recursive Prediction-Error Methods
4 Recursive Pseudolinear Regressions
5 Choice of Updating Step
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
The Recursive Least-Squares
Algorithm
Weighted LS criterion
θt = arg minθ
t∑
k=1
β(t , k)[
y(k) − φT(k)θ]2
where φ is the regressor, has solution
θt = R−1(t)f(t)
R(t) =t∑
k=1
β(t , k)φ(k)φT (k)
f(t) =t∑
k=1
β(t , k)φ(k)y(k)
Uses Z t and θt−1 cannot be directly used, even if closely
related to θt .
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Recursive algorithm
• Suppose the weighting sequence properties
β(t , k ) = λ(t)β(t − 1, k ), 0 ≤ k ≤ t − 1
β(t , t) = 1
⇒ β(t , k ) =t∏
k+1
λ(j)
where λ(t) is the forgetting factor. It implies that
R(t) = λ(t)R(t − 1) + φ(t)φT (t)
f(t) = λ(t)f(t − 1) + φ(t)y(t)
⇒ θt = R−1(t)f(t)= θt−1 + R−1(t)φ(t)[
y(t) − φT (t)θt−1
]
• At (t − 1) we only need to store the information vector
X(t − 1) = [θt−1, R(t − 1)].
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Efficient matrix inversionTo avoid inverting R(t) at each step, introduce P(t) = R−1(t)and the matrix inversion lemma
[A + BCD]−1 = A−1 − A−1B[DA−1B + C−1]−1DA−1
with A = λ(t)R(t − 1), B = DT = φ(t) and C = 1 to obtain
θ(t) = θ(t − 1) + L(t)[
y(t) − φT(t)θ(t − 1)]
L(t) =P(t − 1)φ(t)
λ(t) + φT(t)P(t − 1)φ(t)
P(t) =1
λ(t)
[
P(t − 1) − L(t)φT (t)P(t − 1)]
Note: we used θ(t) instead of θt to account for slight
differences due to the IC.
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Normalized gain version
To bring out the influence of R & λ(t) on θt−1, normalize as
R(t) = γ(t)R(t), γ(t) =
t∑
k=1
β(t , k)
−1
⇒1
γ(t)=
λ(t)
γ(t − 1)+1
It implies that
θt = θt−1 + R−1(t)φ(t)[
y(t) − φT(t)θt−1
]
R(t) = λ(t)R(t − 1) + φ(t)φT (t)
becomes
ǫ(t) = y(t) − φT (t)θ(t − 1)
θ(t) = θ(t − 1) + γ(t)R−1φ(t)ǫ(t)
R(t) = R(t − 1) + γ(t)[
φ(t)φT (t) − R(t − 1)]
• R(t): weighted arithmetic mean of φ(t)φT (t);
• ǫ(t): prediction error according to current model;
• γ(t): updating step size or gain of the algorithm.
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Initial conditions
• Ideally, R(0) = 0, θ0 = θI but cannot be used (R−1)→
initialize when R(t0) invertible: spare t0 samples s.t.
P−1(t0) = R(t0) =
t0∑
k=1
β(t0, k)φ(k)φT (k)
θ0 = P(t0)
t0∑
k=1
β(t0, k)φ(k)y(k)
• Simpler alternative: use P(0) = P0 and θ(0) = θI, whichgives
θ(t) =
β(t ,0)P−1
0 +t∑
k=1
β(t , k )φ(k )φT(k )
−1 β(t ,0)P−1
0 θI +t∑
k=1
β(t , k )φ(k )y(k )
If P0 and t large, insignificant difference.
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Weighted multivariable case
θt = arg minθ
1
2
t∑
k=1
β(t , k)[
y(k) − φT(k)θ]T
Λk−1[
y(k) − φT(k)θ]
gives, similarly to the scalar case
θ(t) = θ(t − 1) + L(t)[
y(t) − φT(t)θ(t − 1)]
L(t) = P(t − 1)φ(t)[
λ(t)Λt + φT (t)P(t − 1)φ(t)]−1
P(t) =1
λ(t)
[
P(t − 1) − L(t)φT (t)P(t − 1)]
and (normalized gain)
ǫ(t) = y(t) − φT(t)θ(t − 1)
θ(t) = θ(t − 1) + γ(t)R−1φ(t)Λt−1ǫ(t)
R(t) = R(t − 1) + γ(t)[
φ(t)Λt−1φT(t) − R(t − 1)
]
Note: can also be used for the scalar case with weighted norm
β(t , k) = αk
t∏
k+1
λ(j), where the scalar αk corresponds to Λ−1k
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Kalman filter interpretation
• The Kalman filter for
x(t + 1) = F(t)x(t) + w(t)y(t) = H(t)x(t) + v(t)
is
x(t + 1) = F(t)x(t) + K(t)[y(t) − H(t)x(t)], x(0) = x0,
K(t) = [F(t)P(t)HT(t) + R12(t)][H(t)P(t)HT(t) + R2(t)]−1
P(t + 1) = F(t)P(t)FT(t) + R1(t) − K(t)[H(t)P(t)HT(t) + R2(t)]KT (t),
P(0) = Π0.
with R1(t) = Ew(t)wT(t), R12(t) = Ew(t)vT(t), R2(t) = Ev(t)vT (t)
• The linear regression model y(t |θ) = φT(t)θ can be
expressed as
θ(t + 1) = Iθ(t) + 0, (≡ θ)y(t) = φT (t)θ(t) + v(t)
Corresponding KF:
(Λt R2(t))
θ(t + 1) = θ(t) + K(t)[y(t) − φT (t)θ(t)],
K(t) = P(t)φ(t)[φT(t)P(t)φ(t) + Λt ]−1,
P(t + 1) = P(t) − K(t)[φT (t)P(t)φ(t) + Λt ]KT (t).
= exactly the multivariable case formulation if λ(t) ≡ 1!
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Resulting practical hints
• if v(t) is white and Gaussian, then the posteriori
distribution of θ(t), given Z t−1, is Gaussian with mean
value θ(t) and covariance P(t);
• IC: θ(0) is the mean and P(0) the covariance of the prior
distribution→ θ(0) is our guess before seing the data and
P(0) reflects our confidence in that guess;
• the natural choice for |Λt | is the error noise covariance
matrix. If (scalar) α−1t = Ev2(t) is time-varying, use
β(k , k) = αk in weighted criterion.
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Time-varying systems
• Adaptive methods and recursive algorithms: time-varying
system properties⇒ track these variations.
⇒ Assign less weight to older measurements: choose
λ(j) < 1, i.e. if λ(j) ≡ λ, then β(t , k) = λt−k and old
measurements are exponentially discounted: λ is the
forgetting factor. Consequently γ(t) ≡ γ = 1 − λ
• OR have the parameter vector vary like random walk
θ(t + 1) = θ(t) + w(t), Ew(t)wT (t) = R1(t)
with w white Gaussian and Ev2(t) = R2(t).
Kalman filter gives
conditional
expectation and
covariance of θ as:
θ(t) = θ(t − 1) + L(t)[
y(t) − φT (t)θ(t − 1)]
L(t) =P(t − 1)φ(t)
R2(t) + φT(t)P(t − 1)φ(t)
P(t) = P(t − 1) − L(t)φT (t)P(t − 1) + R1(t)
⇒ R1(t) prevents L(t) from tending to zero.
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Example: parametrization of the plasma resistivity profile
Consider the time and space dependant η(x , t) (shot 35109),
approximated with the scaling law
η(x , t , θ(t)) eθ1eθ2xeθ3x2
. . .eθNθxNθ−1
where x ∈ RNx and θ = θ(t) ∈ RNθ , then
• the data is processed as
y(x , t) = ln η(x, t)
• the model is
parameterized as
y(t , θ) ln η(x, t , θ(t))
= [1 x x2 . . . xNθ−1]︸ ︷︷ ︸
ΦT∈RNx ×Nθ
θ1(t)θ2(t)...
θNθ(t)
︸ ︷︷ ︸
θ(t)
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Example (2): code for multivariable case
lambda = .5; % forgetting factor
weight = exp(-x.ˆ2./.07); % gaussian distribution
Lambda = inv(diag(weight)); % gauss. weight
P = inv(lambda.*Phi*Phi’); % initial P
for j = 1:Nt % time loop
y_t = y(:,j); % acquire measurement
epsilon = y_t - Phi’*Theta; % prediction err.
Theta_r(:,j) = Theta; % store param. at t
L = P*Phi*inv(lambda.*Lambda + Phi’*P*Phi);
P = (P - L*Phi’*P)./lambda; % update P
Theta = Theta + L*epsilon; % update Theta
y_est(:,j) = Phi’*Theta; % record estimation
cost(j) = epsilon’*inv(Lambda)*epsilon;
end
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Simulation results
5 10 15 20 25
-19
-18.5
-18
θ1
Estimated parameters
5 10 15 20 252.5
3
3.5
4
θ2
5 10 15 20 25
2468
1012
θ3
5 10 15 20 25
-20
-10
0
θ4
time (s)
5 10 15 20 25
10-3
10-2
time (s)
[
y(k)− φT (k)θ] T
Λ−1
k
[
y(k)− φT (k)θ]
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Simulation results (2): effect of λ and Λt
5 10 15 20 25
-19
-18.5
-18
θ1
Estimated parameters
5 10 15 20 250
2
4
6
8
θ2
5 10 15 20 25
-10
0
10
θ3
5 10 15 20 25-20
0
20
θ4
time (s)
λ = .8 and Λt = |EεεT |I
λ = .8
λ = .99
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
The Recursive IV Method
Instrumental Variables (IV)
• Linear regression model: y(t |θ) = φT(t)θ
⇒ θLSN = sol
1
N
N∑
t=1
φ(t)[y(t) − φT (t)θ] = 0
• Actual data: y(t) = φT(t)θ0 + v0(t). LSE θN 9 θ0 typically,
because of the correlation between v0(t) and φ(t):introduce a general correlation vector ζ(t), which elements
are called the instruments or instrumental variables.• IV estimation:
θIVN = sol
1
N
N∑
t=1
ζ(t)[y(t) − φT(t)θ] = 0
=
1
N
N∑
t=1
ζ(t)φT(t)
−1
1
N
N∑
t=1
ζ(t)y(t)
Requires
Eζ(t)φT (t) nonsingular IV cor. with φ,
Eζ(t)v0(t) = 0 IV not cor. with noise
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Choice of Instruments: i.e. ARX
Supposing the true system:
y(t) + a1y(t − 1) + · · ·+ ana y(t − na) = b1u(t − 1) + · · ·+ bnbu(t − nb ) + v(t)
Choose the IV similar to the previous model, while ensuringthe correlation constraints:
ζ(t) = K(q)[−x(t − 1) . . . − x(t − na) u(t − 1) . . . u(t − nb )]T ,
where K is a filter and N(q)x(t) = M(q)u(t) (i.e. N, M from LS
estimated model and K(q) = 1 for open-loop).
⇒ ζ obtained from filtered past inputs: ζ(t) = ζ(t , ut−1)
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Recursive IV method
• Rewrite the IV method as
θIVN = R−1(t)f(t), with R(t) =
N∑
k=1
β(t , k )ζ(k )φT(k ), f(t) =N∑
k=1
β(t , k )ζ(k )y(k )
which implies that
θ(t) = θ(t − 1) + L(t)[
y(t) − φT (t)θ(t − 1)]
L(t) =P(t − 1)ζ(t)
λ(t) + φT (t)P(t − 1)ζ(t)
P(t) =1
λ(t)
[
P(t − 1) − L(t)φT (t)P(t − 1)]
• Asymptotic behavior: same as off-line counterpart except
for the initial condition issue.
• Choice of the IV (i.e. model-dependant):
ζ(t , θ) = Ku(q, θ)u(t) with Ku a linear filter and
ζ(t , θ) : x(t , θ), u(t) with A(q, θ)x(t , θ) = B(q, θ)u(t).
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Recursive Prediction-Error
Methods
Weighted prediction-error criterion
Vt(θ,Zt) = γ(t)
1
2
t∑
k=1
β(t , k)ǫ2(k , θ),
with γ, β as defined above (β(t , k) = λ(t)β(t − 1, k),β(t , t) = 1). Note that
∑tk=1 γ(t)β(t , k) = 1 and the gradient
w.r.t. θ obeys (with ǫ(k , θ) = y(k) − y(k , θ) and ψ ∂y/∂θ):
∇θVt(θ,Zt) = −γ(t)
t∑
k=1
β(t , k)ψ(k , θ)ǫ(k , θ),
= γ(t)
[
λ(t)1
γ(t − 1)∇θVt−1(θ,Z
t−1) − ψ(t , θ)ǫ(t , θ)
]
= ∇θVt−1(θ,Zt−1) + γ(t)
[
−ψ(t , θ)ǫ(t , θ) − ∇θVt−1(θ,Zt−1)]
since λ(t)γ(t)/γ(t − 1) = 1 − γ(t).
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Recursive methodGeneral search algorithm (ith iteration of min. and Z t data):
θ(i)t = θ
(i−1)t − µ
(i)t
[
R(i)t
]−1∇θVt(θ
(i−1)t ,Z t),
Suppose one more data point collected at each iteration:
θ(t) = θ(t − 1) − µ(t)t
[R(t)]−1∇θVt(θ(t − 1),Z t),
where θ(t) = θ(t)t
and R(t) = R(t)t
. Make the induction
assumption that θ(t − 1) minimized Vt−1(θ,Zt−1):
∇θVt−1(θ(t − 1),Z t−1) = 0
⇒ ∇θVt(θ(t − 1),Z t) = −γ(t)ψ(t , θ(t − 1))ǫ(t , θ(t − 1))
along with µ(t) = 1, it gives
θ(t) = θ(t − 1) + γ(t)R−1(t)ψ(t , θ(t − 1))ǫ(t , θ(t − 1))
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Recursive method (2)
• Problem: ψ(t , θ(t − 1)), ǫ(t , θ(t − 1)) derived from
y(t , θ(t − 1)) & y(t , θ) requires the knowledge of all the
data Z t−1, and consequently cannot be computed
recursively.
• Assumption: at k , replace θ by the currently available
estimate θ(k) and denote the approximation of
ψ(t , θ(t − 1)) and y(t , θ(t − 1)) by ψ(t) and y(t).
Ex.1 Finite LPV:
ξ(t + 1, θ) = A(θ)ξ(t , θ) + B(θ)
[
y(t)u(t)
]
[
y(t |θ)ψ(t , θ)
]
= C(θ)ξ(t , θ)
≈
ξ(t + 1) = A(θ(t))ξ(t) + B(θ(t))
[
y(t)u(t)
]
[
y(t)ψ(t)
]
= C(θ(t − 1))ξ(t).
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Ex.2: Gauss-Newton
∂2VN(θ,ZN)
∂θ2≈
1
N
N∑
1
ψ(t , θ)ψT (t , θ) HN(θ), &R(i)N
= HN(θ(i)N),
with the proposed approximation suggests that
R(t) = γ(t)t∑
k=1
β(t , k)ψ(k)ψT (k).
Final recursive scheme
ǫ(t) = y(t) − y(t)
θ(t) = θ(t − 1) + γ(t)R−1(t)ψ(t)ǫ(t)
R(t) = R(t − 1) + γ(t)[
ψ(t)ψT (t) − R(t − 1)]
Together with R(t) from Gauss-Newton example→ recursive
Gauss-Newton prediction-error algorithm.
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Family of recursive prediction-error methods (RPEM)
• Wide family of methods depending on the underlying
model structure & choice of R(t).
• Example: linear regression y(t |θ) = φT(t)θ gives
ψ(t , θ) = ψ(t) = φ(t), the RLS method. Gradient variant
(R(t) = I) on the same structure:
θ(t) = θ(t − 1) + γ(t)φ(t)ǫ(t)
where the gain γ(t) is a given sequence or normalized as
γ(t) = γ′(t)/|φ(t)|2 widely used in adaptive signal
processing and called LMS (least mean squares).
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Example: recursive maximum likelihoodConsider the ARMAX model
y(t) + a1y(t − 1) + · · ·+ ana y(t − na) = b1u(t − 1) + · · ·+ bnbu(t − nb)
+ e(t) + c1e(t − 1) + · · ·+ cnc e(t − nc)
and define θ [a1 . . . ana b1 . . . bnbc1 . . . cnc ]
T . Introduce the vector
φ(t , θ) = [−y(t − 1) . . . − y(t − na) u(t − 1) . . . u(t − nb ) ǫ(t − 1, θ) . . . ǫ(t − nc , θ)]T ,
⇒
y(t |θ) = φT(t , θ)θ, ǫ(t , θ) = y(t) − y(t |θ)ψ(t , θ) + c1ψ(t − 1, θ) + · · ·+ cncψ(t − nc , θ) = φ(t , θ)
The previous simplifying assumption implies that
ǫ(t) = y(t) − φT (t)θ(t)
φ(t) = [−y(t − 1) . . . − y(t − na) u(t − 1) . . . u(t − nb ) ǫ(t − 1, θ) . . . ǫ(t − nc , θ)]T
y(t) = φT(t)θ(t − 1); ǫ(t) = y(t) − y(t)
ψ(t) + c1(t − 1)ψ(t − 1) + · · ·+ cncψ(t − nc) = φ(t)
and the algorithm becomes θ(t) = θ(t − 1) + γ(t)R−1(t)ψ(t)ǫ(t)
⇒ Recursive Maximum Likelihood (RML) scheme.
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Recursive Pseudolinear
Regressions
Very similar to Recursive Prediction-Error Methods except that
the gradient is replaced by the regressor:
y(t) = φT(t)θ(t − 1)ǫ(t) = y(t) − y(t)
θ(t) = θ(t − 1) + γ(t)R−1(t)φ(t)ǫ(t)
R(t) = R(t − 1) + γ(t)[
φ(t)φT (t) − R(t − 1)]
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Choice of Updating Step
How to determine the update direction and length of the step
(γ(t)R−1(t))?
Update direction
1 Gauss-Newton: R(t) approximates the Hessian
R(t) = R(t − 1) + γ(t)[
ψ(t)ψT (t) − R(t − 1)]
2 Gradient: R(t) is a scaled identity R(t) = |ψ(t)|2 · I or
R(t) = R(t − 1) + γ(t)[
|ψ(t)|2 · I − R(t − 1)]
→ trade-off between convergence rate (Gauss-Newton, d2
operations) and algorithm complexity (gradient, d operations)
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Update step: adaptation gain
Two ways to cope with time-varying aspects:
1 select appropriate forgetting profile β(t , k) or suitable gain
γ(t), equivalent as
β(t , k) =t∏
j=k+1
λ(j) =γ(k)
γ(t)
t∏
j=k+1
(1 − γ(j)),
λ(t) =γ(t − 1)
γ(t)(1 − γ(t)) ⇔ γ(t) =
[
1 +λ(t)
γ(t − 1)
]−1
;
2 introduce covariance matrix R1(t) for parameters change
per sample: ր P(t) and consequently the gain vector
L(t).
→ trade-off between tracking ability and noise sensitivity.
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Choice of forgetting factors λ(t)
• Select the forgetting profile β(t , k) so that the criterion
keeps the relevant measurements for the current
properties.
• For ”quasi-stationary” systems, constant factor λ(t) ≡ λslightly < 1:
β(t , k) = λt−k = e(t−k ) ln λ ≈ e−(t−k )(1−λ)
⇒ measurements older than memory time constant
t − k = 11−λ
included with a weight < e−1 ≈ 36% (good if
the system remains approximately constant over t − k
samples). Typically, λ ∈ [0.98, 0.995].
• If the system undergoes abrupt and sudden changes,
choose adaptive λ(t): ց temporary if abrupt change (”cut
off” past measurements).
→ trade-off between tracking alertness and noise sensitivity.
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Conclusions
• Instruments for most adaptation schemes
• Derived from off-line methods by setting a new iteration
when a new observation is performed
• Same results off/on line for specific cases (RLS, RIV) but
data not maximally utilized
• Asymptotic properties of RPEM for constant systems are
the same as off-line: the previous analysis hold
• 2 new important quantities: update direction and gains
• Can be applied to both “deterministic” and “stochastic”
systems
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Homework 6
1 Apply RPEM to a first-order ARMA model
y(t) + ay(t − 1) = e(t) + ce(t − 1).
Derive an explicit expression for the difference
y(t) − y(t |θ(t − 1)).
Discuss when this difference will be small.
2 Consider γ(t) =[∑t
k=1 β(t , k)]−1
and β(t , k) defined by
β(t , k) = λ(t)β(t − 1, k), 0 ≤ k ≤ t − 1
β(t , t) = 1
Show that β(t , k) =γ(k)
γ(t)
t∏
j=k+1
[1 − γ(j)]
Recursive
Estimation
Methods
E. Witrant
Recursive LS
Recursive algorithm
Matrix inversion
Normalized gain
Initial conditions
Multivariable case
Kalman filter
Time-varying systems
Example
IV Method
Choice of Instruments
Recursive IV method
Recursive PEM
Recursive method
Final recursive
scheme
Family of RPEM
Example
Recursive
Pseudolinear
Regressions
Updating Step
Adaptation gain
Forgetting factors
Conclusions
Homework
Reference
1 L. Ljung, “System Identification: Theory for the User”, 2nd
Edition, Information and System Sciences, (Upper Saddle
River, NJ: PTR Prentice Hall), 1999.
Modeling and Estimation for Controland
Optimization and Optimal ControlLab topics
Emmanuel WITRANTFelipe CASTILLO-BUENAVENTURA
University Joseph Fourier,UFR PhITEM,
International Master in Systems, Control & IT
November 26, 2012
Contents
1 Vibration Isolation for Heavy Trucks. 31.1 Perform the three phases of modeling (lab 1) . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . 31.2 Design a simulator (lab 2) . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . 31.3 Bond-graph (lab 2) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . 4
2 Modeling and Observation of the LEGO Mindstorms Robot 52.1 System Description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . 52.2 Robot Programming (lab 1) . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . 62.3 Robot Modeling (lab 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . 62.4 Kalman Filter (lab 2) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . 62.5 Recursive Identification of the LEGO (lab 2) . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . . 72.6 Conclusions (lab 2) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . 7
3 Modeling of a Thermonuclear Plant 83.1 What is thermonuclear fusion? . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . 83.2 Tokamaks and profile control . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . 83.3 System dynamics . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . 9
3.3.1 Resistivity and bootstrap current . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . 103.3.2 Temperature and density profiles . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . 113.3.3 Discretization/Computation issues . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .12
3.4 Model Inputs . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . 123.4.1 Boundary condition . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . 123.4.2 Distributed current deposit from the antennas (LH, ECCD) . . . . . . . . . . . . . . . . . . . . . 123.4.3 Temperature profile modifications . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . 13
3.5 Lab Exercises . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . 143.5.1 Modeling . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . 143.5.2 Simulation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . 14
4 Simulation and Control of an Inverted Pendulum Using Scilab 154.1 Tasks . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . 154.2 Algorithms . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . 15
4.2.1 pendule.dem . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . 154.2.2 simulation.sci . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . 17
5 Identification of an Active Vibration Control Benchmark Us ing Matlab 185.1 System description and preliminary work . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . 185.2 Experiment design and frequency-response analysis . . .. . . . . . . . . . . . . . . . . . . . . . . . . . 195.3 Spectral analysis . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . 195.4 Estimating the disturbance spectrum . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . . . 195.5 Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . 19
6 Analysing the Anthropogenic Impact on the Ozone Layer Depletion 206.1 Data properties (lab 1) . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . 206.2 Transport model and time loop (lab 1) . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . 206.3 Atmospheric scenario (lab 1-2) . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . . 216.4 I/O properties (lab 2) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . 216.5 Conclusions (lab 2) . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . 21
1
7 Optimization: Particle Source Identification in a Tokamak 227.1 Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . 227.2 Simplified Transport Model . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . 23
7.2.1 Problem statement . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . 237.3 Laboratory work . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . 23
7.3.1 State-space description [1h30] . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . . 237.3.2 Optimal on-line identification . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . . . 247.3.3 Use of the experimental data . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . 24
8 Optimization: Flow Control 258.1 System Description . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . . 258.2 Identification . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . . 258.3 Control . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . 268.4 Conclusions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .. . . . . . . . . . . . . 26
2
Chapter 1
Vibration Isolation for Heavy Trucks.
A company has developed a vibration isolation systemfor the use on the back of heavy truck cabs. These cabs arepivoted with stiffmounts on the fronts and the vibration iso-lators are located at the left and right hand side of the cab atthe rear. The isolators consists of fluid filled displacers (pis-ton/cylinder devices) attached by a fluid filled tube (with anorifice) to an air compressed accumulator tank. The dis-placers are about 10 cm in diameter and the tube is about2.5 cm in diameter. The system is shown in Figure 1.1.
1.1 Perform the three phases of modeling (lab 1)
1. structuring the problem:
(a) divide the system into subsystems
(b) define the important variables
(c) for each subsystem, identify the interaction variables
(d) draw a block diagram of the complete system
2. setting up the Basic Equations:
(a) recall the main physical laws that are necessary to solvethis problem
(b) which blocks have a dynamic behavior? which ones are static?
(c) associate an equation to each block
3. forming the State-Space Model:
(a) combine the equations obtained previously to get an algebro-differential set of equation
(b) clearly define the state variables
(c) rewrite the model as a set of first order differential equations
1.2 Design a simulator (lab 2)
Based on the following simplifications:
1. since we are only interested in how the system deviates from equilibrium state and the system is linear, you candisregard the influence from the gravity on the cab and the isolation system, i.e.vin is the only input to the system.This means that the accumulator tank functions like a standard tank and you can denote its fluid capacitance byC f ;
3
Figure 1.1: Vibration isolation system.
2. the wheel dynamics can be disregarded;
3. inertance in the cylinders can be disregarded, i.e. the only inertance in the system is in the tube;
4. friction in the tube can be neglected;
5. the mass in the pistons can be neglected;
the system dynamics writes as
x1 =Am
(
x3 + Rf x2 + L f x2
)
x3 =1
C fx2, x2 = A(vin − x1)
wherex1 is the cab speed,x2 is the flow in the tube andx3 is the tank pressure.Setting arbitrarilyA = π(.05)2, Rf = 10, L f = .01,C f = 5, m = 500 andvin as a noisy sine function with a 50Hz
period:
1. illustrate the dynamics with a block representation inMatlab Simulinkr;
2. obtain the State-Space matrices and simulate the response with aM-File;
3. that provide the system response to a step or noisy input; discuss and illustrate the influence of the design parameters.
1.3 Bond-graph (lab 2)
1. Draw a bond-graph, including causality, for the isolation system model. The velocityvin from the road surface is aninput to the system.
2. Based on the bond-graph description, determine a state-space model for the system. The velocityv of the truck cabis the output.
4
Chapter 2
Modeling and Observation of the LEGOMindstorms Robot
This lab proposes the use of Matlab Simulink and LEGO Mindstorms NXT as a platform for doing theorical and practicalwork in dynamic system modeling, Kalman filter design and parametric identification, respectively. The modeling of theLEGO Mindstorms is proposed to be developed by the students as well as the design of a Kalman filter to estimate theposition of the robot and a recursive parametric identification.
2.1 System Description
In this lab, the LEGO Mindstorms NXT platform will be used. The NXT robot is based on a powerful 32-bits microcon-troller: ARM7, with 256 Kbytes FLASH and 64 Kbytes of RAM. Forprogramming and communications, NXT has anUSB 2.0 port and a wireless Bluetooth class II, V2.0 device. The new control unit has 4 inputs (one of them provides anIEC 61158 Type 4/EN 50 170 expansion for future use) and 3 analog outputs. The new LEGO version offers 4 electronicsensor types: touch, light, sound and distance. In additionof these basic sensors, nowadays a great variety of optionalsen-sors can be bought, as for example vision cameras, magnetic compass, accelerometers, gyroscopes, infrareds searchers,etc. (http://www.hitechnic.com/,http://www.mindsensors.com/).
The last most significant LEGO components are the actuators.In this case, they consist on dc motors that in-corporate encoders sensors integrated. These sensors havea 1-degree resolution, which improves the previous ver-sion encoders resolution. A more detailed description about LEGO Mindstorms NXT motors can be found inhttp://www.philohome.com/motors/motorcomp.htm ([1]).
In this Lab, we are interested on the modeling of the three wheel robot presented in Figure 2.1.
Figure 2.1: Picture and scheme of the LEGO robot with three wheels.
Two of the wheels are independently powered by electric motors and one is left free with the purpose of letting therobot turn. The way of controlling the robot position is to change the speed of the motorized wheels.
• the translation of the robot is produced by the sum of the rotational speed of the two wheels;
• the rotation is produced by the difference between these speeds;
5
Using this information, the student has to find a dynamic model for the LEGO robot applying the three phases ofmodeling seen in class. For this exercice the only availablemeasurements are the encoders sensors integrated in the DCmotors. From this measurements, a wheel angular speed estimator based on a Kalman filter approach has to be conceived.
2.2 Robot Programming (lab 1)
The following functions may be useful for the programming ofthe three wheel robot. Check the Not eXactly C (NXC)Programmer’s Guide for more detailhttp://bricxcc.sourceforge.net/nbc/nxcdoc/NXC_Guide.pdf.
For the servomotors, check:
• ResetTachoCount(port alias)
• MotorTachoCount(port alias)
• MotorActualSpeed(port alias)
• MotorPower(port alias)
• OnFwdReg(port alias,power,...)
For time management
• CurrentTick()
• FirstTick()
• Wait(time)
2.3 Robot Modeling (lab 1)
In this section, it is expected to obtain a model of the robot shown in Figure 2.1 capable of representing the robot positiondynamics.
A-1 Perform the three phases of modeling for the robot dynamics: structuring the problem, setting up the basic equationsand forming the state-space models. Explain the assumptions made for the model conception;
A-2 create the robot model in Matlab Simulink, illustrate the results obtained;
A-3 program in the robot a trajectory and acquire the necessary information from the robot in order to compare the resultswith the simulator previously made. Discuss and illustratethe results.
2.4 Kalman Filter (lab 2)
In this section, it is expected to design a Kalman filter in order to estimate the wheels angular velocitiesωl andωr fromthe encodes of each wheel (θl andθr ). (Read [1])
B-1 write the linear model of the wheel angular velocities and then discretize it. Consider that only the encoders mea-surements are available.
B-2 design a Kalman filter in order to estimate the robot wheels angular velocities. (Consider the measurement andprocess covariance matrices presented in [1]).
B-3 Using Matlab Simulink, illustrate the difference between taking the time derivative of the wheel angleto estimate theangular velocity and the Kalman filter.
B-4 Reconstruct the robot position from the estimated wheelangular speeds using the robot model developed in theprevious section.
B-5 program in the robot the angular speed estimator using the Kalman filter. Implement in the robot the developedmodel in order to get an estimate of the robot position from the wheel angular speeds.
6
2.5 Recursive Identification of the LEGO (lab 2)
In this section, implement a recursive identification of a LEGO robot using the recursive least-square (RLS) algorithm(read second chapter of [2]). Consider:
y(t) = φT ∗ ψ(t) (2.1)
Where the parameter vectorφ is to be estimated from the measurementsy(t). ψ(t) is the regressor (vector of laggedinput-output).
B-1 build the linear difference equation of the dynamic system with output signalθ, and input signalu. In this caseθ isthe measurement given by the encoders. The input signal is the power specified for the electric motor. Take only intoaccount one time step lag,
B-2 which is the advantage of implement the RLS instead of thenon-recursive least square method?;
B-3 state the recursive least-square (RLS) algorithm in other to estimate the parameter vectorφ;
B-4 implement RLS algorithm in the LEGO robot and present theresults obtained;
2.6 Conclusions (lab 2)
Conclude and comment.
7
Chapter 3
Modeling of a Thermonuclear Plant
The aim of thismodeling for control labis to investigate the modeling issues associated with a complex industrial plant,including distributed dynamics and numerous actuators andsensors. A thermonuclear plant, such as a tokamak, providesfor a particularly appropriate choice to fulfill this goal. Indeed, a tokamak is both fascinating as the potential mainsource of energy for future generations and a key challenge for control engineers, as it necessitates advanced controlmethodologies with tight modeling and real-time issues. Inthis lab, the main principles of fusion are first presented,motivating the interest for profile control and dedicated tokamak experiments. We will then focus more precisely on ToreSupra tokamak (CEA-Cadarache, DRFC) and the mathematical description of the magnetic flux dynamics, along with themain actuators. This description will be used to develop an object-oriented model, using thethree phases of modelingpresented in class. Based on an existingMatlabr code and the obtained model, aSimulinkr simulator will be built. Mostof the presented results are detailed in [3].
3.1 What is thermonuclear fusion?
- Excerpts from ITER official websitewww.iter.org-Fusion is the energy source of the sun and the stars. On earth,fusion research is aimed at demonstrating that this
energy source can be used to produce electricity in a safe andenvironmentally benign way, with abundant fuel resources,to meet the needs of a growing world population.
Nuclear fusion involves the bringing together of atomic nuclei. The atom’s nucleus consists of protons with a singlepositive charge and neutrons of similar mass and no charge. The strong nuclear force holds these ”nucleons” togetheragainst the repulsive effect of the proton’s charge. As many negatively charged electrons as protons swarm around thenucleus to balance the proton charge, and the mass of the atomlies almost totally in the nucleus.
The sum of the individual masses of the nucleons is greater than the mass of the whole nucleus. This is becausethe strong nuclear force holds the nucleons together - the combined nucleus is a lower energy state than the nucleonsseparately. The difference, the binding energy (E = m.c2), varies from one element to another. Because of the possibleways that nucleons can pack together, when two light atomic nuclei are brought together to make a heavier one, thebinding energy of the combined nucleus can be more than the sum of the binding energies of the component nuclei (i.e. itis in an even lower energy state). This energy difference is released in the ”fusion” process.
A similar situation occurs when heavy nuclei split. Again the binding energies of the pieces can be more than thatof the whole (i.e. they are in a lower energy state), and the excess energy is released in the ”fission” process. Thesealternatives are shown in Fig. 3.1.
To conclude this short introduction on fusion, an interesting movie describing ITER, the International Tokamak Ex-perimental Reactor, is available at:http://www.iter.org/Flash/iter5minviewing.htm
3.2 Tokamaks and profile control
In the coming years the main challenge in the fusion community will be the development of experimental scenarios forITER. Amongst them, the so-called advanced tokamak steady-state ones will play a significant role, since they will allowto reproduce and study (on a smaller scale), the conditions that are expected to be obtained in a fusion plant of reduced sizeand costs [4]. In these scenarios a particular emphasis is given to the current density profile and to the way of producingthe plasma currentIP. Due to the intrinsic limited availability of magnetic flux in the fusion devices, needed to sustain apurely inductive current,IP will have to be mainly generated by non-inductive sources. In particular, the importance of thereal-time safety factor profile (q-profile) control is emphasized in [5], where an interestingoverview on recent advancesand key issues in real-time tokamak plasma control is provided. A Proportional-Integral (PI) feedback control strategy,
8
Figure 3.1: Fusion and Fission principles
Figure 3.2: A schematic view of Tore Supra toka-mak
based on a simple linear model, is also proposed and its efficiency is illustrated by experimental results, which motivatefurther research developments in this direction.
The control of so-called “advanced” plasma regimes [4, 6, 7]for steady state high performance tokamak operationis a challenge, in particular because of the non-linear coupling between the current density and the pressure profiles. Ina burning plasma, the alpha-particle power will also be a strong function of these profiles, and, through its effect on thebootstrap current, will be at the origin of a large (though ultra-slow) redistribution of the current density. The possibledestabilization of adverse Toroidal Alfven Eigenmodes (TAE) - such as the drift kinetic modes that are anticipated toappear at high values of the central safety factor [8] - as well as potential thermal instabilities due to the ITB dynamicswill further complicate the issue. This motivates the need for further investigation of plasma profiles shaping to guaranteeand control steady-state operation of the plasmas.
Considering the tokamak magnetic configuration, such as theone presented in Fig. 3.3, the plasma can be described asimbricated surfaces. The physical variables of interest for profile control (temperature, density and magnetic flux) can beconsidered as constant on these surfaces. The problem of modeling a plasma in 3-D space can consequently be simplifiedto the modeling of physical variables along the normalized minor radius (1-D problem).
To summarize:
• advanced plasma confinement schemes require further research on current density profile control, which motivatesthe need for a representative real-time model of plasma physics for control synthesis;
• such a model will focus on the magnetic flux dynamics, from which the current density profiles are obtained;
• for a tokamak such as Tore Supra, the distributed inputs are due to the (auto-generated) bootstrap effect, the poloidalcoils (magnetic source), and the lower hybrid (LH), electron cyclotron current drive (ECCD) and ion cyclotron radioheating (ICRH) antennas, as presented in Fig. 3.4.
3.3 System dynamics
In order to simplify the plasma model, we make the following hypotheses:
• the coordinates are supposed to be cylindrical (neglect theimpact of the toroidal shape),
• the diamagnetic effect is neglected.
The magnetic flux dynamics is consequently obtained as [9, 10]:
∂ψ
∂t(x, t) = η∥(x, t)
[
1µ0a2
∂2ψ
∂x2+
1µ0a2x
∂ψ
∂x+ R0 jni(x, t)
]
(3.1)
jφ(x, t) = − 1µ0R0a2x
∂
∂x
[
x∂ψ
∂x
]
(3.2)
whereη∥(x, t) is the resistivity,µ0 = 4π × 10−7 H/m is the permeability of free space,a is the small plasma radius,R0 ismagnetic center location,jni(x, t) = jbs + jLH + jECCD is the non-inductive current source, including both the bootstrapeffect jbs and the microwave current drive (jLH and jECCD), jφ(x, t) is the poloidal current profile (controlled output). Theboundary conditions are given byψ′(0, t) = 0,ψ′(1, t) = f (Ip) or ψ(1, t) = f (Vloop), and the initial conditions are known.
The computations of the resistivity, the temperature and density profiles, as well as non-inductive current sources, arebriefly described below.
9
Figure 3.3: Isoflux maps, particles trajectories and controlled profiles
• Poloidal,
• LH,
• ECCD,
• ICRF.
• j i , q, I i , Vloop,
• l i , βθ, ∆.
Figure 3.4: Input-output representation of Tore Supra
3.3.1 Resistivity and bootstrap current
The plasma resistivity mainly depends on the electron temperature and density profiles,Te(x, t) andne(x, t), respectively,and on the effective value of the plasma chargeZ. We can use the formulae detailed in [11], which can be summarized as
η∥(x, t) = f (Te, ne, Z) (3.3)
where a proportionality onTe−3/2 plays the major role.
The bootstrap current is generated by trapped particles andmay be the main source of non inductive current in specificscenarios (high bootstrap experiments). It is a self induced current (created by the isoflux surfaces) that induces significantnonlinearities in the magnetic flux equation. We consider the model derived in [12]:
jbs(x, t) =eR0
∂ψ/∂x
(
A1Z + αTi (1− αi)
Z− A2
)
ne∂Te
∂x
+A1Z + αTi
ZTe∂ne
∂x
(3.4)
10
with ne = Zni andTi = αTiTe.
3.3.2 Temperature and density profiles
The temperature profiles can be obtained thanks to direct measurement or using the scaling laws proposed in [13], whichare detailed here. The electron temperature profile is estimated with
Te(x, t) ≈α(t)
1+ e−β(t)(x−γ(t))ATe(t) (3.5)
where the normalized shape of the profileTe(x, t)/Te(0, t) is estimated with a sigmoid function defined by its amplitudeα(t), dilatationβ(t) and translation (inflection point)γ(t). The amplitude of the profile isATe(t) and computed from theplasma thermal energy. The shape parameters are set with theswitched model
α, β, γ =
αlh, βlh, γlh if Plh , 0
αω, βω, γω else.
This model then distinguishes the LHCD heating effect from the ohmic and ICRH ones. Selecting the most significantterms, the shape parameters are related to the global and engineering parameters with
αlh = e−0.87I−0.43p B0.63
φ0N∥0.25
(
1+Picrh
Ptot
)0.15
βlh = −e3.88I0.31p B−0.86
φ0n−0.39
e N∥−1.15
γlh = e1.77I1.40p B−1.76
φ0N∥−0.45
(
1+Picrh
Ptot
)−0.54
whereIp is the total plasma current,Bφ0 the toroidal magnetic field (at the center of the plasma),N∥ the parallel refractionindex,Picrh the ICRH power andPtot the total input power. Note that the first three variables areglobal plasma parameterswhile the others correspond to some inputs (the LH powerPlh is included inPtot). For the ohmic and ICRH cases, wehave that:
αω = e−0.37I−0.46p B0.23
φ0n0.22
e
βω = −e1.92I0.38p n−0.33
e
γω = e−0.15I1.03p B−0.51
φ0
(
1+Picrh
Ptot
)−0.46
The dynamics of the temperature profiles is obtained from theplasma thermal energyWth(t)
Wth(t) =3e2
∫
V(1+ αTiαni) neTe dV (3.6)
whereαTi andαni are scaling laws, ande is the electron charge. The temperature profile amplitude isrelated toWth thanksto the relationshipATe(t) = A(t)Wth(t) andWth is estimated with
τth(t) = 0.135I0.94p B−0.15
φ0n0.78
e
(
1+Plh
Ptot
)0.13
Ptot−0.78
dWth
dt= Ptot −
1τth
Wth, Wth(0) = Ptot(0)τth(0)
(3.7)
whereτth(t) is the thermal energy confinement time.The density profiles are obtained from line-averaged measurements or thanks to an interferometer). Considering the
simplest case, the density profile is approximated with
ne0(t) =γn + 1γn
ne(t) and ne(x, t) = ne0(t)(1− xγn) (3.8)
wherene(t) is the line averaged density.
11
3.3.3 Discretization/Computation issues
Main dynamics
Some flexibility in the discretization scheme can be introduced by a variable step spatial discretization:
ψ(xi , t)ex =η∥ i, j
µ0a2
(
d2ψi+1, j − d3ψi, j + d4ψi−1, j
)
+d1η∥ i, j
µ0a2xi
(
ψi+1, j − ψi−1, j
)
+ η∥ i, jR0 jni i, j
= η∥ i, j
(
e1ψi+1, j − e2ψi, j + e3ψi−1, j + R0 jni i, j
)
(3.9)
where the subscripti = 1 . . .N denotes the spatial discretization points andj refers to the time samples. The parameterse1, e2 ande3 can be adjusted to set the desired mesh topography (i.e. it may be interesting to have a more precise resolutionat the plasma center or to use these parameters to ensure the robustness of the discretization scheme).
Similarly, an extra degree of freedom can be introduced in the temporal discretization by using the implicit-explicitscheme: (
ψi, j+1 − ψi, j
δt
)
= h ψ(xi , t)ex+ (1− h) ψ(xi , t)im (3.10)
whereh determines the ratio of explicit to implicit computation.The final computation writes as:
Ai,i+1, j ψi+1, j + Ai,i, j ψi, j + Ai,i−1, j ψi−1, j − Bi,i+1, j ψi+1, j+1 (3.11)
− Bi,i, j ψi, j+1 − Bi,i−1, j ψi−1, j+1 + Si, j = 0 (3.12)
⇔ ψ j+1 = B−1j A j ψ j + B−1
j S j (3.13)
where the transition matrices both depend on the physical model and on the numerical issues.
Boundary conditions
Specific care has to be given for the central point as the magnetic flux gradient becomes in that caseψ′(x1,t)x1= 0
0. One wayto solve this problem is to discretize the equation with:
1x∂
∂x
[
x∂ψ
∂x
]
≈ 1δx2/4
δx22 ψ′1/2
δx22
=4δx2
ψ′1/2 ≈4
δx22
(ψ2 − ψ1) (3.14)
for this specific point.
3.4 Model Inputs
There are three different types of inputs on the previous magnetic field model.
3.4.1 Boundary condition
The boundary input is set by the poloidal coils currentIc(t) (inductive inputs) as [14]:
Lc Ic + MIp + RcIc − Vc = 0 (3.15)
MIc + Lp Ip + Rp
(
Ip − INI
)
= 0 (3.16)
whereRc andLc are the coils resistance and internal inductance,Rp andLp are the plasma resistance and inductance,Mis the matrix of mutual inductances,Vc is the input voltage applied to the coils andINI is the current generated by the noninductive sources. Note that a local control loop setsVc according toIp,re f or Vloop,re f , which are then considered as theactual boundary inputs.
3.4.2 Distributed current deposit from the antennas (LH, ECCD)
This current deposit is approximated thanks to the gaussianapproximation (engineering approach):
jcd/lh(x, t) = ϑ(t)e−(µ(t)−x)2/2σ(t)
12
(Rant,Zant)
(Rabs,Zabs)
φpol
R∗Rc
R0 R
Z
Figure 3.5: ECCD current deposit0 0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1
0
0.5
1
1.5
2
2.5
xwhxrµhxr
Emax
2
Emax
HX
Rem
issi
on measurementgaus. approx.
Figure 3.6: LH current deposit
ECCD current deposit
The total EC current is the sum of several deposits due to several EC beams, which parameters are denoted by the subscriptm. Each current deposit is determined by the position of the steering mirror (Rant,m,Zant,m), its orientation in the poloidaland toroidal plane (φpol,m, φtor,m), and the emission powerPcd,m, as presented in Figure 3.5. The ECCD current deposit islocalized betweenR∗ andRc, and is computed as [15]:
Rc = 0.2044nhI ind
f, (3.17)
R∗m =Rant,m√
2| sinφtor,m|
1−
√
1−4R2
c
R2ant,m
sin2 φtor,m
1/2
, (3.18)
γcd,m =Γ1Te(xabs,m)
Te(xabs,m) + 105×
1− Γ2
[ρm(Rabs,m, φpol,m) + Rabs,m− R0
Rabs,m
]Γ3
, (3.19)
ϑcd,m =γcd,mPcd,m
Rone
[
2πa2∫ 1
0xe−(µcd,m−x)2/2σcd,mdx
]−1
×
sign(φpol,m). (3.20)
LH current deposit
This deposit shape can either be obtained directly from HardX-Rays measurement or computed using the scaling laws:
whxr = α0Bα1φ0
Iα2p nα3Plh
α4 N∥α5 (3.21)
µhxr = β0Bβ1
φ0I β2p nβ3Plh
β4N∥β5 (3.22)
The current drive efficiency is given by [16]:
ηlh(t) = 3.39D0.26n τ0.46
th Z−0.13, (3.23)
with Dn(t) ≈ 2.03− 0.63N∥, and its magnitude is:
ϑlh(t) =1
2πa2∫ 1
0xe−(µhxr−x)2/2σlhdx
ηlhPlh
neR0(3.24)
Summing the bootstrap and antennas current sources, we havethat:
jni(x, t) = jbs(x, t) + jcd(x, t) + j lh(x, t) (3.25)
3.4.3 Temperature profile modifications
The temperature profile modification, due to ICRF and LH antennas, have a direct influence on the magnetic flux throughthe resistivity and bootstrap current.
13
3.5 Lab Exercises
3.5.1 Modeling
Perform the “three steps on modeling” on Tore Supra tokamak (4 points):
1. Structuring the problem
2. Setting up the Basic Equations
3. Forming the State-Space Models
3.5.2 Simulation
1. Download and run theMatlabr files (shot 35109) from the class website.
2. Discretization issues (4 points):
(a) indicate which part of the code should be modified in orderto modify the temporal discretization;
(b) simulate changes in the time step and implicit to explicit ratio.
3. Control architecture (4 points):
(a) run the open-loop and the closed-loop configurations andcompare your results;
(b) analyze the control architecture;
(c) modify the control parameters and comment.
4. Disturbance (3 points):
(a) introduce a disturbance with the ECCD antenna at 15 s and 21 s;
(b) discuss the effect on the dynamics;
(c) investigate the result when a scalar feedback is used.
5. Put a step on boundary control and discuss the effect (3 points).
6. Write a consistent report with an introduction, a conclusion, labelled figures etc. (2 points)
Note: comment and analyze ALL your simulations.
14
Chapter 4
Simulation and Control of an InvertedPendulum Using Scilab
4.1 Tasks
1. StartScilabr and run
?/Demonstrations Scilab/CACSD/Inverted
pendulum
2. Explain the different steps of this demonstration and the illustrated computing capabilities.
3. Find and download in your personal space
\scilab-5.0.3\modules\cacsd\demos\pendulum
4. Relate the demonstration steps to the algorithm (printedin the Appendix).
5. Motivate the equations used for feedback control and observation (Jacobian computation from the nonlinear systemto obtain the linearization, observer formulation and feedback gain design).
6. Modify the control strategy to use an optimal control approach, i.e. with
I =∫ t f
t0
xTdiag(qi)x+ ru2dt
Discuss the impact of the ratiosqi/r on the simulation results. Note: you may use thericcati function.
4.2 Algorithms
4.2.1 pendule.dem
mode(-1)
path=get_absolute_file_path(’pendule.dem’);
getf(path+’simulation.sci’)
getf(path+’graphics.sci’)
//disable displayed lines control
display_props=lines(); lines(0)
my_handle = scf(100001);
clf(my_handle,"reset");
xselect(); wdim=xget(’wdim’)
//mode(1)
dpnd()
//
// equations
//----------
//state =[x x’ theta theta’]
//
mb=0.1;mc=1;l=0.3;m=4*mc+mb; //constants
//
x_message([’Open loop simulation’
’ ’
’ y0 = [0;0;0.1;0];
//initial state [x x’’ theta theta’’]’
’ t = 0.03*(1:180); //observation dates’
’ y = ode(y0,0,0.03*(1:180),ivpd);
//differential equation integration’
’ //Display’
15
’ P = initialize_display(y0(1),y0(3))’
’ for k=1:size(y,2),
set_pendulum(P,y(1,k),y(3,k));end’])
//
y0=[0;0;0.1;0];
y=ode(y0,0,0.03*(1:180),ivpd);
P=initialize_display(y0(1),y0(3));
for k=1:size(y,2), set_pendulum(P,y(1,k),y(3,k));end
//
x_message([’Linearization’
’ ’
’ x0=[0;0;0;0];u0=0;’
’ [f,g,h,j]=lin(pendu,x0,u0);’
’ pe=syslin(’’c’’,f,g,h,j);’
’ // display of the linear system’
’ ssprint(pe)’])
//
mode(1)
x0=[0;0;0;0];u0=0;
[f,g,h,j]=lin(pendu,x0,u0);
pe=syslin(’c’,f,g,h,j);
ssprint(pe)
mode(-1)
//
x_message([’Checking the result’
’ //comparison with linear model computed by hand’;
’’
’ f1=[0 1 0 0’
’ 0 0 -3*mb*9.81/m 0’
’ 0 0 0 1’
’ 0 0 6*(mb+mc)*9.81/(m*l) 0];’
’ ’
’ g1=[0 ; 4/m ; 0 ; -6/(m*l)];’
’ ’
’ h1=[1 0 0 0’
’ 0 0 1 0];’
’ ’
’err=norm(f-f1,1)+norm(g-g1,1)+norm(h-h1,1)+norm(j,1)’])
//
mode(1)
f1=[0 1 0 0
0 0 -3*mb*9.81/m 0
0 0 0 1
0 0 6*(mb+mc)*9.81/(m*l) 0];
g1=[0 ; 4/m ; 0 ; -6/(m*l)];
h1=[1 0 0 0
0 0 1 0];
err=norm(f-f1,1)+norm(g-g1,1)+norm(h-h1,1)+norm(j,1)
mode(-1)
x_message([’Linear system properties analysis’
’ spec(f) //stability (unstable system !)’
’ n=contr(f,g) //controlability’
’ ’
’ //observability’
’ m1=contr(f’’,h(1,:)’’) ’
’ [m2,t]=contr(f’’,h(2,:)’’)’])
//---------
mode(1)
spec(f) //stability (unstable system !)
n=contr(f,g) //controlability
//observability
m1=contr(f’,h(1,:)’)
[m2,t]=contr(f’,h(2,:)’)
mode(-1)
//
x_message([’Synthesis of a stabilizing controller
using poles placement’
’ ’
’// only x and theta are observed:
contruction of an observer’
’// to estimate the state : z’’=(f-k*h)*z+k*y+g*u’
’ to=0.1; ’
’ k=ppol(f’’,h’’,-ones(4,1)/to)’’ //observer gain’
’// compute the compensator gain’
’ kr=ppol(f,g,-ones(4,1)/to)’])
//-------------------------------------------------
//
//pole placement technique
//only x and theta are observed:
contruction of an observer
//to estimate the state : z’=(f-k*h)*z+k*y+g*u
//
to=0.1; //
k=ppol(f’,h’,-ones(4,1)/to)’ //observer gain
// verification //
// norm( poly(f-k*h,’z’)-poly(-ones(4,1)/to,’z’))
//
kr=ppol(f,g,-ones(4,1)/to) //compensator gain
//
x_message([’Build full linear system
pendulum-observer-compensator’
’ ’
’ ft=[f-g*kr -g*kr’
’ 0*f f-k*h]’
’ ’
’ gt=[g;0*g];’
’ ht=[h,0*h];’
’’
’ pr=syslin(’’c’’,ft,gt,ht);’
’’
’//Check the closed loop dynamic’
’ spec(pr.A)’
’ ’
’//transfer matrix representation’
’ hr=clean(ss2tf(pr),1.d-10)’
’ ’
’//frequency analysis’
’ black(pr,0.01,100,[’’position’’,’’theta’’])’
’ g_margin(pr(1,1))’])
//---------------------------------------------
// state: [x x-z] //
mode(1)
ft=[f-g*kr -g*kr
0*f f-k*h];
gt=[g;0*g];
ht=[h,0*h];
pr=syslin(’c’,ft,gt,ht);
// closed loop dynamics:
spec(pr(2))
//transfer matrix representation
hr=clean(ss2tf(pr),1.d-10)
16
//frequency analysis
clf()
black(pr,0.01,100,[’position’,’theta’])
g_margin(pr(1,1))
show_pixmap()
mode(-1)
//
x_message([’Sampled system’
’ ’
’ t=to/5; //sampling period’
’ prd=dscr(pr,t); //discrete model’
’ spec(prd.A) //poles of the discrete model’])
//---------------//
mode(1)
t=to/5;
prd=dscr(pr,t);
spec(prd(2))
mode(-1)
//
x_message([’Impulse response’
’ ’
’ x0=[0;0;0;0;0;0;0;0]; //initial state’
’ u(1,180)=0;u(1,1)=1; //impulse’
’ y=flts(u,prd,x0); //siscrete system simulation’
’ draw1()’])
//-----------------//
mode(1)
x0=[0;0;0;0;0;0;0;0];
u(1,180)=0;u(1,1)=1;
y=flts(u,prd,x0);
draw1()
mode(-1)
//
x_message([’Compensation of the non linear system with’
’linear regulator’
’’
’ t0=0;t1=t*(1:125);’
’ x0=[0 0 0.4 0 0 0 0 0]’’; // initial state’
’ yd=ode(x0,t0,t1,regu);
//integration of differential equation’
’ draw2()’])
;
//--------------------------------------
// simulation //
mode(1)
t0=0;t1=t*(1:125);
x0=[0 0 0.4 0 0 0 0 0]’; //
yd=ode(x0,t0,t1,regu);
draw2()
mode(-1) lines(display_props(2)) x_message(’The end’)
4.2.2 simulation.sci
function [xdot]=ivpd(t,x)
//ydot=ivpd(t,y) non linear equations of the pendulum
// y=[x;d(x)/dt,teta,d(teta)/dt].
// mb, mc, l must be predefined
//!
g=9.81; u=0 qm=mb/(mb+mc) cx3=cos(x(3)) sx3=sin(x(3))
d=4/3-qm*cx3*cx3
xd4=(-sx3*cx3*qm*x(4)**2+2/(mb*l)*(sx3*mb*g-qm*cx3*u))/d
//
xdot=[x(2);
(u+mb*(l/2)*(sx3*x(4)**2-cx3*xd4))/(mb+mc);
x(4);
xd4]
function [y,xdot]=pendu(x,u)
//[y,xdot]=pendu(x,u) input (u) - output (y) function of the pendulum
//!
g=9.81; qm=mb/(mb+mc) cx3=cos(x(3)) sx3=sin(x(3)) d=4/3-qm*cx3*cx3
xd4=(-sx3*cx3*qm*x(4)**2+2/(mb*l)*(sx3*mb*g-qm*cx3*u))/d
//
xdot=[x(2);
(u+mb*(l/2)*(sx3*x(4)**2-cx3*xd4))/(mb+mc);
x(4);
xd4]
//
y=[x(1);
x(3)]
function [xdot]=regu(t,x)
//xdot=regu(t,x) simulation of the compound system
// pendulum - observer - compensator
//!
xp=x(1:4);xo=x(5:8); u =-kr*xo //control [y,xpd]=pendu(xp,u)
xod=(f-k*h)*xo+k*y+g*u xdot=[xpd;xod]
17
Chapter 5
Identification of an Active Vibration ControlBenchmark Using Matlab
As stated inWikipedia, “active vibration control is the active application of force in an equal and opposite fashion to the forces imposedby external vibration. With this application, a precision industrial process can be maintained on a platform essentially vibration-free.Many precision industrial processes cannot take place if the machinery is being affected by vibration. For example, the productionof semiconductor wafers, or for reducing vibration in helicopters, offering better comfort with less weight than traditional passivetechnologies. In the past, only passive techniques were used. These include traditional vibration dampers, shock absorbers, and baseisolation.”
In this lab, you will learn how to useSystem Identificationmethods seen in class in order to model a benchmark on vibration controlrecently proposed byGIPSA-lab[17].
5.1 System description and preliminary workThe active vibration compensation experiment is depicted in Figure 5.1. It can be basically considered as a LTI system with two inputs(the controlled forceu and the disturbance forceuP) and one output: the residual forcey.
Figure 5.1:GIPSA-labvibration benchmark.
In order to prepare the lab properly:
1. Download all the files in the class website and read throughthe system descriptionBenchmarkV1.1.pdf. Check for additionalinformation in [17].
2. Synthesize the information (2 pages summary) that is madeavailable concerning the system properties (number of input andoutput channels, derivative effects), data (sampling frequencies, inputs properties) andsimulation files.
In order to illustrate the key concepts seen in class, do the following.
18
5.2 Experiment design and frequency-response analysis1. Load a data set (i.e.data_prim1.mat) and visualize the input datau:
(a) which kind of signal is it?
(b) superpose this signal everyN = 210 − 1 samples, i.e. usingi = 1:N; plot(i,u(i),i,u(i+N),i,u(i+2*N))
what can you conclude on the input sequence?
(c) estimate its power spectral density (i.e. withpwelch) and conclude on the frequency contents;
(d) discuss the properties and advantages of such an input for system identification.
2. Input/output relationships:
(a) load and comment the functionfreq_resp_cal provided on the website;
(b) plot on the same graph the magnitude of the frequency obtained by spectral analysis for the primary and secondarychannels;
(c) what happens when the PRBS clock frequency is divided by two?
(d) discuss the similarities and differences between the two channels (common poles/zeros, order, etc.).
3. Conclude with some guidelines concerning the model architecture.
Note: if theMatlab Signal Processing Toolboxis not installed on you computer, you may download theoctave-signalfree software athttp://octave-signal.sourcearchive.com/
5.3 Spectral analysis1. Empirical transfer-function estimate (ETFE):
(a) compute the ETFE (i.e. withetfe) of both channels and discuss your results;
(b) smooth the ETFE thanks to Hamming window and vary the window parameterγ (i.e. withspa, getff andbode);
(c) for eachγ, computeM(γ) andW(γ). Relate these parameters to the smoothed ETFEs obtained above.
2. Autocorrelation analysis:
(a) compute the correlation between inputs and outputs for both channels;
(b) what is the periodicity of the peaks (in number of samples)? What is the implication of this observation?
(c) by zooming on the cross correlations (choosing one period), what is the horizon of the deterministic behavior?
(d) what can be concluded on the model order, or the presence of delays, for both channels?
Note: the materialDigital Signal Processingproposed in [18] may be useful for this analysis.
5.4 Estimating the disturbance spectrum1. Identify (withMatlab Identification Processing Toolbox) the decoupled data thanks to a 6th order state-space model with to the
prediction-error method.
2. Using this model, compute the residual and its spectrum.
3. Discuss the model choice and compare it with the other models easily available.
4. Set the experimental inputs as common inputs in the model provided insimulator_bench.mdl and in the previous state-spacemodel, and compare the two outputs with the experimental data.
5.5 ConclusionsSummarize your main results and conclude.
19
Chapter 6
Analysing the Anthropogenic Impact on theOzone Layer Depletion
The aim of this lab is to investigate the anthropogenic impact on the ozone layer deplection thanks to the analysis of a model ofCH3CCl3 transport in firn ice cores. As detailed in [19]:
“1,1,1-Trichloroethane is an excellent solvent for many organic materials and also one of the least toxic of the chlorinated hydro-carbons. Prior to the Montreal Protocol, it was widely used for cleaning metal parts and circuit boards, as a photoresistsolvent in theelectronics industry, as an aerosol propellant, as a cutting fluid additive, and as a solvent for inks, paints, adhesivesand other coatings.
It was also the standard cleaner for photographic film (movie/slide/negatives, etc.). Other commonly available solvents damageemulsion, and thus are not suitable for this application. The standard replacement, Forane 141 is much less effective, and tends toleave a residue. 1,1,1-Trichloroethane was used as a thinner in Correction fluid products such as Liquid paper. Many applications for1,1,1-trichloroethane (including film cleaning) were previously done with carbon tetrachloride, which was banned in YEAR1.
The Montreal Protocol targeted 1,1,1-trichloroethane as one of those compounds responsible for ozone depletion and banned itsuse beginning in YEAR2. Since then, its manufacture and use has been phased out throughout most of the world.
1,1,1-Trichloroethane is generally considered as a non-polar solvent, but since all three electronegative chlorine atoms lie on thesame side of the molecule, it is slightly polar, making it a good solvent for organic matters that do not dissolve into totally non-polarsubstances such as hexane. 1,1,1-Trichloroethane in laboratory analytics has been replaced by other solvents.”
6.1 Data properties (lab 1)Download the fileCH3CCl3.mat on the class website and run:
%% Data loading
load CH3CCl3.mat % Qdirect x0 A B Qatm Qfin x
nt = length(Qatm.time);
nx = length(x);
ts = mean(diff(Qatm.time));
figure(1)
plot(Qatm.time, Qatm.values)
xlabel(’Time (years)’,’fontsize’,14)
ylabel(’Concentration’,’fontsize’,14)
title(’Initial scenario’,’fontsize’,16)
• What are the types of data provided to you?
• From the figure, can you guess YEAR1 and YEAR2 mentioned in the introduction?
6.2 Transport model and time loop (lab 1)The system provided in the data file is the transport model:
x = Ax+ Bu, x(0) = x0
wherex is the state vector of the gas concentrations at the discretized depth andu is the atmospheric scenario. Using a purely implicitscheme, as seen in class, the time evolution of the concentration in the firn can be obtained as:
%% Time loop (remove top and bottom)
I_ts = eye(nx-2)./ts;
A_D = inv(I_ts - A);
20
Qdirect = zeros(nx-2,nt);
Qdirect(:,1) = x0;
B_D = A_D*[B;zeros(nx-3,1)];
A_DCS = A_D*I_ts;
for i = 1:nt-1
Qdirect(:,i+1) = A_DCS*Qdirect(:,i)
+ B_D*Qatm.values(i+1);
end
% Complete the model with atmospheric and bottom
boundary conditions
Qdirect = [Qatm.values’;Qdirect;Qdirect(nx-2,:)];
1. Write the discrete equations corresponding to Euler backward, Euler forward and Tustin discretizations. What wouldbe the truediscretization?
2. Modify the previous code to run your results.
3. Superpose your results on the figure:
figure(2)
errorbar(Qfin.space,Qfin.values,Qfin.error,’x’)
hold on
plot(x,Qdirect(:,end),’b’)
axis tight
xlabel(’Depth (m)’,’fontsize’,14)
ylabel(’Concentration’,’fontsize’,14)
title(’Ice core measurement vs. model’,
’fontsize’,16)
4. Investigate the impact of the time step and the robustnessof the schemes.
6.3 Atmospheric scenario (lab 1-2)1. Modify the atmospheric scenario by adding a white noise and a bias on the atmospheric input: verify that this noise is a discrete
stochastic signal. How can you characterize the model sensitivity?
2. What are the time constants associated with the model (provide for different depths)?
3. What is the frequency content of the atmospheric scenario? Hint: look at the power spectrum, computed thanks tohttp://www.ac.tut.fi/aci/courses/ACI-42070/esiselostus.pdf.
4. Draw the Bode diagram, taking as outputs the concentrations at different depths. Conclude on the seasonality effect of atmo-spheric concentration changes in the deep firn.
6.4 I/O properties (lab 2)Consider fictitious measurements carried on ice below the last firn depthy (i.e. inferred from trace gas measurements in ice and madetime-dependant through the firn sinking speed).
1. What is the I/O cross spectrum?
2. How can it be used to infer the disturbance model?
3. Discuss the choice of the sampling interval of the atmospheric scenario.
6.5 Conclusions (lab 2)Conclude on the main aspects covered in this lab and the key steps to validate the scenario/model adequacy, as well as the inferredphysical properties. Propose some perspectives for the identification and optimization of such models.
21
Chapter 7
Optimization: Particle Source Identificationin a Tokamak
In this work, the problem of particle source identification from distributed electron density measurements in fusion plasmas, such asthe ones obtained in Tore Supra tokamak, is considered. A transport model, suitable for identification purposes, is proposed based on asimplification of classical particle transport models. Youwill first derive a steady-steady state model from the PDE discretization. Theoptimization strategies considered in class will then be applied to determine a feedback scheme that provides an on-line identificationof the source input. You will finally apply your algorithm to dedicated experimental data from Tore Supra.
7.1 IntroductionRecent developments in control theory and controlled thermonuclear fusion research are naturally leading to researchtopics of commoninterest that are particularly challenging for both scientific communities. For example, new modeling and identification tools are neededfor the understanding and analysis of complex physical phenomena that occur in thermonuclear fusion. The representation (qualitativeand quantitative) of particles transport at the plasma edgeis an example of such topics.
Tokamak experiments, such asTore Supraor JET, are equipped with Lower Hybrid (LH) antennas to heat the plasma and createthe toroidal current. The LH waves are recognized as the mostefficient non-inductive current drive sources and their use is forecastedfor ITER experiment. The efficiency of such antennas is strongly related to our ability toensure an appropriate coupling between thewaves on the plasma, which directly depends on the electron density in a region called thescrape-off layer (SOL), located between thelast closed magnetic surface (theseparatrix) and the wall. This region, as well as the key elements discussed in this paper, are depictedin Fig. 7.1. The importance of local density control in the SOL is discussed in [20], where dedicated experimental conditions areestablished for long distance shots (plasma experiments) on Tore Supra. It is further emphasized in [21], where the coupling efficiencyis improved thanks to gas puffing in front of the launcher on JET (reduced reflected power). Afirst attempt to control the couplingbetween LH and the SOL is proposed in [22], based on a scalar model.
The electron density in the SOL is directly influenced by the LH input powerPLH [20]. Indeed, a small but significant part ofthis power is absorbed in the plasma edge during the wave propagation to the core. A possible effect of this absorption is the gasionization, which results in an increased electron density. This suggests thatPLH is a key parameter for the local control of the electron
Figure 7.1: SOL, limiter and LH launcher in atokamak cross section (i.e. Tore Supra).
0
0.2
0.4
0.6
0.8
1 00.1
0.20.3
0.4
0
0.01
0.02
0.03
0.04
0.05
Time (s)Normalized space
Par
ticle
den
sity
Figure 7.2: Distribution from the state-space description.
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density, and consequently for the coupling efficiency, which motivates the development of new modeling tools based on experimentalmeasurements. On Tore Supra, the electron density is measured by using a microwave reflectometer that has both a good spatial(r ≈ 1 cm) and temporal resolution (t ≈ 2 msfor the shot considered).
The aim of this work is to develop an identification method fordetermining thesource term, i.e. the number of electrons createdper unit time and volume, when the high frequency heating is switched on. To achieve this, we derive a particle transport model forthe area of non-confined plasma (SOL) and develop an appropriate parametric identification method for distributed systems, based onthe reflectometer measurements. The resulting algorithm then provides an efficient tool for analyzing the coupling associated with theelectron density in the SOL, even if the detailed physical relationships are unknown.
7.2 Simplified Transport ModelThe aim of this section is to present a model that is suitable to determine the particle transport. Thanks to specific hypotheses associatedwith the physical properties of the SOL, the following transport model was established in [23]:
∂n∂t= D⊥(t)
∂2n∂r2− cs(r, t)
2Lc(r, t)n+ S(r, t) (7.1)
whereD⊥ is the cross-field diffusion coefficient (typically∼ 1 m2s−1), cs is the sound speed,Lc is the connecting length along the fluxtube to the flow stagnation point andS = Sl + SLH reflects the particle source induced by the limiterSl and the LH antennaSLH . Thesound speed can be approximated ascs ≈
√(Te + Ti)/mi , whereTe andTi are the electron and ion temperatures, andmi is the ion mass.
The ratioTi/Te is obtained thanks to the experimental measurements described in [24]. Lc is deduced from the safety factorq (whichtypically varies by 10 % in the SOL) asLc = 2πrq.
A Neumann boundary condition∂n(0, t)/∂r = 0 is set close to the center while a Dirichlet onen(L, t) = nL(t), wherenL(t) is givenby the measurements, governs the plasma edge. Note thatr = 0, L denote relative coordinates with respect to the model domain ofvalidity (i.e. r = 0 at the separatrix location). The data set considered in this paper is characterized by repeated LH impulses, whichhighlight the impact of LH antenna.
7.2.1 Problem statementThe problem considered in this paper is to determineS(r, t) in (7.1) from the given signals ofn(r, t). The transport parameters (assumedconstant according to a supposed quasi-steady state behavior - i.e. these parameters vary slowly in comparison with thestate dynamics)D⊥, cs andLC are obtained from existing models and measurements.
Using the subscriptst andx to denote the time and space derivatives, respectively, where x = r/L ∈ [0, 1] is the normalized radius,the class of systems considered can be described as:
nt(x, t) = α nxx(x, t) − γ n(x, t) + S(x, t),nx(0, t) = nx0(t), n(1, t) = nL(t),n(x,0) = nr0(x),
(7.2)
whereα = D⊥/L2 is the diffusion andγ = cs/(2LC) > 0 is the sink term. For the numerical applications, we will take α = 18.54 andγ = 30.54.
7.3 Laboratory work
7.3.1 State-space description [1h30]1. Based on model (7.2), a central discretization
∂2u∂z2≈
ui+1 − 2ui + ui−1
∆z2
and the use of the boundary conditions as
nx(0, t) ≈n1(t) − n0(t)∆x
= nx0(t), nN+1 = nL(t),
derive aN-dimensional state-space model of the density distribution in the standard form
X = AX+ BU, X(0) = X0
whereX = [n1 n2 . . .nN]T , U = [S1 S2 . . .SN, nx0, nL]T , A ∈ RN×N andB = [ I [a 0 . . .0]T [0 . . .0 c]T ] ∈ RN×N+1. For thesimulations, choosenx0(t) = −0.04 andnL(t) = 0.0005.
2. Choose an arbitrary source term with a gaussian distribution
S(x, t) = S(x) = ϑe−(µ−x)2/2σ
a boundary valuenL and a sigmoid initial condition
n(x, 0) =α
1+ e−β(x−γ)
As starting values, you can chooseϑ = 2, µ = .3,σ = .05,α = 3× 10−2, β = −15,γ = .7.
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3. Write aMatlabr .m file to visualize the eigenvalues ofA as well as the gaussian and sigmoid distributions.
4. Design aSimulinkr model to simulate the density dynamics according to your initial condition and source distribution.
5. With the proposed numerical values, you should obtain a distribution like the one presented in Figure 7.2.
7.3.2 Optimal on-line identificationThe estimation problem is formulated as a tracking control problem based on the following considerations [25]:
• the model is the system which dynamics have to be controlled;
• S(t) acts as the controlled input;
• the measurementnmeas(t) is the tracked reference;
• in order to fit the model with the measurements, our aim is to minimize the modeling errornmeas(t)−nmodel(t) over the simulationtime interval;
• this can be achieved thanks to a LQR tracking controller withintegral action.
Note that the state-space dynamics write asX = AX+ S + ω
whereω(t) is a known exogenous input (containing the boundary conditions in our case). Then:
1. write the dynamics of the extended stateXe = [X ǫI ]T , whereǫI is the integrated estimation error (i.e. ˙ǫI = nmeas(t) − X(t)) as
Xe = AeXe + Be1S + Be2Ω
with Ω(t) = [nx0 nL nTmeas]
T ;
2. based on the cost function
I =12
∫ ∞
0
[
XǫI
]T [
0 00 Qr
]
︸ ︷︷ ︸
Q
[
XǫI
]
+ STR S · dt
show that the optimality necessary conditions lead to
S = −R−1BTe1 (PXe + γ)
with
−Q = PAe + ATe P− PBe1R−1BT
e1P
γ =[
PBe1R−1BT
e1 − ATe
]−1PBe2Ω
3. show that the optimal estimation architecture corresponds to the one depicted in Figure 7.3;
4. interpret the relative roles ofGlqr andGlqr,W in the control architecture;
5. using theMatlabr function lqr to solve the ARE equation set the previous architecture and verify that you get your initialsource distribution (using the previous model to generate the measurements).
Figure 7.3: Optimal tracking of the particle source.
7.3.3 Use of the experimental data1. Download the experimental data filemeasurements.mat and implement your algorithm on it.
2. Illustrate the impact of the relative values ofQr andRon the algorithm efficiency.
3. Discuss your results and propose some future research perspectives on this topic (i.e. as applications of the modeling classmaterial).
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Chapter 8
Optimization: Flow Control
The problem of temperature control systems with time transport of mass is studied in this lab with application to a process-controlsystem for Poiseuille flow. To further investigate the phenomenon of energy transmission in a Poiseuille flow, a specially conceivedsystem has been designed to test and validate advanced control strategies on this subject. In this lab, a model identification procedureand an optimal PID have to be developed for the temperature output control of the device under constant mass flow conditions.
8.1 System DescriptionTo further investigate the phenomenon of energy transmission in a Poiseuille flow, a specially conceived system has beendesigned totest and validate advanced control strategies on this subject. Figure 8.1 shows the diagram of the device.
Figure 8.1: Temperature and humidity flow control scheme
This device is composed primarily by a heating column, a tube, a resistance, two fans, a flow meter and temperature sensors. Thepurpose of this experiment is to control the outlet temperature of the tube by controlling the power dissipated on the resistance fordifferent air mass flows (exogenous inputs generated by the fans)through the tube.
8.2 IdentificationIn this section, it is expected to obtain a model of the deviceshown in Figure 8.1 capable of representing the dynamic relation betweenthe power dissipated in the heating resistor (control input) and the output temperature under a constant mass flow. An identificationprocedure has to be conceived in order to acquire the required information for the model construction.
A-1 Discuss which identification strategy could be suitablefor this application. Design the experiments to carry out onthe device inorder to obtain the necessary information.
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A-2 Using the data-log available in the HMI (human machine interface), capture the results of the experiments. Use Matlab or Excelto process the information acquired and perform the identification procedure chosen in the previous step.
A-3 Build the identified the model in Matlab an compare with the results obtained from the device.
8.3 ControlThe flow control device has already a PID programmed in the PLCfor controlling different quantities on the system. The PID parameterscan be configured directly from the HMI which results practical for control implementation in the device.
B-1 Design an optimal PID control for the system using the model obtained in the previous section. Use a LQR approach.
B-2 Test the PID controller using the model in Matlab.
B-3 Implement the optimal PID in the flow control system and compare the results obtained with respect to the simulation performedpreviously.
8.4 ConclusionsComment and conclude.
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