Wideband Four-Way Out-of-Phase Slotline Power Dividers

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1

Abstract—Two new wideband four-way out-of-phase slotline power dividers are proposed in this paper. The half-wavelength slotlines are employed to construct the presented compact power dividers. Based on the proposed power-dividing circuit, a four-way power divider is implemented with compact size and simple structure. To obtain high isolation among the four output ports and good output impedance matching, another four-way out-of-phase slotline power divider with improved isolation performance is designed by introducing an air-bridge resistor and two slotlines with isolation resistors. The simulated and measured results of the proposed power dividers demonstrate reasonable performance of impedance matching, insertion loss, amplitude balancing, and isolation among the output ports.

Index Terms—Wideband, slotline, out-of-phase power divider, compact, isolation

I. INTRODUCTION HE rapid development of RF and microwave industries

requires various of passive or active RF circuits [1-15, 32, 33]. Among various RF circuits, high-power solid-state power amplifier is important in industrial applications [15, 16, 19, 20, 32, 33], which is widely used in wireless/wireline communications such as industrial systems and consumer electronics. As the output power from an individual solid-state device is rather modest at microwave/millimeter-wave frequencies, the need for power combining becomes evident. This has motivated considerable research activities to develop wideband and efficient power-dividing/combining circuits at these frequencies [1, 10-28]. Moreover, the power dividers /combiners with low cost and compact size [8, 9, 12, 21-28] are widely used in antenna arrays, mixers, phase shifters, and so on. Then, they play an important role in many industrial electronic systems. Recently, various multiple-way power dividers have been presented and developed [1, 10-21]. In these designs, several kinds of waveguide-based wideband power dividers, such as ring-cavity power divider [1], rectangular waveguide

Manuscript received July 7, 2013. Accepted for publication August 9, 2013. Copyright (c) 2009 IEEE. Personal use of this material is permitted.

However, permission to use this material for any other purposes must be obtained from the IEEE by sending a request to [email protected].

Kaijun Song, Yuxia Mo, and Yong Fan is with the EHF Key Laboratory of Science, School of Electronic Engineering, University of Electronic Science and Technology of China, Chengdu 611731, China (e-mail: [email protected], [email protected]).

Quan Xue is with the State Key Lab of Millimeter Waves, Department of Electronic Engineering, City University of Hong Kong, Hong Kong.

power dividers [15-17], radial waveguide power dividers [10-13], conical power divider [14], and coaxial waveguide power dividers [18-20] have been widely investigated and used in microwave and millimeter-wave systems because of their low insertion loss, wide bandwidth, and high power capability. However, these waveguide-based dividers may suffer from problems of low isolation between the output ports of the power divider, and have poor impedance matching at the output ports.

In [21-27], several planar power dividers have been studied, which can achieve good impedance matching at all the output ports and high isolation among the output ports. However, only two- or three-way power-dividing function can be implemented due to the structural limitation, which will greatly limit their application in practical systems. A multilayer substrate integrated waveguide (SIW) four-way out-of-phase power divider is presented in [28], which can implement good performances of impedance matching at all ports and isolation among the output ports.

In this paper, the half-wavelength slotline is employed to construct the novel wideband four-way out-of-phase power dividers. Moreover, a compact four-way out-of-phase slotline power divider operating at C- and X-band has been designed. To further improve isolation performance and impedance matching of the output ports, the air-bridge isolation resistor and two slotlines with isolation resistors are applied to the power divider. The equivalent-circuit model and even- and odd-mode analysis method are also involved to develop the power divider with isolation improvement. Finally, the two four-way out-of-phase slotline power dividers have been implemented. The measured results agree with the simulated ones closely. The measured results indicate that the proposed power dividers have several advantages, such as wide bandwidth, low insertion loss, excellent impedance matching at all ports, good amplitude balance at output ports, and

Wideband Four-Way Out-of-Phase Slotline Power Dividers

Kaijun Song, Senior Member, IEEE, Yuxia Mo, Quan Xue, Fellow, IEEE, and Yong Fan, Member,

IEEE

T

Input Output

0

180o

o

0o

Power amplifier array Out-of-phase power combiner

Out-of-phase power divider

180o

180o

180o

0o

0o

Fig. 1 Schematic illustration of the power-combining amplifier



2

reasonable isolation among the output ports (for the power divider with isolation improvement). Fig.1 shows a schematic illustration of the power-combining amplifier using out-of-phase four-way power divider/combiner. It can be seen that the out-of-phase four-way power divider/combiner can be applied to construct the power -combining amplifier after appropriate arrangement.

II. FOUR-WAY OUT-OF-PHASE SLOTLINE POWER DIVIDER A. Structure and Design

Fig. 2(a) shows the proposed half-wavelength slotline four-way out-of-phase power divider without isolation resistor. It is composed of input/output microstrip feedlines and a half-wavelength (λs/2) slotline resonator (namely d+ls≈λs/4), where λs is the guided wavelength of the slotline at the central frequency. The slotline resonator is bent to reduce the size of the presented power divider. The input microstrip feedline (port 1) is placed at the middle of the power divider, which is an open-circuited microstrip line with a length lm≈λm/4 (λm is the guided wavelength of the microstrip line at the center frequency) from the slot to the open end, while the four output ports with identical sizes are arranged symmetrically on the two sides of the input microstrip feedline. The half-wavelength slotline resonator is etched on the bottom layer of the substrate and crossed with the input and output microstrip feedlines near the center of the slotline. It can also be seen that the presented power divider only has five parameters, which will greatly simplify the analysis and optimization of the presented power divider.

When the RF signal is excited at the input microstrip feedline, the RF power will transmit to the center of the slotline through the microstrip-slotline transition, and then be parallelly divided

into two-equal RF signals, where each transmits to the slotline-microstrip transition of the output microstrip lines through the slotline. Finally, the RF power at the slotline-microstrip transition is equally divided into the two output ports located at the two ends of the same microstrip line.

The electric field and current distributions are drawn in the slotline and microstrip line in Fig. 2(b) (To describe them clearly and directly, the slotline is not bent), respectively, which enables us to have a conceptual understanding of the in-phase and out-of-phase coupling behavior. It can be seen from Fig. 2(b) that port 1, port 2, and port 5 are in-phase because their current directions are the same. Similarly, port 3 and port 4 are also in-phase. However, port 2 and port 3 are out-of-phase because of their opposite current directions. Obviously, port 4 and port 5 have a phase difference of 180°. That is to say, the two output ports located at the two ends of the same microstrip line is out-of-phase when RF power is transmitted through the slotline-microstrip transition from slotline to the microstrip line. Then, the total four-way out-of-phase power divider can be viewed as the combination of a two-way microstrip/slotline in-phase power divider and two two-way slotline/microstrip out-of-phase power dividers. It can be seen that the four output signals are of equal power levels. Apparently, the two ports located at the same microstrip line are out-of-phase, while the two ports, which are located at the same side of slotline and also symmetric with respect to the input microstrip feedline, are in phase.

Fig. 3 shows the transmission-line equivalent circuit of the proposed four-way out-of-phase power divider, where Z0 and Zs are the microstrip-line and slotline characteristic impedances, respectively. θm=2πlm/λm, θs=2πls/λs, and θd=2πd/λs denote the microstrip-line electrical length of the extend portion, the slotline electrical length of the extend portion, and the slotline electrical length between two adjacent microstrip lines at coupling point, respectively. The microstrip-slotline transition can be expressed as an equivalent transformer [29]. The transformer turn ratio n represents the coupling magnitude between the microstrip line and slotline. After making a number of approximations in the analysis, a closed-form expression for transformer turn ratio n is given as [29]

Port 1

Port 3

dZs,θ

dZs,θ

Z0 Z0

MLSLML

Port 2Z0

Port 5Z0

Port 4Z0Z0, mθ

n:1

1:n

1:n

sZs,θ

sZs,θ

Fig. 3 Equivalent circuit of the proposed power divider

lsW s

W m

Port 1

Port 2

Port 3Port 4

Port 5

lm

d

Top layer Bottom layer

(a)

E-field currentPort 1

Port 2

Port 3Port 4

Port 5

(b)

Fig. 2 The proposed power divider (a) structure (b) phase analysis of theoutput ports



3

( ) ( )22

00 2/2/

emes

semmes

kkWkJWkJn

+=

⎥⎦

⎤⎢⎣

⎡+

+−

⋅hkkhkk

kkhkkhkk

kk es

r

rem

1211

12

1112

22

sincossincosεε

(1) where J0(·) is the zeroth-order Bessel function and

remresremesr kkkkk εεεε −−=−−= 0222

01

102 −+= remreskk εε

reses kk ε0= , remem kk ε0=

Here, εrem and εres are the effective dielectric constants of the microstrip line and the slotline, respectively. εres can be determined according to [29].

According to the equivalent circuit (shown in Fig. 3) of the proposed power divider, the input impedance matching and transmission performance can be analyzed and synthesized. The reflection and transmission coefficients of the four-way slotline out-of-phase power divider can be calculated according to its equivalent circuit. Therefore, its frequency response can also be analyzed and optimized. Finally, the optimized dimensions of the four-way slotline out-of-phase power divider can be obtained under the desired frequency response. Fig. 4 shows the comparison of the calculated frequency response from its equivalent circuit with the simulated one by using an EM simulator (IE3D). It can be seen that they agree well with each other, which validates the validity of its equivalent circuit. Moreover, the operation frequency range perhaps can cover the UWB band (3.1 GHz - 10.6 GHz) after increasing the coupling of the microstrip-slotline transition. B. experimental results

According to above analysis method, a compact wideband four-way out-of-phase slotline power divider is designed by using the equivalent circuit and the EM simulation about the

proposed power divider is performed by using a commercially tool (IE3D). The power divider is fabricated on a substrate Taconic TLE-95-0150 with a dielectric constant rε of 2.95, a thickness of 0.38 mm, and loss tangent of 0.0012. Fig. 5 shows the fabricated four-way out-of-phase slotline power divider. As mentioned above, to further reduce size, the slotline etched on the bottom layer of the substrate is bent. Although a low dielectric constant has been used, the fabricated four-way power divider is very compact with a size of 10 mm× 8 mm. The dimensions for the fabricated power divider are found to be (as illustrated in Fig.2): d=1.36 mm, sL =9 mm, mL = 6.5

mm, sW =0.2 mm, Wm= 0.96 mm. The simulated and measured S parameters are shown in Fig.

6 to Fig. 8. It can be seen that the measured results agree with the simulated ones closely over the entire design frequency range. The measured 10-dB return loss bandwidth is from 4 to 9.6 GHz, while 15-dB return loss bandwidth is found to be approximately 4 GHz (from 4.6 to 8.6 GHz). The measured insertion losses S21, S31, S41, and S51 are within 6.5±0.5 dB in a wide frequency range from 4 to 9 GHz. The amplitude imbalance is due to the assembly and fabrication errors.

Fig. 7 shows the simulated and measured isolation among the output ports, while Fig. 8 shows the return loss of the output ports. Obviously, it can’t simultaneously obtain good

(a) (b)

Fig. 5 Photograph of the fabricated power divider: (a) top view, (b) bottom view.

3 4 5 6 7 8 9 10 11 12-50

-40

-30

-20

-10

0

S11(measured) S21(measured) S31(measured) S41(measured) S51(measured)

S11(simulated) S21(simulated) S31(simulated) S41(simulated) S51(simulated)

Mag

nitu

de (d

B)

Frequency (GHz)

Fig. 6 Simulated and measured S parameters of the fabricated power divider without isolation resistors

3 4 5 6 7 8 9 10 11 12

-30

-20

-10

0

S21(calculated) S21(simulated) S11(calculated) S11(simulated)

Mag

nitu

de (d

B)

Frequency (GHz)

Fig. 4 Calculated and simulated frequency response of the proposed power divider ( ml =6.5 mm, sl =9 mm, d=1.36 mm, sW =0.2 mm, mW =0.96 mm)



4

impedance matching at all ports and good isolation among the output ports since the proposed power divider is a lossless structure [28, 30]. The isolation (S32 and S54) between the output ports located at the same microstrip line is less than 4 dB, while the other isolation (S42, S52, S43, and S53) between the output ports located at the different output microstrip lines is greater than 10 dB from 3 to 10.2 GHz, as shown in Fig. 7. The measured return losses of the output ports are around 10 dB from 3 to 11 GHz, as shown in Fig. 8.

Fig. 9 shows the measured phase difference between different output ports. The phase difference between S∠ 21 and

S∠ 31 is about 180°±2° over a wide frequency range from 3.3 to 11.8 GHz, while that between S∠ 31 and S∠ 41 is about 0°±2° from 3 to 11.7 GHz. It can be noted that the port 2 and port 3, which are located at the same microstrip line, are out-of-phase, while the port 3 and port 4, which are symmetric with respect to the input microstrip feedline, are in-phase.

III. FOUR-WAY SLOTLINE OUT-OF-PHASE POWER DIVIDER WITH IMPROVED ISOLATION PERFORMANCE

A. Structure and Analysis As mentioned above, the above proposed power divider

suffers from poor isolation performance between the output ports located at the same microstrip line. To improve the isolation performance, the isolation circuits can be considered. Fig. 10 shows the configuration of the proposed four-way out-of-phase power divider with improved isolation performance. A new air-bridge isolation resistor R is across the input microstrip lines and connected with the two central points of the two output microstrip lines, which can be applied to improve the isolation performance between the output ports located at the different output microstrip lines.

To further improve the isolation performance (S32 and S54) between the out-of-phase output ports (such as port 2 and 3), the isolation resistor embedded in slotline under the common ground can be applied [25, 31]. As shown in Fig. 10, the slotline (with a length lD≈λs/2) with isolation resistor is employed to improve the isolation between the out-of-phase output ports located at the same output microstrip line. The output microstrip line is bent in a ring shape and a slotline with isolation resistor R0 is etched on the bottom layer, as shown in Fig. 10. The length lr/2 is approximately quarter wavelength at the central operating frequency [25] and the isolation resistor

3 4 5 6 7 8 9 10 11 12-14

-12

-10

-8

-6

-4

-2

S32(measured) S42(measured) S52(measured) S43(measured) S53(measured) S54(measured)

S32(simulated) S42(simulated) S52(simulated) S43(simulated) S53(simulated) S54(simulated)

Isol

atio

n (d

B)

Frequency (GHz)

Fig. 7 Simulated and measured isolation among the output ports

3 4 5 6 7 8 9 10 11 12-14

-12

-10

-8

-6

-4

-2

0

S22(measured) S33(measured) S44(measured) S55(measured)

S22(simulated) S33(simulated) S44(simulated) S55(simulated)

Out

put R

etur

n Lo

ss (d

B)

Frequency (GHz)

Fig. 8 Simulated and measured output return loss

3 4 5 6 7 8 9 10 11 12170

175

180

185

190

S31- S41(measured)S21- S31(measured)

Frequency (GHz)

Pha

se V

aria

tion

(deg

ree)

-10

-5

0

5

10

Phas

e Va

riatio

n (d

egre

e)

Fig. 9 Measured phase difference between the output ports

Port 1

Port 2

Port 3Port 4

Port 5

Top layer Bottom layer

R0

R0R

Bottom view

lr

lD

W D

lg

g

d0

Fig. 10 Configuration of the four-way power divider with isolation



5

R0 is located at the center of the slotline. B. Microstrip/slotline in-phase power dividing

Apparently, the even- and odd-mode method can be employed to analyze the presented power divider. Fig. 11(a) shows the even-mode equivalent circuit of the presented power divider for the microstrip/slotline in-phase power dividing. In this case, no current flows through the air-bridge isolation resistor R, since two signals with the same magnitude and phase are applied to each output port. Hence, the circuit element R can be omitted. In the bisected circuit, since the input port impedance (2Z0) is equal to the output port impedance, a simple impedance matching circuit (see Fig. 11(a)) is used. Therefore, we can synthetize and optimize the microstrip-line electrical length θm of the extend portion, the slotline electrical length θs of the extend portion, the slotline electrical length θd between two adjacent microstrip lines at coupling point, the slotline characteristic impedance Zs, and the microstrip-slotline equivalent transformer turn ratio n to obtain reasonable levels of insertion loss, input/output return losses, and isolation between the output ports simultaneously.

Fig. 11(b) shows the odd mode circuit for the microstrip /slotline in-phase power dividing. In this case, two signals with the same magnitude and 180° out of phase are applied to two output ports, and there is a voltage null along the symmetry plane, where the power dividing circuit is symmetry with the input microstrip feedline. To obtain good input/output impedance matching and good isolation between the output ports, the following relationship has to be satisfied

RZZRZ

nj

ZnZnZZj s

s

dsm

dms

0

022

0

20

24cot

)tancot2(/tancot2 −

=−−

+ θθθ

θθ

(2)

Then,

04ZR = (3) Equation (2) provides a simple guideline in selection of the

isolation resistor R. C. Slotline/microstrip out-of-phase power dividing

The odd-mode equivalent circuit for the slotline/microstrip out-of-phase power dividing circuit of the presented power divider is shown in Fig. 12(a). In this case, two signals with the same magnitude and phase are applied to each output port, since no current flows through the common ground plane. Then, the slotline (including the isolation resistors) etched on the common ground plane can be omitted, as shown in Fig. 12(b). According to the odd-mode impedance matching circuit shown in Fig. 12(b), the condition for good input/output impedance matching is given by

ss

s

ZZj

Zn 2cot

0

2

=−θ

(4)

Then, the slotline characteristic impedance Zs can be determined by

02

2 Zn

Zs = (5)

In addition, according to equation (4), the following expression can also be derived as

0cot =sθ Namely,

2ππθ ±= ks , ),2,1,0( ⋅⋅⋅±±=k (6) According to equation (6), the slotline length ls of the extend

portion is about quarter-wavelength, which is compatible with the above analysis (see section II) when d is very short.

ML

1:nZ0

Z0

Output

Output

SL

Input

sZs,θZs

(a)

Input

ML

1:n

Zs/2 Z0

Output

sZs/2,θ

SL

(b) Fig. 12 Odd mode equivalent circuit of the presented power divider for slotline/microstrip out-of-phase power dividing circuit (a) odd-mode analysis, (b) odd-mode impedance matching circuit.

Input

2Z0

MLSLML

2Z0, mθ

:1 dZs,θ 1:n

sZs,θ R/2 2Z0

Output

O.C.

(a)

Input

0

MLSLML

2Z0, mθ

n:1 dZs,θ 1:n

sZs,θ R/2 2Z0

Output

(b) Fig. 11 Equivalent circuit of the presented power divider for microstrip/slotline in-phase power dividing: (a) even-mode circuit, (b) odd-mode circuit.



6

Equation (6) provides a simple guideline in selection of ls. Fig. 13(a) shows the even-mode equivalent circuit of the

presented power divider for the slotline/microstrip out-of-phase power dividing circuit, where two signals with the same magnitude and 180° out of phase are applied to each output port. In this case, no current flows through the input slotline. The isolation resistor 0R is doubled ( 02R ) and the

slot gap capacitor gC is halved ( 2/gC ). Fig. 13(b) shows the

even-mode impedance matching circuit for a bisection of the slotline/microstrip out-of-phase power dividing circuit. Hence, the input impedance ineZ can be given by

grine CRj

Rj

ZZ0

00

12

tan ωθ ++= (7)

where ω is the operating angular frequency and θr=πlr/λm. To obtain good impedance matching between the input impedance

ineZ and the output impedance 0Z , the following condition has to be satisfied:

0 00

0

2tan 1ine

r g

Z RZ Zj jR Cθ ω

= = ++

(8)

Namely,

020

0

)(12 Z

CRR

g

=+ ω

(9)

20

200

)(12

tan g

g

r CRCRZ

ωω

θ +

−= (10)

According to equation (8) and (9), the following expression

can be derived as

)cot1(2

00 r

ZR θ+= (11)

It can be seen that 2/00 ZR = when lr is equal to half wavelength, which provides a guideline in selection of lr. Then, good impedance matching and isolation performance between the output ports can be achieved for the slotline/microstrip out-of-phase power dividing circuit when the above equation (8) to (11) are satisfied. In addition, the parameters of the slotline/microstrip out-of-phase power dividing circuit can also be obtained according the above equations. D. Simulated and measured results of the fabricated power divider with isolation resistors

A wideband four-way out-of-phase slotline power divider with improved isolation performance is designed according to the above design method. It is difficult to simulate the presented power divider with isolation resistors by using an IE3D, which is suitable for the microwave circuit without resistor. So, the commercial software CST MICROWAVE STUDIO is used to simulate and optimize the proposed power divider. The substrate used in the proposed power divider with improved isolation is the same as that in the above power divider without isolation resistors. The fabricated four-way

(a)

(b)

Fig. 14 Photograph of the fabricated power divider with isolation resistors: (a) top view, (b) bottom view.

ML

Cg

Output

Output

R0

rZ0,θ

rZ0,θ

(a)

Cg/2

2R0rZ0,θ

Z0

OutputInput

O.C. Zine

(b) Fig. 13 Even mode equivalent circuit of the presented power divider for slotline/microstrip out-of-phase power dividing circuit (a) even-mode

analysis, (b) even-mode impedance matching circuit.



7

out-of-phase slotline power divider with isolation resistors is shown in Fig. 14, where port 1 indicates input port and ports 2-5 indicate the output ports. Similarly, the fabricated power divider is compact with a size of 14.5 mm×15 mm. The dimensions for the fabricated power divider are found to be (as illustrated in Fig. 2 and Fig. 10): d=1.8 mm, 0d =0.3

mm, g =0.2 mm, sL =8.4 mm, mL =7.3 mm, rL =14.7

mm, gL =1 mm, DL =15 mm, sW =0.2 mm, mW =0.96

mm, DW =0.6 mm. The values for the isolation resistors R and R0 are 200 Ω and 25 Ω, respectively.

The simulated and measured results of the fabricated power divider with isolation resistors are shown in Fig. 15, Fig. 16, and Fig. 17. It can be seen that the measured results show a reasonable agreement with the simulated ones over the operating frequency range. The measured 10-dB return loss bandwidth is found to be approximately 6.4 GHz (from 3.7 to 10.1 GHz) (see Fig. 15), while the simulated 10-dB return loss bandwidth is from 3.8 to 9.6 GHz. The difference between the simulated and measured results is most likely attributed to the fabrication and assembly errors. The measured insertion losses S21, S31, S41, and S51 are within 6.8±0.4 dB in a wide frequency range from 3.9 to 9 GHz, as shown in Fig. 15.

Fig. 16(a) shows the simulated and measured output return losses, while Fig. 16(b) shows the simulated and measured isolation among the output ports. The measured output return losses S22, S33, S44, and S55 are greater than 11 dB in the frequency range from 3 to 10.4 GHz, while the simulated ones are greater than 12.7 dB from 3 to 10.1 GHz, as shown in Fig. 16(a). Meanwhile, the simulated isolation between the output ports is greater than 10.6 dB from 3.7 to 11 GHz, while the simulated isolation S42 and S53 are greater than 15 dB from 3 to 9 GHz. Moreover, the measured isolation between the output ports is greater than 10 dB from 3 to 10.8 GHz, while the measured isolation S42 and S53 are greater than 13 dB from 3.2 to 8 GHz. Compared with the above power divider without

isolation resistors, the designed power divider with isolation resistors has improved isolation performance. The isolation S32 and S54 have been greatly improved from less than 4 dB (see

3 4 5 6 7 8 9 10 11-25

-20

-15

-10

-5

0

S22(measured) S33(measured) S44(measured) S55(measured)

S22(simulated) S33(simulated) S44(simulated) S55(simulated)

Out

put R

etur

n Lo

ss (d

B)

Frequency (GHz)

(a)

3 4 5 6 7 8 9 10 11-20

-15

-10

-5

0

S32(measured) S42(measured) S52(measured) S43(measured) S53(measured) S54(measured)

S32(simulated) S42(simulated) S52(simulated) S43(simulated) S53(simulated) S54(simulated)

Isol

atio

n (d

B)

Frequency (GHz)

(b) Fig. 16 Simulated and measured results of the proposed power divider with isolation resistors: (a) output return losses, (b) isolation among the output ports

3 4 5 6 7 8 9 10 11170

175

180

185

190

S31- S41(measured)

S21- S31(measured)

Frequency (GHz)

Phas

e Va

riatio

n (d

egre

e)

-10

-5

0

5

10Ph

ase

Varia

tion

(deg

ree)

Fig. 17 Measured phase difference between the output ports

3 4 5 6 7 8 9 10 11-40

-30

-20

-10

0

S11(measured) S21(measured) S31(measured) S41(measured) S51(measured)

S11(simulated) S21(simulated) S31(simulated) S41(simulated) S51(simulated)

Mag

nitu

de (d

B)

Frequency (GHz)Fig. 15 Simulated and measured S parameters of the fabricated power divider with isolation resistors



8

Fig. 7) to greater than 10 dB (see Fig. 16(b)) over the designed frequency range. Since the isolation resistors and good impedance matching circuit have been employed, good input/output impedance matching and reasonable isolation between the output ports can be achieved simultaneously.

The measured phase difference between different output ports is shown in Fig. 17. The phase difference between ∠S31 and ∠S41 is about 0°±2° over a wide frequency range from 3 to 10.5 GHz, while that between∠S21 and ∠S31 is about 180°±2° from 3.1 to 10.9 GHz.

IV. CONCLUSION Wideband four-way out-of-phase slotline power dividers

have been presented. The appropriate isolation circuits have been developed to improve its isolation performance and impedance matching of the output ports, while the equivalent-circuit method has been used to analyze the presented out-of-phase power dividers. The simulated and measured results of two four-way out-of phase slotline power dividers (without and with the isolation resistors) have shown acceptable input impedance matching, low insertion loss, and good amplitude balance. Moreover, the out-of-phase slotline power divider with isolation resistors has also demonstrated reasonable isolation among the output ports. It can be seen that the presented slotline power dividers have the advantages of compact size, wide operating bandwidth, good input/output impedance matching, low insertion loss, good amplitude balance at output ports, and reasonable isolation among the output ports, which makes it very competitive in the practical applications.

ACKNOWLEDGEMENT The work for this grant was supported by National Natural

Science Foundation of China (Grant No: 61271026), by the Program for New Century Excellent Talents in University (Grant No: NCET-11-0066), and by the Research Fund of Shanghai Academy of Spaceflight Technology (SAST201243).

REFERENCES [1] K. Song and Q. Xue, “Ultra-Wideband (UWB) Ring-Cavity

Multiple-Way Parallel Power Divider,” IEEE Trans. Ind. Electron., 2013, vol. 60, no. 10, pp. 4737-4745.

[2] X.-Y. Zhang, C.-H. Chan, Q. Xue, B.-J. Hu, “RF Tunable Bandstop Filters With Constant Bandwidth Based on a Doublet Configuration,” IEEE Trans. Ind. Electron., 2012, vol. 59, no. 2, pp. 1257-1265.

[3] E. DiGiampaolo, F. Martinelli, “A Passive UHF-RFID System for the Localization of an Indoor Autonomous Vehicle,” IEEE Trans. Ind. Electron., Oct. 2012, vol. 59, no. 10, pp. 3961-3970.

[4] Y. Jin, F.F. Dai, “Impact of Transceiver RFIC Impairments on MIMO System Performance,” IEEE Trans. Ind. Electron., Jan. 2012, vol. 59, no. 1, pp. 538-549.

[5] J. H. Deng, W. S. Chan, B.-Z. Wang, S.Y. Zheng, and K. F. Man, “A RFID multi-criteria coarse and fine space tag antenna design,” IEEE Trans. Ind. Electron., Jun. 2011, vol. 58, no. 6, pp. 2522–2530.

[6] J.-W. Lee, D. H. Thai Vo, Q.-H. Huynh, S. H. Hong, “A Fully Integrated HF-Band Passive RFID Tag IC Using 0.18-um CMOS Technology for Low-Cost Security Applications,” IEEE Trans. Ind. Electron., June 2011, vol. 58, no. 6, pp. 2531-2540.

[7] T. Thacker, D. Boroyevich, R. Burgos, Fei Wang, "Phase-Locked Loop Noise Reduction via Phase Detector Implementation for Single-Phase Systems,” IEEE Trans. Ind. Electron., June 2011, vol. 58, no. 6, pp. 2482-2490.

[8] H. L. Zhang, B. J. Hu, X.-Y. Zhang, “Compact Equal and Unequal Dual-Frequency Power Dividers Based on Composite Right/Left-Handed Transmission Lines,” IEEE Trans. Ind. Electron., 2012, vol. 59, no. 9, pp. 3464-3472.

[9] J.-L. Li and B.-Z. Wang, “Novel design of wilkinson power dividers with arbitrary power division ratios,” IEEE Trans. Ind. Electron., 2011, vol. 58, no. 6, pp. 2541 –2546.

[10] Y.-P. Hong, D. F. Kimball, P. M. Asbeck, J.-G. Yook, L. E. Larson, “Single-Ended and Differential Radial Power Combiners Implemented With a Compact Broadband Probe,” IEEE Trans. Microw. Theory Tech., 2010, vol. 58, no. 6, pp. 1565-1572.

[11] K. Song, Y. Fan, Z. He, “Broadband radial waveguide spatial combiner,” IEEE Microw. Wirel. Compon. Lett., 2008, vol. 18, no. 2, pp. 73-75.

[12] K. Song, Y. Fan, and Y. Zhang, “Eight-Way Substrate Integrated Waveguide Power Divider with Low Insertion Loss,” IEEE Trans. Microw. Theory Tech., 2008, vol. 56, no. 6, pp. 1473-1477.

[13] A.E. Fathy, S.-W. Lee, and D. Kalokitis, “A simplified design approach for radial power combiners”, IEEE Trans. Microw. Theory Tech., 2006, vol. 54, no. 1, pp.247-255.

[14] D. I. L. de Villiers, P. W. van der Walt, and P. Meyer, “Design of a ten-way conical transmission Line power combiner,” IEEE Trans. Microw. Theory Tech., 2007, vol. 55, no. 2, pp. 302-308.

[15] X. Jiang, S. C. Ortiz, and A. Mortazawi, “A Ka-band power amplifier based on the traveling-wave power-dividing/combining slotted-waveguide circuit,” IEEE Trans. Microw. Theory Tech., 2004, vol. 52, no. 2, pp. 633–639.

[16] N.-S. Cheng, P. Jia, D. B. Rensch, and R. A. York, “A 120-W X-band spatially combined solid-state amplifier,” IEEE Trans. Microw. Theory Tech., 1999, vol.47, no. 12, pp. 2557–2561.

[17] J.P. Becker, and A.M. Oudghiri, “A planar probe double ladder waveguide power divider”, IEEE Microw. Wireless Compon. Lett., 2005, vol. 15, no. 3, pp.168-170.

[18] K. Song, Q. Xue, “Planar Probe Coaxial-Waveguide Power Combiner/Divider,” IEEE Trans. Microw. Theory Tech., 2009, vol. 57, no. 11, pp. 2761-2767.

[19] P. C. Jia, L. Y. Chen, A. Alexanian, etc, “Broad-Band High-Power Amplifier Using Spatial Power-Combining Technique,” IEEE Trans. Mircow. Theory Tech., 2003, vol. 51, no. 12, pp.2469–2475

[20] P. C. Jia, L. Y. Chen, A. Alexanian, and R. A. York, “Multioctave Spatial Power Combining in Oversized Coaxial Waveguide”, IEEE Trans. Microw. Theory Tech., 2002, vol. 50, no. 5, pp.1355-1360

[21] K. Song, Q. Xue, “Novel Ultra-Wideband (UWB) Multilayer Slotline Power Divider with Bandpass Response,” IEEE Microw. Wireless Compon. Lett., 2010, vol. 20, no. 1, pp. 13-15.

[22] A.M. Abbosh, “Design of Ultra-Wideband Three-Way Arbitrary Power Dividers,” IEEE Trans. Microw. Theory Tech., 2008, vol. 56, no. 1, pp. 194-201.

[23] S. W. Wong, L. Zhu, “Ultra-Wideband Power Divider With Good In-Band Splitting and Isolation Performances,” IEEE Microw. Wireless Compon. Lett., 2008, 18, no. 8, pp. 518-520.

[24] K. Song, Q. Xue, “Ultra-wideband out-of-phase power divider using multilayer microstrip-slotline coupling structure,” Microwave and Optical Technology Letters, 2010, vol. 52, no. 7, pp. 1591-1594.

[25] T. Yang, J.-X. Chen, Q. Xue, “Three-way out-of-phase power divider,” Electron. Lett., 2008, vol. 44, no. 7, pp. 482-483.

[26] M. E. Bialkowski and A. M. Abbosh, “Design of a Compact UWB Out-of-Phase Power Divider,” IEEE Microw. Wireless Compon. Lett., 2007, vol. 17, no. 4, pp. 289-291.

[27] L. Chiu, T. Yum, Q. Xue, and C. Chan, “A wideband compact parallel strip 180 Wilkinson power divider for push-pull circuitries,” IEEE Microw. Wireless Compon. Lett., 2006, vol. 16, no. 1, pp. 49–51.

[28] D.-S. Eom, J. Byun, H.-Y. Lee, “Multilayer Substrate Integrated Waveguide Four-Way Out-of-Phase Power Divider,” IEEE Trans. Microw. Theory Tech., 2009, vol. 57, no. 12, pp. 3469-3476.

[29] K. C. Gupta, R. Garg, I. Bahl, and P. Bhartia, Microstrip Lines and Slotlines, 2nd ed. Norwood, MA: Artech House, 1996.

[30] D. M. Pozar, Microwave Engineering, 3rd ed. New York: Wiley, 2005.



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[31] J.-X. Chen, C.H.K. Chin, K.W. Lau, and Q. Xue, “180° out-of-phase power divider based on double-sided parallel striplines,” Electron. Lett., 2006, vol. 42, no. 21, pp. 1229-1230.

[32] A. Al Tanany, A. Sayed, G. Boeck, “Design of Class F-1 Power Amplifier Using GaN pHEMT for Industrial Applications,” German Microwave Conference, 2009, vol. 1, pp. 1-4.

[33] J. S. Przybyla, P. Bromley, R. Bowness, J. Clare, M. Maunsell, D. Briscoe, “Development of a High Power RF Amplifier Using Industrial Power Electronics,” 12th International Power Electronics and Motion Control Conference, 2006. EPE-PEMC 2006, vol. 1, pp. 2156-2161.

Kaijun Song (M’09-SM’12) received the M.S. degree in radio physics and the Ph.D. degree in electromagnetic field and microwave technology from University of Electronic Science and Technology of China (UESTC), Chengdu, China, in 2005 and 2007, respectively.

Since 2007, he has been with EHF Key Laboratory of Science, UESTC, where he is currently a Professor. From 2007 to 2008, he was a postdoctoral research fellow with Montana Tech of the University of Montana, Butte, USA. He was a

research fellow with State Key Laboratory of Millimeter Waves of China, City University of Hong Kong from 2008 to 2010 and a senior visiting scholar in November 2012. He has published more than 80 internationally refereed journal papers. His current research fields include microwave and millimeter-wave/THz power-combining technology; UWB circuits; and microwave remote sensing technologies. He is the Reviewer of tens of international journals, including IEEE Transactions and Letters.

Yuxia Mo graduated in electromagnetic field and microwave technology from the University of Electronic Science and Technology of China (UESTC), Chengdu, China, in 2011 and is now pursing her M.S. Degree. Her research interests include millimeter-wave/THz power-combining technology and microwave /millimeter-wave devices, circuits and systems.

Yong Fan (M’05) received the B.E. degree from Nanjing University of Science and Technology, Nanjing, Jiangsu, China, in 1985 and the M.S. degree from University of Electronic Science and Technology of China, Chengdu, Sichuan, China, in 1992.

He is now with the School of Electronic Engineering, University of Electronic Science and Technology of China, where he is currently a full Professor. His current research interests include electromagnetic theory, millimeter-wave technology, communication and system. He has authored and coauthored over 130

papers. Mr. Fan is a senior member of the Chinese Institute of Electronics. He

received the first award of science and technology of national industry, the second award of science and technology progress of ministry of electronic industry, the third award of science and technology progress of ministry of information industry, and the third award of science and technology progress of Sichuan province (twice).

Quan Xue (M’02-SM’04-F’11) received the B.S., M.S., and Ph.D. degrees in electronic engineering from University of Electronic Science and Technology of China, Chengdu, China, in 1988, 1990, and 1993, respectively.

Since 1999, he has been with City University of Hong Kong, Hong Kong, where he is currently a Professor, the Director of the Information and Communication Research Center, the Deputy Director of the State Key Laboratory (Hong Kong) of

Millimeter Waves of China, and the Assistant Head of the Department of Electronic Engineering. He has authored or coauthored over 180 internationally refereed journal papers and over 70 international conference papers. He is currently an Editor of International Journal of Antennas and Propagation. His current research interests include microwave passive components, active components, antenna, MMICs, radio frequency identification and RFICs, etc. Dr. Xue is currently an Associate Editor of IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES and IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS.

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