Voltage-driven class E amplifier and applications

T. Brabetz and V.F. Fusco

Abstract: A voltage-driven class E power amplifier topology is presented. The operating principleof the circuit is explained, and measurement results for a MIC implementation using an OMMICED02AH 6� 50mm pHEMT as the active device are given. At the nominal operating frequency of870MHz, the MIC achieves 18dB maximum gain, produces 18dBm output power, and amaximum power added efficiency of 93%.

1 Introduction

The class E amplifier is a highly nonlinear highly efficientswitching-class amplifier introduced by N.O. and A.D.Sokal in 1975 [1]. Being a switching amplifier, the class Eamplifier is in theory capable of reaching 100% efficiencyfor maximum output power. In contrast, the class C poweramplifier, also capable of in theory 100% efficiency, canonly achieve such high efficiency levels for low outputpower settings [2]. This makes the class E amplifierattractive for mobile power amplification applications, sinceincreased efficiency at high output power levels meanslonger battery lifetime and less heat dissipation.

Another reason for choosing a switching class amplifier isto make implementation of output power level controleasier, for example with a class E amplifier, power levelcontrol can in principle be achieved by controlling the biasvoltage [3], which has the added advantage that efficiencylevels do not suffer as much as they do for conventionallinear amplifiers when operated in power back-off mode.

2 Working principle

2.1 Conventional class E designFigure 1 shows the conventional current driven class Ecircuit schematic as suggested by Cripps [4]. The activedevice is represented by the switch, its switching actionstimulates an oscillation in the moderate-Q series resonantcircuit LS, CS tuned close to the RF, thus forcing asinusoidal current through the load resistor RL. In parallelwith the device is a shunt capacitance Cp, DC bias is appliedthrough an RF choke of sufficient size to sustain constantcurrent throughout an entire RF cycle. The opening andclosing of the switch is controlled by the RF input signal’szero crossings, and is therefore to a first approximationindependent of the input signal amplitude.

The task of the shunt capacitor Cp (Fig. 1) is to act as atemporary energy store, to allow the series resonant circuitLS, CS to sustain a sinusoidal current while the switch isopen. The sinusoidal output current, iy is either flowing

through the switch, isw while it is closed, or into thecapacitor, iC while the switch is open. If the switch timingand the size of Cp have been chosen correctly, then theshunt capacitor Cp is completely discharged during thenegative half-swing of the output current, and the switchingis taking place while there is no voltage present over theoutput terminals of the active device. Hence powerdissipation is reduced to zero. In a real class E amplifierthe switch timing is set via the gate bias.

In contrast, Sokal’s original design [1] uses the samecircuit as given in Fig. 1, but the LS, CS circuit is used toshape the flanks of the switching waveform rather than theoutput signal waveform itself. Hence, Sokal’s seriesresonant circuit is tuned to a frequency significantlydifferent to that of the RF signal. Since Sokal’s approachrelies on the switch being fast with respect to the outputsignal, its use is restricted to lower frequencies than thoseobtained if operated according to the principles suggestedby Cripps.

2.2 Problems with the conventional designThe class E power amplifier as described in Section 2.1 iswell documented in the literature. However, in spite of itspotential for 100% efficiency, uptake by industry has beenslow so far. This is believed to be the result of a number offactors that will be further examined in this Section.

As mentioned before, the class E power amplifier isinherently nonlinear, since the input waveform controlsonly the timing of the switch by means of its zerotransitions. The switch then stimulates a series circuit toresonate, and consequently the amplitude of the outputsignal to a first approximation does not depend on theamplitude of the input signal. This property prevents theclass E amplifier from being used for systems requiringamplitude modulation. However, since most modern digitalmodulation schemes are phase modulated, this restrictionbecomes irrelevant for a number of state-of-the-art com-munication system applications.

Most significantly, the conventional class E poweramplifier suffers from a major technology problem. Thisoccurs owing to the fact that the conventional design iscurrent based, i.e. current waveforms control circuitbehaviour. Therefore, the conventional class E poweramplifier is inherently sensitive to active device parasiticresistances. The reason for this is that since the currentsflowing through the active device are relatively high, evensmall parasitic resistances cause voltage drops that are largeenough to interfere with the ideal operation of the circuit.

The authors are with the Institute of Electronics, Communications andInformation Technologies, Queen’s University Belfast, Northern Ireland SciencePark, Queen’s Road, Queen’s Island, Belfast, BT3 9DT, UK

E-mail: T.Brabetz@ecit.qub.ac.uk

r IEE, 2005

IEE Proceedings online no. 20045159

doi:10.1049/ip-map:20045159

Paper first received 17th November 2004 and in revised form 10th February2005

IEE Proc.-Microw. Antennas Propag., Vol. 152, No. 5, October 2005 373

As a consequence, the majority of all class E microwavedesigns known to the authors have used HBT’s as the activedevice, since their collector resistances are about a factor of10 smaller than the drain-source resistance of a typical100mm gate width pHEMT. For example, the collectorresistance of a 2� 10mm2 HBT, capable of providingtypically 35mA collector current for about 1.5V collectorvoltage, is about 1.7O [5], while the drain resistance of acomparable pHEMT with 100mm gate width, typicallyproviding about 40mA current for 2V drain voltage [5], isin the range 10–20O. Only for relatively large devices withgate widths in excess of 1mm does the drain resistance ofpHEMTs drop to less than 2O [5].

Furthermore the on-conductance of a pHEMT drainvaries with the gate width [5]. The only pHEMT-based classE designs known to the authors feature relatively largedevices, with total gate widths in the range 1mm [6] to12mm [3]. For these devices the quoted performance isgood (e.g. 90% power added efficiency at 3.25GHz [6]), butyield, reliability, and compliance with normal GaAsfoundry parameters for such large devices have not beenstated.

2.3 Voltage-based class E designTo reduce the impact of active device channel resistance onclass E power amplifier behaviour, we decided to build aclass E amplifier that is controlled by voltage waveformsinstead of current waveforms. A voltage-driven circuit wassuggested by Raab in 1977 [7], but its detailed propertieswere not investigated, and moreover it contained atransformer, which is difficult to build at microwavefrequencies. In addition, we have been unable to find anyreferences to further work done on this type of circuit. Thusthe purpose of this paper is to produce to our knowledge forthe first time practical investigations of the characteristics ofsuch a topology.

The authors therefore suggest a simplified version of thatgiven in [7], the topology shown in Fig. 2. Here, the shunt

capacitor Cp in Fig. 1 has been replaced by a series inductorLp. The series resonant circuit LS, CS, (Fig. 1) has beenreplaced with a parallel circuit. This parallel circuit nowforces a sinusoidal voltage across RL and the switch. If theswitch is open, i.e. isw¼ 0, then this sinusoidal voltage buildsup over the switch, Vsw, since VL ¼ diswðtÞ=dt. If the switchis closed, then Vsw¼ 0, and the sinusoidal voltage builds upover Lp. The sole task of the capacitor CB in series with theparallel circuit LS, CS is to prevent LS from shorting thebias voltage, and the bypass capacitor to decouple the biasvoltage from the RF signals has been replaced with an RFchoke. Hence, the new voltage-based design is the electricaldual of the conventional current-based approach, but doesnot require a transformer at microwave frequencies.

For a voltage-based approach, an important parameterof the active device when used as a switch is its off-resistance, since leakage currents during the off-cycle impaircircuit performance. According to Anholt [5], the off-resistance of a HBT used as a switch is virtually infinite, andhence superior to the channel resistance of a pinched-offpHEMT, which is typically in the order of several kO.Hence, for low frequencies, performance of HBTs may besuperior to that of pHEMTs. However, at the frequenciesdiscussed here, the dominant parameter is the outputcapacitance, which is around 30 fF for small pHEMTdevices, while an HBT has a collector-emitter capacitance ofabout 40 fF. Hence, during the off-cycle, neither device isvastly superior over the other.

The drain-to-source capacitance of a pHEMT in firstapproximation increases linearly with total gate width [5].Hence, with increasing gate width the leakage currentsduring the off-cycle increase, and the advantage of thevoltage-based class E design over the conventionalapproach is mitigated. In first approximation, the voltage-based approach will cease to be superior to the conventionalapproach when the decrease in output capacitance duringthe off-cycle is of similar magnitude to the increased channelresistance during the on-cycle for small devices. Since thechannel resistance of small devices is about ten times higherthat that of large devices, this point is reached when theoutput capacitance has approximately tenfold compared tothe small devices, which happens at around 1.5mm totalgate length [5].

In conclusion, the voltage-base approach presented here,owing to its inherent robustness against channel resistance,should allow the use of reasonably small pHEMT devices tobuild high-efficiency class E amplifiers for low-powerapplications. As total gate width increases, so does thedrain-source capacitance, mitigating the positive effects ofthe voltage-based approach. Owing to these features itappears that the voltage-based approach is particularlysuited for circuits with low RF power requirements whereintended battery life is important, e.g. applications in theISM band at 2.4GHz where EIRP should be less than100mW in Europe [8]. With an 8� 75mm device (notavailable to us at the time of construction) our calculationsshow that at 2.4GHz, 100mW at 80% efficiency should bepossible.

3 Simulation results

3.1 Evaluation using ideal componentsThe new topology was first simulated with Agilent’sAdvanced Design System (ADS) using an ideal switch withan adjustable series resistance Rsw (Fig. 2) replacing theactive device. Since the switch is an extremely nonlinearcomponent, transient simulation was chosen as the mostsuitable simulation technique.

Vout

iq

LS

Lp

CS

CB

RL

VL

isw

Vsw

Rsw

VVDC

Fig. 2 Voltage-driven class E power amplifier circuit schematic

RL

i�

LSCS

iC

Cp

isw

VVDC

Fig. 1 Current-driven class E power amplifier circuit schematic[4]

374 IEE Proc.-Microw. Antennas Propag., Vol. 152, No. 5, October 2005

Figure 3 compares the waveforms of the new structureFig. 3c with those of the original Cripps design, (Figs. 3a, b).Figure 3a shows, from top to bottom, the switch current isw,the shunt capacitor current iC, and the voltage across theshunt capacitor, VC, respectively. For this simulation, theswitch has been stimulated with an internal resistance of0.1O (an ideal 0O resistance lead to numerical problems) inthe closed state.

Figure 3a shows how the sinusoidal output current iycreated by the series resonant circuit LS, CS, Fig. 1, is splitcleanly into current isw, and current iC as long as the switchtiming is precise. The switch series resistance in this case issmall enough to ensure that iy flows exclusively through theswitch while it is closed, ensuring that the shunt capacitorCP is discharged at the moment of switching, bottom graph.

The centre graph shows the same current and voltagewaveforms, but this time with a series resistance of 5O forthe switch in the closed state. As can be seen, the signals aresignificantly distorted. Particularly, owing to the higherresistance of the switch, some current flows into the shuntcapacitor CP even when the switch is closed, and thereforethe shunt capacitor voltage VC is no longer zero at the timeof switching, causing a spike that upsets circuit operation.The two currents isw and iC, when combined, no longer leadto a sinusoidal signal, and the circuit ceases to workproperly.

Figure 3c shows, from top to bottom, the inductorvoltage VL, switch voltage Vsw, and inductor switch currentisw, respectively. The switch has again been set to 5Ointernal resistance. For the new design, the voltages VL andVsw complement each other such that, when overlaid on thesame scale, they result in the required sinusoidal outputsignal. Since no spike is created upon switching that couldinterfere with the circuit timing, the circuit continues tooperate in spite of the presence of switch series resistance.Further stimulation has shown that we can increase Rsw inFig. 2 up to 20O without unduly impairing the voltagedriven waveform shapes. The new circuit has somewhat

different timing characteristics compared to Cripps’s designowing to the fact that voltages rather than currents areswitched.

4 Class E amplifier MIC

4.1 Circuit descriptionTo confirm these results, we build a hybrid MIC version ofthe voltage-driven topology (Fig. 4). The circuit wasconstructed using Taconics TLY-5-0100-CH softboard,mounted on a brass carrier using conductive epoxy toprovide mechanical stability. Standard surface-mountcomponents, whose values are defined in Table 1, wereused for the passive network, while the active device is anOMMIC ED02AH 6� 50mm pHEMT device. The com-ponent values were first obtained using the design equationsderived by Cripps [4], and then optimised in Agilent’sAdvanced Design System (ADS) using transient simulationswith a foundry-supplied large-signal model for thepHEMT.

800

600

400

200

0

−200

i sw

800

600

400

200

0

1086420

−200

−2

i CV

C

i swi C

VC

VL

Vsw

i sw

400

300

200

100

0

400

200

0

765432

−400

−200

01

6420

− 4−6−8

−2

5

10

0

−5

−10

−15

20

15

10

5

0

−540 41 42 40 41 42 40 41 42

a b ctime

A

A

C

C

Fig. 3 Waveformsa Ideal switchb Lossy switchc Suggested circuit with lossy switch

Fig. 4 Photograph of class-E MIC

IEE Proc.-Microw. Antennas Propag., Vol. 152, No. 5, October 2005 375

The active device has been mounted with conductiveepoxy onto a brass pedestal in order to bring its surfacelevel with the TLY-5 substrate, and the electrical connec-tions established using 17.5mm gold wire-bonds. SMAsockets have been soldered directly onto the TLY-5microstrip lines to allow the test equipment to be connected.DC gate bias was supplied through the RFin port using abias tee.

In order to keep the circuit simple, a dedicated outputmatching circuit was not used, i.e. any matching providedby the LS, CS network was not optimised. For themeasurements, the input of the amplifier was matched to50O using Focus Microwaves model 1808 tuners. Owing tothe limited tuning range of the tuners, and the highlyreactive input impedance of the pHEMT, two tuners wererequired to achieve a small signal return loss of 20dB at870MHz. The tuning point of the tuners was establishedempirically to achieve maximum power output for giveninput power level.

4.2 Measurement resultsFigure 5 shows the measurement setup. The input signalwas generated using an Agilent HP 8657B signal generator,capable of providing a maximum of 18dBm input power at870MHz. A drain bias of +2V is directly supplied to theVcc port of the class E amplifier. Finally, the output signal isfed through a second bias tee to an Agilent HP E 4407Bspectrum analyser. A spectrum analyser was chosen toensure that the output signal was of acceptable spectralpurity, and the amplitude readings of the spectrum analyserwere confirmed using an HP 437B power meter. Figure 6shows the results obtained with this setup.

At the nominal input power level of 0dBm, the poweramplifier exhibits 17.9dB gain resulting in an outputpower of 17.9dBm (61.7mW), 95% efficiency and 93%power added efficiency at a frequency of 870MHz. Biasconditions were +2V bias voltage, �0.1V gate bias, and32.6mA bias current resulting in 65.2mW consumed DCpower. Harmonic suppression was 37.9dB for the second

harmonic, and 38.4dB for the third harmonic. For lowerinput power levels, the amplifier achieves higher gain, e.g.27dB for �15dBm input signal level. However, since thebias current remains virtually unchanged as drive powerlevel is varied, efficiencies under those operating conditionsare low at around 24%. Maximum efficiency of 96%, andmaximum power added efficiency of 94% are achieved for1dBm input power level.

These results, shown in Fig. 6, compare well with thepublished literature for other pHEMT devices but used inconventional current driven mode, e.g. a class E PA MICintroduced by Nagle et al. in 2001 [3], which at 900MHzexhibited a gain of 9.5dB, an output power of 31.5dBm,and a PAE of 53%, however using a much larger devicehaving 12mm total gate length compared to the 300mmused here. In 2002, Tayrani [6] introduced another class Eamplifier MMIC, which exhibited 13dB gain and a peakpower added efficiency of 90% at 3.25GHz, but againrequired a device of 1000mm total gate length, and a biasvoltage of 12V, which is not attractive for hand-held mobiledevices. These results together with a number of HBTdesign performances are compared against each other inTable 2 (empty fields mean that the authors did not specifythe respective parameter).

Next, the ability to directly control the output power levelof the class E amplifier via its DC bias voltage wascharacterised (Fig. 7). Increasing the bias voltage from 0.5to 4V, the output power level could be varied by 14dB.Efficiency and power added efficiency (PAE) vary over thisrange, e.g. PAE from about 45–100%; however, thesevalues are still high compared to conventional linearamplifiers, e.g. class A or B, which loose efficiency followingan exponential law when operated in power back-off mode[9]. Figure 7 shows that over the bias voltage range 0.5–4Vwe can control the amplitude of the class E amplifier from6.7 to 20.6dBm without unduly changing its power addedefficiency. The measurements were limited to a maximumbias voltage of 5V owing to the maximum rating of thedrain-to-source RF voltage of 7V for the OMMIC deviceused.

In Fig. 7, for this operating mode of very low powerlevels and very high efficiencies, the resulting DC currentsare becoming very small, and hence hard to measureaccurately, as indicated by the increasing size of the errorbars in Fig. 7, thus the indication is that efficiency isgenerally better that 90%.

Figure 8 shows a plot of efficiency, PAE, and gainagainst frequency. As can be seen, the circuit maintains an

Table 1: Component values used

Component Value

Lp 6.5nH

LS 37nH

CS 1pF

CB 47pF

RF choke 37nH

Switch 6� 50mm pHEMT

Thurlby

sourceDC

Thurlby

sourceDC

bias tees

HP 8657 B HP E 4407 B

gate bias

focustuner

focustuner

drain bias

class E PA MIC

Fig. 5 Measurement setup

0

20

40

60

80

100

−10 −5 0 5 100

6

12

18

24

30

efficiency, %

PAE, %

gain, dB

input power, dBm

dB%

Fig. 6 Gain, efficiency and power added efficiency against inputpower at 870 MHz

376 IEE Proc.-Microw. Antennas Propag., Vol. 152, No. 5, October 2005

efficiency and power added efficiency of more than 50%,and a gain of more than 15dB over 11.5% bandwidth.

5 Conclusions

In this work, what we believe to be the first practicaldemonstration of a voltage-driven class E power amplifiercircuit has been given. A consequence of our work has beento show that voltage-driven class E amplifiers are best suitedfor small devices, hence low-power MMIC implementationfor applications where high efficiency is essential. Theoperation of the new circuit has been explained, and the

concept examined by the means of simulation, andmeasurement.

A prototype hybrid MIC version of the circuit was builtand measured. The circuit exhibited very good performance,with 27dB maximum gain, 21dBm maximum outputpower, 96%maximum drain efficiency, and 94%maximumpower added efficiency. The circuit uses a single 6� 50mmED02AH pHEMT device, and to our knowledge these arethe best results achieved to date with this size of deviceoperated in class E mode.

6 Acknowledgments

The authors gratefully acknowledge TDK UK, TDKIreland, and EPSRC under grant EP/C002083/1 forfunding this project, Alan Black and Dr. Neil Buchananfor help with the assembly of the MIC hybrids, andCeleritek UK Ltd., – Belfast for carrying out most of thewire bonding work. The work described in this paper is thesubject of British Patent Application No. 0404121.6, filed 25February 2004.

7 References

1 Sokal, N.O., and Sokal, A.D.: ‘High-efficiency tuned switching poweramplifier’, U.S. Patent 3,919,656, United States Patent Office,November 1975

2 Razavi, B.: ‘Microelectronics, RF communications engineering andemerging technologies series’ (Prentice Hall, Upper Saddle River, NJ,USA, 1998), Chap. 9, pp. 298–325

3 Nagle, P., Burton, P., Heaney, E., and McGrath, F.: ‘A linear andefficient wide-bandwidth handset RF transmitter’. European Conf. onWireless Technology, 2001, pp. 181–184

4 Cripps, S.C.: ‘RF power amplifiers for wireless communications’(Artech House, Inc., Norwood, MA, USA, 1999)

5 Anholt, R.E.: ‘Electrical and thermal characterization of MESFETs,HEMTs, and HBTs’ (Artech House, Inc., Norwood, MA, USA,1995)

6 Tayrani, R.: ‘A broadband monolithic S-band class-E poweramplifier’. IEEE Radio Frequency Integrated Circuits (RFIC) Symp.Dig., Seattle, WA, USA, June 2002, pp. 53–56

7 Raab, F.H.: ‘Idealized operation of the class E tuned poweramplifier’, IEEE Trans. Circuits Syst., 1977, CAS-24, (12),pp. 725–735

8 ‘Supplement to IEEE standard for IT, part 11’ IEEE Standard802.11b-99, 2005

9 Asbec, P.M., Larson, L.E., and Galton, I.G.: ‘Synergistic design ofdsp and power amplifiers for wireless communications’, IEEE Trans.Microw. Theory Tech., 2001, 49, (11), pp. 2163–2169

10 Grebennikov, A., J.ager, H., and Weigel, R.: ‘High-efficiency class Emonolithic HBT power amplifiers for wireless applications’. 6thEuropean Conf. on Wireless Technology, Munich, Germany, 2003,pp. 313–316

11 Koller, R., Stelzer, A., Abt, K.-H., Springer, A., and Weigel, R.: ‘Aclass-E GSM-handset PA with increased efficiency’. 33rd EuropeanMicrowave Conf., Munich, Germany, 2003, pp. 257–260

Table 2: Comparison of recent class E amplifier designs

QUB Nagle Tayrani Grebennikov et al. Koller et al.

Device pHEMT pHEMT pHEMT HBT HBT

Lg 0.3mm 12mm 1mm n/a n/a

Pout 21dBm 32dBm 23dBm 33dBm1 35dBm1

Gain 27dB 9.5dB 15dB 33dB –

Z 96% 53% – 57% 71%

PAE 94% 49% 90% 47% 63%

Frequency 870MHz 900MHz 3.25GHz 1.8GHz 900MHz

Vcc 2V 3.3V 12V 3.5V 3.6V

Year 2004 2001 [3] 2002 [6] 2003 [10] 2003 [11]

1Multi-stage design

Vg : − 0.1 V

P in: 9 dBm

efficiency, %

PAE, %Pout, dBm

0

5

10

15

20

25

dB

Vcc, V

0.50

20

40

60

80

100

1.5 2.5 3.5 4.5

%

Fig. 7 Efficiency and output power against bias voltage of class Eamplifier

efficiency, %

PAE, %

gain, dB

efficiency > 50%PAE > 50%gain > 15 dB

0

20

40

60

80

100

%

19

17

15

13

11

9

7

5

3

dB

frequency, GHz

0.900.850.800.750.70

Fig. 8 Efficiency and gain against frequency

IEE Proc.-Microw. Antennas Propag., Vol. 152, No. 5, October 2005 377