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69
Chapter 5
SENSORLESS OPERATION OF FOC USING
MODEL REFERENCE ADAPTIVE SYSTEM
WITH SLIDING MODE CONTROL
5.1. Introduction
The position or speed sensor (such as tachometer based speed
sensors, optical incremental sensors or electromechanical resolvers) in
induction motor drives impose many practical problems such as complexity
of hardware, difficulties in application to hostile environment, increased
cost, reduced reliability due to cables and sensors itself, difficulties of
mechanical attachment of sensor to the electric machine, increased axial
length of the machine and electromagnetic noise interference. To solve these
problems, various speed or position sensorless control schemes have been
developed for variable speed AC drives. In sensorless drives, no conventional
speed or position monitoring devices are used, instead the speed or/and
position signal is obtained by using monitored voltages and/or currents and
by utilizing mathematical models [81].
Although several schemes are available for sensorless operation of a
vector controlled drive, MRAS is popular because of its simplicity. SMC is
considered to be the appropriate methodology to replace PI controllers in
MRAS for the robust nonlinear control of induction motor drives due to its
order reduction, disturbance rejection, strong robustness and simple
implementation by means of power converter. SMC is a control strategy in
Variable Structure System (VSS) having a proper switching logic with high
70
frequency discontinuous control actions depending on the system state,
disturbances and reference inputs.
In this chapter, based on the concepts of MRAS and SMC, a
mathematical model of speed estimation system is presented for sensorless
FOC induction motor drive. Based on this, simulation model of sensorless
FOC induction motor drive using MRAS-SMC is developed and validated
with extensive simulation results. Performance of the same is compared with
sensorless FOC induction motor drive model using MRAS-PI.
5.2. Model Reference Adaptive System
5.2.1. Overview of MRAS
The model reference approach takes advantage of using two
independent machine models, reference model and adjustable model for
estimating the same state variable. The estimation error between the
outputs of the two computational blocks is used to generate a proper
mechanism for adapting the speed. The difference between the two
estimated vectors is used to feed a PI controller. The output of the controller
is used to tune the adjustable model, which in turn actuates the rotor
speed. However, PI controllers may drop the performance level due to
the continuous variation in the machine parameters and operating
conditions in addition to nonlinearities contributed by the inverter.
5.2.2. MRAS Design: Hyper Stability Concepts
The adaption algorithm for MRAS can be taken into account the
overall stability of the system and to ensure that the estimated speed will
converge to the desired value with satisfactory dynamic characteristics [82].
The adaptive scheme for the MRAS estimator can be designed based on
Popov’s criteria for hyper stability concept [83], this relate to the stability
properties of a class of feedback systems. In general, a model reference
adaptive speed estimator system can be represented by an equivalent non-
linear feedback system which comprises a feed forward time-invariant linear
subsystem as well as a feedback non-linear time-varying subsystem. In Fig.
5.1, the input to the linear time-invariant system is u, which contains the
71
stator voltage and currents and its output is y, which is the speed-tuning
signal. The output of the non-linear time invariant system is ω, and u = -ω.
The rotor speed estimation algorithm (adaptation mechanism) is chosen
according to Popov’s hyperstability theory, whereby the transfer function
matrix of the linear time invariant system must be strictly positive real and
the non-linear time-varying feedback system satisfies Popov’s integral
inequality. Fig. 5.1 shows the standard non linear time varying feedback
system, which is said to be asymptotically stable if it satisfies the following
two conditions:
i) The transfer function of the feed-forward linear time invariant block
must be strictly positive real.
ii) The error should converge asymptotically.
iii) The non linear time varying block satisfies the Popov’s integral
inequality:
2(0, )
0
tT
t y dtη ω γ= ≥ −∫
(5.1)
where y : Input vector
ω : Output vector of the feedback block
γ 2 : Finite positive constant
Fig. 5.1 Standard Non-Linear Time varying feedback system
5.3. MRAS Based Speed Estimation
MRAS computes a desired state called functional candidate using two
different models. In the rotor flux based MRAS, the rotor flux is used as an
72
output value for the model to estimate the rotor speed. The MRAS scheme is
based on two independent machine models, the reference model and the
adaptive (adjustable) model for estimating the same state variable. Speed
estimation is done by comparing the output of the reference model with the
output of the adaptive model until the error between the two models
disappear. The reference model does not contain the speed to be computed,
which represents stator equation and is usually known as voltage model.
The reference value of the rotor flux components in the stationary frame are
generated from the monitored stator voltage and current components, which
are given by [1] and [34].
( )rr s s s s s
m
LV R i dt L i
Lα α α αψ σ = − − ∫
(5.2)
( )rr s s s s s
m
LV R i dt L i
Lβ β β βψ σ = − − ∫ (5.3)
0
0
r s ss sr
r s ss sm
V iR L pLp
V iR L pL
α α α
β β β
ψ σ
ψ σ
+ = − +
(5.4)
where,
11
(1 )(1 )s r
σσ σ
= −+ +
dp
dt=
The adaptive model contains the estimated rotor speed, which
represents the rotor equation and is usually known as the current model.
The adaptive values of rotor flux components are given by [1] and [34].
1ˆ ˆ ˆ ˆ( )r m s r r r r
r
L i dtα α α βψ ψ ω τ ψτ
= − −∫ (5.5)
1ˆ ˆ ˆ ˆ( )r m s r r r r
r
L i dtβ β β αψ ψ ω τ ψτ
= − +∫
(5.6)
1ˆ
ˆ ˆ
ˆ ˆ1ˆ
rr s rrm
r s rrr
r
iLp
i
α α α
β β β
ωψ ψτ
ψ ψτω
τ
− − = + −
(5.7)
73
The instantaneous angular speed ωr of the rotor flux vector on an
open loop basis can be obtained from the measured voltages and currents.
The rotor flux vector angle and its derivative are expressed as:
1ˆ
tanˆ
r
r
β
α
ψφ
ψ−
=
(5.8)
( ) ( )2 2
ˆ ˆ ˆ ˆ
ˆ ˆ
r r r r
r r
p pp
α β β α
α β
ψ ψ ψ ψφ
ψ ψ
−=
+ (5.9)
Substituting (5.7) in (5.9),
2 2
ˆ ˆˆ
ˆ ˆ
r s r smr
r r r
i iLp
α β β α
α β
ψ ψφ ω
τ ψ ψ
−= +
+ (5.10)
The difference between the two estimated vectors is fed to an adaption
mechanism to generate estimated value of rotor speed which is used to tune
the adaptive model. The adaption mechanism of conventional rotor flux
MRAS is a simple fixed gain linear PI controller. The adaptive scheme for the
MRAS estimator can be designed based on Popov’s criteria for hyper stability
concept [82]. When the rotor flux of the adjustable model is in accordance
with that of the reference model the rotor speed of the adjustable model
becomes the real motor speed. The tuning signal, eω actuates the rotor
speed, which makes the error signal zero. The expression for estimated rotor
speed is given by [34]. MRAS of this kind are extensively used to identify
plant parameters and inaccessible variables. In designing the adaption
mechanism for a MRAS, it is important to consider the overall stability of the
system and to make sure that the estimated quantity will converge to the
desired value with suitable dynamic characteristics. Practical synthesis
technique for MRAS structures based on the concept of hyper stability was
described in [82]. Generally the models are linear time varying systems and
ωr is a variable, but for deriving an adaption mechanism, treat ωr as a
constant parameter of the reference model. State error equations are
obtained by subtracting (5.7) for the adjustable model from the
corresponding equations of the reference model shown in Fig. 5.2 as:
74
Fig. 5.2 MRAS representation as a Non-Linear feedback system
[ ]
1
ˆˆ
ˆ 1
rr r rr
r rr rr
r
r
e ep
e e
α β α
β βα
ωψ τ
ω ωψ
ωτ
− − − = − + −
(5.11)
[ ] [ ][ ] [ ]e A e Iω ω ω= + −&
(5.12)
These equations describe a non linear feedback system since ὣr is a
function of the state error. In this hyper stability is assured provided that
the linear time invariant forward path transfer matrix is strictly positive real
and the non linear feedback satisfies Popov’s criterion for hyper stability.
5.3.1 Selection of Speed Adaption Law
For designing the adaptive laws the Popov’s inequality criterion is
made use of which consists of the following steps:
1. Transform the MRAS into an equivalent system called a non linear time
variable feedback system, which includes a feed forward linear model
and a non linear feedback system.
75
2. Ensure strict positive realness of the feed forward linear model.
3. Design the adaptive laws which can ensure that the non linear
feedback block satisfies the Popov inequality given by (5.1).
4. Transfer the equivalent system back to the MRAS system.
Step 1: Transform the MRAS into an equivalent system
The first step to transform the MRAS into a Non Linear Time (NLT)
variable feedback system is shown in Fig. 5.2. Consider the system given by
(5.12) where the matrix A is constant and hence can be included in the
linear time invariant forward block. Now ὣr is a function of the state error
and is time varying in nature, hence along with the adaption mechanism
can be included in the non linear feedback system.
Step 2: Ensure strict positive realness of the feed forward linear
model
The second step is to ensure strict positive realness of the feed
forward linear model. For that consider a general system represented by,
x Ax Bu= +&
(5.13)
y Cx Du= +
(5.14)
Following Kalman-Yakubovich- Popov lemma (Positive Real Lemma),
Lemma 1: Let Z(s) =C (sI-A)-1B+D be a (p x p) transfer function matrix,
where A is Hurwitz, (A, B) is controllable, and (A, C) is observable. Then Z(s)
is strictly positive real if and only if there exist a positive definite symmetric
matrix P, matrices W and L, and a positive constant ԑ such that
T TPA A P P L Lε+ − = −
(5.15)
T TPB C L W= −
(5.16)
T TW W D D= +
(5.17)
So for the current system given by (5.11), it can be proved that A is
Hurwitz and also it is evident that (A, I) (as B = I) is controllable and (A, I) (as
C = I) is observable. Now for the system (5.16) can be rewritten as in (5.19),
where B = I, C = I and D = 0.
T TPA A P P L Lε+ − = −
(5.18)
76
TP I=
(5.19)
Let ԑ=1 then
1 0
0 1P I
= =
(5.20)
Substituting the value of P from (5.20) and A from (5.11) in (5.18)
gives (5.21) which ensures that LTL is symmetrical positive definite and
hence Z(s) satisfies conditions of (5.18) and is strictly positive real.
2( 1) 0
20 ( 1)
rT T
r
PA A P P L Lτ
ε
τ
− + + − = − =
− +
(5.21)
Step 3: Ensure strict positive realness of the feed forward linear
model
The third step is to design the adaptive laws which can ensure that
the non linear feedback block satisfies the Popov inequality criterion given
by (5.1). For the system represented by Fig. 5.2, (5.1) can be written as:
2(0, )
0
tT
t e dtωη ω γ= ≥ −∫
for all t≥ 0 (5.22)
where γ2 is a positive constant,
eω is the output of the linear system,
ω is the output of the non linear system.
Substituting for eω and ‘ω’ in this inequality Popov’s criterion, the
present system becomes:
2
0
ˆ ˆ ˆ[ ][ ]t
r r r r r re e dtβ α α βψ ψ ω ω γ− − ≥ −∫
(5.23)
Let 1 2
0
ˆ ( ) ( )t
r f f dω τ τ τ= + ∫
(5.24)
then (5.23) becomes,
77
21 2
0 0
ˆ ˆ[ ][ ( ( ) ( ) )t t
r r r r re e f f d dtβ α α βψ ψ ω τ τ τ γ− − + ≥ −∫ ∫
(5.25)
A solution to this inequality can be found through the relation as:
and
2 2 2
0
2
0
1 1. ( ) ( ) ( ( ) (0)) . (0)
2 2
( ) 0, 0
t
t
k f t f t dt f t f k f
f t dt k
= − ≥ −
≥ >
∫
∫
&
(5.26)
by assuming the values of f1(τ) and f2(τ) as given below and Popov’s
inequality criterion can be ensured by the following function as:
1ˆ ˆ ˆ ˆ( ) p r r r r p r r r rf K e e Kβ α α β β α α βτ ψ ψ ψ ψ ψ ψ = − = −
(5.27)
2ˆ ˆ ˆ ˆ( ) i r r r r i r r r rf K e e Kβ α α β β α α βτ ψ ψ ψ ψ ψ ψ = − = −
(5.28)
Hence the adaptive law for ὣr is given by PI adaptive law as:
0
ˆ ˆ ˆ ˆ ˆt
r p r r r r i r r r rK K dβ α α β β α α βω ψ ψ ψ ψ ψ ψ ψ ψ τ = − + − ∫
(5.29)
Step 4: Transfer equivalent system back to MRAS system
The last step is to transfer the equivalent system back to the MRAS
system whose representation is as shown in Fig. 5.3.
Fig. 5.3 Structure of MRAS based speed estimator
78
ˆr p iK e K e dtω ωω = + ∫
(5.30)
It is important to design the adaptation mechanism of the MRAS
according to the hyper-stability concept, which will result in a stable and
quick response system where the convergence of the estimated value to the
actual value can be assured with suitable dynamic characteristics [1].
Popov’s criterion of hyper-stability for a globally asymptotically stable
system is used in deriving the speed estimation relation, which represents
the difference between the reference model and the adjustable model. The
parameter ε should be passed through a PI block that is found to be
satisfactory for the adaptive scheme, it gives the below objective function as:
ˆ ˆr r r reω β α α βψ ψ ψ ψ= −
(5.31)
The classical rotor flux MRAS speed observer, generally gives
reasonable speed estimation in the high and medium speed ranges.
However, at low speed regions, the performance deteriorates due to
integrator drift and initial condition problems and sensitivity to current
measurement noise [84].
Block diagram of MRAS based speed identification system is shown in
Fig. 5.4. The reference model is independent of of rotor speed. The
adjustable model requires stator currents and is dependent on the rotor
speed. The currents, is� and isβ and voltages, vs� and vsβ are taken from
motor terminals for the speed estimation.
Figure 5.5 describes the classical rotor flux MRAS-PI with both
references and adaptive models for rotor speed estimation. The parameter eω
should be passed through a PI block that is found to be satisfactory for the
adaptive scheme.
The speed estimated from MRAS is fed back to a speed controller in a
sensorless drive and is compared with the reference speed to get the
command output. Fig. 5.6 shows the block diagram of sensorless FOC
induction machine with MRAS-PI speed estimator.
79
Fig. 5.4 Block diagram of MRAS based speed identification system
Fig. 5.5 Block diagram of MRAS-PI speed estimator
80
Fig. 5.6 Block diagram of sensorless FOC with MRAS-PI speed estimator
5.3.2 Design of PI for MRAS Based Speed Estimator
The transfer function in (5.32) is obtained from (5.7) through
linearization with respect to a certain operating point [28]:
2
1( )2 2
1( )
1ˆ( )
r
rs
r rs
r
se
G
s
ω
ψτ
ω ω ωτ
+ ∆
= = ∆ − ∆ + +
(5.32)
Figure 5.7 depicts the block diagram of (5.32). Assuming ωs = 0 for
simplicity, damping factor ξ and natural angular frequency ωc can be
specified by using Kp and Ki as follows:
2
12 c
rp
r
K
ξωτ
ψ
−
= (5.33)
( )2
2
c
i
r
Kω
ψ= (5.34)
81
In (5.33), Kp and Ki are fixed and calculated as a function of rotor flux
magnitude, ψr, in the steady state, which is obtained from current model.
Especially in transients and braking mode, it may cause to significant
errors. For the proposed method of work, rotor flux magnitude, ψr, which is
calculated from current model is replaced with that of voltage model. Since
voltage model have been selected as reference model and current model
must follow it, with this replacement, the proposed method will be in
prediction mode. On the other hand, Kp and Ki are on-line tuned with
respect to instantaneous variation of rotor flux, instead of its expected value
in steady state. In this way, rotor flux magnitude unexpected variations, will
be neutralized by on-line tuning of PI parameters and its resulting probable
instability will be prevented. This modification makes the proposed method
adaptive for various conditions. Prediction and adaptation are two
characteristics of the proposed method. This method does not complicate
the algorithm, because rotor flux magnitude, ψr, is continuously calculated
in the MRAS algorithm and thus there is no need for additional
computations.
Fig. 5.7 Speed estimator dynamics
5.4. Sliding Mode Control - Concepts
SMC is basically a non linear control method that alters the dynamics
of a non linear system by application of a high frequency switching control
varying system structure for stabilization. It is the motion of the system
trajectory along a chosen line, plane or surface of the state space. Control
structures are designed so that trajectories always move towards switching
condition and the ultimate trajectory will slide along the boundaries of the
control structures. Sliding mode can be reached in finite time due to
discontinuous control law. SMC is deterministic because only bounds of
82
variations are considered and non linear since the corrective term is not
linear. It is robust since once on the sliding surface the system is robust to
bounded parameters variations and bounded disturbances. Sliding mode
control is an appropriate robust control method for the systems where
modeling inaccuracies, parameter variations and disturbances are present.
Sliding variables converge asymptotically and insensitive only against
matched perturbations and outputs with relative degree first are certain
disadvantages of sliding mode control. Chattering due to implementation
imperfections is the main weakness of SMC. The advantages of sliding mode
control comprise:
i. Low sensitivity to plant parameter uncertainty (bounded uncertainty)
ii. Reduced order modeling of plant dynamics
iii. Finite time convergence
Ability to result in very robust control system is the most important
aspect of the above scheme. In SM, a control law is designed so as to bring
the system trajectory on a predefined surface called the sliding surface.
Sliding phase is that phase where it slides along the predefined surface to
the equilibrium point and reaching phase is the phase before it touches the
sliding surface. A typical sliding mode control phase plot is shown in Fig 5.8,
where the equilibrium point is indicated by ‘O’ and reaching phase is
represented by the trajectory PQ and sliding phase by the trajectory QO. By
choosing appropriate control law, once the phase trajectory touches the
sliding surface, it is maintained in the sliding surface and finally converges
to the equilibrium point, the system dynamics will be governed by the
dynamics of the sliding surface. Then the system remains invariant to
parameter variations. In sliding mode controller, the system is controlled in
such a way that the error in the system states always move towards a
sliding surface. The sliding surface is defined with the tracking error of the
state and its rate of change as variables. The control input to the system is
decided by the distance of the error trajectory from the sliding surface and
its rate of convergence. At the intersection of the tracking error trajectory
with the sliding surface, the sign of the control input must be changed, thus
83
the error trajectory is always forced to move towards the sliding surface. If
the desired error dynamics is:
2 1 0S X CX= + =
(5.35)
Then,
i) The control objective of sliding mode control is to reach S = 0 in finite
time.
ii) The manifold S = 0 is known as the sliding surface
iii) Once in the surface, the control must keep the trajectories “sliding" on
the surface: sliding mode
Fig. 5.8 Phases of Sliding mode control
5.5. Design of Sliding Mode Controller
The sliding mode controller design basically involves two steps:
1. Choose the sliding surface
2. Design a control law that brings the phase trajectory to the sliding
surface and maintains the trajectory on the sliding surface itself and
finally converge it to the equilibrium point.
84
5.5.1. Selection of the Sliding Surface
Here only flux controller sliding surface is chosen and the respective
surface is taken as:
p iS K e K e dtψ ψ ψ ψ ψ= + ∫
(5.36)
ˆrefeψ ψ ψ= −
(5.37)
5.5.2. Selection of Control Law
For designing the control law, the induction machine state model in
stationary reference frame with stator current and stator flux as the state
variables is made use of. The induction machine state model in stationary
reference frame can be expressed as:
1s r r rs s r s s s s s
s r s r s s
R R Ri i i i V
L L L L L Lα α β α α β α
ωω ψ ψ
σ σ σ σ σ= − − − + + +&
(5.38)
1s r r rs s r s s s s s
s r s r s s
R R Ri i i i V
L L L L L Lβ β α β β α β
ωω ψ ψ
σ σ σ σ σ= − − − + + +&
(5.39)
where, 2
1 m
s r
L
L Lσ = −
s s s sV i Rα α αψ = −&
s s s sV i Rβ β βψ = −&
The control law for the sliding mode controllers has to be selected in
such a way that it should be stable. So, in order to prove the stability of
sliding mode controllers appropriate Lyapunov functions are chosen from
which the required discontinuous control law can be derived. Here choosing
a Lyapunov Function as:
21( )
2V S Sψ=
(5.40)
Taking the derivative of (5.40), it gives
( )V S S Sψ ψ= &&
(5.41)
85
Taking the derivative of Sψ
ˆp ref i pS K K e Kψ ψ ψ ψ ψψ ψ= + − && &
(5.42)
Using (5.38) in (5.42) it can be written as:
1ˆ
pS M Kψ ψψ= − &&
(5.43)
where 1 1 1 2ˆ,p ref i p s sM K K e K F D V D Vψ ψ ψ ψ β αψ ψ= + = − + +&&
in which, ( )1ˆs
p s s s s
RF K i iψ α α β βψ ψ
ψ= +
1
ˆ
s
pD Kβ
ψ
ψ
ψ=
2ˆs
pD K αψ
ψ
ψ=
hence can be written as:
1 1 1 2s sS M F DV D Vψ β α= + − −&
(5.44)
which can be written as:
S M F Dν= + −& (5.45)
To ensure Lyapunov stability condition, choose the control law as follows:
1( ( ))v D M F Ks sat sα−= + + +
(5.46)
Substituting (5.46) in (5.45), (5.41) can be written as:
( )( )1( ) ( )V S S M F DD M F Ks sat sα−= + − + + +&
(5.47)
The above equation can be written as:
( )( )V S Ks sat sα= − +&
(5.48)
In this, the first derivative of the Lyapunov equation is negative
definite. Thus from (5.48) it is evident that, the sliding mode controller is
stable as it satisfies the Lyapunov’s stability condition.
86
5.6. Sliding Mode MRAS Speed Observer
The sliding mode is a control technique to adjust feedback by
previously defining surface. The system which is to be controlled will be then
forced to that surface and the behaviour of the system slides to the desired
equilibrium point [84]. Induction machine as a nonlinear system can be
demonstrated as state space model in the canonical form [38]:
( , ) ( , ) ( )x A x t B x t u t= +&
(5.49)
where, [ ]min max, , , ( ( , )) , ,n n mx R A F u R rank B x t m u u u∈ ∈ ∈ = ∈
A time varying surface is defined in the state space by the scalar equation,
( ) 0S x =
(5.50)
The aim is to maintain the system motion on the manifold S(t), which is
defined as:
{ }( ) : ( , ) 0iS x x e x t= =
(5.51)
d
i i ie x x= −
(5.52)
where, e is state error or tracking error, xid is the desired state vector and xi
is the state vector. The control input u has to guarantee that the motion of
the system described in (5.49) is restricted to the manifold S in the state
space. The sliding mode control should be chosen such that the candidate
Lyapunov function, V which is a scalar function of S and its derivative
satisfies the Lyapunov stability criteria as:
21( ) ( )
2V S S x=
(5.53)
( ) ( ) ( )V S S x S x= &&
(5.54)
According to Lyapunov theory, if the time derivative of V(S) along a
system trajectory is negative definite, this will ensure that it constrains the
state trajectories to a point towards the sliding surface S(t) and once on the
surface, the system trajectories remain on the surface until the origin is
87
reached asymptotically. Thus the sliding condition is achieved by the
following condition in (5.55) makes the surface an invariant set.
( ) ( )V x S xη≤ −&
(5.55)
where ‘η’ is a strictly positive constant on outside of S(t). Lyapunov stability
criteria will achieve as in (5.55) states that the squared “distance” to the
surface, measured by e(x)2, decreases along all system trajectories [38]. The
condition also implies that some disturbances or dynamic uncertainties can
be tolerated while keeping the surface an invariant.
Starting from any initial condition, the state trajectory reaches the
sliding surface in a finite time and then slides along the surface
exponentially with a time constant equal to 1/K. The control rule can be
written as:
( ) ( ) ( )eq swu t u t u t= + (5.56)
where, u(t) is the control vector, ueq(t) is the equivalent control vector
and usw(t) is the switching vector and must be calculated so that stability
condition as per (5.57) for the selected control is satisfied.
( ) ( ( , ))swu t sign S x tη=
(5.57)
where,
1
( ) 0
1
sign S
−
= = +
for
for
for
0
0
0
S
S
S
<
=
>
5.6.1. Construction of Sliding Mode Observer
The sliding mode control theory is now applied to the rotor flux MRAS
scheme for speed estimation by replacing the conventional constant gain PI
controller. With reference to dynamic model of induction machine and the
speed tuning signal, the time varying sliding surface S(t) is constructed as:
( , ) 0S x t e Ke dtω ω= + =∫ (5.58)
88
where, K is the switching gain which is strictly positive constant. When the
system reaches the sliding surface, the error dynamics at the sliding
surface, S(t) = 0 will be forced to exponentially decay to zero. Thus,
0S e Keω ω= + =& &
(5.59)
e Keω ω= −&
(5.60)
substituting (5.59) in (5.54),
( ) ( )V S S e Keω ω= +& &
(5.61)
time derivative of (5.31) is
ˆ ˆ ˆ ˆ
r r r r r r r reω β α β α α β α βψ ψ ψ ψ ψ ψ ψ ψ= + − −& && &&
(5.62)
substituting the flux values
1 1ˆ ˆ ˆ ˆ ˆ ˆ ˆ( )m m
r r r r s r r r r s r r r r r r r r
r r r r
L Le i i i
T T T Tω β α α β α β α α β β α β α β β α αψ ψ ψ ψ ψ ψ ψ ψ ψ ψ ω ψ ψ ψ ψ= − + − − + − +& &&
(5.63)
ˆre a bω ω= −&
(5.64)
where,
1 1ˆ ˆ ˆ ˆm m
r r r r s r r r r s r r r
r r r r
L La i i i
T T T Tβ α α β α β α α β β α β αψ ψ ψ ψ ψ ψ ψ ψ ψ ψ= − + − − +& &
ˆ ˆr r r rb β β α αψ ψ ψ ψ= +
substituting in (5.61)
ˆ( ) ( )rV S S a Ke bω ω= + −&
(5.65)
the time derivative of V(S) is negative definite for the following conditions:
0
ˆ( ) 0
0
ra Ke bω ω
<
+ − =>
for
for
for
0
0
0
S
S
S
>
=
<
(5.66)
This can be attained when: ˆr eq swu uω = +
(5.67)
89
where, eq
a Keu
b
ω+=
. ( )swu sign Sη=
The block diagram for adaptive mechanism with sliding mode control
is presented in Fig. 5.9.
Fig. 5.9 Block diagram for adaptive mechanism with sliding mode control
The equivalent control, ueq defines the control action which keeps the
state trajectory on the sliding surface and the switching control, usw depends
on the sign of the switching surface and η is the hitting control gain which
makes (5.61) negative definite, whose main purpose is to make the sliding
condition viable and the value of η should be large enough to overcome the
effect of external disturbance [85], [86]. In the equivalent control, the
function b in the denominator may cause problems when its value
approaches zero. By allowing magnetization of the machine before starting
up and by adding a small positive value, this problem can be solved [87].
90
5.6.2. Chattering Problem
In order to account for the presence of modeling imprecision and of
disturbances, the control law has to be discontinuous across S(t). In
practice, switching is not instantaneous and value of S(t) is not known with
infinite precision. This leads to oscillations in the state vector at finite
frequency, referred to as chattering as shown in Fig. 5.10. It results in high
heat losses in electrical power circuits, high wear of moving mechanical
parts and low control accuracy [38], [39]. The controller given will have
chattering near sliding surface due to the presence of sign function. These
drastic changes of input can be avoided by introducing a boundary layer
with width, φ [85]. By replacing sign(s) with sat(S/φ), it becomes,
( / )swu sat Sη φ=
(5.68)
where,
( / )( / )
( / )
sign Ssat S
S
φφ
φ
=
( / ) 1
( / ) 1
if S
if S
φ
φ
≥
<
a) Ideal b) Practical (with chattering)
Fig. 5.10 Sliding mode
A natural solution to reduce the chattering in the estimated speed is
by means of a Low-Pass Filter (LPF) as in (5.69).
' 1
1sw swu u
sµ=
+ (5.69)
91
The LPF also eliminates the spikes due to the differentiation of the
fluxes in the estimated speed. In this work, LPF is used to reduce the
chattering in the estimated rotor speed.
5.7. Implementation of MRAS-SM in FOC Drive
Simulation model for sensorless FOC induction motor drive using
MRAS-PI is developed based on the block diagram shown in Fig. 5.6. The
induction motor model, induction motor controller, MRAS-SM speed
estimator and the PWM inverter are shown in Fig. 5.11. For control
purposes, the motor model is first transformed into the d-q axis model. The
machine variables can be transformed into d-q components by using the �-β
to d-q transformation. The block calculates the voltages applied in the d-q
axis. The d-q axis voltages are converted into the �-β components. Both of
these transformations require the rotor flux angle, which is calculated as
shown in the block diagram. A non-linear adaption mechanism is proposed
to replace the fixed gain PI controller which is conventionally used for rotor
flux MRAS observer. The scheme is based on SM theory, where speed
estimation adaption law is derived based on Lyapunov theory to ensure
Fig. 5.11 Block diagram of FOC induction motor with MRAS-SM
92
estimation stability with fast error dynamics. To evaluate the performance of
the MRAS with SM scheme in sensorless FOC induction motor drive,
simulation model has been developed using MATLAB/Simulink as shown in
Fig. 5.11. The induction motor is fed by a SVM voltage source inverter
having parameters as mentioned in Appendix-I and the rotor speed is
estimated by MRAS-SM speed observer. The values of sliding mode
parameters for the simulation are: K = 100 and η = 0.1 which are obtained
by trial and error.
5.8. Simulation Results and Discussion
In the simulation, for comparing the performance of torque, the motor
starts from a standstill state with reference speed 140 rad/sec and
application of a load torque, TL = 5 Nm at time, t = 1 sec. Figures 5 .12 and
5.13 show the performance of electromagnetic torque with time for cases
with MRAS-PI and MRAS-SM respectively. The motor torque has a high
initial value in the speed acceleration zone, then the value decreases to zero
and increases to the applied load torque in both cases. In both cases
chattering appears in torque response, which will be filtered out by the
mechanical system inertia due to high frequency changes in the torque.
Fig. 5.12 Torque response for MRAS-PI
0
2.5
5
7.5
10
12.5
15
0.0 0.3 0.5 0.8 1.0 1.3 1.5 1.8 2.0Time (sec)
Torq
ue
(Nm
)
ωref =140rad/sec
93
Fig. 5.13 Torque Response for MRAS-SM
Figure 5.14 shows the speed response for start-up and steady state of
the induction machine for three reference speeds, 50 rad/sec, 100 rad/sec
and 140 rad/sec for sensorless FOC drive using MRAS-PI. Figure 5.15
shows the speed responses for three reference speeds, 50 rad/sec, 100
rad/sec and 140 rad/sec for sensorless FOC drive using MRAS-SM. The
applied load torque, TL = 5 Nm at t = 1 sec for all the reference speeds for
both cases. In both the cases, the estimated speed is close to the actual
speed and tracking the reference in all the three speed ranges. MRAS-SM
shows better performance when reference speed is 50 rad/sec, compared to
the case where reference speed is 150 rad/sec. Thus MRAS-SM can provide
better performance in low speed range compared to MRAS-PI.
In Fig. 5.15, for case-2, chattering, a problem associated with sliding
mode control is more when reference speed is 140 rad/sec compared to 50
rad/sec. However, when a load torque is applied, the chattering is more or
less same in all speed ranges.
0.0
2.5
5.0
7.5
10.0
12.5
15.0
0.0 0.3 0.5 0.8 1.0 1.3 1.5 1.8 2.0
Time (sec)
Torq
ue
(Nm
)ωref =140rad/sec
Fig
Fig.
0
20
40
60
80
100
120
140
160
0.0 0.2
Ro
tor
spee
d (
rad
/sec
)
0
20
40
60
80
100
120
140
160
0.0 0.2
Ro
tor
spee
d (
rad
/sec
)R
oto
r sp
eed
(ra
d/s
ec)
94
Fig.5.14 Rotor speed vs. Time using MRAS
Fig. 5.15 Rotor speed vs. Time using MRAS
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6
Estimated speed
Actual speed
Reference speed
Time (sec)
0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6
Estimated speedActual speedReference speed
Time (sec)Time (sec)
ime using MRAS-PI
ime using MRAS-SM
1.8 2.0
Estimated speed
Actual speed
Reference speed
1.8 2.0
Estimated speedActual speedReference speed
95
The error in rotor speed for all the speed ranges is shown in Fig. 5.16.
From the figures, it is clear that MRAS-SM scheme provides accurate speed
estimation in high, medium and low speed ranges.
a) for ωref = 50 rad/sec
b) for ωref = 100 rad/sec
c) for ωref = 140 rad/sec
Fig. 5.16 Speed error for MRAS-SM
-10
-5
0
5
10
0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
Err
or(
rad
/sec
)
Time (sec)
-10
-5
0
5
10
0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
Err
or(
rad
/sec
)
Time (sec)
-10
-5
0
5
10
0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
Err
or(
rad
/sec
)
Time (sec)
96
Variation of current in stator windings for the three reference speeds are
shown in Fig. 5.17. As in the case of electromagnetic torque, the current
signals in stator windings have high values in transient state. In the steady
state, the motor torque only has to compensate the friction and the load
torque.
a) for ωref = 50 rad/sec
b) for ωref = 100 rad/sec
c) for ωref = 140 rad/sec
Fig. 5.17 Stator current for MRAS-SM
-8.0
-4.0
0.0
4.0
8.0
0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
Sta
tor
curr
ent(
A)
Time (sec)
-8.0
-4.0
0.0
4.0
8.0
0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
Sta
tor
curr
entr
(A)
Time (sec)
-8.0
-4.0
0.0
4.0
8.0
0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
Sta
tor
curr
ent
(A)
Time (sec)
97
Figure 5.18 presents the estimated rectangular components of the
rotor flux in the loaded condition in time range between 1.80 sec and 2.0sec
for all the three speed ranges.
a) for ωref = 50 rad/sec
b) for ωref = 100 rad/sec
c) for ωref = 140 rad/sec
Fig. 5.18 Estimated rotor flux for MRAS-SM
-1.5
-1.0
-0.5
0.0
0.5
1.0
1.5
1.80 1.85 1.90 1.95 2.00
Roto
r fl
ux (
Wb
)
Time (sec)
-1.5
-1.0
-0.5
0.0
0.5
1.0
1.5
1.80 1.85 1.90 1.95 2.00
Roto
r fl
ux (
Wb
)
Time (sec)
-1.5
-1.0
-0.5
0.0
0.5
1.0
1.5
1.80 1.85 1.90 1.95 2.00
Roto
r fl
ux (
Wb
)
Time (sec)
98
The speed tuning signal which is the input to the sliding mode
observer for various reference speeds in all the speed ranges are shown in
Fig. 5.19.
a) for ωref = 50 rad/sec
b) for ωref = 100 rad/sec
c) for ωref = 140 rad/sec
Fig. 5.19 Speed tuning signal of MRAS-SM
-0.10
-0.05
0.00
0.05
0.10
0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
e ω
Time (sec)
-0.10
-0.05
0.00
0.05
0.10
0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
e ω
Time (sec)
-0.10
-0.05
0.00
0.05
0.10
0.0 0.2 0.4 0.6 0.8 1.0 1.2 1.4 1.6 1.8 2.0
e ω
Time (sec)
99
Figure 5.20 shows the circular locus of rotor flux vector when the
reference speed is 100 rad/sec. The estimated rotor flux angle for the
reference speed is 50 rad/sec is shown in Fig. 5.21.
Fig. 5.20 Estimated rotor flux locus for MRAS-SM
Fig. 5.21 Estimated flux angle for MRAS-SM
By comparing and observing all the simulation results, it ensures that
the proposed MRAS with sliding mode observer works well when the
parameters are precisely measured and do not change during operation.
-1.5
-1.0
-0.5
0.0
0.5
1.0
1.5
-1.5 -1.0 -0.5 0.0 0.5 1.0 1.5
ψrd(W
b)
ψrd
(Wb)
-2
-1
0
1
2
1.80 1.85 1.90 1.95 2.00
Flu
x a
ngle
(ra
d)
Time (sec)
100
5.10. Summary
In this chapter, a novel adaption mechanism using sliding mode
control is proposed to replace the classical PI controller in MRAS observer
for SVM fed induction machine. The adaption mechanism is based on
Lyapunov theory to ensure stability with fast error dynamics. The chattering
problem of the estimated speed is effectively reduced by using a low pass
filter. The performance of the developed MRAS sliding mode observer has
been illustrated and the speed and torque responses are compared with that
of PI controller by simulation results. The developed adaption mechanism
shows better performance in a wide range of speed compared with that of PI
controller, for which performance deteriorates at low speed. The speed
estimation by the rotor flux MRAS with sliding mode observer has ensured
very good accuracy in both transient and steady state for all ranges of speed
control.
5.11. Publications Related to this Chapter
International Conference: 1. G. K. Nisha, Z. V. Lakaparampil and S. Ushakumari, “FFT Analysis for Field
Oriented Control of SPWM and SVPWM Inverter fed Induction Machine With and
Without Sensor”, International conference on Advance Engineering
Technology(ICAET’13), Mysore, India, pp. 34-43, 27 January 2013.
2. G. K. Nisha, Z. V. Lakaparampil and S. Ushakumari, “Sensorless Vector Control of
SVPWM Inverter fed Induction Machine using Sliding Mode”, IEEE International
conference on Green Technology (ICGT’12), Thiruvananthapuram, India, pp. 29-36,
18-20 December 2012.
International Journal: 1. G. K. Nisha, Z. V. Lakaparampil and S. Ushakumari, “Performance Study of Field
Oriented Controlled Induction Machine in Field Weakening using SPWM and SVM
fed Inverters”, International Review of Modeling and Simulations, vol. 6, no. 3, pp.
741-752, June 2013.
2. G. K. Nisha, Z. V. Lakaparampil and S. Ushakumari, “FFT Analysis for Field
Oriented Control of SPWM and SVPWM Inverter fed Induction Machine With and
Without Sensor”, International journal of Advanced Electrical Engineering, vol. 2, no.
4, pp. 151-160, June 2013.