Quasi-synchronous digital trunked TETRA performance

708 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. 48, NO. 3, MAY 1999

Quasi-Synchronous Digital TrunkedTETRA Performance

Alejandro Moran, Fernando P´erez Font´an, Member, IEEE,Jose M. Hernando Rabanos,Member, IEEE, and Manuel Montero del Pino

Abstract— In some private mobile radio/public access mo-bile radio (PMR/PAMR) applications, there is a stringent needfor high-coverage locations probabilities. A spectrally efficientapproach in this case is the use of several radio transmittersoperating in simulcast mode. There have been several analogmobile radio systems working in this way up to now, but lessis known about the performance of digital trunked radio systemsoperating in simulcast mode. In this paper, predicted digitalTrans-European Trunk RAdio (TETRA) system performanceresults operating in quasi-synchronous mode are presented. Theseresults were obtained by simulation of such a system under a widerange of operational conditions. A comparison is also presentedwith the European analog standard MPT 1327 currently in oper-ation. It has been concluded that quasi-synchronous techniqueswell known in analog PMR/PAMR can also be successfully usedin digital PMR/PAMR applications.

Index Terms—MPT 1327, overlap coverage areas, simulcast,TETRA.

I. INTRODUCTION

SEVERAL area coverage techniques have been tradition-ally used in mobile radio communication system design

to obtain the wanted coverage area with the required locationsprobability: 90%, 95%, . The most common approach towide area coverage is multichannel or frequency reuse as incellular systems. However, other alternatives are available toradio planners such as voting systems, synchronous and quasi-synchronous schemes, etc. [1]. In some private/professionalmobile radio (PMR) and public access mobile radio (PAMR)applications, for example, security, field services, utilities,etc., large coverage areas may not be required and, in manycases, these networks may even be limited to a single fixedradio station. However, in irregular terrain or in dense urbanareas, adequate local coverage cannot be achieved by usinga single transmitter. In these situations, a spectrally efficientcoverage solution is the use of several transmitters with thesame nominal frequency operating in quasi-synchronous orsimulcast mode. Up to now there is some experience in theoperation of analog quasi-synchronous transmission systems,but less is known about the performance of digital quasi-synchronous systems. In this paper, the quasi-synchronous ap-proach is addressed in the framework of the new pan-European

Manuscript received February 14, 1997; revised June 16, 1997.A. Moran and F. P. Fontan are with the Department of Communications

Technologies, University of Vigo, E-36200 Vigo, Spain.J. M. H. Rabanos and M. Montero del Pino are with the Telecommunications

Engineering School, Polytechnic University of Madrid, Madrid, Spain.Publisher Item Identifier S 0018-9545(99)04021-9.

digital trunked system standard Trans-European Trunk RAdio(TETRA) [2]. Performance characteristics can be obtainedtheoretically by a pure analytical approach, but this is fairlyabstract and complex. Thus, it has been decided to carry outsimulation studies which are easier to implement and give aphysical insight of the simulcast system operation.

II. BACKGROUND

Quasi-synchronous transmission using two or more ra-dio stations is a technique used to improve area coverageprobability when a single transmitter is insufficient. A quasi-synchronous transmission system basically consists of the useof two or more radio stations transmitting the same signalstoward the desired service area using the same radio channels.Fig. 1 illustrates a six-cell radio system using a multifrequencyscheme and where one of the cells is not adequately servedby a single radio station due to signal blockage problems.In this case, in order to fill in the coverage gaps left by themain or master radio station in the cell two more slave quasi-synchronous stations are used to improve coverage quality.

The probability that two or more stations simultaneouslyexperience deep shadowing or blockage is greatly reduced.Small frequency offsets (a few hertz to a few tens of hertz) areallowed between the transmitted radio frequency (RF) carriersat the different coverage radio stations. On the other hand, theuplink (mobile-to-base) operates as a voting system where thebest received signal at the different radio stations is selected.The main aim of a quasi-synchronous system will thus be toachieve macrodiversity as is shown in Fig. 2. The use of thesame radio channels at the different fixed stations provides theadded advantage of frequency economy. No extra frequencyassignments are required for local coverage.

When one of the received signals is greater than the others,no cochannel interference is experienced since this strongersignal dominates over the rest due, for example, to the captureeffect in FM systems. However, in the overlapping zones(Fig. 3), i.e., where the mobile receives approximately thesame signal levels from more than one transmitter, problemsmay appear, and, hence, special care must be taken to ensurelow error rates.

The reception of similar amplitude signals on the samecarrier at the mobile causes interference problems. Added tothe received multipath structure caused by the surroundingclutter there will be another source of multipath due to thetransmission of the same signal on the same nominal RF carrierfrequency from two or more radio stations. Fig. 4 illustrates

0018–9545/99$10.00 1999 IEEE

MORAN et al.: QUASI-SYNCHRONOUS DIGITAL TRUNKED TETRA PERFORMANCE 709

Fig. 1. Multifrequency area coverage scheme combined with a simulcast cell.

Fig. 2. Macrodiversity gain effect for irregular terrain areas in a quasi-synchronous system.

Fig. 3. Capture zones and overlap zones in a quasi-synchronous system.Fig. 4. “Artificial” multipath effect due to the simulcast operation.


Fig. 5. Two-station simulcast system configuration.

the effect of the reception of two equal amplitude carriers onthe same radio channel. Multipath-like fades appear on theoverall received signal giving rise to noise bursts where thebit error rate (BER) is high. Due to the slight frequency offsetallowed between the RF carriers, the deep nulls caused willchange position with time. A static terminal will observe aslowly fading signal.

The quasi-synchronous technique has been successfullyused in numerous applications ranging from simple repeatersystems with subaudio tone signaling [3] to analog trunkedsystems (MPT 1327) [4].

In a quasi-synchronous system, digitized voice and dataare sent from the trunking system controller (TSC) to thetransmitters via land lines or point-to-point radio links. Suchlines may have different lengths as shown in Fig. 5. Theseline length differences give rise to different delays. Similarly,there will be different propagation delays from the radiostations to the mobile receiver as is also shown in Fig. 5if the mobile is closer to one of the radio stations. Bothline delay and radio propagation delay plus other delaysources are accumulated in the phase of the received signal.These delay differences can be critical to the performance ofthe system, thus, appropriate delay equalization is required.Another critical parameter that will be paid attention to isthe offset between base-station RF carrier frequencies.RF carrier nominal frequencies may slightly drift betweenmaintenance adjustment periods. However, a maximum offsetlimit must never be exceeded if adequate system performanceis to be preserved.

In this paper, second-generation European digital trunkedsystem TETRA [5] performance studies are presented forsystems using a quasi-synchronous approach to enhance localcoverage. Both bit and codeword error rates (CWER’s) havebeen assessed for the different logical channels defined in theTETRA standard.

III. B RIEF OVERVIEW OF THE TETRA SYSTEM [5]

TETRA is an European Telecommunications Standardiza-tion Institute (ETSI)-defined system for PMR/PAMR applica-tions with far more enhanced features than the existing analogstandards, i.e., MPT 1327 (UK DTI). Two versions of thesystem have been defined:

1) TETRA voice data;2) TETRA packed data optimized.

The characteristics of the TETRA voice data version arereviewed.

A large number of bearer services and teleservices havebeen defined providing the opportunity to implement a widerange of communication applications.

The system uses a frequency-division multiple-access(FDMA) structure with 25-kHz RF channels both in theuplink and downlink directions. On the other hand, each RFchannel implements a time-division multiple-access (TDMA)structure supporting four logical channels (for voice, data,or signaling). A gross bit rate of 36 kbps and filtered/4-shift-DQPSK modulation are used. In order to adequatelyreject adjacent channel power, limit intersymbol interference,and ease receiver synchronization a raised cosine filter isemployed with a rolloff factor of 0.35.

RF bursts with the general structure shown in Fig. 6 arefitted into each of the four TDMA time slots. The uplinkbursts are preceded by a preamble used for power rampingand power amplifier linearization followed by a postamble forpower ramping. In the downlink transmission is continuous,and this means that no power ramping time is needed andthe available extra time intervals are used to broadcast anadditional training sequence between downlink bursts in orderto improve reception quality in the mobiles.

In the system there exist seven types of bursts which arefitted into time slots making up a frame structure. The bursts


Fig. 6. TETRA system general burst scheme.

Fig. 7. Error protection schemes for the different TETRA logical channels.

types defined in the TETRA standard are the following:

1) control uplink;2) linearization uplink;3) normal uplink;4) normal continuous downlink;5) synchronization continuous downlink;6) normal discontinuous downlink;7) synchronization discontinuous downlink.

Additionally, a multiframe structure of 18 frames is definedwhich allows the introduction of associated control channelstogether with their corresponding traffic channels and a hy-perframe to facilitate the monitoring of adjacent cells by themobile and accommodate a cryptographic scheme.

Two basic types of logical channels have been defined:

1) traffic channels, carrying speech or data in circuitswitched mode;

2) control channels, carrying signaling messages and packetdata.

Different traffic subchannels are defined for speech or dataapplications with several data rates:

1) speech traffic channel (TCH/S);2) speech or data traffic channels

a) 7.2-kbps net rate (TCH/7.2);

b) 4.8-kbps net rate (TCH/4.8);

c) 2.4-kbps net rate (TCH/2.4).

There are five categories of control channels.

1) Broadcast control channel (BCCH), comprising the fol-lowing.

a) Broadcast network channel (BNCH).

b) Broadcast synchronization channel (BSCH).

2) Linearization channel (LCH), with two subchannels.

a) Common linearization channel (CLCH) shared by allthe mobiles in the uplink direction.

b) Base-station linearization channel (BLCH) downlinkused by the base station.


Fig. 8. Detailed TETRA simulcast system simulation layout.

3) Signaling channel (SCH) shared by all the mobiles,which is further divided into three categories dependingon the size and direction of the messages.

4) Access assignment channel (AACH) downlink. It is usedto indicate the assignment of the uplink and downlinkslots.

5) Stealing channel (STCH), which is bidirectional. It isassociated with a TCH and temporarily “steals” a part ofthe TCH capacity in order to transmit control messageswhen fast signaling is required.

In Fig. 7, the coding schemes used for the different channeltypes are summarized.

IV. SIMULATION SCHEME

In order to carry out simulations, the transmitter and receiverblocks were implemented following the TETRA specifications[2]. Error probabilities were computed by means of an error-check block. In order to account for multipath propagationeffects and quasi-synchronous transmission, a channel blockwas introduced in the simulations as is shown in Fig. 8. Twoquasi-synchronous transmitters were simulated and only thedownlink (mobile reception) was studied.

The transmitter block was fed by a PRN sequence generator.Several block and convolutional coders were implementedaccording to Fig. 7. The modulation scheme was/4-shiftDQPSK with a gross bit rate of 36 kbps in a 25-kHz RFchannel bandwidth. A square-root-raised cosine filter with arolloff factor of 0.35 was placed after the modulator.

At the receiver, a pass-band filter was placed right beforethe demodulator. A simple receiver without an equalizer wasimplemented. The phase of the incoming signal was integratedover a symbol period in order to compute its change during

TABLE I

the last s (symbol period). Only four-phase change valuesare allowed in the /4-shift-DQPSK modulation scheme. Inorder to compute the received bits, a decision device withthe appropriate thresholds was placed after the sampler. Idealsynchronization was assumed. In the simulations, a 10-m/sspeed (36 km/h) was considered for the mobile receiver.

Quasi-synchronous operation is defined by three parameters,namely, therelative delays between modulating signals dueto land line and propagation delays, therelative receivedamplitudes, and the frequency offset of the carriers. Thetransmitter output was fed to two channel blocks representingthe signals received from two fixed radio stations. Thesesignals were added at the receiver. Finally, Gaussian noisewas also added.

Narrow-band channel propagation conditions were consid-ered. This assumption is acceptable when the multipath delayspread is much smaller than the inverse of the signal band-width, in this case, BW kHz and 1/BW s. Typicaldelay spread values will hardly reach a few microseconds asshown in Table I [5]. From the table, it is seen that the narrow-band assumption holds at least for rural and typical urbanareas.

The channel amplitude and phase variations were generatedusing a simple geometrical model [6], [7]. The model producesthe complex envelope variations due to multipath. Typically, aRayleigh probability density function will be followed by thereceived amplitude when the direct signal is not available.In the frequency domain, the classical U-shaped Doppler


(a) (b)

Fig. 9. BER versusEb=No. Rayleigh channel (- - - ), simulcast operation:A = 10 dB (- - -), A = 3 dB (- - - -), andA = 0 dB (-�-�-�-�).(a) Delay = 0 �s. (b) Delay= 3:5 �s (6% of Ts).

spectrum is present. If the direct ray is considered, a Ricedistribution will characterize the amplitude variations. Themodel assumes that the mobile is surrounded by a crown ofpoint scatterers with a uniform azimuth distribution. Thesescatterers are illuminated by a distant transmitter. The signalis scattered at each point in the crown before it reachesthe mobile receiver through multiple paths. A direct raymay be considered at will, thus, producing Rice or Rayleighdistributions. The total received signal is the coherent sumof all scattered rays plus the direct ray, if it exists. Channelsamples along the mobile route are produced with a spatialseparation of less than , a typical value would beThis spacing provides sufficiently close samples so that noreceived signal deep nulls are lost in the sampling process.

In order to carry out transmission system performance simu-lations a travelled distance-to-time conversion must be carriedout according to the assumed vehicle speed. Rayleigh andRice fading channels are slowly variant for moderate vehiclevelocities when relatively low binary transmission rates areconsidered. This means that a number of transmitted symbolswill “see” approximately the same channel amplitude andphase. Eight samples per symbol were used in the simulations.In order to match this sample rate, the same number of channel(amplitude and phase) samples per symbol were produced bymeans of simple interpolation out of the simulated channelsamples having a spatial separation of

The simulations presented in this paper were carried outusing the Ptolemy (University of California at Berkeley)simulation package environment. The carrier frequency wasset to 400 MHz close to one of the bands internationallyforeseen for allocation to the TETRA system [5]. The meanamplitude ratio in decibels of the two incoming signals was

Fig. 10. BER versus frequency offset of aEb=No = 35 dB. A = 0 dB (-- - - -), A = 3 dB (- - - -), A = 5 dB (-�-�-�), andA = 10 dB (� � � � �):


Fig. 11. BER versus frequency offset of the 2.4-kbps TCH.A = 0 dB (- - - - -), A = 3 dB (- - -), andA = 10 dB (� � � �):

TABLE II

TABLE III

set to a range of to dB in order to simulatethe system performance in the overlapping coverage areasof two transmitters. RF carrier offsets up to 3000 Hz anddelays up to one half of the symbol period were studied.The assumption was made that the location of the receiverwas midway between the two transmitters. This means thatthe delay difference in the plots is due fundamentally tounequalized line delay differences.

V. RESULTS

Fig. 9 shows BER values for different levels fora TETRA system channel before error protection decodingassuming a Rayleigh channel. Two cases are shown in thefigure [see Fig. 9(a)], where the relative linepropagationdelays of the two received signals are zero and [see Fig. 9(b)],where the relative delay is 3.5s (6% ). The parameter inthe figure is the relation in logarithmic units between the twoquasi-synchronous signals at the input of the receiver. FromFig. 9, it can be observed how, as the ratio grows,the curves corresponding to the Rayleigh channel (dB, i.e., one signal dominates) depart from those belongingto quasi-synchronous operation (10 dB:and dB). The reason for this is that as the trans-mitted power is increased, the Rayleigh effects are partiallymitigated, but not the interference effects due to the simulcastoperation.

For the simulation results shown below, the ratiowas set to a “good” working level of approximately 35 dB,


Fig. 12. BER versus frequency offset of the 4.8-kbps TCH.A = 0 dB (- -- -), A = 3 dB (- - - -), andA = 10 dB (� � � �):

a point where the transmissions are mainly impaired by thequasi-synchronous operation effects. This selection is justifieddue to the fact that interference between the two stations willbe more likely in mutual visibility areas. A Rayleigh modelwas assumed in the simulations since it represents the worstcase situation.

In Fig. 10, results are presented for different delay (Table II)frequency offset combinations. In the figure it is clearlyshown how the BER deteriorates dramatically with increasingfrequency offsets and delays, especially when the two receivedsignals present similar amplitudes ( dB).

A. Traffic Channels (TCH) Results

TETRA traffic channels (TCH’s) use data protectionschemes based on punctured convolutional codes. In aGaussian channel, the performance of a convolutional codecan be obtained as a function of the raw BER (beforechannel coding) and the properties of the code. Althoughthe channel found in a mobile communications environmentis not Gaussian at all, the interleaving scheme employed,allows the use of this approach to quantify BER’s in theTCH’s. The approximation used here was [8]–[10]

Fig. 13. WER versus offset plots for relative delaysT = 0; Ts=16; Ts=8;andA = 0 dB (- - - - -), A = 3 dB (- - - -), A = 5 dB (-�-�-), andA = 10dB (� � � �):

where is a constant depending on the code andis thefree distance of the code. In Table III, the values used areshown.

The results obtained for the 2.4- and 4.8-kbps traffic chan-nels are shown in Figs. 11 and 12 for the delay values inTable II. The solid horizontal line represents the thresholdperformance level set by the TETRA standard for each channeltype (BER and ). A strong influence of therelative delay (line delay) can be observed. Once a givendelay value of approximately 10% of is exceeded, largeBER values were observed no matter what the carrier offsetvalue was. It was also observed that for moderate values ofthe delay the BER performance degrades rapidly when thefrequency offset exceeds 100 Hz.

B. Access Assignment Channel Results

Separate simulations were carried out for the access assign-ment channels (AACH) since, instead of a convolutional code,it uses a Reed–Muller (30,14) block code. This code presentsa minimum Hamming distance of eight, and, thus, it can detectup to seven errors and correct up to three errors.

The word error ratio (WER) was studied first for differentfrequency offsets and delays. Fig. 13 shows three WER versus


Fig. 14. Error probabilities versus number of errors in the 30-b codewords for offsets: 10 Hz (- - - - -), 30 Hz (- - -), 100 Hz (-�-�-�), and 300 Hz(� � �) for the different delay-relative amplitude ratios in Table IV.

TABLE IV

frequency offset plots for different delays and amplituderelation parameter values. It can be observed that for offsetssmaller than 100 Hz, WER values remain practically constant.It can also be observed that the delay is the most sensitiveparameter when setting up a simulcast radio network.

The WER is not, however, a relevant parameters providedthat the number of errors in a codeword do not exceed thedetection capabilities of the block code used. This can be seenfrom Fig. 14 where the error probabilities in a 30-b codewordare given for the parameters shown in Table IV.

From Fig. 14, it can be concluded that as the frequencyoffset increases the probability of having a larger number of

errors in the 30-b codewords decreases. That is, as the offsetgrows the WER also grows, however, these codewords willcontain a smaller number of bits in error.

Fig. 15 clarifies this. In the figure, two error time seriesare presented. These series correspond to the transmission of20 000 30-b codewords for a relative delay of 7s (12.5%

), dB, and frequency offsets of 10 and 300 Hz. Itcan be observed that for a 300-Hz offset, errors are distributedmore uniformly among transmitted codewords, and althoughthe number of erroneous words increases, the average numberof errors per codeword is smaller. In this way, the probabilityof exceeding seven errors in a codeword is smaller for largeroffsets. In the simulated case, the probability of exceedingseven errors is 1.5 10 for a 300-Hz offset while for a10-Hz offset is 5.2 10 .

The transmission performance of the AACH channel mustbe assessed in terms of two error parameters: the messageerasure ratio (MER) or probability of receiving and detect-ing an erroneous message and the probability of undetected


Fig. 15. Error time series forT = 7 �s � 12.5%Ts, A = 10 dB, and frequency offsets of 10 and 300 Hz.

Fig. 16. MER versus offset for relative delaysT = 0; Ts=16; Ts=8; andA = 0 dB (- - - - -), A = 3 dB (- - - -), andA = 10 dB (-�-�-).

erroneous message (PUEM) or probability of receiving anerroneous message and mistaking it for a legitimate one.

Fig. 17. PUEM versus offset for relative delaysT = 0; Ts=16; Ts=8; andA = 0 dB (- - - - -), A = 3 dB (- - - -), andA = 10 dB (-�-�-).

Figs. 16 and 17 show several MER and PUEM versusfrequency offset plots for different delays and amplituderations. It can be observed how the MER and the PUEMpresent opposite behaviors as the frequency offset increases.


Fig. 18. MPT 1327 forward control channel time slot structure and codeword structure.

For offset values below 100 Hz, the MER increases slowlywhereas the PUEM does the opposite. For higher offset values,sharper variations of both parameters are observed.

As for the relative delay, it is again verified that this isthe most sensitive parameter. Values on the order ofcan still be tolerated, but for higher values performance isseriously impaired.

C. MPT 1327 System Forward Control ChannelResults (Analog Trunked System)

It is interesting to compare the performance of a digitalsystem such as TETRA with an analog system such as theMPT 1327 system which is considered as a de facto standard inEurope for trunked radio networks. In order to present similarsimulations to those already presented for the TETRA systems,first a brief overview of the MPT 1327 standard must be made.This is an analog system using FM modulation in the trafficchannels which are designed specially for voice applications.As for the control channels the data flow with a bit rate of 1200b/s undergoes a double modulation process. First, an audiofrequency carrier is frequency-shift-keyed (FSK) modulated(a binary “0” is represented by the frequency 1800 Hz andthe binary “1” by 1200 Hz), and then a FM modulationis followed. Other modulation characteristics can be foundin references such as (UK) DTI’s MPT 1327, MPT 1343,and other associated documents. Two-frequency channels of12.5-kHz bandwidth have been assumed in the simulationssince this is the most common channel separation in thebands around 400 MHz for trunked applications. For TETRA,however, a channel bandwidth of 25 kHz was considered.

The MPT 1327 forward control channel provides the timereference for a slotted Aloha multiaccess scheme. Transmis-sions in the downlink are arranged in 106.67-ms time slots.Two 64-kb codewords are sent in each time slot. The first wordis called control channel system codeword (CCSC) whichidentifies the system to the mobile terminals and provides therequired synchronization for the reception of the second wordin the slot: address codeword (AC) (Fig. 18). Additionally,if required, a time slot may contain two data codewordswhich are used to send short data messages across the controlchannel. Also, in Fig. 18 the codeword structure used in the

TABLE V

MPT 1327 system is presented. Bits 2–48 carry informationwhereas bits 49–64 carry data protection bits generated with a(63, 48) block code using the following generating polynomial:

where the 64 codeword is completed by adding one single bitin order to produce an even parity codeword. The decodingalgorithm is not specified in the standard, however, severaloptions are possible. Table V lists the options studied in thispaper.

Similarly to what was done for the TETRA system, simula-tions are shown where the influence of three major installationparameters are assessed: the frequency offset, the relationbetween the received amplitudes, and the relative delayexpressed as a percentage of the symbol periodwhich,in this case is 833 s.

A preliminary BER study was carried out for differentvalues of the parameter (10, 3, and 0 dB) and the delayparameter ( and and s ). These simulations areshown in Fig. 19, where the Rayleigh case is compared withthree simulcast Rayleigh situations. It can be observed howthe simulcast operation greatly deteriorates the performance ofthe system. As for the TETRA case, the working point used tocarry out the evaluation of the performance reduction effectsfor different offsets, delays and relative received levels was setto approximately 35 dB. Again, it was deemed that simulcasteffects are more important in the mutual visibility areas fromboth transmitters where a high signal level is received fromboth transmitters.

In Fig. 20, the simulation setup for the MPT 1327 analogtrunked system is presented. Before proceeding to analyze the


(a) (b)

Fig. 19. BER versus C/N. Rayleigh channel (- - -), A = 10 dB (- - - - -), A = 3 dB (- - -), andA = 0 dB (-�-�-). (a) Delay T = 0 �s.(b) Delay T = 26 �s (3% Ts).

Fig. 20. Detailed MPT 1327 simulcast system simulation layout.

composite behavior of the modulation plus channel codingscheme used in the forward control channels, row BER studieswere performed. The simplest receiver structure was assumedfor the receiver (Fig. 21). The phase-locked loop (PLL)-basedFM demodulator was modeled by a differential equation [11].After the PLL, a symbol period integrator was implementedproviding at its output the phase increment produced. Acomparison device was placed at the end of the receivingchain. Ideal synchronization recuperation was assumed.

In Fig. 22, results for different offsets and relative ampli-tudes are shown. These results were calculated for the working

Fig. 21. MPT 1327 receiver block diagram.

point selected of C/N dB and for the delays given inTable VI.

From the observation of Fig. 22, it can be concluded thatthe BER is strongly dependent on the frequency offset. Values


Fig. 22. BER versus frequency offset.A = 0 dB (- - - - - -),A = 3 dB (- - - -),A = 5 dB (-�-�-�), andA = 10 dB (� � � � � �) for the delays in Table VI.

TABLE VI

higher than 30 Hz produce a great deterioration of the BERwith values well above 10 As for the delay, the BERmaintains acceptable levels if the delay does not exceed 10%of , and past this value transmission performance is greatlyimpaired.

The BER parameter does not allow a complete understand-ing of the transmission chain performance under simulcastconditions. It is important to study the error distributionwithin each 64-b codeword. Fig. 23 presents the word errordistribution for the following conditions: BER ,

dB, and delay s In spite of the high BERvalue considered, the probability of having more than three

errors in each 64-b codeword is smaller than 6%. For the(64, 48) code used, 3-b error words can be handled withoutdifficulty. The WER was also studied for different conditions.The results are shown in Fig. 24. However, the WER is not asignificant parameter since the capabilities of the decodermust be accounted for. In spite of having large WER values,if the number of errors in each codeword is small enoughoverall performance will be maintained. The next step wasto perform a study including the coder properties. Table Vgives several decoder implementation options. Fig. 25 presentsCWER simulation results for the different coder implementa-tions in Table V and for several simulcast conditions (offsets,delays, and relative signal levels). The left column plots inFig. 25 represent the probability that a codeword in error isdetected and the right column plots represent the probability ofmistaking a codeword in error for a correct one (false alarm).In each plot, four lines are shown presenting the followingoffset-delay combinations: (5 Hz, s ), (5 Hz, s

), and (50 Hz, 26 s) and (50 Hz, 52 s). It can clearlybe observed how as the correcting power of the decoder is


Fig. 23. Distribution of the number of error in a codeword.

(a)

(b)

Fig. 24. WER versus frequency offset for (a)T = 0 �s and (b)T = 52 �s. A = 0 dB (- - - - - -), A = 3 dB (- - - -), A = 5 dB (-�-�-�-),and A = 10 dB (� � � �):


Fig. 25. CWER probability versus A parameter for offsets: 5 Hz (- - -), 50 Hz (- - - - - - -), and delaysT = 26 �s andT = 52 �s.

increased (from top to bottom of the figure), the probability ofwords in error decreases. On the other hand, the probabilityof mistaking an erroneous word for a good one increases.Only for the last case (soft decoding) much more powerfuland complex a good balance between both error conditions isachieved: for example, for an amplitude ratio dB only2% of the transmitted codewords are in error and only 0.5%are mistakenly interpreted as legitimate codewords. It mustbe remembered that these results correspond to nonequalizedconditions. From the analysis of the figure, it is clear that asthe frequency offset increases the error distribution amongsttransmitted codewords becomes more uniformly distributed,that is, although the number of codewords in error increasesthe number of these words that present a number of errorsabove the detection threshold is smaller. Thus, moderate offsetvalues facilitate the detection process. However, it has alsobeen observed that for offset values above 50 Hz, the systemperformance deteriorates rapidly.

Other factors to be borne in mind are the usual conditionsunder which radio transmissions will be performed. If termi-

nals are usually in motion when making calls other factor to betaken into account which increases the relative offset betweensimulcast transmissions is the Doppler shift (that is velocitydependent). The worst case will be when a mobile travels alongthe straight line defined by both stations. Another factor to betaken into account is that FM-modulated voice transmissions(traffic channels) present different sensitivity to delays andfrequency offsets. It has been experimentally observed thatfor equalized paths carrier offsets must never exceed a fewHertz [12]. A possibility is to setup the installation of analogsystems with different offsets for control and traffic channels.

VI. CONCLUSIONS

In this paper, digital pan-European trunked system stan-dard TETRA performance has been evaluated when a quasi-synchronous transmission scheme is used to improve areacoverage probability. Two relevant conclusions may be drawnfrom this study. For one, the delay differences due to differenttransmission line lengths from the trunking system controller


to the simulcast stations severely degrade the performance ofquasi-synchronous systems. This means that adequate delayequalization is required. Maximum relative delays of 12s(20% ) and carrier frequency offsets of 100 Hz were shownto be acceptable.

Second, it has also been shown that an appropriate selectionof the carrier frequency offset value makes it possible tocounteract the adverse characteristics of the fast fading channeland the quasi-synchronous interference. Adequate offsets willwhiten the sequence of errors and facilitate the task of the errorprotection code. A maximum frequency offset of about 100 Hzcan be permitted in typical quasi-synchronous installations.

A similar study was carried out for the forward controlchannel of the European analog MPT 1327 system using 12.5-kHz-bandwidth channels and a double modulation process forthe signaling data: audio subcarrier FSKFM. This systemwas shown to be more vulnerable to simulcast transmissionconditions than the TETRA system. Delays exceeding 10%

or carrier frequency offsets exceeding 50 Hz produceunacceptable system performances.

It can be concluded that quasi-synchronous techniqueswhich have been successfully implemented in the past foranalog PMR applications including analog trunked systems,i.e., MPT 1327, can also be applied to second-generationdigital pan-European trunked systems in order to benefit fromthe improved coverage probabilities and enhanced spectralefficiency obtained by this macrodiversity scheme.

REFERENCES

[1] R. J. Holbeche, “Area coverage techniques,” inLand Mobile RadioSystems, R. J. Holbeche, Ed. (IEEE Telecomms. Series no. 14), ch.2, 1995.

[2] Trans-European Trunk Radio (ETSI), “Documents 05.01, 05.02, 05.03,05.04, 05.05, and 05.08,” Nov. 1993.

[3] G. D. Gray, “The simulcasting technique: An approach to total-arearadio coverage,”IEEE Trans. Veh. Technol., vol. VT-28, pp. 117–125,May 1979.

[4] BS 770 MPT 1327 VHF-UHF Repeater-Base Station, Bosch, 1996.[5] F. Gourgue, “Air interface of the future European fully digital trunk

radio system,” inIEEE Veh. Technol. Conf., 1993, pp. 714–717.[6] F. Perez-Fontan, A. V. Castro, and J. P. V. Poiares Baptista, “A

simple numerical propagation model for nonurban mobile applications,”Electron. Lett., vol. 31, no. 25, Dec. 1995.

[7] F. P. Fontan, J. Pereda, M. J. Sedes, M. A. V. Castro, S. Buonomo,and P. Baptista, “Complex envelope three-state Markov chain simulatorfor the LMS channel,”Int. J. Satel. Commun., vol. 15, no. 1, pp. 1–15,Jan./Feb. 1997.

[8] S. Benedetto, E. Biglieri, and V. Castellani,Digital TransmissionTheory. Englewood Cliffs, NJ: Prentice-Hall, 1987.

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Alejandro Mor an was born in La Coruna, Spain,in 1972. He received the Telecommunications En-gineer degree from the University of Vigo, Vigo,Spain, in 1996. He is currently working towardsthe Ph.D. degree in the field of multicarrier spread-spectrum communications at the University of Vigo.

Fernando Perez Fontan (M’96) was born in Vi-lagarcia de Arousa, Spain, in 1959. He receivedthe Ph.D. degree from the Polytechnic Universityof Madrid (UPM), Madrid, Spain, in 1992.

He is a Lecturer at the Telecommunications En-gineering School, University of Vigo, Vigo, Spain.

Jose M. Hernando Rabanos (M’94) received thePh.D. degree in telecommunications engineeringfrom the Polytechnic University of Madrid (UPM),Madrid, Spain.

He is a Professor in the Signals, Systems andRadiocommunications Department, UPM. He haswritten two textbooks on radio transmission andmobile communications. His professional activity isdevoted to teaching and research in radiocommuni-cations. He led research projects in radio planningand radio propagation. He has published several

technical papers in international journals and cooperates with radio propa-gation and mobile services ITU-R study groups.

Manuel Montero del Pino was born in Lillo-Toledo, Spain, in 1936. He received the Telecommu-nications Engineering degree and the Ph.D. degreefrom the Polytechnic University of Madrid (UPM),Madrid, Spain.

He was with Iberia Airlines as a CommunicationsDepartment Head and with Telefonica as SystemsDepartment Director. He is the author of numeroustelecommunications and control articles, books, andpatents. He has participated in several ITU-R andITU-T working groups. Currently, he is a Professor

at the Telecommunications Engineering School, UPM.