Progress in optical modulation formats for high-bit rate WDM transmissions

IEEE JOURNAL OF SELECTED TOPICS IN QUANTUM ELECTRONICS, VOL. 12, NO. 4, JULY/AUGUST 2006 469

Progress in Optical Modulation Formats forHigh-Bit Rate WDM Transmissions

Gabriel Charlet

(Invited Paper)

Abstract—With the progress of optical communication systems,and especially the constraints brought by wavelength division mul-tiplexing (WDM) transmissions and increased bit rates, new waysto convert the binary data signal on the optical carrier have beenproposed. It appears clearly now that several of the methods pro-posed by research laboratories will be applied into commercialproducts soon due to the large improvements generated. This pa-per intends to summarize some of the most interesting proposedmodulation formats for high bit rate (especially 40 Gb/s) WDMtransmissions.

Index Terms—Amplitude shift keying, optical communication,optical modulation, phase shift keying.

I. INTRODUCTION

A T THE beginning of optical communications, binary elec-trical data were converted into optical data by a simple

way: a binary electrical “0” was converted into a low-intensityoptical signal, whereas a binary electrical “1” was convertedinto a higher intensity optical signal. In order to cope with thecontinuous increased bandwidth requirement, two main meth-ods have been used. The first one was to increase the channelbit rate from 155 to 622 Mb/s, 2.5, 10 Gb/s, and now 40 Gb/s byusing the progress in time division multiplexing (TDM) tech-nologies, and the second one was to increase the number ofwavelengths transmitted over a single fiber from one to sev-eral tens or hundreds by using wavelength division multiplex-ing (WDM) technologies. However, with the increase of datarates, more advanced ways to encode data into optical signal,i.e., more advanced modulation formats, have been proposed.But few of them have been really deployed in optical trans-mission systems except over some ultralong-haul submarineroutes.

Current installed optical communication systems use WDM,i.e., several wavelengths, each of them carrying different infor-mation multiplexed into one single optical fiber and transmittedas depicted in Fig. 1, where four wavelengths are modulatedby electrical data, before being multiplexed. After transmissionthrough a single optical fiber, the optical signal is demultiplexed(each wavelength is sent to a specific output fiber) before beingdetected by four receivers (one for each wavelength).

The most widely deployed backbone systems are workingwith wavelengths modulated at a bit rate of 10 Gb/s, with thesimplest modulation format, i.e., non-return-to-zero (NRZ).

Manuscript received December 15, 2005; revised April 6, 2006.The author is with Alcatel Research and Innovation, 91460 Marcoussis,

France (e-mail: [email protected]).Digital Object Identifier 10.1109/JSTQE.2006.876185

Fig. 1. Principle of WDM transmission.

Up to approximately 100 wavelengths can be multiplexed intoa single fiber with a channel spacing of 100 or 50 GHz. Thiscorresponds to an information spectral density from 0.1 to0.2 b/s/Hz. In the coming years, the bit rate per wavelengthis expected to increase from 10 to 40 Gb/s on the networkof telecommunications operators to fulfill the capacity re-quirement while keeping the same channel spacing. This willintroduce new constraints, and new modulation formats couldbe useful to successfully surmount them.

As various applications can be targeted for 40-Gb/s transmis-sions, various modulation schemes are likely to be introduced.We could split the optical transmission market into at least threesegments, each of them having various constraints. The firstone is very short reach (VSR), i.e., transmission distance below2 km. For this application, the main constraints on the transmitterand receiver are low cost, low power consumption, and reducedfootprint. The second one is metropolitan and regional market.These systems are used to dispatch or collect data around largecities or between neighboring cities and thus explain the moder-ate reach encountered, between 50 and 600 km, typically. One ofthe main requirements of such networks is to be able to dynam-ically allocate the resources, especially by adding or droppingwavelengths at the various nodes (located within each main city)of the network. Here, WDM techniques are used in conjunctionwith optical amplifiers, as for the last segment which is long-hauland ultralong-haul systems. These systems interconnect majorcities and metropolitan/regional networks. Here, the transmis-sion reach can be much longer, between 500 km up to severalthousand kilometers (and even 12 000 km for some submarinecables).

Of course, system constraints are completely different. Forreduced reach systems, transponder cost is one of the majorconcerns. But the cost of the transponder is less sensitive forregional and even less for ultralong-haul terrestrial or submarinesystems (where the line cost can become predominant), and thusmore complex technologies could be used in the transpondersto surmount the technical challenges.

1077-260X/$20.00 © 2006 IEEE

470 IEEE JOURNAL OF SELECTED TOPICS IN QUANTUM ELECTRONICS, VOL. 12, NO. 4, JULY/AUGUST 2006

Fig. 2. Impact of chromatic dispersion on pulses’ integrity.

II. OPTICAL COMMUNICATION LIMITATIONS

In order to understand the requirements on the modulationformats, it is important to know the current limitations of opticalcommunication systems.

The objective of this section is to describe the various phe-nomena that limit the transmission reach and to explain thestrategies applied to compensate or mitigate them.

A. Linear Effects

The most challenging limitation encountered in optical trans-mission in the 1980s (or even before) was the attenuation ofoptical fibers. Thanks to the progress of fiber manufacturing, theattenuation of fibers has gone down to slightly below 0.2 dB/kmat 1.55 µm (the wavelength, where the attenuation is nearly atits minimum). After 100 km, the loss is 20 dB, i.e., the powerhas been divided by 100.

Optical amplifiers are therefore, mandatory when propagatingover long distances. Optical amplifiers based on erbium-dopedfibers have been developed at the end of the 1980s [1] and arenow widely deployed within terrestrial and submarine systems.They provide high gain, large optical bandwidth, and low-noisefigure (NF), and several tens of erbium-doped fiber amplifiers(EDFA) can be cascaded.

Once periodical amplification is implemented, the impact offiber chromatic dispersion [2] arises as one of the most limitingimpairments. Chromatic dispersion refers to the phenomenonby which the various spectral components of the signal do nottravel at the same speed and corresponds to the wavelength (orfrequency) dependence of the fiber refractive index. Owing tothis effect, the pulses broaden, leading to bit-to-bit overlaps andthe information after detection may be corrupted because of theintersymbol interference. If we consider a conventional 10-Gb/ssignal (spectrum width ∼0.15 nm), the maximum transmissiondistance over [standard single mode fiber (SSMF), which has achromatic dispersion of 17 ps/nm/km], does not exceed 100 km.The impact of chromatic dispersion on a sequence of data isschematized in Fig. 2.

The optical transmitter converts an electrical binary sequenceinto optical power (upper part of Fig. 2). Due to the fiber chro-matic dispersion, each “1” broadens and “0” and “1” are in-creasingly difficult to distinguish, as shown in the lower part ofFig. 2.

Fig. 3. Impact of DGD on the pulse broadening and on the eye diagram.

When moving from 10 to 40 Gb/s, the constraints due tochromatic dispersion increase by a factor 16(42), and the trans-mission reach could be limited to a few kilometers only. Thisresulting limitation on distance can be overcome in two ways.The first one, which had been used in the first transoceanic ca-bles, was to reduce the chromatic dispersion of the fiber itselfto around 0 ps/nm/km, thanks to a redesign of the fiber indexprofile. The second one, which is now exclusively used, is tocompensate for the chromatic dispersion by concatenating thetransmission fiber with another fiber exhibiting a dispersion ofopposite sign.

Another physical phenomenon in the fiber can degrade highbit rate transmissions: due to the imperfections of fiber manu-facturing (namely nonperfectly circular fiber core, constraintswithin the fiber, stress induced birefringence), the light velocitywithin the fiber can change slightly depending on the polariza-tion of light. This effect is called polarization mode dispersion(PMD) [3]. It may cause pulse broadening as shown in Fig. 3 bygenerating a differential group delay (DGD) between the twoparts of the pulse, one aligned along the slow axis and the otheralong the fast axis. Contrary to chromatic dispersion, the PMDchanges quickly with time (seconds to milliseconds range) [4].A good way to explain the phenomenon is to model the fiber asa concatenation of very small, birefringent fiber sections. Thismeans that the optical index, or the speed of the light, is notstrictly identical whether the light is aligned along any of the twoorthogonal axes, called principal axes. Light traveling in any ofthese sections splits into two subpulses (the so-called modes),which get time-delayed by a quantity called DGD, unless it isaligned exactly along one of the principle axes. However, suchan alignment cannot be maintained in a real system. In addition,the principal axes of each birefringent section rotate randomlywith respect to the axes of the previous and the followingsections, as time changes. Hence, the DGD is not constant. ThePMD value of a fiber is defined as the average value of the DGDover time. The impact of PMD is difficult to apprehend, as mostof the time it is small but suddenly turns the system bit error rate(BER) below acceptable limits during a few minutes per year.PMD constraints will be challenging to surmount at 40 Gb/s,especially when current 10-Gb/s systems will be upgradedto 40 Gb/s.

Some techniques have been proposed to mitigate this impair-ment. The most performing ones are optical methods, which useone or several sections of birefringent elements which have tobe dynamically adjusted per channel to mitigate the impact ofthe PMD. Nevertheless, the cost associated with such optical

CHARLET: PROGRESS IN OPTICAL MODULATION FORMATS FOR HIGH-BIT RATE WDM TRANSMISSIONS 471

PMD compensators appears too large and they have not beenused yet. Electrical methods have been proposed also as theirassociated cost is expected to be lower.

The best choice can be sometimes to replace fibers havingpoor PMD properties by new ones with better specifications or toelectrically regenerate the optical signal before the degradationsintroduced by PMD are too large. Another possibility is to selecta modulation format that is more tolerant to PMD impairments.

B. Noise and Nonlinear Effects

When fiber loss is compensated for by optical amplifiers andchromatic dispersion problems are solved, new problems arisewhich cannot be fully compensated as the previous ones. Due tothe cascade of optical amplifiers, the optical signal to noise ratio(OSNR) degrades along the transmission path. When the OSNRbecomes too low, it is no more possible to recover the informa-tion, which means that the BER worsens beyond acceptablelimits.

These limits are defined by the correction capabilities of theforward error correction (FEC) code used; a signal having aBER of 10−3 can be corrected to a quasi error free stream(BER better than 10−13), thanks to modern FEC installed. Alimited overhead is required to implement the FEC, it is usu-ally around 7%, leading to a bit rate of 43 Gb/s instead of40 Gb/s. Nevertheless, for simplicity and clarity purposes inthe next parts of the manuscript, 40 Gb/s will be used to de-scribe a signal with the 7% FEC overhead. The more straight-forward way to improve the OSNR is to increase the signalpower at the input of the optical amplifiers. This can be doneby increasing the signal power at the output of the previousamplifiers.

This leads to new limitations called nonlinear effects [2].Because the transmission length is very long (typically severalhundreds or thousands of kilometers), and the light is confinedwithin a very small core (characterized by the effective areaAeff∼50–80 µm2), a moderate power level P at the input of eachspan (∼1 mW, i.e., 0 dB ·m), leads to a modification of the fiberrefractive index with the signal power according to n = n0 +n2P/Aeff , which can degrade the signal quality through thetight interplay with chromatic dispersion even if the nonlinearcoefficient of the silica n2 is very small (2.7 × 10−20 m2/W).

Nonlinear effects are often categorized into two sets of ef-fects, those resulting from the propagation of a single channeland those resulting from the interactions between WDM chan-nels. Single-channel nonlinear effects manifest mainly throughself phase modulation (SPM), whereby each channel altersits own phase. SPM translates into pulse distortions throughchromatic dispersion. WDM nonlinear effects are often splitinto cross phase modulation (XPM), whereby the phase ofeach channel is modified by the power of the neighboringchannels, and four wave mixing (FWM), whereby three chan-nels interact to transfer a fraction of their energy to a fourthone.

Tradeoffs on optical power between too much noise and toomuch nonlinear effects have to be found for optimal systemperformance, i.e., minimal BER.

C. Upgrade Constraints

Operators have built wide WDM optical networks workingat a bit rate of 10 Gb/s between years 2000 and 2005, and mostof them do not intend to build a specific network for 40-Gb/sapplications from scratch.

This will generate some constraints when moving to 40 Gb/sas the channel spacing used for a majority of long haul andultralong haul networks is fixed at 50 GHz. As the spec-trum is naturally four times broader at 40 Gb/s than at10 Gb/s, when the same modulation scheme is applied, thesimplest modulation format, which is NRZ, cannot be useddirectly. For this reason, new generation of modulation for-mats have been pushed to handle this specific bandwidthlimitation.

The modulation formats proposed at 40 Gb/s can nearly besplit into two families, the ones which are designed for a rel-atively wide 100-GHz spacing (even if some formats couldexhibit better performances when a wider channel spacing isconsidered) and the other ones which have been proposed toanswer to the 50-GHz channel spacing constraints.

D. Impact of Bit Rate on Modulation Formats

This paper deals with modulation formats at 40 Gb/s. Evenif some characteristics are bit rate independent, some crucialparameters as the nonlinear effects are particularly affected as afunction of the bit rate [5], [6].

In this paper, we would like especially to highlight that po-tentially higher benefit could be obtained by using advancedmodulation formats at 40 Gb/s than at 10 Gb/s, and justifythe reason why so much developments are expected to encode40-Gb/s signals, while they have not been applied before for10-Gb/s systems.

The first reason is the origin of nonlinear degradation. Whereit comes mainly from adjacent channels carrying unknown infor-mation at 10 Gb/s and thus generating unpredictable distortions,the degradation at 40 Gb/s comes largely from intrachannel dis-tortion (i.e., due to the information of the previous or followingbits). Here, by optimizing the modulation format, it will beshown how largely the performance can be improved.

The 40-Gb/s bit rate seems also better suited for ultra highinformation spectral density. If we consider a spectral densityof 0.8 b/s/Hz, it will require an ultra narrow channel spacing of12.5 GHz, when working at 10-Gb/s channel rate. In this case,large WDM nonlinear penalties will be observed and the con-straints on the optical filters to extract the wavelengths will bealso extremely stringent. At 40 Gb/s, the channel spacing willbe 50 GHz at the same information spectral density, a standardchannel spacing widely used at 10 Gb/s and where optical fil-ters are commonly available. The constraints on the laser wave-length stability are also divided by 4, when moving from 10 to40 Gb/s.

We will first discuss the evolutions proposed to increase thetransmission length at 40 Gb/s, to achieve nearly the same reachthan that of 10-Gb/s systems, and in the second part, we willdiscuss about the ways explored to narrow the spectrum widthbelow 50 GHz, while keeping good transmission properties.


Fig. 4. MZM used to generate NRZ.

III. MODULATION FORMATS DESIGNED FOR

ULTRALONG-HAUL TRANSMISSION

The first 40-Gb/s experiments were using optical time di-vision multiplexing (OTDM). Several sequences working ata lower bit rate (usually 10 Gb/s) composed of narrow opti-cal pulses were combined to generate a sequence at 40 Gb/s.This limited the variety of modulation formats available. It wasmainly return-to-zero (RZ), even if the phase and the polar-ization of the various sequences were not always accuratelycontrolled. At the very end of the 1990s, high-speed electronicsbecame available and 40-Gb/s electrical time division multi-plexer has been used to combine several electrical stream into a40-Gb/s stream [7]. It then opens the road to advanced modula-tion formats at 40 Gb/s.

When several low bit rate streams are electrically multiplexedup to 40 Gb/s, the signal voltage amplitude is usually low (sev-eral hundreds of millivolts) and has to be amplified by a driverup to ∼5 V to drive an electro-optic modulator as described inFig. 4. The amplitude of the electrical signal has to be compat-ible with the characteristic of the modulator and especially theVπ of this one which characterizes the voltage required to shiftthe phase in one arm of the Mach–Zehnder modulator (MZM)compared to the other one by π. The interferences between thetwo arms of the modulator will change from the constructivestate to the destructive one, and the signal will thus pass from a“1” to a “0” as depicted in Fig. 4. The optical NRZ signal willapproximately reproduce the binary electrical signal.

NRZ is likely to be used for VSR application and also perhapsfor metro market. But as the cost and footprint are key driversin these areas, MZM could be replaced by electro-absorbtionmodulators (EAM). These modulators could even be integratedwith the laser to reduce even more the footprint. A last advantageof EAM is the low driving voltage required, approximately 2 Vcompared to 5 V with standard MZM. This leads to a reducedpower consummation of the system.

Fig. 5. Intrachannel XPM.

Nevertheless, NRZ format does not appear to be sufficientlyefficient for several applications. Other formats have been pro-posed to allow optical transmission over longer distances be-cause of a better tolerance to nonlinear effects, to optical noise,or to be more tolerant to chromatic dispersion variation, to PMDdistortion, or also to exhibit a narrower spectrum width. Noneof the modulation format can handle perfectly all these con-straints. Some compromises have to be found, depending on theapplication, on the priorities, and on the complexity allowed.

In this section, we will focus our attention on modulationformats designed for ultralong haul transmission. The most im-portant criteria for this application is the tolerance to nonlineareffects (the longer the distance, the larger the number of spans,and the larger the accumulated nonlinear effects are) and also tothe optical noise (the longer the distance, the larger the numberof optical amplifiers and the higher the optical noise level is).

A. Intrachannel Nonlinear Effects

One of the first objectives for advanced modulation formatwas to reduce the detrimental impact of nonlinear effects. Non-linear electronic (also called Kerr) effects translate into threemajor phenomena: SPM which corresponds to the modifica-tion of the phase of each channel by its own intensity, XPM,and FWM, which both results from interactions between WDMchannels. Over a given fiber, theoretical considerations indicatethat the higher the bit rate, the lower the relative impact of WDMeffects. In contrast to most WDM systems at 10- and 40-Gb/stransmissions are mainly impaired by SPM. To obtain better in-sight on SPM, the phenomenon itself has been categorized intothree basic symptoms, the so-called intrachannel effects [8].

For the sake of clarity (but without loss of generality), wedepict next the optical data stream carried by each channel as aseries of optical pulses, i.e., assuming RZ format. However, theconsiderations detailed next apply to any modulation format.Each pulse (symbol) of the data stream can have its phase af-fected proportionally to its own power profile by the SPM effect,a phenomenon simply referred to as iSPM (for intrapulse SPM).When undergoing chromatic dispersion, the pulses broaden andoverlap the neighboring pulses as schematized in Figs. 5 and 6,which gives rise to intersymbol nonlinear interactions. Theseinteractions can be separated into two families [8], named aftertheir similarities with WDM nonlinear effects. The first one isintrachannel cross phase modulation (iXPM), which designatesthe phase modulation of one symbol proportionally to the powerprofile of the neighboring dispersed symbols.

The second one is intrachannel four-wave mixing (iFWM).iFWM designates a power exchange between different symbols,


Fig. 6. Explanation of the origin of iFWM.

which occurs when three regularly spaced frequencies withinthree different symbols interact to generate energy at a fourthfrequency, every time all three frequencies coexist within thesame small time slot. In Fig. 6, we have schematized the spectralcontent of each pulse, spreading over a finite bandwidth beforebeing fed into a fiber span. Through chromatic dispersion (herepositive), the leading edge of each pulse experiences a shifttoward higher frequencies, whereas the trailing edge experi-ences a shift toward lower frequencies. When they overlap,the three consecutive pulses in Fig. 6 have different spectralcontent, which is the seed for four-wave mixing. At the end ofthe span, dispersion compensators are used to shrink the pulsesback to their original width but cannot undo the interpulseenergy exchanges caused by iFWM. The consequences aresome power fluctuations of the “1” pulses and the generationof so-called ghost pulses at the location of “0” bits.

At 10 Gb/s, intrachannel effects are often negligible. Theybecome dominant at 40 Gb/s because, in a given dispersive fiber,the broadening ratio of pulses normalized to the bit duration issixteen times stronger than at 10 Gb/s. A way to contain them isto select a fiber with lower local dispersion [9] and to avoid largeaccumulated dispersion. This solution is difficult to apply, asvery low dispersion fiber usually has a dispersion slope, whichis extremely difficult to compensate over a wide wavelengthband. In addition, the use of a very low dispersion fiber wouldreinforce detrimental WDM nonlinear effects.

Several millions of simulation runs have been performed [6]to assess the impact of intrachannel effects as a function ofbit-rate and dispersion. One remarkable conclusion is that intra-channel effects cause the exact same performance degradationin a system having a bit rate B and a dispersion parameter β2 (re-lated to the dispersion D in picosecond per nanometer-kilometerthrough D = −2πcβ2/λ2) and in another system having a bitrate 2 B and a dispersion parameter β2/4. It was found thatβ2B

2Leff is a good parameter for assessing the impact of in-trachannel effects, where Leff = (1 − exp(−αL))/α stands forthe effective length (L being the span length and α the attenua-tion coefficient). When β2B

2Leff � 1, pulse-to-pulse overlap isvery limited and intrachannel penalties remain low. Conversely,

Fig. 7. Relative impact of iSPM, iXPM, and iFWM on the performance.

when β2B2Leff � 1, pulses overlap so many of their neigh-

bor pulses that bit-to-bit information exchanges are scrambledand intrachannel impairments are also limited. The maximumpenalty is achieved when β2B

2Leff∼1 [6].Another preferred solution to reduce transmission penalties

at 40 Gb/s is to choose a modulation format that is more re-sistant to intrachannel effects and especially iFWM. In order todetermine the relative impact of iSPM, iXPM, and iFWM, a spe-cific simulation tool has been proposed [10], which accuratelydiscriminates these effects, while resorting to the conventionalSplit-step Fourier transform. Using this tool, the impact of thedifferent intrachannel effects on the system performance in var-ious 43-Gb/s line configurations is computed.

Here the simulated link is composed of 15 spans of 100 kmof SSMF or NZDSF (having a dispersion of 4 ps/nm/km). Thedispersion map has been optimized (especially the amount ofdispersion located within the transmitter, i.e., the precompen-sation and also the amount of dispersion within the receiver,i.e., the postcompensation) to obtain the best performance. TheQ-factor is plotted in Fig. 7 as a function of the channel inputpower into each span for the three modes, iSPM, iSPM, andiXPM, and iSPM, iXPM and iFWM. We observe, on the rightfigure, first that almost no penalty is measured when iSPM oriSPM and iXPM are considered up to 5-dB ·m input power perspan. When the impact of iFWM is added, the maximum perfor-mance is obtained for 2-dB ·m input power. It has to be notedthat the conclusions are slightly different when lower dispersionfiber is considered (left part of the figure). The overall systemperformance is better but the relative impact of iXPM, which isalmost negligible over high dispersion fiber, becomes larger.

A promising technique against iFWM is to add some specificphases on each pulse to produce destructive interferences ratherthan constructive ones even if the expected benefits could bereduced over fiber infrastructure with a low local dispersion.

B. Periodical Phase Shifting

One of the first tentative has been to resort from a modula-tion format called carrier suppressed RZ (CS-RZ). In fact, theoptical phase of every other bit is shifted by π, this leads to thedisappearing of the carrier as the mean amplitude of the signalis now null (same power with a phase of “0” and with a phaseof “π”) and justify the name of the format. The objective is toreduce the nonlinear interactions between the adjacent pulses.The method to generate CS-RZ, is to first encode the data withan electro-optic modulator in order to generate a NRZ stream,


Fig. 8. CS-RZ generation method.

Fig. 9. Intrachannel FWM in case of CS-RZ (the phase of the bit is indicatedbelow them by “0” or “π”).

then, a second MZM is used to carve the pulses and to generatethe π phase shift of every other bit. It is done by biasing the mod-ulator at the minimum of transmission and by driving it with aclock signal at half the data rate (i.e., 20 GHz for 40 Gb/s) asdepicted in Fig. 8. In order to minimize the insertion loss of themodulator, it is preferable to have a amplitude of 2 Vπ on theclock signal. The Vπ of MZM is usually around 5 V, the voltagerequired here to drive properly the CS-RZ modulator is around10 V. Due to this generation method, the pulse width of CZ-RZis wider (duty cycle ∼66%) than for standard RZ (duty cycle∼50%) generated by a clock signal working at the bit rate.

As already explained, the main nonlinear limitation occursfrom the interactions between adjacent bits. Some specific se-quences are more impaired than others and are at the originof the majority of the errors generated within the transmission.The case of an isolated “0” between a long sequence of “1”is clearly one of the most limiting sequence for on–off keying(OOK) modulation formats.

In the following example schematized in Fig. 9, we will con-sider for simplicity a sequence of two “1” followed by an isolated“0” and then two “1.” The phase jump generated by CS-RZ willchange mainly the intrachannel four wave mixing (iFWM), asiFWM is a nonlinear effect sensitive to the phase of the differentspectral components involved in.

By iFWM, energy can be transferred within time slots 3× 4combinations only [11] when a five bit long sequence is consid-ered, one of them is represented in Fig. 10 and the four cases arelisted below. Ai corresponds to the amplitude of the bit i at thetime of the interaction whereas A∗

i corresponds to the complexconjugate of the amplitude of the i th bit. The energy transfer isproportional to the product of the three terms in each case

Combination (a): A22A

∗2

Combination (b): A24A

∗5

Combination (c): A1A4A∗2

Combination (d): A2A5A∗4.

Fig. 10. Calculated spectrum for (left) NRZ, (center) RZ, and (right)CS-RZ, and estimated spectrum width associated.

We will now look at the phase of the intermodulation productgenerated by these four components

Combination (a): 0 + 0 − π = 0[2π]Combination (b): 0 + 0 − π = 0[2π]Combination (c): π + 0 − 0 = 0[2π]Combination (d): π + 0 − 0 = 0[2π].

As can be seen, the four combinations, which generate a ghostpulse within the time slot 3, are all in phase. That means thatconstructive interferences will occur and that the π-phase shiftis not efficient in this configuration, which is one of the mostdetrimental. But if CS-RZ is not so efficient to mitigate nonlin-ear impairment, nevertheless, it has the advantage to exhibit anarrower spectrum than standard RZ as shown in Fig. 10. It isdue to: 1) a change within the spectrum shape due to the π-phaseshift, and 2) to the slightly wider pulse duration compared tostandard RZ (having a ∼50% duty cycle). This has the advan-tage to reduce the linear crosstalk due to WDM transmission,when a 100-GHz spacing is considered.

To improve the tolerance to nonlinear effects, other phasecombinations have been suggested to suppress more effectivelythe iFWM. It has especially been proposed to alternate the phaseof RZ pulses between 0 and π/2. This format is called π/2alternate-phase RZ (π/2 AP-RZ) [12], [13], and is particularlyeffective to reduce iFWM.

If we consider again the configuration with 2 consecutive “1”followed by an isolated “0” and 2 “1” again, we will calculatethe phase of the generated ghost pulse within the third time slot

a) 0 + 0 − π/2 = −π/2[2π]b) 0 + 0 − π/2 = −π/2[2π]c) π/2 + 0 − 0 = π/2[2π]d) 0 + π/2 − 0 = π/2[2π].As it can be seen, two iFWM components have a certain

phase (−π/2) while the two otherones have a π-phase differ-ence (π/2). This will generate destructive interferences andminimize the ghost pulse construction as seen in Fig. 11, whereπ/2 AP-RZ is compared with CS-RZ. Two identical sequencesare computed in a single channel configuration after a transmis-sion through 15 km× 100 km SSMF with a 5-dB ·m channelpower.

The ghost pulse appears very large in the case of CS-RZ,whereas it is negligible in the case of π/2 AP-RZ; the powerfluctuations of the “1” are also dramatically reduced.


Fig. 11. Impact of π/2 periodic phase modulation on reduction of iFWM.

Fig. 12. Sensitivity improvement by DPSK.

C. Phase-Modulated Signal

The most common way to encode the data on an opticalsignal is to vary the intensity of the light to code a “0” or a “1.”Nevertheless, it has also been proposed to modulate the phaseof the light in order to transmit the data. Phase shift keying(PSK) has been proposed more than 20 years ago but a renewedinterest has been observed since 2002 [14]. In order to avoidthe use of complex coherent receiver that is able to detect thephase of the encoded signal, a differential approach has beenproposed in conjunction with a balanced receiver. One of themain advantages of differential phase shift keying (DPSK) isthe 3-dB improved tolerance to optical noise, also called OSNRsensitivity, brought by the balanced receiver [15].

A simple way to explain this tolerance is to draw the constel-lation diagram for both formats, OOK and DPSK (see Fig. 12),i.e., the electrical field at the expected locations of “0” and “1”symbols in the complex plane, assuming a normalized intensity.Considering OOK, a “0” symbol has a nearly null amplitude andtherefore falls in the center of the circle, whereas a “1” symbolhas an amplitude normalized to 1. “1” symbols can be locatedanywhere on the circle, depending on their phase. Phase doesnot matter in our explanation, and we choose to represent theOOK symbols with a null phase, i.e., along the real axis. WhenDPSK is considered, both “0” and “1” symbols have the sameamplitude (

√2/2), in order to maintain the same average power

than for OOK format, but a phase difference of π. Again, theiractual phase does not matter here, and we may locate DPSKsymbols about along the real axis, about the circle center. Foreach symbol of the bit stream, any noise perturbation adds to the

Fig. 13. DPSK generation method.

Fig. 14. DPSK demodulation method.

electrical field and shifts its coordinates in the complex planeoff the value represented in Fig. 12. Since, the distance between“0” and “1” is larger for DPSK than for OOK, it takes a largeramount of noise to have a “1” mistaken for a “0” or the reversefor DPSK than for OOK. This intuitively enlightens why theOSNR sensitivity of DPSK is better than that of OOK. Theseconsiderations hold only if DPSK is detected with a specialreceiver that can distinguish between the phases of symbolswithout power loss, as in a balanced configuration.

Contrary to what could be thought, DPSK formats are usuallynot generated by using a phase modulator but by using a Mach–Zehnder one. The main reason is that the phase level is preciselydefined when an MZM is considered. In the case of a phasemodulator, all the imperfection during the generation will leadto phase inaccuracy which will degrade the performance. Theconventional generation method of DPSK using an MZM isshown in Fig. 13. We observe some similarities with the CS-RZ generation, and here the bias is also set to the minimumtransmission point of the modulator while the data amplitude isadjusted to 2 Vπ. At Each phase transition, the optical signalpass by the minimum of transmission of the modulator and thusthe eye diagram at the transmitter side looks like an inversed RZone even if the information is located within the phase of thesignal, i.e., within the upper rail and not within the transitionsbetween the bits.

In Fig. 14, we schematize a DPSK receiver. The spectrum atthe transmitter side and the eye diagram are shown in the upperpart on the left of the figure. The phase modulated signal is thendemodulated by a Mach–Zehnder interferometer exhibitinga one-bit delay (approximately 25 ps at 40 Gb/s) between


Fig. 15. Numerical comparison of performance of RZ and RZ-DPSK.

both arms. One output of the interferometer is called theconstructive port whereas the other one is called the destructiveport. As differential detection is used to detect the signal,the information (optical phase 0 or π) has to be differentiallyprecoded at the transmitter side. This is done by an electricalcomponent called precoder, which will be described in moredetails in Section IV-B.

We can observe that the eye and the spectrum on the con-structive port and on the destructive port are different. A “1”appears on the constructive port when two consecutive bits havean identical phase, on the contrary a “1” is generated on the de-structive port when the phase of two consecutive bits are shiftedby π. This variation of phase is generated at the transmitter sideby passing from one side of the transmission function of theMZM to the other side through a zero transmission level. Dur-ing the transitions, no power is emitted. This transition effect isobserved on the destructive port and explains the RZ shape ofthe destructive eye.

But the advantages of DPSK formats are not limited to anenhanced OSNR sensitivity. The tolerance to nonlinear effectsis also improved [16], [17]. Single-channel simulations havebeen performed to estimate the performance (measured here inQ-factor) obtained with RZ and RZ-DPSK over 15 km× 100 kmspans of SSMF. The channel power has been gradually in-creased and the Q-factor has been computed for each case.When the channel power increases, the OSNR at the endof the link increases proportionally. This explains why theQ-factor improves first with the channel power. When higherlevels are reached, the signal becomes degraded by nonlineareffects and the Q-factor drops. Fig. 15 clearly shows that theRZ-DPSK outperforms RZ for two reasons; the first one is theimproved performance obtained at low power due to the im-proved OSNR sensitivity and the second one is its higher toler-ance to nonlinear effects. This makes it possible to send highersignal power into the fiber, and hence bridge longer distanceswhile staying within the customer requirements in terms of BER(or Q-factor).

Nevertheless, these conclusions do not apply at 10 Gb/s,where the hierarchy of the impairments is not the same. OOKmodulation formats, like RZ, do not suffer a lot from iFWM at10 Gb/s. When WDM nonlinear effects remain still low (i.e.,if the channel spacing is wide enough or if the value of fiberchromatic dispersion is high enough), 10-Gb/s channels are im-paired by nonlinear effects for a channel power much largerthan the nonlinear threshold of 40-Gb/s channels, consideringthe same OOK modulation format. In this case, DPSK suffers

Fig. 16. APol RZ-DPSK transponder scheme and typical sequence.

from another source of degradation, which is called nonlinearphase noise or Gordon–Mollenauer phase noise [18]. The opticalnoise generated by the optical amplifiers (ASE noise) changesthe power of each bit. Then, the phase of the bit is modifiedduring the propagation as the nonlinear phase noise added isproportional to the intensity of the bit. The nonlinear interactionbetween noise and signal will modify the phase level of each bitof the DPSK signal and thus corrupt the transmission [19]. Thiseffect is reduced at 40 Gb/s where optical pulses spread quicklyand overlap with adjacent pulses [20]. Due to this nonlinearinteraction between noise and signal, the tolerance to nonlineareffects can be reduced by several decibels, when using DPSKinstead of OOK at 10 Gb/s [21].

The main interest of DPSK at 10 Gb/s lies most likely inhigh information spectral density transmission, where WDMnonlinear effects become predominant and where OOK formatsbecome more impaired. As the energy within DPSK channel isnearly pattern independent (contrary to the case of OOK format),the impact of XPM generated by one DPSK channel on theother one is quite low. The highest information spectral densityreported (0.65 b/s/Hz) within a 10-Gb/s channel over ultralonghaul distance (here more than 10 000 km) has been obtainedusing DPSK and a channel spacing of 16 GHz only [22].

In order to further increase the tolerance to nonlinearities, thereduction of the interactions between the adjacent bits is of majorimportance. A way to reduce them is to use the two polarizationsof the light to code the data. The polarization of every other bit isrotated by 90◦, thus generating alternate polarization RZ-DPSK(APol RZ-DPSK) [23].

This can be realized by using a polarization modulator drivenby a clock signal working at half the bit rate as shown in Fig. 16.At the receiver side, the optical demodulator has to be slightlymodified, in order to compare the phase of two bits havingthe same polarization. It is done by considering a 2-b delaydemodulator. This implies also a modification of the electricalprecoder at the transmitter side.

When chromatic dispersion of the fiber leads to a large pulseoverlap, half of the energy is on one polarization, wherein theother half is on the other polarization. It has been observedby simulation and experimentally that the method of alter-nating the polarization of every other bit allows to increasethe nonlinear threshold by approximately 3 dB, as illustratedin Fig. 17.


Fig. 17. Experimental improvement brought by alternate polarization on RZ-DPSK, as measured in the conditions of the experiment [24].

Fig. 18. Tolerance to DGG of RZ-DPSK and APol RZ-DPSK.

Fig. 19. Recorded Q-factors after 9180 and 11 220 km in the experiment [24].

No difference is noticeable at low power (when nonlinear ef-fects are still low), but APol RZ-DPSK clearly outperforms RZ-DPSK at high power. The maximum performance is obtainedfor a channel power 3.5-dB higher when alternate polarizationscheme is used. One disadvantage of APol RZ-DPSK, whichseems an excellent candidate for ultralong haul transmission,is its PMD tolerance that is slightly lower than RZ-DPSK. Areduction of the tolerance to DGDs in back-to-back has beenmeasured and found reduced by nearly 20% compared to RZ-DPSK as shown in Fig. 18 and confirmed by numerical simula-tions [25].

Nevertheless, if the limitations induced by PMD are lowenough or can be mitigated, APol RZ-DPSK modulation formatcan be used to cross ultralong distances. Recently, a 1.6-Tb/s(40 Gb/s× 40 Gb/s) transmission has been demonstrated over atranspacific distance (9180 km), while keeping more than 3-dBmargin compared to the FEC limit (typical margin required bysystem designers to take into account the degradation of thesystem performance during the 25 years lifetime) [24] as shownin Fig. 19.

As various telecom operators intend to upgrade their 10-Gb/snetwork at 40 Gb/s, it is quite interesting to compare whatperformance (or transmission distance) could be achieved withcurrent deployed 10-Gb/s channels (using mostly NRZ modu-lation format) with the modulation format which gives the bestperformance at 40 Gb/s for ultralong haul transmission, namelythe APol RZ-DPSK. This comparison has been done by usingstandard EDFA and over a fiber having a medium dispersion

Fig. 20. Q-factor performance at 3200 and 4400 km with 10-Gb/s NRZ orwith 40-Gb/s APol RZ-DPSK and a doubled capacity.

(TeraLight, D = 8 ps/nm/km). At 10 Gb/s, the channel spacingwas set at 50 GHz, whereas the channel spacing was 100 GHzfor 40-Gb/s transmission. Hence, the overall system capacityis doubled. The system was optimized for 10-Gb/s bit rate andit has been found a nearly similar performance after 3200 km(with approximately 3-dB margin) [26] and 4400 km (withoutmargin compared to the FEC threshold) as shown in Fig. 20.

It suggests that 10-Gb/s systems could be upgraded to40-Gb/s channel rate using APol RZ-DPSK, while keeping ap-proximately the same reach if the system PMD is low enough.

IV. ULTRA NARROW BANDWIDTH MODULATION FORMAT

During the 1990s, the overall capacity of optical systemsincreased drastically for several reasons. First, the channel rateincreased from 622 Mb/s at the beginning of the decade upto 10 Gb/s, ten years later, thanks to the progress of TDMtechnologies. On the other side, WDM technologies have beenused and the bandwidth of optical amplifiers increased from afew nanometers to more than 30 nm, while the channel spacingdecreased during the same time from 200 GHz down to 100 GHzand then 50 GHz.

In order to further increase the system capacity, the opticalbandwidth and the information spectral density could be in-creased. It appears challenging to increase the bandwidth ofEDFA wider than 40 nm (even if dual-band systems using twoamplifiers in parallel have been proposed, they do not appearvery cost effective), so the main direction is to improve the in-formation spectral density, which is currently set at 0.2 b/s/Hzfor a 10-Gb/s channel with 50-GHz spacing.

A. Optical Filtering

The best way to increase the information spectral densitywould be to reduce the spectrum width of each channel. Severalsolutions have been proposed. As the two sidebands of an op-tical NRZ spectrum contain redundant information, it has beenproposed [27] to filter out one of them to decrease the spectrumwidth. A vestigial sideband format (VSB) is thus generated. Ithas been proposed to use VSB filtering only at the receiver inconjunction with a specific frequency allocation scheme. The40-Gb/s WDM channels are alternatively spaced by 75 and50 GHz, as illustrated in Fig. 21. In the receiver, a given channelis selected with a very narrow filter (30 GHz at 3 dB), tunedoff the channel central frequency toward the 75-GHz-spacedneighboring channel.


Fig. 21. Channel spectrum within (top) the transmission line, (middle) afterleft side VSB filtering, and (bottom) right side VSB filtering.

Fig. 22. Optimal optical filter detuning.

Fig. 23. Asymmetric spectrum of 50-GHz-spaced channel after VSB filteringat the transmitter side.

The optimum frequency shift and the associated eye diagramare described in Fig. 22. An information spectral density of0.64 b/s/Hz thus can be obtained. One drawback of the solu-tion is the nonfull compatibility with the standard ITU grid(100-GHz spacing, 50-GHz spacing or 25-GHz spacing).

In order to solve this problem, a filtering at the transmitterside also has been proposed [28]–[30]. VSB filtering at thetransmitter and at the receiver side was proposed to reduce thespectrum width and thus to be compatible with the 50-GHz ITUgrid [29]. Thus, a 50-GHz periodical asymmetric spectrum isgenerated as depicted in Fig. 23.

Then the selection filter at the receiver side is also tuned offby 10–20 GHz to optimize the BER performance, generating aneven more asymmetric spectrum, as shown in Fig. 24.

The same type of method has also been applied to CS-RZ,where one of the two carriers is filtered out to generate a kindof NRZ signal having modified spectral contents. It has beendemonstrated that the tight VSB filtering can be applied several

Fig. 24. Spectrum (gray) before channel selection and (black) after channelselection at the receiver done by VSB filter.

times [31] and thus potentially allows transmission in trans-parent networks through several reconfigurable optical add anddrop multiplexers (ROADM), each of them having a filteringfunction narrower than 50 GHz at 3 dB [32].

B. Duobinary/PSBT Modulation Format

The previous methods used a modulation format having arelatively wide spectrum, which was then narrowed by opticalfiltering. Another possibility can be explored: the generation ofa naturally narrow spectral width modulation format. Contraryto NRZ or RZ modulation formats, which resort only froman intensity modulation, a phase modulation can be applied toreduce the spectrum width. The proposed format is known asduobinary or phase-shaped binary transmission (PSBT) [33]. Ifwe consider duobinary, the 40-Gb/s binary electrical signal isdelayed by one bit and summed with the same sequence (thisfunction can also be realized by a transversal electrical filter).This generates a three level electrical eye, depending whetherthe two consecutive data are 1, 1 (generating a 2), 1, 0 or 0, 1(generating a 1) or 0, 0 (generating a 0). The three-level signal isamplified to feed an MZM biased at the minimum transmissionlevel. The electrical level 2 becomes an optical “1” with a opticalphase “0.” The electrical level 1 becomes an optical “0” as themodulator is biased at the minimum transmission level. Theelectrical level 0 becomes an optical “1” with a phase of “π.”In this case, the narrowband filtering is done in the electricaldomain and generates a modulation format using phase jump toreduce its spectral width. The optical receiver used for duobinaryis a standard NRZ receiver which detects only the intensity andnot the phase. It appears clearly that the received data are notthe same than the binary electrical data sent to the transmitterdue to the duobinary encoder (delay and add filter plus theMZM). In order to recover the same data at the receiver, anelectrical precoder is mandatory. It will modify the electricalbinary sequence before the encoding. The same precoder is alsoused for DPSK modulation format.

This precoding function can also be performed at a lower bitrate within a parallel mode (on the 16 tributaries at 2.5-Gb/sfor example). If the precoding function is realized directly at40 Gb/s, the most common approach is to use a TFF and a AND

gate [38], [39] depicted in Fig. 25. It can also be understood bythe following way: at each time the binary sequence contains a“0,” the state of the output sequence is changed (from “1” to “0”or from “0” to “1”); otherwise, the state of the output sequencestays at the same level than the previous bit.


Fig. 25. Possible PSBT precoder scheme.

Fig. 26. Generation of Duobinary based on DPSK.

A simple example of sequence precoding (assuming that theprecoded sequence begins with a “1” or a “0”), of the encodingthrough the electrical filter and then the optical modulator isshown in the lower part of Fig. 25.

It can be observed that the intensity of the optical signal isthe same as the original binary electrical sequence, thanks to theprecoder. This explains why a simple NRZ receiver can be usedto detect the duobinary.

Another method of generation has been proposed based onDPSK [34]. An optical demodulator splits the signal in twoparts; one is delayed by 1 b before being recombined withthe other part. Two signals are available at the output of theinterferometer as shown in Fig. 26, the so-called constructiveport corresponds to the modulation format described previously:duobinary.

But a practical way to generate the duobinary is the use ofa low-pass electrical filter (usually a 12-GHz 3-dB bandwidthBessel filter) to replace the delay and add function described forthe duobinary. A Bessel filter is a relatively good approximationof the add and sum filter required for duobinary, as shown inFig. 27 and is quite easy to implement.

By using such a filter, the modulation format does not haveexactly the same properties than duobinary, hence a specificname has been chosen to describe this format: PSBT [33].

In the following experiments described [35]–[37], we will usePSBT naming and not duobinary as low-pass Bessel filters havebeen used for the format generation.

The generation method of the PSBT is described in Fig. 28.Here again, the MZM is biased at the minimum of the transmis-

Fig. 27. Comparison of filter shape used to generate Duobinary and PSBT.

Fig. 28. Generation method of PSBT.

Fig. 29. Tolerance to CD of NRZ, duobinary, and PSBT.

sion, and the amplitude of the electrical signal is set to 2 Vπ. Butthe electrical signal has been precoded by a digital circuit (andgate and tff for example) and then been encoded by a low-passBessel filter where the bandwidth is around 12 GHz and by theMZM.

One of the properties that is often associated with a narrowspectrum width is the large tolerance to chromatic dispersion.Nevertheless, we will see that the link is not straightforward.Fig. 29 shows numerical simulation of Q-factor penalty forthree modulation formats, NRZ, duobinary, and PSBT when theresidual chromatic dispersion varies from −200 to +200 ps/nm.We clearly see here that PSBT is much more tolerant than theduobinary which, despite its spectrum width nearly twice nar-rower than standard NRZ, does not exhibit a much larger tol-erance. The superior tolerance of PSBT is mainly due to theresidual energy between two consecutive “0” which will gener-ate destructive interferences with neighboring “1” when somechromatic dispersion is accumulated.


Fig. 30. Beneficial impact of a narrow 0.3-nm optical filter on PSBT.

Several parameters should be taken into account when select-ing a modulation format for a specific application. The toleranceto optical noise is one of the most important. This point is con-sidered as the main drawback of PSBT. A penalty around 2 dBis often considered compared to standard NRZ at 10 Gb/s.

Nevertheless, at 40 Gb/s, it is possible to take advantageof optical filters (to select the channel at the receiver), whichhave a width nearly comparable to the PSBT one (∼40 GHz).Thus, the modulation format can be slightly modified by thisnarrow filtering resulting from a wider eye opening as shown inFig. 30, and in the same time a reduction of the optical noisesent on the receiver. Both phenomena are positive and lead toan improvement of the OSNR sensitivity of PSBT.

The first experimental demonstration of 40-Gb/s transmis-sion has been done in 2001 [35], where 80 channels spaced by50 GHz were transmitted over 3 km× 100 km. In order to mini-mize linear and nonlinear crosstalk, the adjacent channels werecross-polarized.

In a second step [36], the performance obtained was largelyimproved in terms of transmission distance. Ultralong haultransmission based on PSBT has then been realized due partlyto the 7% FEC overhead that was applied. On the one hand, itgives more flexibility as the target BER is around 10−3 insteadof 10−9. On the other hand, the linear crosstalk is enhancedby the 7% broader spectrum. The OSNR sensitivity was alsoimproved by using narrow optical filtering at the receiver sidewith sharper shape. Fig. 30 shows the improvement of the eyeopening obtained using a 0.3 nm (3-dB bandwidth) optical filter.This allows to improve experimentally the tolerance to OSNRby 2 dB at 10−5 target BER. The transmission distance wasincreased up to 2100 km, while a key feature of PSBT, the largetolerance to chromatic residual dispersion variation, was pre-served. In a third experiment [37], the state of polarization ofall the channels was aligned at the transmitter side to representa worst case in terms of linear and nonlinear interactions. Themean Q-factor degradation was limited to 0.3 dB only comparedto a case when the adjacent channels are cross-polarized. The1-dB Q-factor tolerance to residual dispersion excursion wasalso recorded after 17 km× 100 km transmission over SSMFto be around 200 ps/nm, which is much larger than the usualchromatic dispersion tolerance observed at 40 Gb/s.

Fig. 31. Schematic of a DQPSK modulator with parallel architecture.

C. Differential Quaternary Shift Keying

DPSK has been widely used for ultralong haul transmissionexperiments due to its large tolerance to optical noise (3-dBimprovement brought by the balanced receiver) and to opti-cal nonlinearities. Nevertheless, it appears rapidly that DPSKwas not suitable for ultra dense WDM transmission (i.e., with50-GHz channel spacing at 40 Gb/s) especially, when the chan-nel state of polarization was not controlled at the transmitterside due to the large linear crosstalk experienced.

In order to get benefit from the improvement brought by bal-anced receiver and to have the compatibility with 50-GHz chan-nel spacing, differential quadrature phase shift keying (DQPSK)[40] appears as a promising solution. The information is no moreencoded within two phase states (0 or π) but within four phaselevels (π/4, 3π/4,−π/4 and −3π/4). Each symbol now con-tains 2 b. If 40 Gb/s are transmitted, the symbol rate is only 20 Gsymbol/s. The symbol duration is doubled compared to DPSK,which leads to a reduction by a factor 2 of the spectrum width,making it clearly compatible with 50-GHz channel spacing.

Several transmitter architectures have been proposed to gen-erate DQPSK. Two are based on a DPSK transmitter (usingeither a phase modulator or an MZM to generate the 0 and πlevels). Then, another phase modulator is used to generate a0 or π/2 phase modulation. These methods have at least twodrawbacks. The first one comes from the practical difficulty toproduce an accurate phase shift. Each electrical mismatch is di-rectly transferred into a phase mismatch. The other one comesfrom the reduced eye opening obtained by using this modula-tion method (due to the transition between the various phaselevel) [41]. Nevertheless, this impact is reduced if an RZ pulsecarver is used.

The preferred method, which required a specific modulator,is based on a Mach–Zehnder superstructure as schematized inFig. 31. Along each arm of the structure, an MZM is insertedand is fed with a specific data pattern at 20 Gb/s. This resultsin two optical DPSK optical streams at 20 Gb/s. One is phase-shifted by π/2 and combined with the other into the two-arminterferometric structure.

The phase level at the output of the modulator is equal to

ϕ =[(ϕ1 + ϕ2)

2

]√2 cos

(ϕ1 − ϕ2

2

)


Fig. 32. Typical DQPSK eye diagram with two possible transitions.

Fig. 33. Tolerance to PMD of RZ-DQPSK compared to a binary modulationformat.

where ϕ1 and ϕ2 are the phases encoded within each arm of thesuperstructure Mach–Zehnder

ϕ1 = 0or π

ϕ2 = +/ − π/2.

The eye diagram of a DQPSK signal is slightly different fromthe one of DPSK. Especially three kinds of transition have tobe considered between two adjacent bits. Either the phase isconstant from 1 b to the next one, in this case the intensity staysconstant at a “1” level, or the phase difference between twoconsecutive bits is π (as for DPSK), here the intensity of thesignal passes by a “0” level, or the phase difference betweenthe two bits is π/2 and the intensity of the signal during thetransition is set at “1/2” as shown in Fig. 32, where a DQPSKeye diagram is represented.

The receiver is composed of two balanced receivers, eachcontaining an optical demodulator with free spectral range of20 GHz followed by a balanced photodiode. Contrary to DPSKreceiver, each optical demodulator is detuned by π/4 for the firstone and by–π/4 for the second one in order to demodulate thesignals in phase and in quadrature, respectively. This detuninggenerates imperfect interferences at the receiver side and leadsto a degradation of the OSNR sensitivity compared to DPSKmodulation format.

One of the most straightforward impacts of the doubling sym-bol duration compared to binary modulation format is the in-crease by approximately a factor 2 of the tolerance to PMD [42].This is a quite important parameter at 40 Gb/s, where the PMD ofthe link could be a limitation to the transmission performancesin a lot of cases. The tolerance to PMD is nearly doubled asshown in Fig. 33, when the modulation format evolves from abinary one (DPSK here) to a quaternary one (RZ-DQPSK).

D. Coherent Receiver and Advanced Modulation Formats

It has been clearly seen that year after year, more complexmodulation formats have been proposed and demonstrated ex-

perimentally. But even more, it appears that the developmentsare launched by several manufacturers to propose such complexmodulation formats on the market soon.

What are the future trends? It seems that optical communi-cations try now to resort of some of the methods used for yearsby radio communications. The challenge here is to handle anoptical carrier around 200 THz and very high channel data rateof 40 Gb/s. There is nearly five orders of magnitude higher thanwhat is considered in radio communication.

Coherent receivers have been proposed recently using ad-vanced digital signal processing methods [43] to detect QPSKsignal (as the signal is no more differentially encoded, the “D” isno more needed). But it could be envisaged to encode even morecomplex modulation formats as quadrature amplitude modu-lation (QAM). QPSK corresponds to 4QAM, but 16 or even64QAM could be transmitted and detected.

The main drawback of coherent receivers is the associatedcost, in terms of optical components (four balanced receiversare required to detect the signal on both polarizations) and alsoin terms of complexity of real time signal processing associatedwith.

V. EQUALIZATION AND ELECTRICAL ADVANCED PROCESSING

Several signal processing methods have been proposed toenhance the transmission performances and especially the tol-erance to some specific impairments. These techniques could beused for various modulation formats even if the quantificationof the improvement is format dependant.

The most straightforward are equalization techniques thatcould be applied at the receiver side. Equalization can be per-formed either in the optical domain [44], or in the electrical do-main [45]. The optimum performance is obtained with opticalequalization as the phase information is preserved; neverthe-less, electrical equalization appears more cost effective and theintegration within an optical system easier to perform.

The simplest method used for optical or electrical equalizeris called feed forward equalizer (FFE) [45], [46]. The signalis split into several paths; each of them is delayed by a de-termined duration (a bit duration or half a bit usually). Thepower associated within each path can be varied and the sig-nals are then combined to be detected by a decision elementto select if the bit is a “0” or a “1.” Electrical equalizer al-lows to improve the tolerance to residual dispersion (between50% and 100%, approximately). It can also be used to miti-gate the impact of PMD by a small amount (improvement byaround 50%). If larger tolerance is required, a more complexsolution can be used: Maximum likelihood sequence estimation(MLSE) [47].

MLSE is more complex but is one of the most powerfulequalization techniques. The optical signal is converted into anelectrical signal by a photodiode and the electrical signal is thendigitalized by using analog–digital converter (ADC) working at2 samples/b. A processor using a Viterbi algorithm is then usedto select the most likely sequence associated with the detectedsignal. This method has not yet been applied at 40 Gb/s but isalready commercially available at 10 Gb/s.


Another method proposed recently [48] at 10 Gb/s is to syn-thesize the optical signal at the transmitter side in order to pre-compensate the distortions, which will occur during the propa-gation. The optical modulator synthesizes the intensity and thephase of the optical signal thanks to a Mach–Zehnder super-structure. The chromatic dispersion is of course the main im-pairment which could be compensated. But single channel non-linear effect can also be calculated and thus precompensated.Nevertheless, the requirements on the transmitter are high, acomplex modulator is associated with digital–analog converterworking at 2 samples/b and a powerful digital signal processorcapable to synthesize the signal is mandatory. Due to techno-logical constraints, this method has not yet been used 40 Gb/sand the possibility to do it at relatively short term is question-able. Another drawback of the solution is the optical transmis-sion regime where high-accumulated dispersion is encountered.This leads to a large impact of nonlinearities. The single chan-nel impairments can be partially compensated but the WDMimpairments seem also severe [49] and cannot be compensatedeasily.

VI. CONCLUSION

We have presented recent progresses observed in the field ofmodulation formats used for 40-Gb/s optical transmission. Asseveral applications are targeted, several modulation formatsappear as good candidates for industrialization and field de-ployment. Amplitude, phase and polarization codings are nowenvisaged in optical transmissions to maximize the system per-formance.

We observe a first trend towards modulation formats havinga better OSNR sensitivity and nonlinear tolerance like DPSK tobe able to achieve transmission distances approximately com-parable with what is done at 10 Gb/s.

A second trend is the search for modulation formats compati-ble with high information spectral density constraints like PSBTand DQPSK. These formats should be able to pass through acascade of narrow optical filters and to be compatible with the50-GHz ITU grid.

It is also likely that the amount of electrical processing willincrease in the following years, either for equalization purposeor electrical precompensation or coherent detection.

ACKNOWLEDGMENT

The author would like to be grateful to M. Lefrancois, for hiscontribution to this work, and also to S. Bigo for his help andprecious comments.

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[17] A. Klekamp, R. Dischler, and W. Idler, “Impairments of bit to bit alternate-polarization of nonlinear threshold, CD and DGD tolerance of 43 Gb/sASK and DPSK formats,” presented at the Opt. Fiber Commun. Conf.,Anaheim, CA, Mar. 6–11, 2005, Paper OFN3.

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[19] H. Kim and A. H. Gnauck, “Experimenal investigation of the per-formance limitation of DPSK systems due to nonlinear phase noise,”IEEE Photon. Technol. Lett., vol. 15, no. 2, pp. 320–322, Feb.2003.

[20] H. Kim and P. Winzer, “Nonlinear phase noise in phase-coded transmis-sion,” presented at the Opt. Fiber Commun. Conf., Anaheim, CA, Mar.6–11, 2005, Paper OThO3.

[21] T. Mizuochi et al., “A comparative study of DPSK and OOK WDMtransmission over transoceanic distances and their performance degrada-tions due to nonlinear phase noise,” J. Lightw. Technol., vol. 21, no. 9,pp. 1933–1943, Sep. 2003.

[22] L. Becouarn et al., “3 Tb/s transmission (301 DPSK channels at 10.709Gb/s) over 10,270km with a record efficiency of 0.65 b/s/Hz,” presented atthe Eur. Conf. Optical Commun., Rimini, Italy, Sep. 21–25, 2003, PaperTh. 4.3.2.

[23] A. H. Gnauck, J. Leuthold, C. Xie, I. Kang, S. Chandrasekhar,P. Bernasconi, C. Doerr, L. Buhl, J. D. Bull, N. A. F. Jaeger, H. Kato,and A. Guest, “6 Gb/s× 42.7 Gb/s transmission over 10 200 km EDFAamplified SSMF spans using polarization alternating RZ-DPSK,” pre-sented at the Opt. Fiber Commun. Conf., Los Angeles, CA, Feb. 22–27,2004, Paper PDP35.

[24] G. Charlet, R. Dischler, A. Klekamp, P. Tran, H. Mardoyan, L. Pierre,W. Idler, and S. Bigo, “WDM bit-to-bit alternate-polarisation RZ-DPSKtransmission at 40 Gb/s× 42.7 Gb/s over transpacific distance with largeQ-factor margin,” presented at the Eur. Conf. Optical Commun., Stock-holm, Sweden, Sep., 5–9, 2004, Paper Th4.4.5.

[25] J. K. Fischer, S. Randel, and K. Petermann, “PMD outage probabili-ties of optical fiber transmission systems employing bit-to-bit alternatepolarization,” IEEE Photon. Technol. Lett., vol. 17, no. 8, pp. 1647–1649,Aug. 2005.


[26] G. Charlet et al., “Upgrade of 10 Gb/s ultralong haul system to 40 Gb/sthanks to Apol RZ-DPSK modulation format,” presented at the OECC’05.Seoul, Korea, Paper PDP02.

[27] S. Bigo et al., “5.12 Tb/s (128 Gb/s× 40 Gb/s WDM) transmission over3 × 100 km of TeraLight fibre,” presented at the Eur. Conf. Optical Com-mun., Munich, Germany, 2000, Paper PD1.2.

[28] T. Miyakawa et al., “2.56 Tb/s (40 Gb/s× 64 WDM) unrepeatered 230km transmission with 0.8 b/s/Hz spectral efficiency using low noise fiberraman amplifier and 170 µm2 Aeff fiber,” presented at the Opt. FiberCommun. Conf., Anaheim, CA, Mar. 17–22, 2001, Paper PD26.

[29] G. Charlet et al., “3.2 Tb/s (80 Gb/s× 42.7 Gb/s) C-band transmission over9 × 100 km of TeraLight fiber with 50 GHz channel spacing,” presentedat the OAA’02, Vancouver, BC, Canada, Jul. 14–17, 2002, Paper PDP1.

[30] D. F. Grosz et al., “5.12 Tb/s (128 Gb/s× 42.7 Gb/s) transmission with0.8 b/s/Hz spectral efficiency over 1280 km of standard single-mode fiberusing all-Raman amplification and strong signal filtering,” presented at theEur. Conf. Optical Commun., Copenhagen, Denmark, Sep. 8–12, 2002,Paper PD4.3.

[31] A. Agarwal, S. Chandrasekhar, and R.-J. Essiambre, “VSB-CSRZ forspectrally efficient optically-routed network,” presented at the Eur. Conf.Optical Commun., Stockholm, Sweden, Sep. 5–9, 2004, Paper We3.4.4.

[32] G. Raybon, A. Agarwal, S. Chandrasekhar, and R.-J. Essiambre, “Trans-mission of 42.7 Gb/s VSB-CSRZ over 1600 km and four OADM nodeswith a spectral efficiency of 0.8 b/s/Hz,” presented at the Eur. Conf. OpticalCommun., Glasgow, Scotland, Sep. 25–29, 2005, Paper Mo 3.2.7.

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[34] D. Penninckx, H. Bissessur, P. Brindel, E. Gohin, and F. Bakhti, “Op-tical differential phase shift keying (DPSK) direct detection consideredas a Duobinary signal,” presented at the Eur. Conf. Optical Commun.,Amsterdam, The Netherlands, 2001, Paper We.P.40.

[35] H. Bissessur et al., “3.2 Tb/s (80 Gb/s× 40 Gb/s) C-band transmissionover 3 km× 100 km with 0.8 b/s/Hz efficiency,” presented at the Eur.Conf. Optical Commun., Amsterdam, The Netherlands, Oct. 1–4 2001,Paper PD.M.1.11.

[36] G. Charlet et al., “6.4 Tb/s (159 Gb/s× 42.7 Gb/s) capacity over 21km× 100 km using bandwidth-limited phase-shaped binary transmis-sion,” presented at the Eur. Conf. Optical Commun., Copenhagen, Den-mark, Sep. 8–12, 2002, Paper PD4.1.

[37] , “Cost-optimized 6.3 Tb/s capacity terrestrial link over 17 km× 100km using phase-shaped binary transmission in a conventional all-EDFASMF-based system,” presented at the Opt. Fiber Commun. Conf., Atlanta,GA, Mar. 23–28, 2003, Paper PD25.

[38] C. Rasmussen, T. Fjelde, J. Bennike, F. Liu, S. Dey, B. Mikkelsen,P. Mamyshev, P. Serbe, P. V.-D. Wagt, Y. Akasaka, D. Harris, D. Gapont-sev, V. Ivshin, and P. Reeves-Hall, “DWDM 40 G transmission over trans-pacific distance (10 000km) using CSRZ-DPSK, enhanced FEC and all-Raman amplified 100 km UltraWaveTM fiber spans,” presented at the Opt.Fiber Commun. Conf., Atlanta, GA, Mar. 23–28, 2003, Paper PD18.

[39] G. Charlet, E. Corbel, J. Lazaro, A. Klekamp, R. Dischler, P. Tran, W. Idler,H. Mardoyan, A. Konczykowska, F. Jorge, and S. Bigo, “WDM transmis-sion at 6Tb/s capacity over transatlantic distance using 42.7 Gb/s diffeen-tial phase-shift keying without pulse carver,” presented at the Opt. FiberCommun. Conf., Los Angeles, CA, Mar. 22–27, 2004, Paper PDP36.

[40] R. A. Griffin, R. I. Johnstone, R. G. Walker, J. Hall, S. D. Wadsworth,K. Berry, A. C. Carter, M. J. Wale, J. Hughes, P. A. Jerram, and N.J. Parsons, “10 Gb/s optical differential quadrature phase shift keying(DQPSK) transmission using GaAs/AlGaAs integration,” presented at theOpt. Fiber Commun. Conf., Anaheim, CA, Mar. 17–22 2002, Paper FD6.

[41] M. Ohm and T. Freckmann, “Comparison of different DQPSK transmit-ters with NRZ and RZ impulse shaping,” presented at the IEEE/LEOSWorkshop Advanced Modulation Formats, San Francisco, CA, Jul. 1–2,2004, Paper ThB2.

[42] G. Charlet et al., “151 Gb/s× 43 Gb/s transmission over 4080 km basedon return-to-zero differential quadrature phase-shift keying,” presented atthe 31st Eur. Conf. Optical Commun., Glasgow, U.K., Sep. 25–29, 2005,Paper Th 4.1.3.

[43] S. Tsukamoto, D. S. Ly-Gagnon, K. Katoh, and K. Kikuchi, “Coherentdemodulation of 40 Gb/s polarization-multiplexed QPSK signals with16GHz spacing after 200-km transmission,” presented at the Opt. FiberCommun. Conf., Anaheim, CA, Mar. 6–11, 2005, Paper PDP29.

[44] C. R. Doerr, S. Chandrasekhar, P. J. Winzer, L. W. Stulz, and A.R. Chraplyvy, “Simple multi-channel optical equalizer for mitigatingintersymbol interference,” presented at the Opt. Fiber Commun. Conf.,Atlanta, Georgia, Mar. 23–27, 2003, Paper PD11.

[45] H. Buelow, F. Buchali, W. Baumert, R. Ballentin, and T. Wehren, “PMDmitigation at 10Gb/s using linear and nonlinear integrated equalizer cir-cuits,” Electron. Lett., vol. 36, no. 2, pp. 163–164, Jan. 2000.

[46] B. Franz, D. Rosener, R. Dischler, F. Buchali, B. Junginger, T. F. Meister,and K. Aufinger, “43 Gb/s SiGe based electronic equalizer for PMDand chromatic dispersion mitigation,” presented at the Eur. Conf. OpticalCommun., Glasgow, U.K., Sep. 25–29, 2005, Paper We 1.3.1.

[47] A. Farbert, S. Langenbach, N. Stojanovic, C. Dorschky, T. Kupfer,C. Schulien, J.-P. Elbers, H. Wernz, H. Grisser, and C. Glingener, “Per-formance of a 10.7Gb/s receiver with digital equalizer using maximumlikelihood sequence estimation,” Eur. Conf. Optical Commun., Stockholm,Sweden, Sep. 5–9, 2004, Paper Th4.1.5.

[48] D. McGhan, C. Laperle, A. Savchenko, C. Li, G. Mark, and M. O’Sullivan,“5120 km RZ-DPSK transmission over G652 fiber at 10 Gb/s with nooptical dispersion compensation,” presented at the Opt. Fiber Commun.Conf., Anaheim, CA, Mar. 6–11, 2005, Paper PDP27.

[49] R.-J. Essiambre and P. J. Winzer, “Fibre nonlinearities in electronicallypredistorted transmission,” presented at the Eur. Conf. Optical Commun.(ECOC’05), Glasgow, U.K., Sep. 25–29, 2005, Paper Tu 3.2.2.

Gabriel Charlet was born in Rueil Malmaison,France, in 1976. He received the engineering degreefrom Ecole Superieure d’Optique, Orsay, France, in1999.

In 2000, he joined Alcatel Research and Innova-tion, Marcoussrs, France, where he has been workingon WDM transmission systems and realized severalmultiterabit/s transmission records. He is the authorof 9 postdeadline papers published in major confer-ences (OFC, ECOC, OAA, and OECC) and more than20 patents.

Documents

Progress in optical modulation formats for high-bit rate WDM transmissions