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Berkeley
Power Amplifiers for Communications
Prof. Ali M. Niknejad
U.C. BerkeleyCopyright c© 2014 by Ali M. Niknejad
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Power Amplifier Specifications
Peak Output Power
Efficiency
Power Gain
Amplifier Linearity
Stability over VSWR
Ability to transmit into anunknown/varying load
Power Control
Step size,range
High efficiency at back-off
Heat
Pin PL
Pdc
Zin = 50!
Zout != 50!
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Peak Output Power
The peak output power determines the range for two-waycommunications. When we hit sensitivity limits, the only wayto increase the range is to increase the Tx power.
The peak power is specified at the 1-dB compression point orthe maximum output power – the “clipping” point (makes abig difference).
∼1W for cellular handsets ( 1 km distance)∼100mW for W-LAN (100 m)∼10mW for W-PAN (Bluetooth) (1-10 m)∼1mW for body area networks.
In practice, the average power transmitted may be much lowerthan the peak output power due to “back-off”, to obtainlinearity for the amplitude modulation (fast time scale) or forpower control (slow time scale)
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Peak Efficiency
Power Added Efficiency (PAE) is a popular metric.Pout is the output power, Pin is the input power,and Pdc is the DC power consumption of the PA
For high power gain systems (Gp), the efficiencyapproaches the drain
drain efficiency (ηd), or for a BJT, the “collector”efficiency, or simply the efficiency of the last stage
The efficiency of the PA is an important measure ofthe battery life of the wireless transceiver. Since thePA power dwarfs the power consumption in thereceiver, it is usually the most importantspecifications.
For lower power systems (below 10mW), the powerof the entire transmitter chain is important andshould be taken into consideration.
ηPAE =Pout − Pin
Pdc
ηPAE =Pout − Pout
Gp
Pdc
=Pout
Pdc(1− G−1
p )
ηPAE = ηc (1− G−1p )
≈ ηc
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Average Efficiency with Power Control
For a constant envelope signal (phase/frequency modulation),the average efficiency is equal to the average efficiency atpeak power.
Due to power control, though, we must take into account thestatistics of the transmitted signal. Modern systems usepower control to minimize the impact of a transmitter onnearby systems (interference) and hence only use as muchpower as needed to achieve low error communication with thebase station.
Thus the actual average efficiency depends on how theefficiency varies with output power
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Power Statistics
ηav =
∫ ∞−∞
η(P)g(p)dP
Given the distribution of power levels, or the PDF g(P), wecan calculate the expected value of the efficiency
Unfortunately, for most PAs the efficiency drops at low power.
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Envelope Statistics
For signals with amplitude modulation, the average efficiencydepends not only on the desired power level, but also on thestatistics of the envelope.
The amount of power variation is usually captured by thePAR, or the Peak-to-Average Radio.
The PAR is a strong function of the type of modulation.Systems with the highest PAR are OFDM systems employingmultiple carriers.
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Linearity
f1 f22f2-f12f1-f2 3f2-2f13f1-2f2
IM3IM5
2f1 2f2
HD2 HD3
dB
f
transmit filter mask
f1 f22f2-f12f1-f2 3f2-2f13f1-2f2
IM3IM5
2f1 2f2
HD2 HD3
dB
f
transmit filter mask
The traditional way to characterize narrowband systemlinearity is with IM3. Since the system may be driven into astrongly non-linear regime, all odd order harmonics should becarefully taken into account to ensure that excessive spectralleakage does not occur.
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Sources of Non-Linearity
PAs exhibit nonlinear distortion in amplitude and phase. For amulated signal, both sources of distortion are significant.
The dominant sources are AM-to-AM and AM-to-PM.
Amplitude distortion: AM-to-AM conversionPhase distortion: AM-to-PM conversion
For input: x(t) = A(t) cos(ωt + φ(t))
Corresponding output:y(t) = g [A(t)] cos(ωt + φ(t) + ψ[A(t)])
AM-to-AM conversion dominated by gm non-linearity beforeclipping
AM-to-PM conversion dominated by non-linear capacitors(phase delay)
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AM-AM and AM-PM Non-Linearity
For a narrowband signal, we can partition the non-linearityinto an amplitude-amplitude (AM-AM) component and anamplitude-phase (AM-PM) component
This behavioral model can be used to run system levelsimulations to see the effect of non-linearity on a modulatedwaveform
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Adjacent Channel Power (ACP)
885kHzACP=P1(dBm)-P2(dBm)
P1
P2
1.25MHz
30 kHz
For modern communication systems, the IM specificationsleave a lot to be desired since they are only based on two-toneexcitation. Increasingly, the actual modulation waveformneeds to be tested with the PA.
To ensure proper etiquette, the amount of power leaking intoan adjacent channel is carefully specified.
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Transmit Mask Specs
2.4 Performance Metrics 27
spectral emission is set by each radio standard as a spectral mask requirement. While
transmitting a signal, the emission levels must fall below the limits set by spectral mask.
The spectral mask requirements affect both in-band and out-of-band emissions.
Unwanted emissions are caused by a number of factors including non-linearity in the
system, noise resulting from interference with other circuits or spurious tones created by
clocks or frequency synthesizers. Because these non-idealities affect the in-band signal
they can also have an effect on the modulation accuracy. The in-band spectral mask
requirement for DCS1800 is illustrated in Figure 2.6.
Typically, one of the most difficult portions of the spectral mask requirement is
close to the carrier at. For example, Figure 2.7 shows the same GSM spectral mask at
relatively low offset frequencies. In addition a GMSK modulated signal has also been
illustrated in the same plot. Notice that the modulated signal is always below the spectral
0 1 2 3 4 5 6 7-80
-70
-60
-50
-40
-30
-20
-10
0
10
Frequency Offset (MHz)
Rel
ativ
e P
ower
(dB
c)
Figure 2.6 GSM in-band spectral mask requirement.
Chapter 2 Transmitter Fundamentals 28
mask. Non-idealities in the transmitter, which distort the signal, will bring the modulated
signal higher and thus closer to violating the spectral mask. These non-idealities will be
discussed in more detail in the following section.
In addition to in-band requirements, out-of-band spectral emissions requirements
also have an affect on transmitter design. While the out-of-band emissions don't affect
the modulation accuracy or other transmitters in the same band, the limits are often lower
than the in-band limits, making them more difficult to satisfy. For example, one of the
most difficult requirements in the DCS1800 standard is the emissions requirement that
falls in the DCS1800 receive band.
-400 -200 0 200 400
-60
-40
-20
0
Frequency (KHz)
PS
D
Figure 2.7 GSM spectral mask at low offset frequencies with GMSK modulated signal.
Every standard therefore has a transmit mask specificationthat must be met. This limits spectral regrowth, noise, andother spurious transmissions in the band and in nearby bands.Above examples are for GSM.
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FCC Limts
1 1.5 2 3 4 5 6 7 8 9 10
-40
-45
-50
-55
-60
-65
-70
-75
-80Frequency (GHz)
EIRP
Spe
ctra
l Den
sity
(dBm
/MH
z)
Frequency EIRP(MHz) (dBm/MHz)
960-1610 -75.31610-1990 -53.31990-3100 -51.33100-10600 -41.3> 10600 -51.3
While the transmit mask is standard specific, every transmittermust comply with FCC limits (in the US). The above mask isfor an unlicensed device meeting part 15 requirements.
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Error Vector Magnitude
Chapter 2 Transmitter Fundamentals 26
For certain types of modulation the difference is measured by the error vector
magnitude (EVM) while in other systems only the phase error is the critical metric.
Generally, phase and frequency modulation schemes, which are constant envelope, use
phase error as the metric and NCE modulation schemes use EVM. For example for
GMSK modulation used in DCS1800 cellular phones, the standard requires an RMS
phase error of less than 5 degrees.
2.4.2 Spectral Emissions
In addition to modulation accuracy, it is critical that a transmitter only emit a specified
amount of radiation so as not to interfere with other devices both in the same system and
in other systems. Ideally, the transmitter would only transmit a perfectly modulated
signal with no other undesired spectral emissions. However, this is rarely the case and
thus limits must be set for the levels of unwanted spectral emissions. The limit of the
Ideal Signal
Actual Signal
T Phase Error
Error Vector
I
Q
Ideal Phase
Figure 2.5 Graphical representation of the error vector and phase error.
Chapter 2 Transmitter Fundamentals 32
Another common and useful graphical representation of accuracy of a transmitted
signal involves altering the baseband input. Typically modulated data is applied to the
baseband but in this test, the baseband inputs are low frequency sinusoids given by
)2cos()( tftI bbS (2.14)
and
)2sin()( tftQ bbS . (2.15)
It might be expected that the up-converted spectrum would contain two frequencies, but
in the ideal case either the negative or positive sideband is rejected, depending on the sign
of I(t) and Q(t). When the modulator contains errors, the unwanted sideband is present
and can be used as a measure of the accuracy of the modulator. For example the SSB test
output of a modulator with two degrees of phase error is illustrated in Figure 2.11. The
upper, desired sideband is, is located at the carrier frequency plus the baseband frequency
cos(S/4-T/2)
sin(S/4+T/2)
sin(S/4-T/2)
cos(S/4+T/2
S/4
Ideal
Phase Error
T/2 T/2
Figure 2.10 Quadrature phase error constellation.
While the ACP is a good way to measure how much the PAsignal will deteriorate a neighboring channel signal, the EVMis a measure of how much the PA interferes with itself.
The EVM measures the systematic deviation of theconstellation points from the ideal positions due to amplifiernon-linearity.
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Example: GSM Transmitter Non-IdealityChapter 2 Transmitter Fundamentals 42
In addition to affecting the modulation accuracy, phase noise also changes the
modulated spectrum as illustrated in Figure 2.21. Much like the case with thermal noise,
phase noise can raise cause spectral mask violations. In this example the phase noise is
large enough to cause spectral mask violations close to the carrier. However, unlike
thermal noise, phase noise decreases as the frequency of interest moves away from the
LO frequency. Even with this property, phase noise is still a major concern at large offset
frequencies because the spectral mask often decreases faster than the phase noise. As a
result phase noise at large offset frequencies is often problematic in transmitter design.
-1 -0.5 0 0.5 1
-1
-0.5
0
0.5
1
I
Q
IdealPhase noise
Figure 2.20 GMSK constellation diagram with phase noise.
Chapter 2 Transmitter Fundamentals 40
In addition to thermal noise, flicker noise can also negatively impact a transmitted
signal. Flicker noise is present in CMOS transistors and the mean-square drain current is
given by
ffIKiaD
d ' 2 (2.18)
where K is a constant for a given device, Id is the drain bias current, and a is a technology
dependent constant. Due to the inverse relationship between the spectral density and the
frequency, flicker noise is a more significant problem at low offset frequencies.
Therefore flicker noise can impact the modulation accuracy of the transmitted signal but
generally will have little impact on unwanted spectral emissions.
-1 -0.5 0 0.5 1
-1
-0.5
0
0.5
1
I
Q
IdealThermal Noise
Figure 2.18 GMSK constellation diagram with thermal noise.
The first example shows the impact of phase noise whereasthe second plot shows the impact of noise (both amplitudeand phase).
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Transmitter Noise
2.5 Non-Idealities 39
Shown in Figure 2.17 is a DCS1800 spectral mask and the power spectrum of two
GMSK signals: one with added thermal noise and one without. The two spectra are
shown together to illustrate the effects of the noise. At low offset frequencies the two
spectra are indistinguishable and overlap one another. However, for frequency offsets
above approximately 300 kHz and below -300 kHz, the noise is clearly evident leading to
a spectral mask violations.
The effects of thermal noise on the constellation of a GMSK signal are shown in
Figure 2.18. Thermal noise will clearly cause error in the magnitude and phase of the
transmitted signal.
-600 -400 -200 0 200 400 600-80
-60
-40
-20
0
Frequency offset (KHz)
PS
D (
dB)
W ith noise
W ithout noise
Figure 2.17 GMSK signal with thermal noise.
Transmitter noise is important for two reasons. First the noiselevel should not significantly impact the EVM/BER of thetransmitter itself. More importantly, the noise leaking intoother bands must meet specs. This is especially problematicin FDD systems that transmit and receive at the same time.
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Digital and Analog Modulation
Both digital and analog modulation schemes involve amplitudeand/or phase modulation: Vo(t) = A(t) cos(ωt + φ(t))
Linearity specs of PA determined by envelope variation
Most spectrally efficient modulation schemes have largeenvelope variations
Analog FM (AMPS) uses constant envelope ⇒ can useefficient non-linear power amplifiers (60%-70%)
GMSK (GSM) uses constant envelope as well
π/4 DQPSK (IS54/136) has 3dB peak to average ratio (PAR)
QPSK (CDMA base station) has 10dB of PAR, OQPSK(CDMA handset) has 3dB of PAR
802.11g OFDM at 54 Mbps (52 sub-carriers) has about 6-8dB of PAR
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Modulation Schemes
Key specifications are the peak-to-average radio, the peakpower, and the power control range.
Constant modulation schemes much easier (GSM, AMPS).
WiFi uses OFDM, which is the hardest ! LTE up-link uses asingle carrier to ease the PA back-off requirements.
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Switching versus “Linear” PA
Two general classes of PA: Linear andNon-Linear
“Linear PAs” preserve amplitude andphase information while “Non-linear PAs”only preserve phase mod. Typically (notstrictly), linear PAs employ transistors ascurrent sources (high Z), non-linear PAsemploys transistors as switches (low Z)
Linear PAs can drive both broadband andnarrowband loads.
Non-linear PA usually drive a tunedcircuit load
Amplitude information in a non-linear PAcan be recovered by:
Oversampling, duty cycling, or varyingthe supply voltage
vi(t) io(t)
vi(t) = A(t) cos(ωt + φ(t))
io(t) = GmA(t) cos(ωt + φ(t))
−A0 cos(ω0t + φ(t))
A0 cos(ω0t + φ(t))
VDD
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Clipping: A “Linear” PA is Impossible
All amplifiers eventually clip,that is the output cannot behigher than some multiple ofthe power supply. Note thatthe peak amplifier outputcan be arbitrarily large, butthe average output powerwill limit.
If we “back-off” sufficientlyfrom the peak so that theamplifier never clips, then wecompromise the efficiency.
We can generally make acompromise and choosesufficient back-off to meetthe EVM specs.
Ideal ClippingPoint
Limited Output
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Power Back-off
In applications requiring a linear PA due to PAR, we mustback-off from the peak power point to avoid clipping thewaveform.
For 10 dB of PAR that means operating the PA at 10 dBlower power (or power back-off).
An OFDM 802.11g system that needs 20 dBm at antenna andhas a PAR of about 17 dB. That means to transmit 20 dBmaverage power, the PA should be capable of transmitting 37dBm !!!
In practice the peak amplitude is a rare event and the PAshould be allowed to clip. A 6-7dB back-off is typical.
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Power Control
Most wireless systems have power control. Power control isimportant to limit transmit power to the lowest possiblesetting. This saves battery power and limits the amount ofinterference to other nearby users.
There are two power control loops to consider: (1) Mobilepower control loop and (2) Basestation to mobile powercontrol loop
The mobile unit must transmit a given output power with acertain resolution. In GSM the output power can be off by±2dB.
in CDMA systems, the noise level is actually set by thisinterference so all users are required to back-off to make thesystem as a whole more efficient.
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Closed Loop Power Control
The mobile power control loop can be a closed loop or openloop system. In an open loop system, the power of the outputpower of the hand-held is measured and calibrated for eachDAC setting. Then an open loop system is used estimatedbased on a one-time calibration.
In a closed-loop system, the output power is estimated using adirectional coupler, a voltage measurement, or a currentmeasurement.
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Stability over VSWR
The PA generally must be able todrive a varying load. The ability todrive a given range of loads isspecified as the VSWR, e.g. aVSWR of 3:1
A system with a VSWR of 3:1 candrive any load with magnitude aslarge as 3× 50Ω or as small as50Ω÷ 3 = 17Ω.
On the Smith Chart any load lyingon a constant VSWR circle is avalid load, or any impedance suchthat
SWR−1 ≤ |x + jy | ≤ SWR+1
0.1
0.1
0.1
0.2
0.2
0.2
0.3
0.3
0.3
0.4
0.4
0.4
0.50.5
0.5
0.60.6
0.6
0.7
0.7
0.7
0.8
0.8
0.8
0.9
0.9
0.9
1.0
1.0
1.0
1.2
1.2
1.2
1.4
1.4
1.4
1.61.6
1.6
1.81.8
1.8
2.02.0
2.0
3.0
3.0
3.0
4.0
4.0
4.0
5.0
5.0
5.0
10
10
10
20
20
20
50
50
50
0.2
0.2
0.2
0.2
0.4
0.4
0.4
0.4
0.6
0.6
0.6
0.6
0.8
0.8
0.8
0.8
1.0
1.0
1.01.0
90-9
085
-85
80-8
0
75-7
5
70-7
0
65-6
5
60-6
0
55-55
50-50
45
-45
40
-40
35
-35
30
-30
25
-25
20
-20
15
-15
10
-10
0.04
0.04
0.05
0.05
0.06
0.06
0.07
0.07
0.08
0.08
0.09
0.09
0.1
0.1
0.11
0.11
0.12
0.12
0.13
0.13
0.14
0.14
0.15
0.15
0.16
0.16
0.17
0.170.18
0.18
0.190.19
0.20.2
0.210.21
0.220.22
0.23
0.230.24
0.240.25
0.25
0.26
0.26
0.27
0.27
0.28
0.28
0.29
0.29
0.3
0.3
0.31
0.31
0.32
0.32
0.33
0.33
0.34
0.34
0.35
0.35
0.36
0.36
0.37
0.37
0.38
0.38
0.39
0.39
0.4
0.4
0.41
0.41
0.42
0.42
0.43
0.43
0.44
0.44
0.45
0.45
0.46
0.46
0.47
0.47
0.48
0.48
0.49
0.49
0.0
0.0
AN
GLE O
F TRAN
SMISSIO
N C
OEFFIC
IENT IN
DEG
REES
AN
GLE O
F REFLECTIO
N C
OEFFIC
IENT IN
DEG
REES
—>
WA
VEL
ENG
THS
TOW
ARD
GEN
ERA
TOR
—>
<— W
AVE
LEN
GTH
S TO
WA
RD L
OA
D <
—
IND
UCT
IVE
REA
CTAN
CE C
OMPO
NENT (+jX/Zo), O
R CAPACITIVE SUSCEPTANCE (+jB/Yo)
CAPACITIVE REACTANCE COMPONENT (-j
X/Zo), O
R INDUCTI
VE SU
SCEP
TAN
CE (-
jB/Y
o)
RESISTANCE COMPONENT (R/Zo), OR CONDUCTANCE COMPONENT (G/Yo)
VSWR = 3 Circle
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Power Control Loops
Directional Coupler
Power Amplifier
Filter
Power Detector
Reference
AnalogPower
Control
V + ≈ V +
V −
εV −εV + Z0
A directional coupler is one of the more accurate methods tomeasure the power delivered to the load (antenna). Thepower reflected from the antenna due to a mismatch is notcomputed. But the directionality of the coupler is key.
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RF Power Transistors
In a BJT there is a direct trade-off between the breakdownvoltage and the
fT of the device. Some people define this as a metric for thetransistor.
Thus we should employ the lowest tolerable fT device for ourPA. That’s because such a device can swing the largestvoltage.
Unfortunately, the trend in technology is the opposite, mostlyfor digital and RF applications, giving us over 100 GHz fT andonly 1-2 volts to operate with. This is good for digital.
CMOS devices also require a large Cox (small Tox)
(gate control) for short channels, and thus gate oxidebreakdown is a big issue. Punchthrough is another breakdownmechanism.
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BJT Device Power Gain
For 300mA of current need20,000 mm2 of Si area
Operating at frequencies∼ fT/10 (say 2.5 GHz in 25GHz process)
Device parasitics dominateimpedance levels. Don’tforget Temp!!!
Package parasitics limit gainby providing feedback
Gain determined by (M.Versleijen et. al.):
GP =(ωTω
)2 1
1 + ωTRLCBC
RL
ωTLE + RE + RB(1 + ωTRLCintBC )≈
1CBCLE
ω2
CCjc
rb
RC
RE RS
Ccs
LGLE
B
C
E S
Cjc
rb
RC
RE RS
Ccs
LGLE
Typical Terminal Impedance Levels (CE)
Zin = 40 + j -50 Zout = 3 + j -30
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CMOS Device Losses
For each finger of the CMOS device, we must carefully extractthe capacitance and resistive parasitics. The layout will have alarge impact on these parasitics.
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Power FET Layout
sou
rce
sou
rce
drain
gate
g3 d2d2
d3d3 g2
g1 g1
g2
g3g3
g2
g1
d1 d1
GATE
DRAIN
Power FETs are typically very large (millimeter size) and thelayout is broken into sub-cells. Each sub-cell is broken intomulti-fingered transistors and the gate/drain lines are delayequalized.
The layout metals introduce significant resistance andcapacitance, which needs to be carefully modeled.
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CMOS Device Power Gain
Need a good model of device, especiallyresistive paraistics. To predict power gainneed to know the gate/source/drainresistance, including the interconnectparasitics (vias, metal, poly), and the gateinduced parasitics.
Substrate parasitics will also limit powergain and needs to be extracted accurately.
Include inductance for high frequencies orvery large “distributed” device.
Note that the point the device looksdistributed depends on the slow wavevelocity due to large gate and drain cap.
G
B
S D
CjdCjsR1 R2
R4
R3
R5
Gate Bulk
Drain
Source
Lg Rg
Ld
Ls
Rd
Rs
Cgdext
Cgsext
Cdb
Csb
Rdb
Rsb
Rbb
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Substrate Parasitics
perpenicular body contact vertical
contact
R‖ ∝ 1
W
R⊥ ∝ W
W
Notice that the placement of the substrate contacts has animportant impact on the overall substrate loss (the outputimpedance of the device). Increasing the device finger widthW will decrease the “parallel resistance” component butincrease the “perpendicular resistance”.
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MOS CV Curve
Classical
Quantum
Poly Depletion
Cgg
Cox
VGBVFB VTH
Capacitors need to be large signal accurate over voltageswing. Voltage swings maybe negative and move device intoaccumulation (due to inductors).
Make sure the CV model is accurate (BSIM capmod=2).Include quantum effects and poly depletion.
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Typical Multi-Stage PA
Po~1W=30dBm
Power Gain ~ 13dB
On-chip Matching NetworksPower Loss ~ 2dB
Off-chip Output Matching Networks
1W 100mW 10mW
1mW
Need 1W at antenna and about 30 dB of power gain
Each amplifier stage has about 13 dB of power gain
Interstage matching networks have an insertion loss of about2 dB
If you cannot afford loss at output stage, must off-chipcomponents (preferably in the package to keep them close andparasitics minimal).
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Count your dBs
In RF receiver design we throw around a lotof dB’s without giving it much thought. Forinstance, you may put in a margin 3dB inyour design. But for a PA, this is not so easy! 3dB is a factor of 2 in power !!
Likewise, any loss in the signal path can hurtthe PA efficiency considerably.
Consider designing a 1W PA with anefficiency of 65%. But due to a customerdemand, you have to budget up to 1dB ofextra loss at the output.
That means your PA efficiency canpotentially drop to 52%!
dB Linear 0.1 0.977 0.3 0.933 0.5 0.891 0.7 0.851 0.9 0.813 1.0 0.794 1.2 0.759 1.4 0.724 1.6 0.692 1.8 0.661 2.0 0.631
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What PA Needs to Drive
chip
bou
ndar
y
chip boundary
Diplexer
DCS/PCS BPF: PCS
BPF: DCS
BPF: GSM
LPF: PCS/DCS
Diode/Switch
RX
GSM
TX Isolator
Isolator
Coupler
Diode/Switch
LPF: GSM
Coupler
RX
TX
Need SAW filter to eliminate out of band emissions.Directional coupler measures output power.In a half duplex system, a switch is used for RX and TX. In afull duplex system, a duplexer is used to isolate the TX andRX. In the extreme case, a circulator can be used as well.Typical cell phone PA that needs to put out 0.5W to theantenna (LTE). Due to loss in output matching network,coupler, duplexer/diplexer, and SAW filter, need to put out anadditional 3 dB.
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Parasitic Coupling
VCC
Signal throughESD cap
Package mutual inductance/capacitance
Thermal feedbacksubstrate coupling
Self heating; Intrinsicshunt/series FB
Ground Bounce
Supply Bounce
Signal thoughmatching andbias network
Package: ESD, bias, pins, bond wires
Substrate: Devices (passive and active), thermal
Maximum safe power gain 30 dB
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Ground Bounce
Since the load current is large (amps), and it flows out of thechip to the external load, there is considerable “bounce” inthe ground and supply lines
Vbounce = LdI
dt
Besides limiting the voltage swing (efficiency), for on-chipsignals referenced to the internal ground, this is not a bigissue. But for any external signals referenced to the cleanboard ground, this ground bounce is a problem (it cansubtract or add from the input signal, for instance)
For this reason, the output stage ground is often separated tomitigate this coupling effect.
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Packaging Issues
package lead
bond wire chip
bond pad
package leadbond wire
The emitter/source inductance is a major problem as it limitsthe device swing, reducing the efficiency of the amplifier. Italso is a big source of ground bounce that can lead toinstability.
Use as many bondwires to reduce this inductance. If possible,use a package with an exposed paddle to reduce the bondwirelength.
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Downbond
package lead
bond wirechipdown bond
exposed "paddle"
To reduce the inductance to “gnd”, we can use an exposedpaddle style package, where the chip is glued to a groundplane which is directly soldered to the board.
The bond wire to ground is a downbond, and it is shorter andthus the overall inductance for the ground can be reducedsubstantially.
Leadless packages are also preferred (such as QFN).
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Flip-Chip Package
bump
chip via
pads
Flip chip packages are more expensive but allow very lowinductance bumps (< 100pH) to the package ground. Thiseliminates both the bond wire inductance and the packagelead inductance.
Another option, the entire PA can be constructed withlumped components in the package by utilizing high qualitypassives. This is more of a module than an integrated PA.
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Stability Issues
groundnoise
local ground
systemground common
mode noise
The input signal comes from an off-chip source (driver amp orVCO buffer). The local ground is bouncing due to the PAoutput stage. To reduce the effects of this ground bounce, afully differential source can be employed. If not available, atransformer can help isolate the two grounds.
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Balanced/Differential Operation
+ vn −
+ vn −
Go differential / balanced to reduce common mode coupling.
Transformer at input helps to isolate input/output.
Watch out for parasitic oscillations (see next slide).
Bypass capacitors (big and small) to cover multiple frequencybands. Big caps are usually MOS varactors.
Plan the package layout early in design.
Spend at least as much time on ground/VDD/bypass issuesas the circuit design !!
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Parasitic Oscillator
Zin
Ld
Cgd
iµ
+vi
−
+vo
−Lg
Cgd
Ld
Consider the medium frequency equivalent circuit for outputstage. Due to the large device size, the gate-to-draincapacitance is substantial. The gate inductance is for biasingor to tune out the input cap.
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How Big?
The amount of power that we can extract from a PA device islimited by the output impedance of the device. As the deviceis made larger to handle a higher DC current (withoutcompromising the fT ), the lower the output impedance.
For a current source style of PA, eventually the device is solarge that power is lost in the device rather than the load.This is the attraction of a switching PA.
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Power Combining (cont)
Lossy Power Combiner
But for a non-switching PA we must perform some powercombining to use more than one device. This way we cantransform the load into a higher impedance seen by each PA.
The power combining networks are lossy and large. We’llcome back to them later.
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Can we “wire” PAs together?
Note that we cannot simply “wire” PAs together since theimpedance seen by each PA increases by N if we connect N inparallel:
RPA =VL
IL/N= NRL
This means that each PA delivers less power for a fixed swing
PPA =V 2swing
2RPA
There is also “load pulling” effects if the sub-PAs are notperfectly in phase
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