Polarisation-diversity phase-encoded patch-antenna transponder

V.F.Fusco

Indexing terms: Transponders, Polarisation diversity, Phase encoding

Abstract: Application of phase encoding to a received R F signal, and retransmission of the signal phase encoded at the same frequency using polarisation diversity, is used to design a simple two-port microstrip patch antenna which will permit single-frequency polarisation-diversity operation. The antenna is characterised in terms of its far-field radiation pattern and its two-port scattering parameters. An injection-locked oscillator is coupled to an adapted passive quasicirculator to perform the phase-encoding function. Here control of the direct tuning voltage to the injection-locked oscillator effects the degree of phase change obtained. The operation of this circuit is discussed and its inclusion with the dual- port patch antenna to form a phase-encoding transponder is described.

1 Introduction

Many applications exist for simple architectures which allow an R F carrier to have its relative phase directly modulated by a control voltage, such as a DC signal for a steered-antenna application, or by a digital data stream for a communication link or transponder appli- cation. Many applications for encoded transponders exist, such as vehicular tagging, channel markers, friendly-fire abatement etc. In this paper, an antenna architecture is described which has minimum complex- ity and which has the potential for fulfilling the objec- tives stated above.

2 Antenna design

A simple antenna is required which will allow polarisa- tion-diversity operation, for this reason a microstrip patch was selected. The patch-antenna dimensions are chosen using the design equations presented in [l]. To allow identical receive and transmit frequencies to be used similtaneously, the length and width of the antenna are made equal. In this case they are chosen to be 70mm, i.e. the antenna length required for reso- nance at 1 GHz when constructed on 1 .bmm-thick FR4 0 IEE, 1997 IEE Proceedings online no. 19971 194 Paper first received 28th October 1996 and in revised form 3rd February 1997 The author is with the High Frequency Electronics Laboratory, The Queen’s University of Belfast, Ashby Building, Belfast BT9 5 A H , N. Ire- land, UK

printed-circuit board. This leads to an antenna width W which is smaller than that prescribed in [l] for best radiation efficiency. Since radiation conductance is related directly to antenna width, the antenna-radiation efficiency will be reduced with respect to a linearly polarised rectangular patch antenna designed for use at the same frequency and on the same material.

70 mm 4 +

70 mn

20 mm -

chip capacitor

port 2

27mmI r - port 1 f.; Fig. 1 Dual-polarised square antenna

The 5062 tap-in points on orthogonal sides of the antenna are first located approximately by a transmis- sion-line model [2] and then verified by an MOM simu- lation [3]. The positioning of these ports requires simultaneous iterative adjustment of the location of ports 1 and 2. The design objective used here is to locate 50Q matched tap-in points at port 1 and port 2 while achieving better than -1 5 dB circuit isolation between these ports. Here an edge-fed configuration was selected in preference to internally located coaxial tap-in points. This was so that the antenna geometry could be kept planar to facilitate the integration of the additional circuit components needed to enable the phase-encoder circuit to be constructed. This constraint means that some port isolation is sacrificed when com- pared with that which could be obtained from a coaxi- ally fed antenna. Ultimately, this port positioning strategy leads to nonidentical port positionings, as shown in Fig. 1. The theoretical intrinsically matched impedances realised are, 48 + $0 for port 1, 55 + j83 for port 2 at 1GHz; on both ports the reactive part is neutralised using a 1.8pF capacitor. For the antenna, S12 and S,, provide the figure of merit for circuit isola- tion between these two ports. High isolation is part of the objective in designing the antenna such that unde- sired feedback of the same frequency retransmitted sig- nal is minimised. The antenna design is completed with the inclusion of small square pads used for the posi-

201 IEE Proc-Microw. Antennas Propag., Vol. 144, No. 3, June 1997

tioning of the matching capacitors and the coaxial probe connections.

From the measured return loss for SI1, antenna reso- nance occurs at 1.0032GHz. At this frequency the input impedance is measured to be 42 + j2.0Q on port 1 and 50 -j2.0 on port 2, S,, = SI2 = -18dB measured, -21 dB simulated.

The patch is connected at port 1 for reception and at port 2 for transmission. This results in the principal excitation-current component for each port being directed towards its respective opposite edge of the patch. Thus, for this antenna, since the feed position is off-centre, current is not distributed evenly along the edges of the patch. This, in turn causes the radiation pattern to be offset yielding a boresight cut for maxi- mum radiation of 55" measured anticlockwise from the x axis in Fig. 1. A value of 59" was simulated for this quantity using commercial software [3].

2 70 Fig. 2 Measured copolar radiation pattern for square microstrip antenna - - 0 - - E plane copolarisation - - e - - H plane copolarisation

90 ..

270 Fig. 3 antenna - - 0 - - E plane copolarisation - - A - - H plane copolarisation

Measured cross-polar radiation pattern for square microstrip

Figs. 2 and 3 show the measured antenna far-field patterns results when excited at port 1 with port 2 ter- minated in 50Q, as measured through the 55" cut; here good isolation between measured copolarisation and crosspolarisation radiation patterns is observed. Similar

202

results are obtained for port 2 excitation. For the meas- ured radiation patterns shown in Figs. 2 and 3, the measured H-plane beamwidth is 100". This is simulated by commercial software [3] as 92" and computed using Bahl [l] as 115". The measured antenna gain was found to be 5dB, and its computed value was 6dB.

3 Phase encoding

Phase encoding is implemented by attempting to vary the frequency of operation of an injection-locked oscil- lator. The retransmitted phase-encoded signal is meas- ured by a test antenna and its phase measured relative to a known reference signal phase using a microwave transition analyser (MTA) configured as a phase bridge, [4]. Here phase shifting is achieved through the

~ ~~

use of a voltage-control splitter as shown in Fig.

oscillator (VCO) and a power 4.

voltaqe-control oscilGtor 8

I amplifier

injection of RFsignal

modulated signal U Fig.4 Block diagram of the phase encoder circuit

The power splitter is arranged to act as a quasicircu- lator, [5] . The R F signal is applied to the input of the amplifier where it is amplified and injected into the VCO through a 3dB Wilkinson power splitter. Signal power from the VCO is isolated from the input antenna connection by high amplifier-reverse-path iso- lation. The presence of the power splitter provides a further 3dB attenuation into the reverse path from the VCO to the antenna-input connection point. This arrangement ensures that minimal signal is fed back to the input. The amplifier also amplifies the received R F signal, thereby permitting, typically, a 100 MHz locking bandwidth for the VCO [6]. In this circuit the injection- locked VCO [7] has two functions. First it acts as a limiting amplifier producing a large output signal for a weak incoming locking signal. This occurs because once it is injection-locked, the VCO output remains at a constant amplitude which is effectively a highly amplified version of the weak incoming R F signal which caused the injection-locking phenomenon to occur [6]. Secondly, it acts as the phase-shifting or encoder element in the following manner.

Without any signal fed to the VCO, this oscillator behaves like a free-running oscillator with its frequency determined by the direct tuning voltage (VT). When an input RF signal is injected into the VCO, the oscillator acquires lock and becomes an injection-locked oscilla- tor, locking to the external R F signal. Once lock is obtained, phase variation can be achieved simply by altering the DC tuning voltage of the VCO. This action will result in a shift of the phase of the VCO signal rel- ative to the injection-locking signal since, if the injected R F frequency is within the locking bandwidth, the VCO oscillation frequency will try to maintain lock and stay in phase with the R F injection signal. How- ever, with variation of V,, the VCO oscillation fre- quency will attempt to break lock but, since it is still within the locking range, only a phase change can

IEE Proc -Microw Antennas Propag , Vol 144, No 3, June 1997

occur [SI. In this way, direct phase modulation of the carrier relative to the injection-locking signal is achieved.

4 Polarisation-diversity phase-encoded transponder

Here the two-port square patch described above is used to achieve reception and retransmission of an encoded R F signal in an orthogonal polarisation mode. A polarised signal is transmitted from antenna A and is received at port 1 of the square patch antenna. Here antennas A and B are linearly polarised rectangular patch antennas designed to operate at 1GHz. The injected power available to the phase shifter measured at port 1 is set for low-level injection, in this case -7dBm, and the separation d between the transmit and receive antennas is set for far-field operation (Fig. 5).

I - el dual-polar i sed antenna

reference

Fig.5 ponder

Phase measurement using a single patch antenna as a trans-

Once received at port 1, the incoming signal is ampli- fied by 11 dB and fed via the Wilkinson power splitter to the phase encoder. The incoming signal locks the VCO which phase encodes it as described above. The encoded signal is then fed to port 2 for retransmission with orthogonal polarisation with respect to the received signal. The overall gain for the phase-encoder circuit is about 7dB. The injection-locking signal is used as the phase reference for the microwave transi- tion analyser (MTA) which measures the relative phase deviation introduced into the return path as measured at port B relative to port A after the nominal system path length has been 'zeroed out' at the MTA. With this arrangement, 0.35V DC bias control to the VCO gives a resultant phase variation of 76" (Fig. 6).

80 -

60 .-

01 .. 0 U

$ 4 0 - .c Q ..

20 --

13.0 13.1 13.2 13.3 tuning voltage,V

Fig. 6 Phase-variation range

Here the receiving antenna is aligned for best polari- sation match with the square antenna. Note that leak- age of the signal owing to nonperfect isolation between

IEE Proc-Microw. Antennas Propag.. Vol. 144, No. 3, June 1997

antenna ports leads to a reduction in phase variation (Figs. 7 and 8). If the transmitted signal from the phase shifter is +7dBm and the antenna port isolation is 18dB, then, using vector addition as in Fig. 8, a the- oretical phase reduction of approximately 3" will occur.

resultant

RF signal from port 1

L signal

4

4 4

Fig. 7 Phase-variation reduction: VCO signal

new resultant RFsignal introduced by port 2 \ leakage

signal

Fig.8 Phase-variation reduction: Oc < 0

5 Conclusions

This work indicates that a CW carrier signal can be simultaneously amplified and phase encoded relative to a carrier signal. The circuit arrangement used is based on a quasicirculator combined with an amplifier for isolation and an oscillator tuned to operate in injec- tion-locked mode. Further, it has been shown that it is possible, using a two-port intrinsically matched patch antenna, to acquire a weak RF signal and retransmit it after triple exploitation of the circuit presented here for signal locking, amplification and phase encoding under only the influence of direct voltage tuning to the oscil- lator. Once the signal has been received, it is retrans- mitted with an orthogonal linear polarisation applied at the same frequency as the incoming signal. In this way, a simple phase-encoding transponder whose fre- quency is stabilised by the incident injection-locking

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signal and whose baseband signal is encoded onto the R F carrier by direct DC tuning control has been fre- quency stabilised by the incident injection-locking sig- nal and whose baseband signal is encoded onto the R F carrier by direct DC tuning control has been demon- strated. In addition, if the phase-encoder circuitry pre- sented here were duplicated, then an array constructed from identical elements could be made to perform beam steering under DC control of the VCO elements.

I thank L.K. Chong and S. Karode for carrying out some of the experimental work presented in this paper. The work was sponsored by the UK Engineering and Physical Science Research Council.

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7 References

1 BAHL, I J ‘Build imcrostrip antennas with paper-thin dimen- sions’, Mzcrowaves, 1979, 18, pp SO-63

2 VAN DE CAPELLE, A ‘Transmission line model for rectangu- lar mcrostrip antennas’ m JAMES, J R , and HALL, P S (Eds ) ‘Handbook of microstrip antennas’ (Peter Peregrinus, 1989), Chap 10

3 SONNET Software Inc ‘User’s manual, version 2.4’ 4 HP71 S00A Microwave Transition Analvser. Hewlett Paclcard

Reference Manual 5 KOTHER, D., HOPF, B., SPORKMANN, T., WOLFF, I., and

KOSLOWSKI. S.: ‘MMIC circulators covering the frequency range from LW-band’. Proceedings of 1995 European Microwave conference, Bologna, pp. 1186-1 190 KUROKAWA, K.: ‘Microwave solid state oscillator circuits’ in HOWES, M.J., and MORGAN, D.V. (Eds.): ‘Microwave devices, device circuit interactions’ (Wiley Interscience, 1976), Chap. 5

7 ‘Mini circuits, voltage controlled oscillators, Mode1 POS 1060’. PO Box 350166, Brooklyn, New York, USA

8 DREW, S., and FUSCO, V.F.: ‘A phase modulated active antenna’, Electron. Lett., 1993, 29, (lo), pp. 835-836

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IEE Proc -Microw Antennas Propag , Vol 144, No 3, June 1997