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IEEE SENSORS JOURNAL, VOL. 7, NO. 12, DECEMBER 2007 1749

Dual-Frequency Microwave Moisture Sensor Basedon Circular Microstrip Antenna

Mohamed Mustafa Ghretli, Kaida Khalid, Ionel Valeriu Grozescu, Mohd. Hamami Sahri, and Zulkifly Abbas

AbstractA dual-frequency sensor was developed to measure

moisture content of lossy liquids. The experiment is based on mea-

surements of far-field reflection at two frequencies in the X-band,8.48 and 10.69 GHz. The replacement of the conventional open

horn antenna with microstrip radiating patches makes the sensor

more versatile and compact. The sensor is interfaced to a note-

book computer using a multipurpose data acquisition board and

few custom made circuits. Data collection, errors correction and

calibration modules were written using LabVIEW programming

language. Three calibration methods were used to obtain an em-

pirical equation, which is applied to obtain the moisture content

of the rubber latex samples. Moisture content can be determined

with a standard error 0.49%.

Index TermsDual microwave frequency sensor, microstrip an-tenna, moisture content, rubber latex.

I. INTRODUCTION

IN THE LAST few decades, microwave dielectric-based sen-

sors were developed for moisture or water content determi-

nation in different materials [1]. The fact that these sensors rely

on measurement of dielectric properties, which are also depen-

dent on other physical properties of the material, i.e., bulk den-

sity, moisture content, temperature, etc., makes them potential

multiparameter sensors provided that appropriate correlationsare established between the measured dielectric properties and

the physical properties of interest [2].

Temperature variation causes errors in all indirect moisture

measurement methods. A common method to eliminate tem-

perature errors employed in industry involves measurement

the sample temperature and compensates for temperature

variations. To increase the accuracy of moisture content de-

termination and to reduce the influence of temperature, a

two-parameter measurement can be used. For example, in

many cases, using the magnitude ratio of reflected waves at two

microwave frequencies in X-band region (8.48 and 10.69 GHz)

Manuscript received January 22, 2007; revised August 16, 2007; acceptedSeptember 3, 2007. This work was supported in part by the Ministry of Sci-ence, Technology and Innovation, Malaysia, and in part by the University PutraMalaysia, Selangor, Malaysia. The associate editor coordinating the review ofthis paper and approving it for publication was Prof. Robert Trew.

M. M. Ghretli was with the Department of Physics, University PutraMalaysia, Serdang, Selangor, Malaysia. He is now with the High Institute ofComputer Technology, Tripoli, Libya (e-mail: [email protected]).

K. Khalid, I. V. Grozescu, and Z. Abbas are with the Department of Physics,University Putra Malaysia, Serdang, Selangor, Malaysia (e-mail: [email protected]; [email protected], [email protected]).

M. H. Sahri is with Faculty of Forestry, University Putra Malaysia, 43400UPM, Serdang, Selangor, Malaysia (e-mail:[email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/JSEN.2007.908920

Fig. 1. View of the instrument.

as a function of temperature, it is possible to obtain a tempera-ture independent behavior. This kind of setup does not involve

phase measurement and the cost will be drastically reduced.

The principle of the sensor is based on measuring reflection

of the mature fields in the far -field region from the sample

interface. This new computer controlled dual-frequency mois-

ture meter is developed for special applications, where the water

content of various lossy liquids and food products is an impor-

tant process parameter. A predetermined calibration equation

is used, avoiding the use of the expensive vector network ana-

lyzers. Three calibration schemes are described.

In this paper, we deviate from the open-ended coaxial line as

a common sensor for moisture measurements and we introducean alternative approach that uses a disk patch microstrip antenna

operating at two different frequencies in the X-Band at 8.48 and

10.69 GHz.

The advantages of the dual-frequency sensor in terms of mea-

surements accuracy are discussed as well. This sensor was tested

with Hevea rubber latex.

Although, the dual-frequency sensor was developed for tem-

perature independent measurements of the moisture content,

it can be operated in single-frequency mode when necessary.

Here, we focus in the design and the development of the dual-

frequency sensor only. An application of the sensor for temper-

ature independent measurement of the moisture content will be

presented somewhere else.

II. DEVELOPMENT OF MOISTURE METER

A general view of the dual-frequency microwave moisture

meter for the lossy liquids is shown in Fig. 1(a). In Fig. 1(b), the

circular microstrip antennas became visible. The test liquid con-

tainer is placed on the top of the microstrip antennas assembly.

The sensor system is interface to a notebook computer

through a data acquisition board (PCI NI-6024-E) having a

maximum sampling rate of 200 Ksamples/s and a 12 bit reso-

lution. This multipurpose board has 16 analog input channels

and 8 digital I/O lines. Six analog input channels are used

in a differential mode to measure the negative output signals

1530-437X/$25.00 2007 IEEE


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1750 IEEE SENSORS JOURNAL, VOL. 7, NO. 12, DECEMBER 2007

Fig. 2. Schematic diagram of microwave reflection system principle.

obtained from the two Schottky detectors and to monitor the

regulated supply voltage. The software interface was written

using LabVIEW programming language. The software incor-

porates beside the data acquisition module, an error correction

module and a system calibration module.

A. Theory

In this work, the microwave reflection system uses a bistatic

air reflection method to determine the water content in rubber

latex samples. The method employs measurements of mi-

crowave reflection magnitudes, from the sample interface, in

a region where the angular field distribution of the antenna is

essentially independent of the distance from the antenna. This

region is known as the far-fields or Fraunhofer region [3]. If the

source has a maximum overall dimension D and the wavelength

, the far-field region is commonly taken to exist at distances

greater than 2D from the source.

Schematic diagram of the microwave reflection system isshown in the Fig. 2. An incident microwave originating from

a transmitter situated the bottom of the sample experiences

in a series of reflections and transmissions from and through

the interfaces of a multilayer system. The reflection signal

arriving at the microwave receiver, situated in the same plane

with the transmitter, consists of a series of reflections which

occur at all system interfaces (air-perspex, perspex-sample, and

sample-air).

The four-layer system (air-perspex-sample medium-air) can

be visualized as a cascaded combination of two port networks.

The total reflectance of the system can be obtained from the flow

graph shown in Fig. 3 using Masons nontouching loops rule [4].

The ratio of the complex wave amplitude at the nodes (b) and(a) can be obtained as

(1)

where

(2)

, and represent the thickness, absorption coefficient, and

reflection coefficient of the medium, respectively. The sub-script letters and corresponds to air, perspex, and sample

Fig. 3. Signal flow graph for the four-layer system.

Fig. 4. (a) Schematic diagram for the microwave sensor with connectors.(b) Top view of the sensor.

medium, respectively. Perspex is a synthetic polymer com-monly called acrylic glass. This thermoplastic and transparent

plastic is sold by the tradenames Plexiglas, Perspex, Plazcryl,

Limacryl, Acrylex, Acrylite, Acrylplast, Polycast, and Lucite.

One must note that all the propagation constants, reflection

coefficients and impedances shown in (1) and (2) are complex

variables.

B. Sensor Design

Fig. 4(a) shows schematic diagram of the dual-frequency

sensor consisting of four circular microstrip patch antennas.

Fig. 4(b) displays the top view of the sensor. Two patches (T1

and T2) serving as transmitters are fed by two dielectric res-onant oscillators operating at the nominal frequencies of 8.48

and 10.69 GHz, respectively. The other two opposite microstrip

patches (R1 and R2) serving as the receivers, are connected to

Schottky wideband coaxial power detectors (MDS 1087-S).

The basic circular antenna geometry comprises a thin con-

ducting patch of radius a on a dielectric substrate with the di-

electric constant and the thickness , backed by a ground

plane. The conducting patch and the ground plane form the elec-

tric walls of a cavity. The resonator is excited by a line current

through a coaxial probe fed from the ground plane at the dis-

tance from the centre of the circular patch. Fig. 5 shows the

schematic design of a single circular microstrip antenna.

The four antenna elements are fabricated on a 1.59-mm-thicksubstrate of polytetrafluoroethylene RT Duroid, supported by a


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GHRETLI et al.: DUAL-FREQUENCY MICROWAVE MOISTURE SENSOR BASED ON CIRCULAR MICROSTRIP ANTENNA 1751

Fig. 5. Schematic design for a single circular microstrip antenna.

5-mm-thick aluminum disk to add strength to the assembly and

to hold the SMA coaxial connectors to each element.

Based on the cavity model [5], the resonant frequencies of

TM modes of a circular patch antenna are given by

(3)

where is the speed of light in free space, is the th root

of the first derivative of the -order Bessel function .

Due to the geometrical consideration, the dominant mode

for our antenna elements is . Consequently, the reso-

nant frequency can be obtained from (3) by introducing

. This resonant frequency does not take into ac-

count the effect of fringing along the edge of the resonator.

This effect makes the patch to look electrically larger. In 1995,

Kumprasert et al. [6] have introduced a correction replacing the

actual radius with an effective radius given by

(4)

Therefore, the dominant resonant frequency of the mode

should be modified, as shown in (5)

(5)

The actual radius of the circular patch antenna corre-

sponding to a desired resonant frequency, in our case 8.48 and

10.69 GHz, can be numerically obtain from (5). Table I shows

the properties and design parameters of circular microstrip

antennas.

When designing a number of antenna elements on a single di-

electric substrate mutual coupling between the elements has to

be taken in consideration. The detector connected to an antenna

element can receive radiation from a neighboring radiating el-

ement even in the absence of a reflecting sample. The mutual

coupling between the antenna elements is discussed in the nextsection.

TABLE I

MATERIALS PROPERTIES AND DESIGN PARAMETERS OF CIRCULAR

MICROSTRIP ANTENNA OPERATING AT 8.46 AND 10.69 GHZ

Fig. 6. Measured mutual coupling between two coaxial-fed circular microstripantennas, for E-plane coupling (dotted line) and H-plane coupling (continuousline).

C. Antennas Mutual Coupling

In general, mutual coupling is primarily attributed to the

space waves fields. It is dependent on the separation distances

between antenna elements. Another important coupling factor

is the surface waves. Surface waves exist and propagate within

the dielectric, and their excitation is a function of the thickness

of the substrate. For thin substrates, the contribution of the

surface wave to the mutual coupling can be neglected [3].Fig. 6 shows E- and H-plane coupling coefficients as a func-

tion of the distance between the radiating edges for a circular

patch of 4 mm radius, 157.5 m thickness, dielectric constant

of 2.23, and resonant frequency of 10.69 GHz. It can be ob-

served that for an edge separation satisfying the E-

and H-coupling coefficients have close values. is the reso-

nant wavelength of the antenna. In the range the

H-plane coupling configuration have a much lower coupling co-

efficient than the E-plane coupling configuration. Increasing the

distances between the radiating elements will further reduce the

coupling factor, but this increases the overall size of the sensor.

E-plane coupling configuration was chosen for the present de-

sign. The edge separation chosen was 0.89 that correspondsto a mutual coupling of dB. It satisfies one of our design


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Fig. 7. (a) S parameter (return loss) for the circular microstrip antenna T2with a resonant frequency of 8.388 GHz. (b)

S

parameter (return loss) for

circular microstrip antenna R2 with resonant frequency of 8.510 GHz.

requirements that no more than 2% of the radiated power is al-

lowed to be coupled. The separation distance of each pair is

different since each pair operates at different frequency. The

distances from the edges of the circular patches to the edge of

the substrate is about , as a normal practice by many re-

searchers to reduce the side loops of the antennas radiation pat-

tern and increase the gain.

D. S-Parameters Measurements for 8.48 GHz and 10.69 GHz

Antenna Pairs

The two ports of the Virtual Network Analyzer (VNA) are

connected to 8.48 GHz antenna pair denoted by T2 and R2 in

Fig. 4(b). Port 1 is connected to antenna T2 and Port 2 is con-nected to antenna R2. Marker 2 in Fig. 7(a) shows a peak return

loss of 15.06 dB at 8.388 GHz. For the intended frequency of

8.48 GHz, the return loss is 10.29 dB.

Using antenna T2 as a transmitter fed by 8.48 GHz oscillator,

the radiated power will beless by 6% thanif it is driven by 8.388

GHz oscillator. Similar, if antenna T2 is used as a receiver, a 6%

reduction of power is detected at antennas output port.

To optimize the performance, it is a good idea to fabricate theantenna element and test it for its resonant frequency before or-

dering the oscillator. Dielectric resonant oscillator (DRO) has a

mechanical tuning to adjust its output frequency within a speci-

fied range. Unfortunately, 8.388 GHz is not within the mechan-

ical tuning range of the 8.48 GHz DRO used and about 6% of

radiated output power is lost.

The parameter for antenna R2 is shown in Fig. 7(b).

The spectrum has a peak at 8.48 GHz with the return loss of

dB. This means that about 96% of the input power is ra-

diated and/or lost by the antenna and only 4% reflected back to

the source, being an improvement over the performance of T2.

Similar S-parameters measurements have been done for an-

tennas T1 and R1. The return loss at 10.69 GHz for T1 and R1was 7.06 and 15.2 dB, respectively [see Fig. 8(a) and (b)].

E. Sample Level

Another parameter to be considered is the liquid (sample)

height, (Fig. 1). It is desired that the microwave reflection

signal amplitude to be independent of the sample height. Equa-

tion (1) can be used to predict the minimum value of the sample

height. Fig. 9 shows the normalized reflection power of the mi-

crowave signal as function of height for a sample of distillated

water and different thickness of the perspex substrate (1, 3, 4,

and 5 mm). The perspex constitutes the bottom of the samples

container.Distilled water has the attenuation constant of 139 m

and skin depth of 7.19 mm. It can be calculated that for dis-

tillated water sample having the height of or 28.76 mm, the

propagating waves are attenuated in a ratio of 99.965% before

reaching the sample-air interface.

To ensure a total attenuation of the microwave signal through

the liquid sample, we use a minimum 50 mm height for the

rubber latex samples.

F. Sample Holder Plate Thickness

The bottom of the sample holder was sought to be a material

with very low attenuation constant for the microwave region ofinterest. Due to the dielectric (skin depth of 8 m) and mechan-

ical properties (rigidity and scratch resistance), the perspex is an

ideal material for the present design.

The thickness of the perspex layer, (Fig. 1) is another sig-

nificant parameter of the design.

The considerations about the perspex layer thickness are not

based on the total attenuation of the microwave signal through

the sample as seen in the case of the sample height. They are

based on the microwave phase change as propagates though the

perspex medium. The superposition of the incident and reflected

depends on the phase relation between them. Fig. 10 shows the

dependence of the microwave normalized reflected power of

perspex layer thickness, corresponding to a water column heightof 50 mm at 10.69 GHz.


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Fig. 8. (a)S

parameter (return loss) for circular microstrip antenna T1 withresonant frequency of 10.64 GHz. (b)

S

parameter (return loss) for circularmicrostrip antenna R1 with resonant frequency of 10.82 GHz.

The lowest ratio of the reflected power (36.3%) occurs for

perspex plate thickness of 5.1 mm corresponding to a quarter

wavelength of a wave traveling in perspex medium at a fre-

quency of 10.69 GHz. The highest ratio of the reflected power

corresponds to a half wavelength of the perspex plate (10.2 mm).

Based on the availability of the materials at this time, the per-

spex plate thickness was chosen to be 3 mm, which corresponds

to the reflected power level of dB.

G. Distance Between Antenna and the Bottom of the Sample

Holder

The distance from the antennas plane and the bottom of thesample holder (perspex) must be optimized for the maximum

Fig. 9. Normalized power of the microwave reflected signal versus watercolumn height for various perspex thicknesses.

Fig. 10. Normalized reflected power amplitude(continuous line) andreflectioncoefficient phase (dotted line) versus perspex plate thickness.

reflected power and to satisfy the far-field condition. This dis-

tance is represented in Fig. 2 by thickness of the air layer .

Fig. 11 shows experimental reflection power as a function of air

layer thickness, corresponding to the perspex layer of 3 mm and

the semi-infinite water layer at 10.69 GHz.

The position of first maxima (16.5 mm) was chosen as the

optimum distance between the sample holder bottom and the

antennas plane. This distance satisfies as well the far-field con-

dition that requires a distance bigger than 8.72 mm for the patch

antenna of radius 6.21 mm (8.48 GHz) and bigger than 6.65 mm

for the patch antenna of radius 4.83 mm (10.69 GHz).

H. Oscillators Warm-Up

To ensure the reliability of the measurements, the oscillators

must be warmed up for at least 4 min before any measurements

are performed. Fig. 12 shows the voltage output of the receivingpatches, while the oscillator warmed up.


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Fig. 11. Experimental reflection power versus distance between sample inter-face and antenna corresponding to 3 mm perspex layer thickness and semi-infi-

nite water layer at 10.69 GHz.

Fig. 12. Output voltage of the receiving patches as the oscillators warm-up.

III. MEASUREMENTS

The sensor was tested using samples of diluted concentrated

rubber latex. The samples were divided into two groups. The

first group, consisting of 12 samples with moisture content (MC)

ranging from 38.7% to 95.0% was used to calibrate the sensor

system. The second group, with the same number of samples,

but with random moisture content was used for the test.

A. Single-Frequency Calibration

After the oscillators have stabilized their output, the signals

corresponding to open air and a sample of distillated water are

recorded. The output of the receivers is sampled at a rate of 1000

sample/s. A total of 10 000 points are recorded and averaged for

each measurement. This attenuated substantially the white noise

present in the system.

The power of the reflected microwave signal from the inter-

face of each diluted latex solution sample (Pi) is normalized bythe value corresponding to the distillated water sample (Pw). In

Fig. 13. Calibration curves for diluted Hevea rubber latex at 8.48 and10.69 GHz.

TABLE II

POLYNOMIAL COEFFICIENTS OF THE CALIBRATION CURVES FOR THE

MOISTURE CONTENT SENSOR AT 8.48 GHZ AND 10.69 GHZ

the following discussion, for convenience, the normalized re-

flected microwave power (P/Pw) is simply denoted by .

It has been found that a third-order polynomial, as shown in

(6), can describe accurately the moisture content-reflected mi-crowave power characteristic of the sensors for both frequencies

(6)

Fig. 11 shows calibration graphs for distillated Havea rubber

latex at 8.48 and 10.69 GHz. The polynomial coefficients in-

volved in (6) are shown in Table II together with the goodness

of fit .

B. Dual-Frequency Calibration and Measurements

Validity of the calibration equations at 8.48 and 10.69 GHz

for the moisture content sensor was examined with a series of

12 samples with MC unknown (Fig. 13). In the beginning, themoisture content of each sample was determined for both fre-

quencies using (6) with the corresponding polynomial coeffi-

cients shown in Table II. One should note that the MC of the

samples could be determined using only any one of the two fre-

quencies. Ideally, the two values of the MC for a given sample

obtained at 8.48 and 10.69 GHz should be identical.

In the next sections, three methods used to improve the accu-

racy of moisture content determination are described.

1) Average Method Prediction: The predicted moisture con-

tent can be slightly improved using the average of the moisture

contend at 8.48 GHz and 10.69 GHz , as shown

in (7)

(7)


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2) Weighted Average Method Prediction: The accuracy

of MC determination could be further improved by using a

weighted average of and instead of normal average

(8)

where the weighting coefficients and satisfy the following

constrain:

(9)

The weight coefficients and can be calculated from the

standard deviation of the power reflection measurements at 8.48

GHz and 10.69 GHz , as shown below

(10)

with

(11)

is the -sample at first frequency (8.48 GHz), is the cor-

responding average value, and is total number of recorded

samples.

3) Multivariate Regression Calibration Method: Another

method used for moisture content prediction is based on a

multivariate model. One must recall that the system uses two

reflected signals and at two different operating frequen-

cies. The signals and represent two dependent variables

in this analysis. The independent variable is the MC of the

sample, obtained using standard oven technique.

The MC can be predicted using both reflection measurementsand through a third-order polynomial expression shown

in (12)

(12)

For the below discussion, it is useful to introduce the following

vectors:

(13)

(14)

(15)(16)

where is the polynomial coefficients vector, and

are the vectors corresponding to the observations at 8.48 and

10.69 GHz, respectively, is the standard moisture content

vector and is total number of observations. The reflected

power matrix, can be constructed as

(17)

TABLE III

POLYNOMIAL COEFFICIENTS FOR THE REGRESSION MODEL

Fig. 14. Residue comparison for the MC determination methods.

Now, the multivariate calibration equation can be written in a

matrix form as

(18)

By solving simultaneously the set of equations described by

(18), the unknown polynomial coefficients vector can be de-termined. Table III shows the obtained values of the polynomial

coefficients in (12).

Therefore, using (12), the moisture content of any unknown

rubber latex sample can be obtained. Fig. 14 shows the residues

(in standard deviations) of the three approaches described

above.

The standard error for the average method was 0.65%, for the

weighted average method was 0.49% and for the multivariate

regression method was 0.50%. It can be observed that despite

its increase complexity, multivariate regression model does not

produce better results. This is also visible in Fig. 14, where the

residues corresponding to the three methods show similar be-havior. For this reason, the present instrument implements the

weighted average method.

IV. CONCLUSION

In this paper, a dual-frequency microwave sensor based on

circular microstrip antenna has been described. Although the

sensor was tested on Hevea rubber latex solutions, tests on other

lossy dielectric liquids such as Soya milk, water-based paints,

and coconut milk are currently under study.

Three calibration schemes are described and tested. The

moisture content of the rubber latex sample could be de-

termined with a standard error of 0.49%. The sensor can

be used also for temperature independent moisture contentmeasurements.


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REFERENCES

[1] A. W. Kraszewski, Microwave Aquametry. Piscataway, NJ: IEEE

Press, 1996, pp. 2532.[2] E. Nyfors and P. Vainikainen, Industrial Microwave Sensors. Nor-

wood, MA: Artech House, 1989, pp. 8186.[3] C. A. Balanis, Antenna Theory: Analysis and Design. New York:

Wiley, 1971, pp. 764767.[4] F. L. Warner, Microwave Attenuation Measurement (IEEE Mono-

graph Series No 19). Stevenage, Herts: Peter Peregrinus, 1977, pp.272277.

[5] I. J.Bahland P.Bhartia, Microstrip Antennas. Norwood, MA: ArtechHouse, 1980, pp. 8591.

[6] N. Kumprasert and W. Kiranon, Simple and accurate formula for res-onant frequency of the circular microstrip disk antenna, IEEE Trans.

Antenna Propagat., vol. 43, no. 11, pp. 13311332, Nov. 1995.[7] B. K. Khalid, The application of microstrip sensors for determination

of moisture content in rubber latex, J. Microw. Power Electromagn.Energy, vol. 23, no. 1, pp. 4552, 1988.

[8] F. Daschner and R. Knochel, 2003 A new Transmission-Line Sensorfor Measuring the Composition of Foodstuffs Using Microwave, inProc. 5th Int. Conf. Electromagn. Wave Interaction with Water and

Moist Substances, Rotorua, New Zealand, Mar. 2326, 2003, pp. 524.

Mohamed Mustafa Ghretli was born in Tripoli,

Libya, in 1965. He received the B.S.E. (Hon) degreein applied physics from McMaster University,Hamilton, ON, Canada, in 1984, the M.Sc. degree

in applied optics from Windsor University, Windsor,ON, in 1992, and the Ph.D. degree in microwavephysics from the UniversitiPutra Malaysia, Selangor,in 2005.

Currently, he is a Lecturer at the Higher Collegefor Electronic Technology, Tripoli, Libya. His re-search interests include microwave moisture sensors,

microwave dielectric properties of materials, and computational physics.

Kaida Khalid was born in Terengganu, Malaysia, in1952. He received the B.Sc. degree in physics fromthe National University of Malaysia, CITY LO-CATION, in 1976, the M.Sc. degree in solid-statephysics from the University of London, London,

U.K., in 1978, and the Ph.D. degree in electronicand electrical engineering from the University ofBirmingham, Birmingham, U.K., in 1986.

Currently, he is a Professor of Microwave Physicsand Techniques with theDepartment of Physics, Uni-versiti Putra Malaysia, Serdang. His research inter-

estsinclude microwave moisture sensors, microwave dielectric properties of ma-terials, and high-power microwave applications. Most of his work is primarilyengaged with the development of moisture sensors for hevea rubber latex andoil palm fruit.

Ionel Valeriu Grozescu was born in Prahova,Romania, in 1965. He received the B.Sc. degree inphysics and the M.Sc. degree in optics and laserstechnology from Bucharest University, Bucharest,Romania, in 1989 and 1990, respectivley, and thePh.D. degree in photothermal phenomena from theUnivesity Putra Malaysia, Selangor, Malaysia, in1994.

He was with the Institute of Atomic Physics,Romania in 1990 involved with R&D activitiesconcerning the laser technology and numerical

simulations of optical nonlinear effects. He is currently an Associate Professorwith the Department of Physics, University Putra Malaysia. His research

activities includes modeling of laser induced thermal phenomena, design, andconstruction of instruments for measurement of thermal properties of liquids,remote control, and automation systems.

Mohd. Hamami Sahri was born in Batu PahatJohor, Malaysia, in 1953. He received the Bachelorof Forestry Sciences degree from the the UniversitiPertanian Malaysia (UPM), now known as theUniversiti Putra Malaysia, Serdang, in 1978, theM.S. degree in wood products engineering from theState University of New York (SUNY), College ofEnvironmental Science and Forestry, Syracuse, in

1981, and the Ph.D. degree in wood anatomy fromUniversiti Kebangsaan Malaysia, Bangi Selangor, in1995.

He started his teaching career with UPM in 1981. He was appointed as Headof theDepartment of ForestProduction, UPM, from1996 to 1999, andpromotedto Associate Professor in 1997. He is currently the Professor of Wood Anatomyand Quality and the Dean of the Faculty of Forestry, UPM. His research inter-ests include characterization of forest plantation and less-used timber species ofMalaysia. Lately, he conducted more research works on lignocellulosic mate-rials from agricultural and forestry resources.

Zulkifly Abbas was born in Alur Setar, Malaysia,

in 1962. He received the B.Sc. degree (Hon) inphysics from the University of Malaysia, KualaLumpur, in 1986, the M.Sc. degree in microwaveinstrumentation from the Universiti Putra Malaysia(UPM), Serdang, in 1994, and the Ph.D. degree

in electronic and electrical engineering from theUniversity of Leeds, Leeds, U.K., in 2000.

Currently, he is a Senior Lecturer with the Depart-ment of Physics, UPM, where he has been a facultymember since 1987. He is also the Head of the Labo-

ratory of Mathematical Sciences and Applications, Institute for MathematicalResearch, UPM, where he is the Leader of the Wave Propagation ResearchGroup. His main personnel research interest is in the theory, simulation, andinstrumentation of electromagnetic wave propagation at microwave frequenciesfocusing on the developmentof microwavesensors for agricultural applications.

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Mustafa Ieee Sensor