Microstrip filters for wireless communications using high-temperature superconducting thin films

Pergamon

Applied Superconductivity Vol. 3, No. 7-10, pp. 483496, 1995 Copyright 0 1996 Elsevier Science Ltd

0964-1807(95)00070~ Printed in Great Britain. All rights reserved 0964-1087/95 $9.50 + 0.00

MICROSTRIP FILTERS FOR WIRELESS COMMUNICATIONS USING HIGH-TEMPERATURE SUPERCONDUCTING THIN

FILMS

DAWEI ZHANG, G.-C. LIANG, C. F. SHIH, M. E. JOHANSSON, Z. H. LU and R. S. WITHERS

Conductus, 969 W. Maude Ave., Sunnyvale, CA 94086, U.S.A.

Abstract-Microstrip filter structures for 900 MHz and 2 GHz wireless communications are presented. The issue of the compactness of the filter layout using YBa2Cu307 (YBCO) high-temperature superconducting (HTS) thin films with limited wafer size is addressed. Compact forward-coupled filters with up to nine poles on 50-mm-diameter wafers, and 19 poles on 75-mm-diameter wafers were realized at 900 MHz. Micro&p lumped-element HTS filters are also presented. Ultra-narrowband microstrip filters with 0.27% bandwidth were demonstrated using a novel frequency-transformation design technique. Power-handling issues for HTS applications are discussed and filters with over 30-W power- handling capability are presented.

1. INTRODUCTION

The wireless communication industry represents a rapidly expanding commercial marketplace. With the development of technology and high customer expectations, carriers are faced with demands for increased capacity and improved call quality. To that end, the performance of the RF front end of the base-station is very critical.

In conventional preselect filters in cellular communication, cavity filters have to be used to achieve the required unloaded Q values. These cavity filters are large, especially when a large number of poles has to be used in the filter to achieve the required rejection skirts. With the invention of high-temperature superconductors, superconducting planar filters have been shown to provide very low insertion loss and very sharp skirts [l-l 11. Unloaded Q-factors of the order of 40,000 have been demonstrated in the planar superconducting microstrip configuration at 77 K [12, 131. With such high resonator Q values, very narrowband filters can be realized in planar circuit form [14, 151. For the same reason, filters with a large number of poles (e.g. 19 poles) can also be achieved in a compact, planar circuit configuration [16].

In low frequency applications, such as the present cellular communication frequencies near 900 MHz, it is difficult to fit a conventional microstrip filter within the limited size of currently available superconducting wafers. For example, a half-wavelength microstrip resonator at 900 MHz on a 0.5-mm-thick LaA103 (LAO) substrate (E =24.1) is over 43 mm long. It is impossible to fit a nine-pole filter with such resonators on a single 50-mm-diameter superconducting wafer using the popular quarter-wavelength parallel-coupled approach.

In this paper, we present HTS thin film microstrip filters for application to wireless communications, from the point of view of the microwave filter designs. Emphasis is placed on accurate design and modeling of these high-performance devices, rather than on details of high- temperature superconductivity itself. All filters presented here were fabricated usi;lg double-sided superconducting thin films (i.e. for microstrip and ground plane) to reduce the filter insertion loss. Two design approaches are presented to resolve the size issue of the filter layout on to limited size wafers. The first approach is a forward-coupled distributed-element filter structure, which consists of a parallel array of fully-aligned half-wavelength resonators. Using this approach, filters with up to nine poles on 50-mm-diameter wafers, and up to 19 poles on 75 mm-diameter wafers were

483

484 DAWEI ZHANC et al.

realized using double-sided HTS thin films on LaA103 substrates at 900 MHz. The second approach is the microstrip lumped-element filter approach. Using this approach, a nine-pole HTS filter at 900 MHz was realized on a single 50-mm-diameter LaAlOs substrate, using HTS thin films for both microstrip and ground plane. In addition, a frequency transformation technique is described. Using this technique, ultra-narrowband filters were realized in a compact microstip configuration. Microstrip filters with low-impedance resonators, i.e. with wide-line resonators, are also presented for high-power applications.

2. FORWARD COUPLED MICROSTRIP FILTERS

The coupling between two parallel, identical half-wavelength planar resonator lines is a function of the length of the overlap section. For stripline resonators, which have symmetric dielectric substrates and ground planes on both the top and bottom sides of the circuit, the coupling is at a null when the two resonator lines are fully overlapped. This is because the even and odd mode phase velocities of the coupled stripline resonators are identical (pure TEM propagation mode). For microstrip lines, which have a dielectric substrate and ground plane only on the bottom side of the circuit, the phase velocities of the even and odd modes are slightly different due to its quasi-TEM propagation mode. This difference in phase velocities results in a finite coupling even when the resonator lines are fully aligned. In this case, the backward coupling between two resonator lines is zero. The design approach of such forward-coupled microstrip filters is thus based on the coupling due to this even and odd mode phase velocity difference of fully aligned half-wavelength microstrip lines [ 1 l-l 3, 16, 171.

For two parallel coupled half-wavelength microstrip lines, the admittance inverter Job and resonator susceptance Bk can be found as

wt3,csc(~,) - z3ocwm Jab = Jkvk+l = z& + Z& - 2Z,,Z,,(cot(&_) cot(&) + csc(e,)csc(e,))

& = &+I = -wL, cow + z&w,)>

Z& + z& - 2Zo,Z,,(cot(e,) cot(8,) + csc(&>csc(8,>> ’

where Z,, and Z,, are the even- and odd-mode impedances of the coupled lines,

e,=$&,de,=$p E,, are the electrical lengths corresponding to even and odd mode

propagation, and E,& and s:r are even and odd mode effective dielectric constants. The resonator slope parameter is then

bk = bk+, = ?!! 2 do w=w,

and the coupling coefficient can finally be calculated as

‘b,k+l = ;j$&. The length of each resonator section can be chosen from

0, + 0, ___ = rc for half-wavelength 2

resonators. The spacing S between the adjacent resonators can be solved by equating the above coupling coefficient k k,k+ 1 to the desired coupling values from the low-pass prototype. To determine the proper resonator spacing required for a specific coupling coefficient, standard microstrip solutions can be applied, such as in Hammerstad and Jensen’s functional approximation of numerical results [18]. Some of the dimensions may have to be fine tuned using electromagnetic full-wave simulation tools due to the limited accuracy and parameter ranges of the empirical models.

Figure 1 shows the mask layout of a nine-pole, 900 MHz, 2.8% fractional bandwidth forward- coupled bandpass filter on a 50-mm-diameter, 0.5-mm-thick LaAlOs substrate. It can clearly be

Microstrip filters for wireless communications 485

Fig. 1. Layout of a nine-pole, 900 MHz, 2.8% bandwidth forward-coupled filter on a single 50-mm- diameter 0.5-mm-thick LaAlOs substrate.

seen that without using the forward-coupled structure, it would be very difficult to fit this nine- pole filter on a single 50-mm-diameter wafer in a distributed-element configuration.

The measured response of this filter using double-sided YBCO on a LaA103 substrate at 77 K is shown in Figs 2 and 3. Figure 2 shows the transmission response of the filter in a coarse scale. The extra transmission zero at the upper band-edge is the result of the extra coupling of non-adjacent resonator lines. This coupling helps to make the filter upper-band rejection skirt even sharper. Figure 3 shows the transmission response in a fine scale and the measured return loss of the filter.

Frequency (GHz)

Fig. 2. Measured response at 77 K of the nine-pole, 2.8% bandwidth, 900 MHz filter shown in Fig. 1, fabricated using double-side coated YBCO thin films on a 0.5~mm-thick LaA103 substrate.

486 DAWEI ZHANG et al.

Fig has

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i c

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f *. ._ ; ; .* :: -. ; 6 : -20 4 : . : :: . : :: * i .* i

I!! -6yc

: : : : : : i ;t :i .; :i

: : i :: s : : i ; : .: : ::

r- -7: ; i : : : . >- -25 8

. :; :: ; : \i ; ,J : ; :

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-10 lllu’ll~l”‘ll’llll’llll~ 40 0.88 0.89 0.90 0.91 0.92 0.93

Frequency (GHz)

. 3. Measured transmission and return loss responses at 77 K of the filter shown in Fig. 1. This less than 0.27 dES insertion loss, less than 0.1 dB passband ripple, and better than 16 dB return

0

-1

-2

s-i‘s s

4’ :I -5

!!v c -7

-8

9

-10

: film#lf t film #2

r

t I

i

filter loss.

0.88 0.89 0.90 0.91

Frequency (GHz)

0.92 0.93

Fig. 4. Comparison of measured transmission response at 77 K between two forward-coupled filters with identical design, fabricated on two separate LaA103 substrates.

This filter exhibits 0.27 dB insertion loss, less than 0.1 dB passband ripple, and better than 16 dB return loss. The unloaded resonator Q-factor of this filter, estimated from the measured dissipation loss of the filter, is 40,000.

It is noteworthy that the measured filter data is obtained right after the photolithographic patterning processing without any tuning process. The measured low passband ripple and high


Fig. 5. Mask layout of the 19-pole, 900 MHz forward-coupled bandpass filter on a 75-mm-diameter, O.%nrn-thick LaA103 substrate.

0

-80

0.80 0.85 0.00 0.05 1.00

Frequency (GHz)

Fig. 6. Measured transmission response of the 19-pole, 900 MHz bandpass filter at 77 K. The dissipation loss in the passband is 0.5 dB. The measured return loss of this filter is better than 15 dB.

return loss shows an excellent design accuracy. The substrate twinning, which we assume creates dielectric constant variations across the LaAlOs substrate, is not an issue in this case. The resonator lines are so long that any local variations of the dielectric constant are averaged over the resonator length. This was varified by another filter fabricated with the identical design on a different wafer, which showed the same passband ripple and return loss level. The less than 0.05% shift in center frequency between the two filters is due to substrate thickness variation from one wafer to another. The comparison of the response of the two filters is shown in Fig. 4 on a fine scale.


Because of the demonstrated high accuracy in design and high unloaded Q values, it is possible to design filters with many more poles, making the filter rejection skirt extremely sharp. Figure 5 shows the layout of a 19-pole, 900 MHz filter on a 75-mm-diameter, 0.5 mm-thick LaAlOs substrate. The compactness of this forward-coupled distributed-element filter is clearly demonstrated in this case. The filter was fabricated on a 75-mm-diameter, double-side YBCO coated LaAlOs substrate grown by MOCVD technique. The measured response of the filter at 77 K is shown in Fig. 6. This filter exhibits very sharp skirts because of the large number of poles used in the filter. Three extra transmission zeroes were located at the upper band-edge, making the upper band-edge rejection sharper. Again, these transmission zeros were accomplished by cross- coupling of non-adjacent resonator lines. The 45 dB out-of-band rejection point is only 0.8 MHz from the upper band edge. Within the passband, the dissipation loss of the 19-pole filter is only 0.5 dB, indicating an average unloaded resonator Q value of around 10,000 across the entire 75 mm-diameter wafer. The return loss of the filter is over 15 dB.

3. LUMPED-ELEMENT MICROSTRIP FILTERS

The lumped-element microstrip filter structure is another candidate for compact filters for cellular applications. In this approach, resonators do not have to be a half-wavelength long. Instead, lumped-element inductors and capacitors are used to form resonant structures [19, 201. The flexibility of realizing these lumped elements gives us another dimension in the compact filter layout.

One of the most popular circuit prototypes for a planar microstrip lumped-element bandpass filter realization is the “tubular” circuit structure. In this circuit (Fig. 7) the number of inductors represents the order of the bandpass filter, or the number of poles of the filter. A rc-capacitor network is inserted between each pair of adjacent inductors. Similar rc-capacitor networks are also used at the beginning and end of the circuit to match the input and output impedance. For design simplicity, all the inductors are transformed to the same inductance value. Using this circuit, the parasitic grounding capacitance of the inductors can be absorbed by the neighboring n-capacitor networks, so that the circuit prototype synthesized from the ideal filter response can be accurately realized in a planar microstrip configuration.

For HTS thin film passive applications, via-holes and cross-over layers should be avoided to eliminate any possible weak links generated at the boundaries of cross-over layers, to avoid any additional losses due to the via-hole, and to simplify the processing and reduce the cost [21]. Without the use of spiral inductor structures, the realizable range of inductors in the microstrip configuration is limited due to the proximity of the ground plane. To minimize the layout size, filter circuit impedance levels usually have to be transformed to lower values. This process changes the values of the inductors and capacitors correspondingly without changing the circuit response. Inductors were realized in microstrip form first. Their parasitic grounding capacitors were then subtracted from the corresponding neighboring rc-capacitor networks. The n-capacitor networks were then realized. By cascading all the individual components together, a final lumped- element filter design was then completed.

Fig. 7. Topology of the “tubular” lumped-element bandpass filter circuit.


I 1 I I I I 1 I I ., - -40 0.75 0.76 0.77 0.76

Frequency (GHz) Fig. 8. Measured response at 77 K of a five-pole lumped-element YBCO microstrip filter. This filter has less than 0.2 dB insertion loss, less than 0.05 dB passband ripple, and better than 20 dB return loss.

For lumped-element microstrip filters on LaAlOs substrates, the measured response indicates that the dielectric constant of a LaAlOs substrate is not uniform across the wafer. This result seems to be found for most devices made on LaA103 wafers, and we assume that the variations in dielectric constant are the result of the extensive twinning that is easily seen by eye in such substrate. However, we do not know of any direct measurements of dielectric constant across the twin boundaries. This localized effect of the dielectric constant will inevitably change the resonant frequency of each resonator and the coupling strengths between the resonators from the original design. In this case, external filter tuning is required to offset the dielectric constant variations and other design and fabrication accuracy limitations. This tuning process is especially important for narrow-band filters on LaAlOs substrate which demand high accuracy in filter design. Figure 8 shows the measured response of a five-pole lumped-element filter fabricated using double-sided

Fig. 9. Mask layout of a nine-pole, 900 MHz lumped-element bandpass filter on a 50-mm-diameter, 0.5 mm-thick LaAIOs substrate.


0.80 0.85 0.90 0.95 1.00

Frequency (GHz)

Fig. 10. Measured transmission response at 77 K of the nine-pole, 900 MHz lumped-element microstrip filter shown in Fig. 9, fabricated using double-sided coated YBCO thin films on a OS-mm-

thick LaAlOs substrate.

YBCO thin films on a 0.5-mm-thick LaAlOs substrate at 77 K. The filter exhibits an insertion loss of less than 0.2 dB, a passband ripple of less than 0.05 dB, and a return loss of over 20 dB.

Figure 9 shows the mask layout of a nine-pole lumped-element microstrip filter at 900 MHz on a 50 mm-diameter LaAlOs substrate. The measured response at 77 K of this filter fabricated using double-sided YBCO is shown in Figs 10 and 11. Figure 10 shows the transmission response of the filter in a coarse scale. The extra transmission zero at the lower side of the passband is caused by the non-adjacent resonator coupling, which helps sharpen the rejection skirt at the lower side of the band edge. Figure 11 shows a blow-up of the transmission response and the measured return loss of the filter. Again, this filter has less than 0.3 dB insertion loss, less than 0.1 dB passband ripple, and better than 17 dB return loss. Pseudo-elliptical filters with transmission zeroes on both

0

-15 g

I -20 s

5 -25 z

0.89 0.90

Frequency (Gt-k)

Fig. 11. Measured transmission and return loss response at 77 K of the filter shown in Fig. 9. This filter has less than 0.3 dB insertion loss, less than 0.1 dB passband ripple, and better than 17 dB return loss.


-10

B -20

s

tj -30

:5 E -40 t g _gJ

-80

-5

-10

m^ -15 %

3

-20 5

a”

-25

-30

0.80 -85

0.82 0.84 0.88 0.88 0.80

Frequency (GHz)

Fig. 12. A typical measured response at 77 K of a pseudo-elliptical bandpass filter which has transmission zeroes on both sides of the band edge, fabricated using double-side coated YBCO thin

films on a 0.5~mm-thick LaA103 substrate.

sides of the passband have also been realized using the microstrip lumped-element approach. Figure 12 shows a typical measured response of such filters at 77 K using YBCO microstrip and ground plane on a LaAlOs substrate.

Unlike stripline, coupling between microstrip resonators decreases very slowly as a function of the resonator separation. Therefore, it is difficult to realize narrowband filters (e.g. < 0.1% fractional bandwidth) in a reasonably-sized microstrip configuration due to the very weak coupling coefficient values required. Microstrip resonators usually must be placed very far apart to achieve the very weak coupling for narrowband filters. In this section, we present a class of lumped-element filters that uses capacitively-loaded inductors to give frequency-dependent inductance values. A novel frequency-transformation technique is used in the design process. Using this approach, strong coupling capacitors can be used in narrowband filter designs. The frequency-dependent inductance transforms the filter to a narrower bandwidth than the original circuit prototype, and does not require small capacitance values for coupling capacitors. Design is accomplished by starting with a moderately narrow-band filter architecture and applying a frequency transformation to the circuit. This method results in very narrowband filters that are difficult to realize using other conventional microstrip circuits, while still using strong coupling capacitor values in the design to keep the resonator sections close together in a compact form [14, 151.

4. ULTRA-NARROWBAND MICROSTRIP FILTERS

For simplicity, consider the lumped-element bandpass filter circuit in Fig. 7 again. In each rc-capacitor network, the series capacitor, Cc,i, is the coupling capacitor between adjacent resonators. Narrowband filters always have very weak coupling between resonators, and require very small values of the coupling capacitors, Cc,i. In microstrip realizations of this circuit, small Cc,i values mean larger resonator separations, that are often difficult to realize due to the limited wafer size.

In this section, we show that we can realize very narrowband filters without using weak coupling capacitance, if a frequency-dependent inductor, L’(o), is used to replace the original


a’((?-)) pure inductors L. By using the inductor slope parameter, - do ’

to govern the bandwidth of the

filter, we can still use easy-to-realize strong coupling capacitors from the relatively wideband filter circuit prototype. As shown in ref. [15], the real filter fractional bandwidth, after using the frequency transformation technique, will be

where 00 is the filter center frequency, Aoc is the filter bandwidth in the circuit prototype, and L’(w) is the frequency-dependent inductance, with L’(w,,) = L. From this bandwidth transformation relationship, it can be seen that to achieve a narrowband filter, we can actually design a broader bandwidth circuit prototype filter with easily realizable coupling capacitors, and then

choose an appropriate inductor slope parameter, dL’(o) -, to achieve the narrowband filter performance. do

The key to the frequency transformation technique described above is the slope of the inductor values as a function of frequency. In the usual transmission line realization of inductors, the

inductor slope parameter, k = dL’(o) -, has a negative value because of the parasitic capacitance to

ground. Thus, the bandwidth odf”the lumped-element filters designed using the conventional transmission line approximation of inductors will always have a broader bandwidth than the original design. (This phenomenon has been observed in ref. [20].) In order to achieve a positive k value, which transforms the bandwidth to the narrower side, other L’(o) mechanisms have to be introduced into the circuit.

One simple realization of L’(w) with a positive k is a single capacitor C in parallel with an inductor Lo. From the resultant impedance Zeq,

1 1 - - +jwc

A -.i~L0 Z

Z eq = joL’(m),

we calculate the inductance c.$ =&as

= Lo

- w2L&

Lo is inductance of the inductor and C the capacitance of the in parallel the inductor. The parameter,

k=dL’(w) = 2ooL;c

do 00 (1 - oz,L&)2

has a positive value. This parallel L-C component can be easily realized using a half-loop of inductor in parallel with an interdigital capacitor.

To demonstrate the frequency transformation technique, we made a five-pole bandpass filter at 900 MHz with 0.27% fractional bandwidth using double-sided YBCO HTS thin films, operating at 77 K. The layout of the filter is shown in Fig. 13. The measured response of this five-pole 0.27% filter is shown in Fig. 14. This filter was designed with coupling capacitors of a 1% filter and then transformed to 0.27% bandwidth using an appropriate inductance slope parameter. To our knowledge, this filter has the narrowest bandwidth reported in the microstrip configuration.


Fig. 13. Mask layout of a five-pole, 0.27% bandwidth lumped-element bandpass filter at 900 MHz on a 50-mm-diameter, 0.5~mm-thick LaAlOs substrate. The ultra-narrowband filter is achieved by using capacitively-loaded inductors which transform the filter bandwidth from the filter circuit prototype

0

-70

-70

-80

bandwidth of 1% to the measured 0.27% bandwidth.

I ,,,I,,

0.87 0.88 0.89 0.90 0.91

Frequency (GHz)

Fig. 14. Measured transmission response at 77 K of the filter shown in Fig. 13, fabricated using double- side coated YBCO thin films on a 0.5 mm-thick LaAlOs substrate. This filter has a measured fractional bandwidth of 0.27%, with a dissipation loss of 1.2 dB in the passband. The return loss of this filter is

better than 20 dB.

5. MICROSTRIP FILTERS FOR HIGH POWER APPLICATIONS

In transmitter applications in wireless communication, filters placed after power amplifiers need to handle reasonably large amounts of RF power. For HTS filters, this is challenging both in materials and design. To enable microstrip resonators to handle more power, the linewidth of the resonators should be increased so that the current density in the cross-section of a microstrip resonator line can be reduced. In addition, superconducting wafers should be bonded to a thermally matched package to maximize the thermal contact to the cold head, which, in turn, minimizes the heat localization in the YBCO films [22-241.

Figure 15 shows the layout of an eight-pole bandpass filter at 2 GHz with 10-R internal resonator impedance on a 75mm-diameter, 0.5~mm-thick LaA103 substrate. By reducing the line impedance from 50 Sz to 10 R, the linewidth is increased from 0.17 mm to 3 mm, an l&fold increase. The measured transmission response of this filter fabricated with double-sided niobium thin films on a LaAlOs substrate is shown in Fig. 16. It should be pointed out that the ultimate


Fig. 15. Mask layout of an eight-pole. 1.2% bandwidth microstrip bandpass filter at 2 GHz, with 1 0-n ointemal resonator impedance, on a 75mm-diameter, 0.5~mm-thick LaAlOs substrate.

1.90 1.95 2.00

Frequency (GHz)

2.05 2.10

Fig. 16. Measured transmission response at 4.2 K of the filter shown in Fig. 15, fabricated using double-side coated Nb superconducting thin films. This microstrip filter has over 85 dB measured out-

of-band rejection.

rejection of this microstrip filter is over 85 dB, demonstrating that even using microstrip structures, one can still reach a very high out-of-band rejection.

Figure 17 shows the third-order-intermodulation measurement at 45 K of a five-pole, 10-Q microstrip filter at 2 GHz using double-sided YBCO on a LaA103 substrate. The third-order intercept is 62 dBm with a best linear fit. The upper curve is the fundamental mode output, and the lower curve is the third-order output. All measured powers are per tone. Figure 18 shows the insertion loss increase of a similar filter measured at 45 K with the increase of the input power at the upper band edge. It can be seen that with over 30-W input power to the filter, the insertion loss increase is only 0.15 dB, which shows the high power-handling potential of microstrip filters [23, 241.


-20 0 20 40

INPUT POWER (dam)

60

Fig. 17. Third-order-intermodulation measurement at 45 K of a five-pole, 1.2% bandwidth microstrip bandpass filter at 2 GHz. The third-order intercept is 62 dBm with best linear fit. The upper curve is the fundamental mode output, and the lower curve is the third-order output. All measured powers are per

tone (measured by D. E. Oates and A. C. Anderson, MIT Lincoln Lab).

o.20 I

INPUT POWER (dBm)

Fig. 18. Insertion loss increase of the filter in Fig. 16 at 45 K with the increase of the input power at the band edge (at 2.012 GHz) [measured by D. E. Oates and A. C. Anderson, MIT Lincoln Lab].

6. CONCLUSION

In conclusion, we have presented both designs and measurements of HTS microstip filters for cellular and PCS/PCN wireless communications. A compact forward-coupled filter approach, and a lumped-element filter approach were demonstrated using double-sided YBCO on LaA103 substrates. Ultra-narrowband filters with 0.27% fractional bandwidth were also achieved in microstrip configuration. Issues for filter power-handling capability were discussed and filters with over 30-W power-handling capability were demonstrated. lipSUP 3-7/10-J


Acknowledgements-The authors wish to thank L. Lui-Wong, A. Delta Cruz and A. Barfknecht for filter

processing, R. Yu for computer data acquisition, T. Luong and Q. Thai for filter tuning, J. Scohor for help in

packaging, and A. C. Anderson and D. E. Oates at MIT Lincoln Lab for high power measurement. This work is

partially supported by ARPA under the auspices of the Consortium for Superconducting Electronics (CSE).

REFERENCES

1.

2.

3.

4.

5.

6.

7.

8.

9.

10.

11.

12.

13.

14.

15.

16.

17.

18. 19.

20.

21.

22.

23.

24.

G. L. Matthaei and G. L. Hey-Shipton, Novel staggered resonator array superconducting 2.3 GHz bandpass filters, in 1993 IEEE Microwave Symp. Digest, pp. 1269-1272, Atlanta, Georgia (June 1993). G. L. Matthaei and G. L. Hey-Shipton, High-temperature superconducting 8.45-GHz bandpass filter for the deep space network, in 1992 IEEE Microwave Symp. Digest, pp. 1273-1276, Atlanta, Georgia (June 1993). W. G. Lyons, R. S. Withers, J. M. Hamm, A. C. Anderson, P M. Mankiewich, M. L. O’Malley, R. E. Howard R. R. Boneni, A. E. Williams and N. Newman, High-temperature superconductive passive microwave devices, in 1992 IEEE Int. Microwave Symp. Digest, pp. 1227-1230, Boston, Massachusetts (June 1991). S. H. Talisa, M. A. Janocko, C. Moskowitz, J. Talvacchio, J. F. Billing, R. Brown, D. C. Buck, C. K. Jones, B. R. McAvoy, G. R. Wagner and D. H. Watt, Low- and high-temperature superconducting microwave filters, IEEE Trans. Microwave Theory Tech. Vol. 39, pp. 1448-1454 (Sept. 1991). R. R. Mansour, Design of superconductive multiplexers using single-mode an dual-mode filters, IEEE Trans. Microwave Theory Tech. Vol. 42, No. 7, pp. 1411-1418 (July 1994). R. R. Bonetti and A. E. Williams, Preliminary design steps for thin-film superconductor filters, in 1990 IEEE Int. Microwave Symp. Digest, pp. 273-276, Dallas, Texas (May 1990). J. A. Curtis and S. J. Fiedziuszko, Miniature dual mode microstrip filters, in 1991 IEEE Int. Microwave Symp. Digest, pp. 443446, Boston, Massachusetts (June 1991). A. Fathy, D. Kalokitis, V Pendrick, E. Belohoubek, A. Pique and M. Mathur, Superconducting narrow band pass filters for advanced multiplexers, in 1993 IEEE Int. Microwave Symp. Digest, pp. 1277-1280, Atlanta, Georgia (June 1993). H. S. Newman, D. B. Chrisey and J. S. Horwitz, Microwave devices using YBa2Cus07--6 films made by pulsed laser deposition, IEEE Trans. Magnetics, Vol. 27, No. 2, pp. 2540-2543 (Mar. 1991). S. J. Hedges and R. G. Humphreys, Extracted pole planar elliptic function filters, Proc. ESA/ESTEC Workshop on Space Applications of High-temperature Superconductors, pp. 97-106, Noordwijk (Apr. 1993). M. S. Schmidt, R. J. Forse, R. B. Hammond, M. M. Eddy and W. L. Olson, Measured performance at 77 K of superconducting microstrip resonators and filters, IEEE Trans. Microwave Theory Tech. Vol. 39, pp. 1475-1479 (Sept. 1991). D. Zhang, G. -C. Liang, F. F. Shih, R. S. Withers, M. E. Johansson and A. Dela Cruz, Compact forward-coupled superconducting microstrip filters for cellular communication, 1994 Applied Superconductivity Conference, Boston, Massachusetts (Oct. 1994). D. Zhang, G. -C. Liang, C. F. Shih, M. E. Johansson and R. S. Withers, Accurate design of compact forward-coupled microstrip filters using an integral-equation field solver. ht. J Microwave and Millimeter-Wave Computer-Aided Engng. Vol. 5, No. 5, pp. 324-330 (1995). D. Zhang, G. -C. Liang, C. F. Shih and R. S. Withers, Narrowband lumped-element microstrip filters using capacitively- loaded inductors, in 1995 IEEE Int. Microwave Symp. Digest, pp. 379-382, Orlando, Florida (May 1995). D. Zhang, G. -C. Liang, C. F. Shih, M. E. Johansson and R. S. Withers, Narrowband lumped-element microstrip filters using capacitively-loaded inductors. IEEE Trans. Microwave Theory Tech. Vol. 43, No. 12 (Dec. 1995). D. Zhang, G. -C. Liang, C. F. Shih, Z. H. Lu and M. E. Johansson, A 19-pole cellular bandpass filter using 75-mm-diameter high-temperature superconducting thin films, IEEE Microwave and Guided Wave Letters, Vol. 5, No. 11 (Nov. 1995). G. -C. Liang, D. Zhang, C. F. Shih, R. S. Withers, M. E. Johansson, W. Ruby, B. F. Cole, M. Krivoruchko and D. E. Oates, High power HTS microstrip filters for wireless communication, in 1994 IEEE bit. Microwave Symp. Digest, pp. 183-186, San Diego, California (May 1994). Reimnut K. Hotfmann, Handbook of Microwave Integrated Circuits, Artech House, Norwood, Massachusetts (1987). D. G. Swanson, Thin-film lumped-element microwave filters, in 1989 IEEE Int. Microwave Symp. Digest, pp. 671-674, Long Beach, California (June 1989). D. G. Swanson, Jr, R. Forse and B. J. L. Nilsson, A 10 GHz thin film lumped-element high temperature superconducting filter, in 1992 IEEE Int. Microwave Symp. Digest, pp. 1191-l 193, Albuquerque, New Mexico (June 1992). S. Ye and R. R. Mansour, Design of manifold-coupled multiplexers using superconducting lumped-element filters, in 1994 IEEE Znt. Microwave Symp. Digest, pp. 191-194, San Diego, California (May 1994). G. -C. Liang, D. Zhang, C. F. Shih, M. E. Johansson, R. S. Withers, D. E. Oates and A. C. Anderson, High power HTS microstrip filters for cellular base-station application, 1994 Applied Superconductivity Conference, Boston, Massachusetts (Oct. 1994). G. -C. Liang, D. Zhang, C. F. Shih, M. E. Johansson, R. S. Withers, A. C. Anderson, D. E. Oates, P Polakos, P Mankiewich, E. de Obaldia and R. E. Miller, High-temperature superconducting microstrip filters with high power-handling capability, in I995 IEEE Int. Microwave Symp. Digest, pp. 191-194, Orlando, Florida (May 1995). G. -C. Liang, D. Zhang, C. F. Shih, M. E. Johansson, R. S. Withers, A. C. Anderson, D. E. Oates, P Polakos, P. Mankiewich, E. de Obaldia and R. E. Miller, High-temperature superconducting microstrip filters with high power-handling capability. IEEE Trans. Microwave Theory Tech. Vol. 43, No. 12 (Dec. 1995).

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Microstrip filters for wireless communications using high-temperature superconducting thin films