IEEE TRANSACTIONS ON PLASMA SCIENCE, VOL. 42, NO. 10, OCTOBER 2014 3023

Low-Ripple and High-Precision High-VoltageDC Power Supply for Pulsed Power Applications

Suk-Ho Ahn, Hong-Je Ryoo, Ji-Woong Gong, and Sung-Roc Jang

Abstract— This paper describes the design and implemen-tation of a three-phase resonant converter with low rippleand high control accuracy. Based on a three-phase LCC-typeresonant converter—which has advantages of low ripple, high-efficiency, and high-power density compared with a single-phaseconverter—a high-voltage power supply with low ripple (<0.1%)was designed. In addition to the general merits of an LCC-typeresonant converter operating at continuous conduction mode—including soft switching, low conduction loss, and current sourcecharacteristics—the proposed scheme uses only one phase undera light-load condition by having different leg designs of the gatedrive circuit and snubber parameters. This allows the design toovercome the operational constraints of the general LCC-typeresonant converter. The distinctive design of the three-phaseconverter structure provides high efficiency and low ripple notonly during rated operation, but also under light-load conditions.In order to analyze the high performance of the proposedscheme from no load to rated load, a PSPICE simulation wascarried out. Comparison results with a conventional LCC-typeresonant converter based on a single-phase structure are analyzedfrom the viewpoints of output ripple, losses, and operable loadrange. Using the proposed converter, a 20-kV, 20-kW high-voltagedc power supply design and implementation was presentedwith a superior gate drive circuit. Finally, the superiority ofthe proposed converter was verified through a simulation andexperimental results. It was experimentally confirmed that thedeveloped power supply achieves high performance in terms ofefficiency (98%), operable load range (0.5–20 kV), and low ripple(0.05%), with a high power density.

Index Terms— High-voltage power supply, LCC resonant con-verter, three-phase resonant converter.

I. INTRODUCTION

IN THE field of pulsed power applications, such aslinear accelerators, the importance of a low output

voltage ripple and a high-precision, high-voltage dc powersupply as the primary source has been emphasized becausethe output voltage ripple affects system performance [1]–[20].For example, research on a capacitor charger with a lowvoltage ripple has been carried out to improve the stability ofklystron-modulators [16]. In another example, the solid-statemodulator for the electron-positron compact linear colliderhas been studied to enhance system efficiency by improving

Manuscript received October 31, 2013; revised January 20, 2014 andApril 27, 2014; accepted June 12, 2014. Date of publication August 26, 2014;date of current version October 21, 2014.

S.-H. Ahn and J.-W. Gong are with the Energy Conversion Technology,University of Science and Technology, Changwon 641-120, Korea (e-mail:whiteyan@keri.re.kr; whiteyan@ust.ac.kr).

H.-J. Ryoo and S.-R. Jang are with the Electric Propulsion Research Center,Korea Electrotechnology Research Institute, Changwon 641-120, Korea(e-mail: hjryoo@keri.re.kr; scion10@keri.re.kr).

Color versions of one or more of the figures in this paper are availableonline at http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPS.2014.2333813

its flat-top stability and pulse reproducibility [17]. For thatapplication, the low output voltage ripple of the high-voltagedc source is considered to be an important factor for designingthe modulator system.

To control the high-precision output of a high-voltagecapacitor charger, the following method has been suggested:a subcharger is used in parallel, operating with relatively lowrated output but with high switching frequency, to be in chargeof voltage control in the maximum charging voltage alongwith the main charger, for the purpose of rapidly charginga capacitor with a large capacity to a lower level (between5% and 10% of the maximum charging voltage) [22]. In astudy to develop a power supply with low-pulse voltage droopand high-voltage pulse reproducibility, the vernier and bouncermethods were suggested. For these applications, low rippleand high precision control are necessary in the high-voltagedc power supply [23]–[30]. To lower a ripple, a method thatincreases the switching frequency or output filter capacityvalue can be used. However, as the capacity of the powersupply increases, it limits the increase in switching frequencyowing to a switching loss that is generated by semiconductordevices. On the other hand, raising the output filter capacitorallows the output voltage ripple to be minimized, but a loadis still vulnerable to an electric arc that can be generatedfrequently in the application field when using an RF tube,such as a magnetron, gyrotron, or radar. In other words, thepower supply at the period of arc generation is tripped witha failure condition, but all of the energy saved in the outputfilter capacitor is transmitted to the load side, and the loadcan get damaged [31]–[33]. To troubleshoot this problem, amethod is now being used that compulsively blocks arc energyby connecting the switch with a fast switching operationcharacteristic between the power device output and load. Thismethod, however, is rather inefficient with regard to systemvolume and cost [5], [34]–[36].

This paper describes a multiphase resonance-type converterthat satisfies the requirements of the power supply, includingoutput voltage ripple, control precision, and low arc energy.

This resonance-type converter can be designed to reduceripples through multiphase operation and zero voltage (ZV)turn-ON. This converter can also have a reduced turn-OFF loss,which can increase the switching frequency. To control lowvoltage and power, a switching leg can be designed. By usingmultiphase switching legs that are independently driven, theswitching frequency increases, and losses occur only at asingle leg. This results in operation with high efficiency inthe full-load control area. Furthermore, the disadvantages ofa resonance-type converter—operating with a relatively low

0093-3813 © 2014 IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission.See http://www.ieee.org/publications_standards/publications/rights/index.html for more information.

3024 IEEE TRANSACTIONS ON PLASMA SCIENCE, VOL. 42, NO. 10, OCTOBER 2014

switching frequency in the operational area, with a light load ormore through multiphase operation—can be lessened and highcontrol precision and low ripple can be achieved in the full-load area, by using an independent high-frequency switchingoperation for a switching leg, separately designed at a lightloading area.

A detailed discussion of the analysis, design, and imple-mentation of the developed 20-kV, 20-kW high-voltage powersupply is presented. This discussion focuses on the powersupply’s efficiency, load range, low ripple, and reliability. Thesimulation and experimental results verify the superiority ofthe proposed converter.

II. ANALYSIS OF THE PROPOSED CONVERTER

A. Single-Phase LCC-Type Resonant Converter

The LCC-type resonant converter has a number of advan-tages, including a current source characteristic and intrinsicvoltage boost-up function. This type of converter is widelyused for designing high-voltage power supplies. Let uscompare a converter in the form of LCC (that has a parallelresonant capacitor with relatively small capacitance) to a seriesresonant capacitor at the secondary side of a series resonantconverter (SRC) operating in continuous conduction mode(operating above resonance). The LCC-type converter hasthe same basic characteristics as the SRC, and can reduceconduction losses by lowering the crest factor. This is done byimproving the resonant current waveform under the influenceof a parallel resonant capacitor. This can reduce the switchingloss by effectively increasing the capacitance of a losslesssnubber capacitor that is used to reduce switching losses.

Using the following analysis of the operation mode, thecharacteristics of the single-phase LCC-type resonant con-verter are examined. Under the steady state, the designedsingle-phase LCC-type resonant converter can be divided intoeight modes within each operating cycle, as shown in Fig. 1.

In mode 1, switch S1 is turned ON, and the input currentflows through the resonant tank. Resonance current flowswhile charging the parallel resonant capacitor. The resonantcurrent rises rapidly as it is influenced by the parallel resonantcapacitor, which has relatively small capacitance values. Theincrease in energy stored in the resonant inductor owing tothe resonant current rising rapidly in mode 1, compared withthe SRC, will lead to an increase in the free-wheeling energythat sustains mode 3 and mode 4. Mode 2 begins as thecharging of the parallel resonant capacitor is completed, andthe resonant current flows through the load.

Switch S1 is turned OFF, and mode 3 begins. In this mode,the lossless snubber capacitors are charged and dischargedto reduce turn-OFF losses. The lossless snubber capacitorsare connected in parallel. Switch S1 is charged by thefreewheeling energy, and the lossless snubber capacitor con-nected in parallel to switch S2 is discharged.

In mode 4, the charging and discharging of the losslesssnubber capacitor are completed, and the resonant currentbegins to flow through the diode connected in parallel to theswitch. In this mode, the voltage across switch S2 becomesalmost zero. Switch S2 is turned ON under a ZV condition, and

Fig. 1. Operation mode diagrams of LCC-type resonant converter.

then a switching half-cycle begins. Modes 5–8 can be analyzedusing a similar procedure to the one mentioned previously. Theresonant inductor current and capacitor voltage have a polaritythat is opposite to that of mode 1–mode 4. The operationwaveform based upon each mode is shown in Fig. 2.

It is clear that the turn-OFF loss—which depends on thetime duration of mode 3 and mode 7—can be reduced byincreasing the value of the snubber capacitance. However, thesnubber capacitor energy stored before turning ON the switchshould be discharged in order to achieve ZV turn-ON. Thus,increasing the snubber capacitance is a constraint because theenergy stored in the resonant tank is also limited, especiallyunder light-load conditions.

By having ZV turn-ON and small turn-OFF losses, the single-phase LCC-type resonant converter is effectively applied toa high power supply using an IGBT with a small saturationvoltage compared with a MOSFET. The LCC-type convertercan achieve high-frequency power supply with high efficiency.However, the single-phase LCC-type resonant converter haslosses that occur as the switching frequency increases undera light-load condition. In addition, when increasing the

AHN et al.: LOW-RIPPLE AND HIGH-PRECISION HIGH-VOLTAGE DC POWER SUPPLY 3025

Fig. 2. Operation waveform of designed LCC-type resonant converter.

Fig. 3. Scheme of proposed asymmetric three-phase LCC resonant converter.

capacitance of the output filter capacitor to reduce the ripple,another constraint arises: it is difficult to protect the load fromthe arc that may frequently occur in high-voltage applications,such as a gyrotron, radar, magnetron, or linear accelerator.

B. Proposed Asymmetric Multiphase LCC-TypeResonant Converter

In order to resolve the constraints of the single-phaseLCC-type resonant converter described earlier, this paperproposes an asymmetric multiphase converter, as shown inFig. 3. This proposed converter uses a number of switchinglegs, including one particular leg for the light-load operationcondition.

To reduce the ripple of the output voltage without increasingthe components of the output filter, a method for drivingthe multiphase converter in parallel is typically used. It isclear that the single-phase LCC-type resonant converter canalso reduce the ripple of the output voltage by using a sym-metric switching leg in multiphase. However, as mentionedpreviously, this has the disadvantage of creating a switchingloss in many switching elements owing to the increase in the

switching frequency in each switching leg under a light-loadcondition. In addition, all switching legs operate in the full-load range, and light-load operation becomes difficult owingto the increase in the controllable minimum operating voltage.It is possible to reduce the value of the snubber capacitorto handle the range of load operation. However, this causesthe efficiency of the converters to decrease by increasing theturn-OFF loss of the semiconductor devices. The basic conceptof the asymmetric multiphase converter proposed in this paperis shown in Fig. 3. It becomes possible for one particularleg, which operates separately under a light-load operationin a multiphase switching leg, to operate separately byemploying smaller values of the snubber capacitor (C1, C4),unlike other switching legs. In other words, with the loadthat the multiphase converter can control efficiently, everyswitching leg, including one particular leg, operates with aphase difference to control the large output and to reducethe ripple. During light-load operation, when the simultaneousmovements of multiphase switching legs are not controllable,the output voltage can be controlled by the isolated operationof one particular, separately designed leg.

In general, all switching legs operate with a phase delayfor reducing the output ripple, and the switching frequencyincreases as the load decreases. In case of a less than specificlight-load condition, the behavior of the switching legs—withthe exception of the switching leg responsible for the lightload—stops. The switching leg responsible for the light loadtakes charge of the load, so the switching frequency of theswitching leg responsible for the light load decreases, whichadjusts the output. By contrast, as the load increases again,the switching frequency of the switching leg responsible forthe light load is reduced. And, if the load is more than thespecified light load, all of the switching legs work again, andthe entire switching frequency increases again and adjusts theoutput. As previously described, the proposed converter wasdesigned to independently operate one switching leg at a lightload. Therefore, if low voltage and power are controlled, thelosses as the switching frequency increases occur in only oneleg, enabling the converter to operate with high efficiency inthe full-load control area.

In addition, unlike the other switching legs, because thesnubber capacitor of a particular leg with a small value canbe charged and discharged via a small resonance current inlight-load operation, ZV switching is possible under the full-load condition. Further, because the control characteristic ofthe resonant converter increases the switching frequency as theload decreases, it is possible to maintain a low output voltageripple even when one particular leg operates separately duringlight-load operation.

As an example, assume that the output voltage has a rippleof 120 kHz when the three-phase converter operates with aswitching frequency of 20 kHz at rated current, and that oneparticular leg operates separately from 60 kHz and higher inlight-load operation. In this case, the output voltage wouldhave a ripple of 120 kHz and higher. Therefore, it is clearthat there will be no large increase in the output ripple.Even three phases can convert into a single phase with afixed output filter. Therefore, even if high-precision output

3026 IEEE TRANSACTIONS ON PLASMA SCIENCE, VOL. 42, NO. 10, OCTOBER 2014

Fig. 4. Operation mode diagrams of proposed gate drive circuit.

control is required—in devices, such as a gyrotron, klystron,magnetron, or radar power supply—high-precision control ispossible without increasing the size of the filter capacitor. Thiscan effectively protect the load because the energy of the filtercapacitor passed to the load side can be minimized in caseof an arc occurrence, while keeping the ripple of the outputvoltage small.

Without complex control, this paper proposed to changethe behavior of the converter only by changing the designof the capacitance of the lossless snubber capacitor of thelegs responsible for light load operation and the componentof gate drive circuit by proposing a gate drive circuit withflexible dead time. In the high-frequency switching of light-load operation, relatively small lossless snubber capacitorswere selected, compared with the capacitors of other switchinglegs, for soft switching. In addition, a method for configuringa more reliable system through hysteresis band control isproposed, and the features and advantages of the proposedpower supply are verified through various experiments.

C. Gate Drive Circuit

As mentioned earlier, the operation of the proposed con-verter can be implemented in the form of a blocking gatesignal from the controller in a particular area by detectingthe power supply’s operating load area. However, this paperproposes a superior gate driver circuit with flexible dead timein the simple structure, as shown in Fig. 4, and implements theoperation of the proposed converter by using the characteristicsof flexible dead time.

The proposed gate drive circuit has several advantages,including implementing flexible dead time by sensing the ZVcondition of the switch, and a transformer for gate isolationthat is easily designed by entering the operating switchingfrequency of half-duty.

By analyzing the following operation mode, the operationalprinciple of the proposed gate drive circuit is described.

At mode 1, a turn-ON signal (+Vpulse) is applied to thegate drive circuit. The voltage across the switch is not ZV.Capacitor C1 is gradually charged through series resistor R1and parallel resistor R2 and, as a result, the turn-ON signal

TABLE I

NUMERICAL SPECIFICATIONS OF 20-KV, 20-KW HIGH-VOLTAGE

DC POWER

of the gate drive circuit does not turn ON the switch.Mode 2 is the case where the turn-ON signal (+Vpulse) isinput in the gate drive circuit. The voltage becomes zero.Diode D2 is forward-biased, and capacitor C1 is chargedthrough resistance R5 and diode D2. Compared with resis-tance R1, resistance R5 is designed to have a very smallvalue so that MOSFET M1 is quickly turned ON under aZV condition. If the voltage of capacitor C1 is charged tomore than the threshold voltage of MOSFET M1, then M1 isturned ON and gate voltage is applied through resistance R6and diode D3 so that the switch is turned ON.

Mode 3 is a turn-OFF mode. If the turn-OFF signal(−Vpulse) is input in the gate drive circuit and MOSFET M2is turned ON, the switch is turned OFF while quickly dis-charging the charged gate capacitor voltage. At this time,MOSFET M1 is turned OFF as well. Further, by beingrecharged to minus voltage through resistance R3 anddiode D1, the voltage of capacitor C1 can conduct theoperation of mode 1 in the next switching cycle.

As shown above, by having the operation wait until thevoltage across the switch of mode 1 becomes zero—thesituation where current flows in the diode connected in parallelat both ends of the switch of mode 4 of the converter operatingmode (that is, the operation of mode 2 enables the turn-ON ofthe switch in a condition that switch both ends becomes zerovoltage)—flexible dead time functionality is achieved. Becauseof these modifications—making the switching leg responsiblefor the light load, having a lossless snubber capacitor, andsetting the predetermined value of R1 to determine maximumdead time in the gate drive circuit differently from otherswitching legs—the operation of the proposed converter isimplemented by letting only the switching leg responsible fora light load do the switching operation under the light-loadcondition.

III. DESIGN AND SIMULATION OF THE PROPOSED

CONVERTER AND GATE DRIVE CIRCUIT

The proposed converter in this paper is implemented asa power supply of a gyrotron, radar, magnetron, or linearaccelerator. Table I lists the specifications for these

AHN et al.: LOW-RIPPLE AND HIGH-PRECISION HIGH-VOLTAGE DC POWER SUPPLY 3027

Fig. 5. Scheme of the proposed asymmetric three-phase LCC resonant converter.

TABLE II

SUMMARY OF DESIGN PARAMETERS

applications, and Table II lists the design parameters accord-ing to the specifications in Table I. The entire schemeof the 20-kV, 20-kW dc power supply, with the proposedconverter and gate drive circuit implemented, is shownin Fig. 5.

A. Structure of 20-kV, 20-kW Asymmetric Three-PhaseLCC-Type Resonant Converter

The proposed converter is designed with a three-phasestructure that takes into account the ripple rate and outputvoltage and power. As mentioned earlier, the LCC-typeresonant converter is composed of a series of resonant

inductor (Lr) and a capacitor (Cr) with a parallelcapacitor (Cp).

In the designed power supply, the additional parallel capaci-tor is not used, as shown in Fig. 5. Capacitors (Cp1-n–Cp4-m)that are connected in parallel with a series of stacked high-voltage diodes (Dhv1-n–Dhv3-m) are designed not only forvoltage balancing, but also to play the role of a parallelcapacitor. When the high-voltage diode stack commutates froma reverse to a forward bias, the transformer secondary windingcurrent flows in order to discharge the voltage across theparallel-connected capacitor. At this transition interval, theseries combination of (Cp1-n–Cp4-m) affects the resonantfrequency in the same way as a parallel capacitor (Cp).

Because of the voltage-boosting property of a parallelcapacitor, the LCC-type resonant converter facilitates thedesign of a high-voltage transformer by reducing the turn ratio.In addition, the converter helps prevent distortion, owing tothe effect of parasitic capacitance. As shown in Fig. 5, threetransformers are connected with the voltage-doubled rectifiers,which consist of a series of stacked diodes (Dhv1-n–Dhv3-m),voltage-balancing capacitors (Cp1-1–Cp4-6) that are also oper-ated as the parallel capacitor (Cp), and filter capacitors(Cf1–Cf6). In addition, a high-voltage sensing circuit isdesigned that uses resistors (Rs1–Rsm) and a capacitor(Cs1–Csm) divider.

B. Control of 20-kV, 20-kW Asymmetric Three-PhaseLCC-Type Resonant Converter

As shown in Fig. 5, the designed 20-kV, 20-kW dc powersupply has two PI controllers that have separate PI gains—onefor the output voltage and the other for the output current.The controllers are connected through diodes. This designis accompanied by the limitation of a single output, withone output controlling the other. Depending on the value ofthe control signal, the frequency of the switching signals is

3028 IEEE TRANSACTIONS ON PLASMA SCIENCE, VOL. 42, NO. 10, OCTOBER 2014

Fig. 6. Simulation model diagram of proposed 20-kV, 20-kW dc power supply using asymmetric three-phase resonant converter.

Fig. 7. Simulation waveforms at rated operation (20 kV, 20 kW) with 20-kωresistor [switching frequency: 20 kHz, ripple (p-p): 10 V (0.05%)].

modulated with a 50% duty cycle. After the signal is phaseshifted, it is transferred through the AND gate, and it is appliedto the gate transformer (Tr-gate1–Tr-gate3) for switching. Theprotection signal generated from the protections—includingthe over-current, over-voltage, and over-temperature—is alsotransferred through the AND gate to the stop gating signal foremergency stop.

For higher reliability, the switching frequency hysteresisband control can be added. As the switching frequency, thehysteresis band control block can choose from the single-phase operation and the three-phase operation. A detailedexplanation is presented in the experimental results part ofthis paper.

C. Simulation of 20-kV, 20-kW Asymmetric Three-PhaseLCC-Type Resonant Converter

In order to verify the design parameters and to evaluatethe characteristics, a PSPICE simulation of the converterwas performed. The PSPICE simulation model of the 20-kV,

Fig. 8. Simulation waveforms at light load condition with 20-kω resistor[switching frequency: 120 kHz, ripple (p-p): 0.5 V (0.07%)].

20-kW dc power supply, with the converter and gate drivecircuit implemented, is shown in Fig. 6. Simulation resultsof the converter performance with switching frequencies of20 kHz and 120 kHz, under the condition of a rated loadof 20 k�—to validate the performance of the 20-kV, 20-kWdc power supply with the converter and gate drive circuitimplemented—are shown in Figs. 7 and 8, respectively. Theoutput voltage in Fig. 7 is 20 kV with all three phasesworking. The output voltage in Fig. 8 is 700 V with onlyone switching working, in one leg that is responsible forthe light load. Fig. 9 shows the simulation results of theswitching loss per switching frequency. A smaller losslesssnubber capacitor was used for soft switching in a leg thatis responsible for light loads compared with other legs (theother legs only work with loads that are greater than lightloads, in consideration of the loss efficiency). The resultconfirmed that the switching loss in the light-load leg wasrelatively large. However, only the light-load leg was operatedin the frequency range of ∼80–100 kHz. The losses in other

AHN et al.: LOW-RIPPLE AND HIGH-PRECISION HIGH-VOLTAGE DC POWER SUPPLY 3029

Fig. 9. Comparison of switching loss.

TABLE III

SUMMARY OF SIMULATION RESULTS

legs were confirmed to be close to 0. Hard switching wasobserved in the light-load leg in a weak frequency above140 kHz with an output voltage of 500 V. Therefore, theswitching frequency range of the converter is found to bebetween 20 and 130 kHz, with output voltages from 500 Vup to 20 kV, while the converter performed normally. Addi-tional simulation of a single-phase LCC-type resonant con-verter was conducted to validate the design and superiority ofthe proposed converter of this paper. Table III shows the designparameter and results. The single-phase LCC-type resonantconverter designed to be compared was also designed to haveidentical output voltage and power and high efficiency asthe converter of this paper. Simulation results—per outputvoltage according to the switching frequency changes in boththe single-phase LCC-type resonant converter and the three-phase LCC-type resonant converter of this paper under thecondition of a rated load of 20 k�—are shown in Fig. 10.In the single-phase structure, hard switching was observedwith a 70-kHz switching frequency and <6-kV output voltage.The output was limited to <6 kV. On the other hand, theconverter developed by this paper was able to work over awide range of output voltage, from 500 V to a maximumof 20 kV.

IV. EXPERIMENTAL RESULTS

Fig. 11 shows the resonant current waveform and outputvoltage in rated output at a rated load of 20 k�, and underthe condition of a maximum output voltage of 20 kV. Theresonant peak current was measured as 65 A, and the switching

Fig. 10. Controllable output voltage simulation results with 20-k� resistor.

Fig. 11. Resonant current waveform and output voltage at rated operationwith 20-k� resistor (time scale: 10 μs/div; resonant current, 50 A/div; outputvoltage, 10 kV/div).

Fig. 12. Inverter output voltage waveform at rated operation with 20-k�resistor (time scale: 10 μs/div and inverter output voltage, 100 V/div).

frequency was 20 kHz. Fig. 12 shows each inverter outputvoltage waveform of a leg, for light load and for other legs.The inverter output voltage slop of a leg for a light load is quitedifferent from the slop of other legs because of the smallerlossless snubber capacitor used for the light-load condition.It was confirmed that there is no hard-switching condition witheach slop.

Figs. 13 and 14 show the output voltages under the con-ditions of maximum switching frequency under a 20-k�-rated-load condition and no-load condition. The minimum

3030 IEEE TRANSACTIONS ON PLASMA SCIENCE, VOL. 42, NO. 10, OCTOBER 2014

Fig. 13. Minimum controllable output voltage measurement test result atrated load of 20 k� (time scale: 20 μs/div; resonant current, 20 A/div; outputvoltage, 1 kV/div).

Fig. 14. Minimum controllable output voltage measurement test result underno-load condition (time scale: 10 μs/div; resonant current, 5 A/div; outputvoltage, 100 V/div).

output voltage was measured as 500 V under a 20-k�-rated-load condition, and the minimum controllable output voltagewas measured as 700 V under a no-load condition. Thiswas found to have satisfied the condition of less than thedesign specification (the minimum output voltage of 1 kV,as explained earlier). For the control patent experiment ofthe converter implemented, the output voltage and switchingfrequency when the proposed converter operates (single oper-ation of the single-phase leg in the light-load area, operationof all three-phase legs when operated at a load more than thelight load) and when a single-phase leg operates independently,were compared under conditions of a 20-k� rated load andno load. The results are shown in Figs. 15 and 16. Theswitching frequency where the converter changes from three-phase operation to single-phase operation under a 20-k�-rated-load condition (using the component of the gate drivecircuit and the converter implemented) is ∼90 kHz. It wasfound that operation changes at ∼80 kHz under a no-loadcondition. That is, the point where the implemented converterchanges from three phase to single phase, and single phase tothree phase, occurs at a switching frequency of 80–90 kHz,depending on the load.

Under a nonlinear load condition showing severe loadfluctuations, the proposed converter can repeat the change

Fig. 15. Control characteristic test results at rated load of 20 k�.

Fig. 16. Control characteristic test results under no-load condition.

Fig. 17. Hysteresis band control characteristic test results at rated load of20 k�.

of three-phase operation and single-phase operation at∼80–90 kHz. For higher reliability, a switching frequency hys-teresis band control can be selected. In this paper, we carriedout the experiment by changing the switching frequency fromthree-phase operation to single-phase operation at 60 kHz, andchanging the switching frequency from single-phase opera-tion to three-phase operation at 40 kHz. The experimentalresults of the switching frequency hysteresis band and controlcharacteristic, under a 20-k�-rated-load condition and no-loadcondition, are as shown in Figs. 17 and 18. In the area thathas a load more than a light load, all switching legs operate.As the load decreases, the switching frequency increases.As shown in Fig. 17, under a 20-k�-rated-load condition,

AHN et al.: LOW-RIPPLE AND HIGH-PRECISION HIGH-VOLTAGE DC POWER SUPPLY 3031

Fig. 18. Hysteresis band control characteristic test results under no-loadcondition.

Fig. 19. Hysteresis band control characteristic test results under no-load.

a light load—that is, <5-kV condition (with a switching fre-quency of 60 kHz or higher)—the operation of the remainingswitching legs, except for the switching leg responsible for thelight load, is stopped. Because only the switching leg respon-sible for the light load takes charge of the load, the switchingfrequency of the switching leg responsible for the light loadis reduced again, to ∼50 kHz, which adjusts the output.By contrast, under a condition of a load more than a lightload, where output increases—that is, under a condition ofmore than 7 kV (with a switching frequency of 40 kHzor less)—all switching legs operate again, and all switchingfrequencies increase again, up to ∼50 kHz, which adjusts theoutput. As shown in Fig. 18, the conversion of all switchingleg actions and the three-phase single leg action occur atoutput voltages of ∼16 and 13 kV, respectively, under a no-load condition. Fig. 19 shows the result of measuring theoutput voltage by switching the frequency after applying thehysteresis band control under a 20-k�-rated-load condition.The output range is wide, from a minimum output voltage of500 V to a maximum output voltage of 20 kV, as the switchingfrequency varies from 20 to 128 kHz. Fig. 20 shows the resultsof measuring the efficiency while adjusting the output. A highefficiency of up to 98% is achieved, with 90% efficiency from∼50% load of ∼9 kW.

V. CONCLUSION

This paper presented an asymmetric multiphase reso-nant converter applicable to a high-voltage power supply.The proposed converted demonstrated high-performance withrespect to output voltage ripple and control precision focusing

Fig. 20. Efficiency measurement depending on the load.

on applications, such as the gyrotron, radar, magnetron,and linear accelerator. The proposed scheme overcomes theconstraints of the single-phase LCC-type resonant converterby decreasing the output ripple, having a wide operableload range, and having high efficiency for the overall loadrange. The operational principle of the proposed converter andgate drive circuit was explained. This explanation includedasymmetric switching leg operation, flexible dead time con-trol, and hysteresis band control. The superiority of theasymmetric multiphase resonant converter was verified, com-pared with the single-phase converter. Based on the analysisof the proposed converter, the design and implementation of a20-kV, 20-kW high-voltage power supply using an asymmet-ric three-phase LCC-type resonant converter were presented.Finally, the developed high-voltage power supply was testedwith a resistor load. The results showed a maximum effi-ciency of 98% and a controllable load range from 500 V to20 kV with 0.05% output ripple. In addition to the switchingfreuency modulation, the hysteresis band control method wassuggested to improve reliability, depending on the frequencycharacteristic of the LCC-type resonant converter. This papershowed the application of the three-phase resonant converterwith regard to power supplies. One advantage of this proposedconverter is that it can be structured into a multiphase typein order to increase the performance and output power withregard to the ripple. This can be done without increasing thefilter capacity, which is related to the arc energy.

REFERENCES

[1] K. Kajiwara, Y. Oda, A. Kasugai, K. Takahashi, and K. Sakamoto,“Progress of the development of gyrotron and gyrotron system forITER,” in Proc. IEEE 13th Int. Vac. Electron. Conf. (IVEC), Apr. 2012,pp. 107–108.

[2] S. Illy, S. Kern, I. Pagonakis, M. Thumm, and A. Vaccaro, “Impact ofgyrotron power modulation on the collector of the 2 MW, 170 GHZgyrotron for ITER,” in Proc. IEEE Int. Conf. Plasma Sci. (ICOPS),Jul. 2012, p. 4B-2.

[3] K. Kajiwara, K. Hayashi, Y. Oda, K. Takahashi, A. Kasugai, andK. Sakamoto, “5 kHz modulation of 170 GHz gyrotron with anode-cathode short-circuited switch,” in Proc. IEEE/NPSS 24th Symp. FusionEng. (SOFE), Jun. 2011, pp. 1–5.

[4] T. Bonicelli et al., “The high voltage power supply system for theEuropean 2 MW electron cyclotron test facility,” in Proc. 13th Eur.Conf. Power Electron. Appl. (EPE), Sep. 2009, pp. 1–10.

[5] J. F. Tooker, P. Huynh, and R. W. Street, “Solid-state high-voltagecrowbar utilizing series-connected thyristors,” in Proc. IEEE PulsedPower Conf. (PPC), Jun./Jul. 2009, pp. 1439–1443.

3032 IEEE TRANSACTIONS ON PLASMA SCIENCE, VOL. 42, NO. 10, OCTOBER 2014

[6] K. D. Papastergiou and D. Aguglia, “Power system design compromisesfor large-scale linear particle accelerators,” in Proc. 15th Eur. Conf.Power Electron. Appl. (EPE), Sep. 2013, pp. 1–9.

[7] M. Jaritz and J. Biela, “Optimal design of a modular 11 kW seriesparallel resonant converter for a solid state 115-kV long pulse mod-ulator,” in Proc. 19th IEEE Pulsed Power Conf. (PPC), Jun. 2013,pp. 1–6.

[8] M. N. Nguyen, C. P. Burkhart, B. K. Lam, and B. Morris, “Recentupgrade of the klystron modulator at SLAC,” in Proc. IEEE PulsedPower Conf. (PPC), Jun. 2011, pp. 1386–1391.

[9] M. Tsuneoka, “Development of high efficiency and crowbar switchlessdc power supply for 1MW CW CPD gyrotron,” Int. J. Electron., vol. 86,no. 2, pp. 233–243, 1999.

[10] F. C. Schwarz and J. B. Klaassens, “A radar power supply with-out a voltage droop,” IEEE Trans. Ind. Electron., vol. IE-31, no. 4,pp. 377–384, Nov. 1984.

[11] R. A. Gardenghi and R. C. Houlne, “Power supply considerationsfor pulsed solid-state radar,” in Proc. IEEE Conf. Rec. 19th PowerModulator Symp., Jun. 1990, pp. 146–152.

[12] V. Yaskiv, O. Shabliy, A. Alpatov, and O. Gurnik, “Development ofswitch power supplies for radar applications,” in Proc. Int. Conf. Radar(CIE), 2001, pp. 851–855.

[13] E. E. Bowles, S. Chapelle, G. X. Ferguson, D. S. Furuno, andM. Marietta, “A high power density, high voltage power supply fora pulsed radar system,” in Proc. 21st Int. Power Modulator Symp.,Jun. 1994, pp. 170–173.

[14] M. J. Bland, J. C. Clare, P. Wheeler, and R. Richardson, “A 25 kV,250 kW multiphase resonant power converter for long pulse applica-tions,” in Proc. 16th IEEE Int. Pulsed Power Conf., vol. 2. Jun. 2007,pp. 1627–1630.

[15] N. C. Jaitly et al., “MV long pulse generator with low ripple andlow droop,” in 8th IEEE Int. Pulsed Power Conf. Dig. Tech. Papers,Jun. 1991, pp. 161–165.

[16] J. S. Oh, S. S. Park, Y. J. Han, I. S. Ko, and W. Namkung, “Designconsiderations for the stability improvement of klystron-modulatorfor PAL XFEL,” in Proc. Particle Accel. Conf. (PAC), May 2005,pp. 1165–1167.

[17] S. Blume and J. Biela, “Design procedure of an active bouncer for anultra precise long pulse solid state modulator,” in Proc. 19th IEEE PulsedPower Conf. (PPC), Jun. 2013, pp. 1–6.

[18] S.-R. Jang, H.-J. Ryoo, S.-H. Ahn, J. Kim, and G.-H. Rim, “Devel-opment and optimization of high-voltage power supply system forindustrial magnetron,” IEEE Trans. Ind. Electron., vol. 59, no. 3,pp. 1453–1461, Mar. 2012.

[19] A. Staprans, E. W. McCune, and J. A. Ruetz, “High-power linear-beamtubes,” Proc. IEEE, vol. 61, no. 3, pp. 299–330, Mar. 1973.

[20] B. D. Blackwell and K. A. Walshe, “Medium power pulsed power supplyexperience,” in Proc. Pulsed Power Symp., 2005.

[21] W. J. Mulligan, S. C. Chen, G. Bekefi, B. G. Danly, andR. J. Temkin, “A high-voltage modulator for high-power RF sourceresearch,” IEEE Trans. Electron Devices, vol. 38, no. 4, pp. 817–821,Apr. 1991.

[22] J. H. Choi et al., “Overview of superconducting magnet power supplysystem for the KSTAR 1st plasma experiment,” Nucl. Eng. Technol.,vol. 40, no. 6, p. 459, 2008.

[23] F. C. Magallanes, D. Aguglia, P. Viarouge, C. de Almeida Martins, andJ. Cros, “A novel active bouncer system for klystron modulators withconstant AC power consumption,” in Proc. 19th IEEE Pulsed PowerConf. (PPC), Jun. 2013, pp. 1–6.

[24] F. C. Magallanes, D. Aguglia, P. Viarouge, C. de Almeida Martins, andJ. Cros, “A novel active bouncer system for klystron modulators withconstant AC power consumption,” in Proc. IEEE Int. Conf. Plasma Sci.(ICOPS), Jun. 2013, p. 1.

[25] S. Maestri, R. G. Retegui, P. Antoszczuk, M. Benedetti, D. Aguglia,and D. Nisbet, “Improved control strategy for active bouncers used inklystron modulators,” in Proc. 14th Eur. Conf. Power Electron. Appl.(EPE), Aug./Sep. 2011, pp. 1–7.

[26] D. Gerber and J. Biela, “Bouncer circuit for a 120 MW/370 kV solidstate modulator,” in Proc. IEEE Pulsed Power Conf. (PPC), Jun. 2011,pp. 1392–1397.

[27] D. Aguglia, “2 MW active bouncer converter design for long pulseklystron modulators,” in Proc. 14th Eur. Conf. Power Electron. Appl.(EPE), Aug./Sep. 2011, pp. 1–10.

[28] H.-J. Eckoldt et al., “Test of a bouncer modulator with pulse cable atflash,” in Proc. IET Eur. Pulsed Power Conf., Sep. 2009, pp. 1–4.

[29] T. Tang, C. Burkhart, and M. Nguyen, “A vernier regulator for ILCMarx droop compensation,” in Proc. IEEE Pulsed Power Conf. (PPC),Jun./Jul. 2009, pp. 1402–1405.

[30] K. Macken et al., “Towards a PEBB-based design approach for a Marx-topology ILC klystron modulator,” in Proc. Particle Accel. Conf., 2009.

[31] S. G. E. Pronko et al., “The performance of the 8.4 MW mod-ulator/regulator power systems for the electron cyclotron heatingfacility upgrade at DIII-D,” Fusion Technol.-Illinois, vol. 39, no. 2,pp. 1111–1115, 2001.

[32] J. Lohr et al. “Practical experiences with the 6 gyrotron system onthe DIII-D tokamak,” in Proc. 20th IEEE/NPSS Symp. Fusion Eng.,Oct. 2003, pp. 228–231.

[33] D. Fasel et al., “Design study of a test stand for ITER gyrotron,” FusionEng. Des., vols. 66–68, pp. 555–560, Sep. 2003.

[34] H. Do et al., “Test result of 5 GHz, 500 kW CW prototype klystron forKSTAR LHCD system,” Fusion Eng. Des., vol. 86, no. 6, pp. 992–995,2011.

[35] I. H. Song, H. S. Ahn, Y. K. Kim, H. S. Shin, C. H. Choi, andC. Moo-hyun, “Development of the 120 kV/70 a high voltage switchingsystem with MOSFETs operated by simple gate control unit,” in Proc.IEEE 33rd Annu. Power Electron. Specialists Conf. (PESC), vol. 3.Jun. 2002, pp. 1181–1185.

[36] S. G. E. Pronko and D. S. Baggest, “The 8.4 MW modulator/regulatorpower systems for the electron cyclotron heating facility upgrade atDIII-D,” in Proc. 18th Symp. Fusion Eng., 1999, pp. 194–197.

Suk-Ho Ahn (M’11) received the B.S. degree inelectrical engineering from Incheon National Univer-sity, Incheon, Korea, in 2009. He is currently pursu-ing the M.S. and Ph.D. degrees with the Universityof Science and Technology, Daejeon, Korea.

His current research interests include soft-switchedresonant converter applications and battery chargersystems.

Hong-Je Ryoo received the B.S., M.S., andPh.D. degrees in electrical engineering fromSungkyunkwan University, Seoul, Korea in 1991,1995, and 2001, respectively.

He was with the Wisconsin Electric Machinesand Power Electronics Consortium, University ofWisconsin-Madison, Madison, WI, USA, from 2004to 2005, as a Visiting Scholar for his post-doctoralstudy. Since 1996, he has been with the Korea Elec-trotechnology Research Institute, Changwon, Korea,where he is currently a Principal Research Engineer

with the Electric Propulsion Research Division and a leader of the PulsedPower World Class Laboratory. He has also been a Professor with theDepartment of Energy Conversion Technology, University of Science andTechnology, Deajeon, Korea, since 2005. His current research interests includepulsed-power systems and their applications, and high-power and high-voltageconversions.

Dr. Ryoo is a member of the Korean Institute of Power Electronics and theKorean Institute of Electrical Engineers.

AHN et al.: LOW-RIPPLE AND HIGH-PRECISION HIGH-VOLTAGE DC POWER SUPPLY 3033

Ji-woong Gong received the B.S. degree in electri-cal engineering from Chonnam National University,Gwangju, Korea, in 2012. He is currently pursuingthe M.S. degree with the University of Science andTechnology, Daejeon, Korea.

His current research interests include soft-switchedresonant converter applications and high-voltagepulsed-power supply systems.

Sung-Roc Jang was born in Daegu, Korea, in1983. He received the B.S. degree from KyungpookNational University, Daegu, in 2008, and the M.S.and Ph.D. degrees in electronic engineering fromthe University of Science and Technology, Deajeon,Korea, in 2011.

He has been a Senior Researcher with theKorea Electrotechnology Research Institute, Elec-tric Propulsion Research Center, Changwon, Korea,since 2011. His current research interests includehigh-voltage resonant converters and solid-state

pulsed-power modulators, and their industrial applications.Dr. Jang was a recipient of the Young Scientist Award at the 3rd Euro-Asian

Pulsed Power Conference in 2010, and the IEEE Nuclear Plasma ScienceSociety’s Best Student Paper Award at the IEEE International Pulsed PowerConference in 2011.