Laboratory Test 8 - - TU Kaiserslautern · The laboratory test deals with on-state and switching behaviour and with the on-state and ... Triac) 3. fully controlled semiconductors,

Laboratory Tests for Energy and Automation Technique, Laboratory Test 8 1

Laboratory Test

8

Analysis of Power Semiconductors 1 Introduction

2 Description of different power semiconductors 2.1 Bipolar transistor 2.2 Gate turn-off thyristor (GTO) 2.3 Metal-oxide-semiconductor field-effect transistor (MOSFET) 2.4 Insulated-gate bipolar transistor (IGBT) 3 Calculation of losses 4 Doing the laboratory test 5 Laboratory test setup 6 Starting the laboratory test 7 Evaluation of the laboratory test 8 Questions for preparing the laboratory test 9 Literature Appendix: Data sheets of power semiconductors


1 Introduction The laboratory test deals with on-state and switching behaviour and with the on-state and switching losses of power semiconductors. In common, power semiconductors are defined as semiconductors designed for high off-state voltage (≥ 100 V) and for high current (≥ 10 A). These semiconductors are produced since the mass production of monosilicon started (about 1960). The semiconductors used in power electronics can be divided in the three types: 1. semiconductors without control (diode, Diac) 2. controlled semiconductors, controlling the switching on but not the switching off

(thyristor, Triac) 3. fully controlled semiconductors, controlling the switching on and off (bipolar transistor,

GTO, MOSFET, IGBT) In the laboratory experiment the fully controlled power semiconductors are tested. These are, as defined above, bipolar transistor, gate turn-off thyristor, metal-oxide-semiconductor field-effect transistor and insulated-gate bipolar transistor. They are used in all application ranges of the modern power electronics (rectifiers, frequency converters, switching-mode power supply, servo drives, industrial drives, traction). In this laboratory test the basic circuit for power semiconductor test is the step-down converter with ohmic-inductive load, as shown in figure 1.

Fig. 1 Step-down converter with ohmic-inductive load If the ohmic-inductive load is completed with a dc voltage source, the circuit corresponds to a dc motor in a stationary working point. The semiconductors can also be divided by the charge carriers which are used for the current flow. The MOSFET is a unipolar device using only one type of charge carriers for the current flow. In power electronics almost exclusively N-channel MOSFETS are used, so that the charge carrier types are electrons. The MOSFET is a voltage controlled device which needs only a small power for switching on. Because of the great mobility of the electrons and the resulting very short switching times switching frequencies up until 100 kHz and above are possible. A disadvantage of these devices is that a high current capability cannot be realized together with a high off-state voltage capability. So, the MOSFETs are used in power ranges less than 10 kVA.

Vd

RL

DF

LL

VSt


The other three devices are bipolar semiconductors. This means, that both charge carrier types, the majority carriers and the minority carriers, are used for the current flow. Based on the physical switching behaviour the switching frequencies are limited to about 1 kHz for the GTO, to about 5 kHz for the bipolar transistor and to about 20 kHz for the IGBT. But with special doping of the corresponding semiconductor layer these power semiconductor devices can be built for high power ranges (bipolar transistor, IGBT) and for very high power ranges (GTO). 2 Description of the different power semiconductors 2.1 Bipolar transistor The bipolar transistor is the oldest device in this laboratory test. It was developed 1948 in the Bell Laboratories by Shockley, Bardeen and Brattain. Its circuit symbol is shown in figure 2.

Fig. 2 Circuit symbol of a bipolar transistor Used as a switching device, in opposite to the use as a linear amplifying device, the transistor has only two characteristic working points in the output characteristic diagram. These are the “On” state and the “Off” state. It depends on the load on which working curve the “On” or the “Off” state is reached (Figure 3).

Fig. 3 Working curves for different loads The load current, which is the same as the collector current IC, is proportional to the control current. This control current is the same as the base current IB. The connection between these currents is the dc amplification factor B = IC/IB. Compared with small signal bipolar transistors the power transistors have a relatively small dc amplification factor. B is in the range of 20…100. Based on this there are very high control currents necessary for a high

E

B

C

I C

VCEVB

Off

On

I B


load current. To reduce the control currents mostly Darlington transistors are used in power electronics. The principle of the Darlington circuit is shown in figure 4.

Fig. 4: Principle of the Darlington circuit In figure 4 a transistor T1 with the dc amplification B1 is connected in common-collector circuit with the transistor T2. With this the resulting dc amplification factor B is [3]: B = B1 + B2 + (B1 • B2) (1) B ≈ B1 + B2 (2) For the laboratory test a Darlington transistor is used. Hereby, both transistor T1 and transistor T2 are on the same substrate. Such transistors are called monolithic integrated Darlington transistors. Besides, in most cases a free-wheeling diode is integrated on the substrate. In figure 5 the construction of a monolythic integrated Darlington transistor can be seen.

Fig. 5: Construction of a monolythic integrated Darlington transistor In figure 5 the base layer is not scaled. The connections KD1 and AD1 are an integrated speed up diode. This diode reduces together with the resistors R1 and R2 the switching time of the transistor. Besides this the resistors R1 and R2 care for the tail current of the transistor T1 (Ic1 in the off state) not being amplified by the transistor T2. This reduces also the total tail current [3]. The resistors result on one hand from doping the base layer p and the emitter region (R2) and on the other hand through the groove of separation between T1 and T2 not being totally drawn to the collector region. So, a small p-channel remains which is the resistor R1. Figure 6 shows the equivalent circuit of the construction in figure 5.

B

E

C

E

B

C

=̂

E1

B E

B1 SIO2 KD1 AD1 B2 E2

p+

p

n-

n+

C


Fig. 6: Equivalent circuit for figure 5 The bipolar transistor is a current controlled device. The base current IB controls the collector current IC. So, with the base current the transistor is set in on or in off state. Simple switching off leads to a long storage time of the charge carriers. This causes high losses during the switching off. Therefore, the power transistor is controlled by a negative base current during switching off. This current sucks the electrons out of the base region and thus reduces the switch off time and by this the losses. The different switching times are defined as: td: delay time tr: rise time ts: storage time tf: fall time In data sheets frequently ton and toff are given. These are defined as: ton = td + tr (turn on time) toff = ts + tf (turn off time) The curves of the collector current and the base current are shown in figure 7. Besides this the characteristic parts of the switching times of a bipolar transistor are shown.

B

R1

C

ER2

T1

T2

"D1"


Fig. 7: Collector current and base current for a bipolar transistor 2.2 Gate turn-off thyristor (GTO) The GTO is a thyristor using a gate-cathode with a fine distribution. The GTO can be switched off using a negative control current pulse. The GTO can be used for very high power (more than 3000 kVA). Figure 8 shows the circuit symbol of a GTO.

Fig. 8: Circuit symbol of an GTO The characteristic curve of the GTO is similar to the characteristic curve of a standard thyristor. The GTO differs from a standard thyristor only by a controlled switching off. In figure 9 the typical three regions are shown. These are the negative non-conducting zone, the positive non-conducting zone and the conduction region.

C

G

A

iC

90%

tr

td

ts

collector current

10%

tf

toff

ton

t

t

90%

10%

iB

-iB

base current


Fig. 9: Characteristic curve of a GTO (same as standard thyristor) The negative non-conducting zone is not used during the laboratory test. The device works on the positive non-conducting zone if a positive anode-cathode voltage UAK is used but the ignition (= switch on) has not been done. This is the so called off-state. If the GTO is ignited by a positive current pulse into the gate the working point on the conducting voltage-current characteristic is adjusted by the load. As shown in figure 10 gate and cathode have a fine distributed finger structure. By igniting the GTO in opposite to the standard thyristor the whole cathode area conducts at the same time. With this the GTOs have smaller switch-on times and smaller rates of rise of forward current compared with standard thyristors. The charge carriers can be sucked out of the cathode because of the gate-cathode structure also during switching off the GTO with the negative gate current.

Fig. 10: Construction of a GTO The gate and the cathode tapes have a typical width of 100 μm to 400 μm. In spite of the microstructure an inhomogeneous current distribution in the device cannot be avoided. This can result in a local thermical overload in the current conducting zones which can destroy the

VRRM

VT0

VT0 VDRM

vD

vT

VA

IA

A

GC

p

p

n-

n+

n+ n+ n+ n+ n+

p n+ p n+n+p p


device if there are high external voltages. Therefore, a limiting of the voltage increase as announced in the data sheet is absolutely necessary. The protection circuit is a RCD network and reduces the voltage increase du/dt during switching off the GTO. The constant load current flows for a short time through the RCD network and charges the capacitor Cs. The time constant of the RCD network is defined only by the capacitor, because the protecting resistor is bridged by the diode Ds. The discharge current of the capacitor is defined by the resistor Rs and it is superposed to the load current of the GTO. The resistor Rs must be dimensioned in a way that the inrush current will not be too much. In common, the RCD protecting circuit reduces the maximum switching frequency and the efficiency of the power stage. Figure 11 shows the RCD network for a GTO.

Fig. 11: RCD network for a GTO Switch on and switch off of a GTO can be explained by using an equivalent circuit based on the two-transistor model (Shockley). In this model the GTO (and similar the thyristor) is understood as a series-/parallel connection or a pnp- and a npn-transistor (figure 12).

Fig. 12: Equivalent circuit for a GTO For the ignition of a GTO only a relative low current pulse is necessary (typical IGT ≥ 400 mA), as in the positive non-conducting area only the junction J2 (transistor T2) is locked. For a GTO being conductive there only charge carriers must be injected in the area P2 (base current of T2). By the rising of the collector current IC2 and the base current of T1 the system reaches the conductive state by themself. Simultaneous the collector current of T1 is the base current of T2. If there is a negative control-current pulse connected with the gate (base

RS

L

CS

DS

G

A

C

T1

T1T2

N

P

N

P

A

C

G N

P

G

A

C

N

P

N

P

a) b) c)


current of T2) the collector current IC2 is reduced. Simultaneous this current is the base current of T1. This feedback process continues until the current IT through the GTO becomes zero. In the conductive state of the GTO the whole centre zone is flooded with charge carriers. So for sucking out the charge carriers a high negative current pulse is necessary. This current pulse is in the range of 20 % to 35 % of the thyristor current IT (= load current). Figure 13 shows the curve of the ignition current for switching on and switching off a GTO.

Fig. 13: Ignition current of a GTO The negative voltage VGK rises the blocking ability of the GTO. 2.3 Metal-oxide-semiconductor field-effect transistor (MOSFET) The MOSFET is the only unipolar device in this laboratory test. The field-effect transistor was described by Shockley in 1952. But it takes a long time after this to construct FET which could fullfil the demands of practical use. The circuit diagram of a MOSFET is shown in figure 14.

Fig. 14: Circuit diagram of a n-channel MOSFET In the power electronics exclusively self-locking MOSFETs and almost exclusively n-channel types are used. The self-locking n-channel MOSFETs are non-conducting without a positive gate-source voltage VGS. The device keeps in the off-state until the voltage VGS reaches the threshold voltage VGSth (BUZ 21: typ. VGSth = 3 V). With a rising control voltage VGS the current flow through the FET rises in dependance of its transfer characteristic (figure 15). MOSFETs are completely blocked for voltages below the threshold voltage. The drain residual-current has a maximum of 4 mA.

tw1

diGdt iGMiG

0,1 iGM

gate currentgate voltage

0,1·iGO

0,5·iGO

0,1·iGM

diGdt

iGO

tw3

tw2

VGO Vgr

VGR

D

S

G


Fig. 15: Transfer characteristic of a MOSFET (BUZ 21) With negative values of VGS the ability of blocking cannot be rised. So, with a unipolar control voltage the whole output characteristic ID = f(VDS) can be passed. Figure 16 shows the output characteristic of a MOSFET.

Fig. 16: Output characteristic of a MOSFET (BUZ 21) The construction of the MOSFET used for this laboratory test (BUZ 21, SIPMOS family) based on a drain metalization followed by a n+-substrate and a n--epitaxial layer. The

0 5 10VGS/V0

5

10

15

20

25

I D/A

0 2 12

VDS

0

10

20

40

I D

4 6 8 10 V

30

A

4,0 V4,5 V

5,0 V

5,5 V

6,0 V

6,5 V

7,0 V

7,5 V

VGS = 8,0 V10,0 V

20,0 V

PD =75 W


thickness and the doping of this n--layer is dependend on the off-state voltage. The n+-polysilicon gate laying above this n--layer is embedded in an isolating silicon dioxide (SiO2). The source metalization (aluminium) covers the whole structure and connects the single transistor cells in parallel. The construction is shown in figure 17.

Fig. 17: Construction of a MOSFET Besides the MOSFET itself the equivalent circuit shows among the MOSFET itself its parasitic elements (figure 18).

Fig. 18: Equivalent circuit for a MOSFET with its parasitic elements Because of the simplified model the capacitances in figure 18 cannot be measured in single. Under neglecting the channel resistor RK and the n--epitaxial resistor RD the following capacitances, which can be found in the datasheet, are defined:

D

S

GRG

RD

CGD

CGS

CDS

RK

inverteddiode

S

D

G isolation

n-channel Sn

p

n


Ciss ≈ CGS + CGD Coss ≈ CDS + CGD Crss ≈ CGD The capacitance CGD is also named Miller capacitance. The combination of the resistors RK and RD roughly equals the drain-source resistor RDSon in the on-state of the MOSFET: RDson ≈ RD + RK The diode in the equivalent circuit represents the pn junction from the p-source area to the n-

-epitaxial layer. This layer is the drain connection via n+-substrate. This results in an inversion diode between drain and source. The characteristic values of this diode can be found in the datasheet. MOSFETs can be controlled almost without power. Energy is only needed for switching on. It is needed primarily for the charging of the input impedance Ciss. The capacitance Ciss has to be charged very quickly by switching on the MOSFET. The necessary currents have to be delivered by the gate drive circuit. As the MOSFET is a transistor the dv/dt and the di/dt are not specified. The protection of the semiconductor is limited on the maximum values of VGsmax, VDcmax, IDmax and ϑJmax. The voltage VDS is limited by using a Z-diode (DZ1 in figure 19). Its value has to be between the voltage VDS delivered from the gate drive circuit and the maxium for VGS which is VGsmax = ±20 V. The gate is protected by the series connector RG. In connection with the input capacitance RG is also protecting the MOSFET from highfrequency oscillations.

Fig. 19: Protection circuits for a MOSFET If inductive loads are switched negative voltage (Uind) has been induced. In this case high voltages occur. For these high voltages the blocking ability of the semiconductor must be defined. The Z-diode DZ2 protects the MOSFET against these too high voltages VDS. It is connected between the gate and the drain connection. The value of this diode must lay between the highest voltage at the device (UD + Uind) and the maximum value of VDsmax (BUZ 21: typ. VDsmax = 100 V). If there is a voltage higher than the Z-diode voltage during the off-state of the MOSFET, the MOSFET switches on and the high voltage decreases. 2.4 Insulated-gate bipolar transistor The IGBT was developed at the beginning of the 1980s and it is available since the end of the 80s. It is a combination of a bipolar transistor and a MOSFET. This also gave the name to it. The lower output stage is realized by a bipolar transistor which is controlled by a

Vd

RL

DF

LL

VSt

RG

DZ1

DZ2


MOSFET. So the control characteristic of an IGBT is similar to the control characteristic of a MOSFET. The control electrode is named equivalent to this as gate. Figure 20 shows the circuit diagram of an IGBT.

Fig. 20: Circuit diagram of an IGBT The construction of an IGBT is similar to the construction of a MOSFET. In case of the MOSFET a n+-substrate and a n--substrate follow after the drain metallization. In opposite to this the IGBT has a pure p-substrate. For this the IGBT belongs to the group of the minority carriers. Here, by controlling the gate the charge carriers are injected from the highly doped emitter area into the meagerly doped collector area. With this the conductivity of the meagerly doped area rises up for several ten powers (conductivitcy modulation). The construction of an IGBT is shown in figure 21.

Fig. 21: Construction of an IGBT The self blocked n-channel MOSFET is the driver for the pnp bipolar transistor. For this the whole voltage loss VCE at the IGBT can be represented as the sum of the voltage losses of the bipolar transistors pn diode and the voltage loss of the MOSFET driver. So the on-state voltage loss of an IGBT can, in opposite to the voltage loss of a MOSFET, never be less than the threshold voltage of a diode. That can also be seen in the output diagram of an IGBT (figure 22).

G

C

E

n+ n+

p+n+

p+

n-

emitter

collector

gate


Fig. 22: Output diagram of an IGBT The IGBT is like a MOSFET a voltage controlled device which can be controlled almost without power. The controlling of the IGBT has the same features as the controlling of a MOSFET. The protection of the IGBT is limited on the values VGE and VCE. This protecting is similar to the protection of a MOSFET. Because VCemax of the IGBT lies in the range of the bipolar transistors a protection against high collector-emitter voltages is not necessary. 3 Calculation of the switching losses and the on-state losses The total losses of a power semiconductor switch are composed by several parts: switch-on losses Pon switch-off losses Poff on-state losses Pos reverse losses PR driver losses PD As an example the calculation of the losses of a bipolar transistor is shown. Hereby the driver losses PD are neglected. Figure 23 shows the definition of the switching times.

0 5 10VCE/V

0

50

100

I C/A

1 2 3 4 6 7 8 9

150

200

250

300

PC = 600W

6V

7V

8V

10V

12V15VVGE = 20VTC = 25V


Fig. 23: Definition of the switching times With respect to figure 23 the losses are:

∫ ∫ ∫∫ ⋅⋅+⋅⋅+⋅⋅+⋅⋅=1

0

3

2

4

3

2

1

t

t

t

t

t

tCCECRCE

t

tCCEC)sat(CE

*VAV dtiv

T1dtiv

T1dtiv

T1dtiV

T1P

In the equation above are: f = T-1: switching frequency t0 – t1: conducting phase t2 – t1: switch off time t3 – t2: non-conducting phase t4 – t3 : switch-on time vCE: collector-emitter voltage iC: collector current VCEsat: saturation voltage iCR: reverse current Due to the low reverse current the reverse losses PR can be neglected. They are:

∫ ⋅⋅=3

2

t

tCRCE

*R dtiv

T1P

The simplified calculation uses linear current curves and linear voltage curves during switch on and switch off as shown in figure 24. With this assumption the following equation for the calculation of the losses is used:

( ) ⎟⎠

⎞⎜⎝

⎛ ⋅⋅⋅+⎟⎠

⎞⎜⎝

⎛ ⋅⋅⋅+⋅⋅⋅=2

tvi

T1

2t

viT1tVi

T1P on

CECoff

CECflw)sat(CEC*VAV

vCE

VCE(sat) iCR

iC

iC

vCE

t1 t2 t3 t4 t


In the equation above are: iC: collector current VCEsat: average saturation voltage vCE: maximum collector voltage tflw = t1: conduction phase toff = t2 – t1: switch off time ton = t4 – t3: switch on time

Fig. 24: Simplified linear current curves and voltage curves for the bipolar transistor The driver losses PD can be estimated as about 10 % of the collector losses P*

AV. With this the total average losses are:

*VAVVAV P1.1P ⋅≈

In the laboratory test the times above announced can be read on an oscilloscope. The measured switch-on time ton can be compared with the theoretical value ton in the datasheet. The switch off-time corresponds essentially to the fall time tf, so that toff ≈ tf. The duration of the conductive phase is calculated to:

2t

2t

Tat offonflw −−⋅=

2tt

Tat fonflw

+−⋅≈

The factor a in the equations above is defined as duty cycle (figure 25).

VCE

iC

ton

td

ttflw toff

90%

10%

tflwtr ts tf t


Fig. 25: Definition of the duty cycle The switch-on and switch-off times are equally divided between both switching conditions. With the simplification shown above the on-state losses can be calculated with:

flwC)sat(CE*D tiV

T1P ⋅⋅⋅=

The switching losses result in:

6tt

ivT1PPP offon

CCE*off

*on

*S

+⋅⋅⋅=+=

For the MOSFET and the IGBT the driver losses can be neglected. So the losses are:

*VAVVAV PP ≈

To calculate the losses of the GTO the equation above has to be used. Likewise the driver losses can be neglected. For the theoretical comparison the on-state voltage of the GTO VF, the switch-on time ton and the switch-off time toff have to be considered. The fundamental frequency f = 1/T.

( ) ⎟⎠

⎞⎜⎝

⎛ ⋅⋅⋅+⎟⎠

⎞⎜⎝

⎛ ⋅⋅⋅+⋅⋅⋅=2

tVi

T1

2t

viT1tvi

T1P off

AKToff

AKTflwFT*VAV

In the equation above are: vF: on-state voltage iT: on-state current VAK: maximum anode-cathode voltage tflw: conduction time toff: switch off time ton: switch on time

iB

ton

t

toff

T

aT (1-a)T


4 Realization of the laboratory test The maximum switching frequencies of the power electronic devices can be found in the respective data sheets. These values may not be exceeded during the laboratory test. 4.1 Measurements with constant duty cycle and variable frequency The following quantities have to be measured: 1. frequency f 2. load current id using the voltage vshunt; here 1 V = 2 A 3. voltages at the power transistors bipolar transistor and IGBT: vCE MOSFET: vDS GTO: vAK 4. switching times For the bipolar transistor and the GTO the measurements have to be done in steps of 50 Hz. For the IGBT and the MOSFET steps of 500 Hz should be used. 4.2 Measurements with variable duty cycle and constant frequency The following quantities have to be measured: 1. duty cycle a 2. load current id using the voltage vshunt; here 1 V = 2 A 3. voltages at the power transistors bipolar transistor and IGBT: vCE MOSFET: vDS GTO: vAK 4. switching times The duty cycle should be changed in steps of 10 %. 5 Construction of the laboratory test The next page shows the breadboard of the laboratory test. All connections in this figure are already done. Only the measuring devices have to be connected. There is in the left part the general supply, the fuse protection and the supply for additonal device mounted. The bigger right part contains the parts transformer, load, power electronic devices and puls-width-modulation generator. Also, there are the push-button switches to switching on and off the electronics and power electronics of the laboratory test. The laboratory test board is switched on with the push-button switch “Elektronik EIN”. The adjusted frequency of the puls-width modulator is shown. The load circuit is set into operation with the push-button “Leistungsteil EIN”. This is displayed by the control lamps of the general supply. The voltage Vd is generated using a bridge rectifier and a capacitor bank. The ohmic-inductive load can be connected to the different power electronic devices by a stepping switch. The appropriate control lamp of the semiconductor device is lighted.


Fig. 26: Breadboard of the laboratory test (english version)

Fig. 27: Breadboard of the laboratory test (german version) Important: Before changing the semiconductor device the load circuit must be separated from the supply with the push-button “Leistungsteil AUS”. The new power electronic device can now be chosen and the load circuit is reconnected (“Leistungsteil EIN”). The supply of the electronic parts must not be disconnected.

mainsfuses

fuses

load circuit

elektronicssockets

L1L2L3

PWMPWMPWM PWM

vdvshunt

DF

LL RL

id

bipolartransistor GTO IGBT MOSFET

emergency

OFF

ON ON

OFF OFF

load circuit elektronics

signal PWM duty cycle, frequency

duty cycle/frequency

EinspeisungAbsicherung

Gesamt-absicherung

Leistungsteil

ElektronikSteckdosen

L1L2L3

PWMPWMPWM PWM

udushunt

DF

LL RL

id

Bipolar-Transistor GTO IGBT MOS-FET

NOT

AUS

EIN EIN

AUS AUS

Leistungsteil Elektronik

Signal PWM Tastgrad,Frequenz

Tastgrad/Frequenz


Using semiconductor specific drivers the control signal with variable frequency or variable duty cycle is connected to the control electrodes (gates) of the power semiconductors. The signal generator can be switched between variable frequency and constant duty cycle or constant frequency and variable duty cycle (“Tastgrad/Frequenz”). The respective switch is found above the LED display on the breadboard of the laboratory test. Both, duty cycle and frequency can be adjusted with the multiturn potentiometer on the right side of the LED display. For measurement the control signal can be tapped at the BNC jack “Signal PWM”. The LED display shows the frequency of the control signal. 6 Starting and course of the test 1. Check whether all automatic circuit breakers are switched on. 2. Switch on electronic supply with the push-button switch “Elektronik EIN”. 3. Choose the power semiconductor device with the stepping switch. The appropriate

device is displayed by a control lamp. 4. Switch on the load circuit with the push-button “Leistungsteil EIN”. The on-state is displayed with the control lamps of the transformer. If the supply for

the electronic circuit is not ready the load circuit cannot set into operation. 5. The measurements described in the instrucion have to be done. 6. Change to another power semiconductor: 1. Switch off load circuit (“Leistungsteil AUS”) 2. Choose the power semiconductor (stepping switch) 3. Switch on load circuit (“Leistungsteil EIN”) Attention: Change power semiconductor only in the off state of the load circuit! The “NOT-AUS” button disconnects the complete laboratory test from mains. It should be used only in emergency cases. Starting after pushing the “NOT-AUS” begins with point 1. (see above). 7 Evaluation of the test The losses of the several semiconductor devices have to be calculated with the measured values. Hereby, the switching losses and the on state losses have to be separated. Further theoretical reference calculation with the values from the datasheet have also to be done. The primary data of PV = function(f) and PV = function(a) have to be transferred in one diagram for each power semiconductor. The differences between theory and practice have to be discussed. 8 Questions for preparing the test 1. Using the equivalent circuit please explain the internal feedback mechanism of a

GTO during switch on and during switch off! 2. Why and how can a GTO be switched off though it is a four-layer device like a SRC? 3. Why must a GTO, against a SCR, not unconditionally be protected against a di/dt too

high? 4. what is the difference regarding the structure of the layers between MOSFET and

IGBT in general?


5. Mark the working characteristics in the output characteristic IC = f(UCE) of a bipolar transistor!

6. The IGBT can be announced as a bipolar power transistor with MOSFET controlling.

Which consequences has this? 7. In which parts can the total losses of a power semiconductor be divided? Hereby

which parts can be neglected? 8. A three-phase transformer is described with 400V/42V. Calculate the output voltage

Ud of a B6 bridge with and without a smoothing capacitor. Hereby the three-phase transformer works in Yd circuit.

9. Calculate the losses in an ohmic load with RL = 3.2 Ω, if with Ud (see question 8) a

step-down chopper with a duty cycle of 30 % works. Neglect the losses in the semiconductor.

10. Show all switching times of a bipolar transistor in an appropriate iC(t) and iB(t)

diagram. 11. Describe the function of the reducing network of a GTO. 9 Literature [1] Kahlen, H.: Material of the course „Leistungselektronik I“. University of Kaiserslautern, 4th edition 1997 [2] Tietze, U.; Schenk, C.: Halbleiter-Schaltungstechnik. Springer-Verlag Berlin, Heidelberg, New York, 11th edition 1999 [3] Macek, O.: Schaltnetzteile, Motorsteuerungen und ihre speziellen Bauteile. Hüthig-Verlag, 1992 [4] Jäger, R.: Leistungselektronik, Grundlagen und Anwendung. VDE-Verlag, Berlin 1977 [5] Heumann, K.: Grundlagen der Leistungselektronik. Verlag B.G. Teubner, 6th edition 1996

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Laboratory Test 8 - - TU Kaiserslautern · The laboratory test deals with on-state and switching behaviour and with the on-state and ... Triac) 3. fully controlled semiconductors,