IN5240 Fundamentals of RF Circuit Design Part 3...Institutt for Informatikk IN5240 Fundamentals of RF Circuit Design Part 3 Sumit Bagga*and Dag T. Wisland** *Staff IC Design Engineer,

Institutt for Informatikk

IN5240 Fundamentals of RF Circuit Design

Part 3

Sumit Bagga* and Dag T. Wisland**

*Staff IC Design Engineer, Novelda AS**CTO, Novelda AS


Outline

• Wireless communication systems

• Performance metrics of a wireless receiver

• RF building blocks

IN5240: Design of CMOS RF-Integrated Circuits,

Dag T. Wisland and Sumit Bagga


Modern RF Front-End (RF-FE)

Fundamentals of Modern RF Receivers© 2015 IEEE International Solid-State Circuits Conference 42 of 63

State of the art modern RF front-end

This is the starting point for the modern wireless receivers

IN5240: Design of CMOS RF-Integrated Circuits, Dag T. Wisland and Sumit Bagga

[Liscidini, ISSCC, 2015]


RF Blocks: Current Mode

• RF amplification: low-noise transconductance amplifiers (LNTAs) à power-to-current

• Frequency translation à mixer, filter and transimpedance amplifier (TIA)– TIA à current input to voltage output at IF/BB



LNA or LNTA?


• LNA– Power-to-voltage conversion à !",$"% → ∞

• High linearity voltage-driven mixer

• LNTA– Power-to-current conversion à !",$"% → 0

• Current mode mixer ß single-stage amplifier


CG Power-to-Voltage Amplifier

• Inductive loading à tuned

parallel resonance

– Frequency selectivity to remove

out-of-band interferers

– Current magnification (! " #$)– No voltage drop %& ≈ 1/*+




Feedback & Feedforward (1/2)


Reactive feedback or feed-forward à !" = $% with no added noise!

[Liscidini, 2006][Li, 2005]


Feedback & Feedforward (2/2)


• [Li, 2005] – !"-boosted CG-LNA with transformer feedforward loop– Feedforward factor is (1 + &'), where n is the turns

ratio and k is the coupling coefficient

• [Liscidini, 2005] – CG-LNA with positive transformer feedback loop– Feedback factor is (1 − &/')


Inductive Degeneration

Narrowband amplifier (! ≫ 1) with gate ($%) and source ($&) inductors à impedance matching at '(



Feedback: Inductive degeneration [4]

nf ≈1+ γQ2gmRs

Gm= IoutVin

=Q ⋅gmQ = 12ω0CgsRs

ω0 = 1(Lg+Ls)Cgs

• This LNTA has a narrowband transfer function

• This LNA requires 2 inductors (at GHz frequencygenerally Lg is external)



Noise-Cancelling

• 2 parallel stages, CG & CS à active balun• Set # = %&'( à noise from CG-stage is mitigated



Feed-forward: Noise cancelling [6]

Ioutp= 12Rs

(Vs−Vn)

Ioutm= −gm2(Vs+Vn)

Iout= Vs2

kRs

+gm§

©¨

·

¹¸+Vn − k

Rs+gm

§

©¨

·

¹¸

• For k=gmRs, the noise used to implement Rs is cancelledhaving

• The noise introduced by the LNTA can be loweredchoosing gm>>1/Rs

nf ≈1+ γgmRs

Gm= gm



LNTA Overview



LNTA comparison

LNTA Bandwidth nf Transconductance Gain

InductiveDegenerated Narrow

Boosted common gate Wide

NoiseCancelling Wide

nf ≈1+ γQ2gmRs

Gm=Q ⋅gm

Gm=1/Rsnf ≈1+ γ1+A

nf ≈1+ γgmRs

Gm= gm

• Inductive degenerated LNTA has the best NF (with Q>1)but is narrow-band

• Noise canceling and inductive degenerated have sameNF if Q=1



Passive Current Mixer & LPF

• Parasitic capacitance (!"#$) at the o/p of LNTA àmixer o/p impedance

– ↑ !"#$ à ↑ TIA noise


Mixer input impedance [8]

• Since the transistors are in triode, the voltage developedby the RF input current in baseband is up-converted atthe input

• Hence, in first approximation, the ZBB is reported at theinput up-converted around the LO frequency



Mixer output impedance [7]

• The real mixer can be modeled as a combination of anideal mixer and a switched capacitor circuit

• From the baseband the switched capacitances are seenas a resistor that also models the noise of the switches




Transistor Parameters and Biasing



Unity Gain Frequency

• "# is used for characterizing the achievable GBW product

• For a CS-stage,– %&(() ≈ +,(%-(()./)– 01(/)

02(/)≈ 34

/(5678561)

– 01(/)029(/)

= 1, and substituting ( = =>, ?@ ≈ ABCD(EAF8EAG)



CMOS transistor in strong inversion saturation àsquare law relationship

– "# =%&'()

*

+

,(-./ − -12)*(1 + 6-#/)

– 89 =:;<

:=>?≈

%&'()

*

+

,(-./ − -12) ≈ 2BCDEF

+

,"#

– 89 ≈ 2"#/(-./−-12) ≈ 2"#/-EH

– IJK ≈ LMN/OP

– IJK ≈LQ

RSTJU

MN

V

Transistor Biasing



!" vs #$

0.001

0.01

0.1

1

10

0.00001 0.00010 0.00100 0.01000 0.10000 1.00000 10.00000

g m [m

A/V]

Id [mA]



!"/$% or &$%/!" Method

• SPICE modeled

• Device parameters (', ), *+) are plotted in terms

of ,-/./ or 2.//,-– For a given value of ,-/./, find values for ',), *+

• Low ,-/./ à strong inversion

• High ,-/./ à weak inversion




Transistor Efficiency


[Boser, 2011]


Speed-Efficiency Product


Figure of merit is the product of !" and #$/&' to find the optimal value for ()* = (,-- (.

0

200

400

600

800

1000

1200

1400

1600

1800

2000

0.1 0.2 0.3 0.4 0.5 0.6 0.7 0.8 0.9 1

g m/I d

*f T [G

Hz/

V]

Vgs-Vt [V]


Current Density, !"/$

• Square law is only accurate over a narrow region

of strong inversion

• Test-bench comprises single device (common-

source) self-biased with ideal current source

– Change current density by changing %, while keeping

&' fixed for constant power consumption and

() (1/(,&'))




Current Density Device Operating Mode

• Weak inversion (subthreshold)– High !" efficiency

– $%& ≳ 100 mV for saturation

• Strong inversion– Highest *+, but poor !" efficiency

– $%& > ($.&−$0) ≳ 200 mV for saturation

• Moderate inversion– Compromise between weak and strong inversion



!"#, $% and &' vs ()/+

• -./ vs 01/2– -./ ≈ 56

789:;(01/2)

• >? vs 01/2– >? ≈ 789:;

5@6 (-AB − -D)

• ED vs 01/2– ED ∝ @

@



! Varying, Constant "#↓ %&


0

5

10

15

20

0.01 0.1 1

g m [m

A/V]

Id/W [mA/µm]


! Varying, Constant "#↑ %&


100

200

300

0.01 0.1 1

f T [G

Hz]

Id/W [mA/µm]


! Varying, Constant "#↑ %&'


-100

0

100

200

300

400

500

600

700

800

900

1000

0.01 0.1 1

V ov [

mV]

Id/W [mA/µm]


Strong vs Weak Inversion


! varying, constant "#• Strong inversion à ↑ %& ß ↓ ()* (≈ 2"#/()*)

– Reducing ↓ "#/Wà ↓ ()* à ↑ %&• Weak inversion à %& maxima

– %& ≈ "#/0(1• %& ⊥ !, L


Swing and !"#


• Overdrive voltage, $%& is defined as the minimum $'( required to keep a transistor in saturation– Rule of thumb: $'( = $%& + 50 mV à ↑ .%

• Larger $%& à reduced signal swing– w/ limited supply voltage (sub-1 V) à no headroom for

CG-stage (cascode) à ↓ stage gain à ↑ input referred noise à ↓ SNR


Trade-Offs

↑ "#$ ↓ "#$↑ Bandwidth

↑ Linearity

↑ Matching

↓ Noise ß &'( devices

↑ Power efficiency

↑ Signal swing

↓ Noise ß RF devices

"#$~*. , ∗ "..




Summary


Parameter Impact Factors

!"#Current efficiency, $%/W

()/$% or 2$%/(), +,, noise, headroom, power dissipation

- ß ., $% and à bandwidth (012, 01%)

3 +,, noise and intrinsic device gain


Key References

1. A. Liscidini, Fundamentals of Modern RF Receivers, ISSCC 2015

2. B. E. Boser, Analog Design Using !"/$% and &' Metrics, 20113. X. Li, S. Shekar and D. J. Allsot, Gm-boosted common-gate

LNA and differential Colpitts VCO/QVCO in 0.18 µm CMOS, 2005

4. A. Liscidini, et. al., Common Gate Transformer Feedback LNA in a High IIP3 Current Mode RF CMOS Front-End, 2006


Documents

IN5240 Fundamentals of RF Circuit Design Part 3...Institutt for Informatikk IN5240 Fundamentals of RF Circuit Design Part 3 Sumit Bagga*and Dag T. Wisland** *Staff IC Design Engineer,