IEEE TRANSACTIONS Oh’ 4. NOVEMBER A Single-Chip MSK ...read.pudn.com/downloads215/doc/fileformat/1010861/... · 158 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. VT-34. KO. 4

IEEE TRANSACTIONS Oh’ VEHICULAR TECHNOLOGY. VOL. VT-34. NO. 4. NOVEMBER 1985 157

A Single-Chip MSK Coherent Demodulator for Mobile Radio Transmission

Abstract-Design and performance of a single-chip minimum shift keying (MSK) coherent demodulator fabricated by complementary metal oxide semiconductor-integrated circuit (CMOS-IC) technology is described. The demodulator consists of a phase detection circuit, carrier recovery circuit, data recovery circuit, and timing-clock recovery circuit. For the carrier recovery circuit, three types of Costas-loop are reviewed from the viewpoint that MSK modulation format has a close relationship to binary phase shift keJing (BPSK) and quadrature phase shift keying (QPSK). Among these loops, a loop of center-frequency locking scheme modified from a BPSK Costas-loop, termed MSK Costas-loop in this paper, is adopted for IC implementation. Digital IC design techniques are next described. Utilizing the sample-and-hold operation of the digital devices, a divided-frequency locking scheme of the quadrature coherent demodulation is proposed. Finally, IC demodulator performance is experimentally shown in the static and Rayleigh fading environments. The bit error rate performance and error-burst characteristic are measured. It is concluded that the single-chip coherent demodulator is suitable for digital mobile radio application.

E I. INTRODUCTION

VEN FOR MOBILE radio communication, digital transmission technology is necessary to transmit

high-speed data, obtain a high-quality voice transmission, and maintain communication security. This has, in recent years, led to an investigation of various modulation techniques in order to determine the most suitable digital mobile radio communication system. One of the most effective candidates is continuous-phase frequency shift keying (CP-FSK) with a 0.5 modulation index. This keying consists of minimum shift keying (MSK) [ 13. and its related modulation schemes such as tamed frequency modulation (TFM) [2] and Gaussian baseband-filtered MSK (GMSK) [3]. This modulation family has the following features: 1) constant envelope. 2 ) narrow-band spectral occupancy. and 3) good bit error rate (BER) performance. Therefore. this modulation family is effective in a mobile radio channel. in which two primary communication resources, the transmitting power and channel bandwidth, are efficiently utilized.

The MSK signal can be detected not only by frequency discriminator detection but also by phase detection with differential or coherent detectors. Coherent detection is superior to the others in BER performance; equivalent to the

>fanuscript recsi\ed August 10. 1983: revised AUgUSt 1. 1985. H. Suruki and Y. Yarnao are aIth the Radio Cornrnunicatlon Netaorks

Laboratorlsz. Nlppon Telegraph and Telephone Public Corporation. 1-2356. Yokosuha-$hi. Kanaga\\a-ken. 238-03. Japan.

H . Kihuchi IS \\ith the Atsuei Electrical Communication Laboratories. Slppon Telegraph and Telephone Public Corporation. 3-9-1 I . 1Lfidori-cho. Musashino-shl. Tokho. 180. Japan.

antipodal signal when the maximum Doppler frequency is not excessively high [4]. Furthermore, BER performance is impaired little, although the narrow-band techniques to obtain TFM or GMSK are applied to MSK.

This paper gives an overview of carrier recovery for coherent detection especially on the Costas-loop. Next, its single-chip integrated circuit (IC) application and performance are described.

11. COSTAS-LOOP APPLICATION TO CARRIER RECOVERY OF MSK

A . Carrier Recovery of MSK Signal The MSK signal s ( t ) can be represented by the complex

form [5]:

s(t) = Re [S(t)eJz7fc‘] (la)

S( t ) = Aejocr), (lb)

$(t)=a, --+ICn, for nT 5 t<(n+ l )T (IC)

an= [ - 1, for space (“sa’) ( 1 4

$,,= [ O’ (modulo 27r) (le)

where Re [ * I , S ( t ) , f,, and n denote the real part of [ . I . complex envelope of s(t) , carrier frequency, and an integer. respectively. Envelope S(t) . has constant amplitude A , and modulated instantaneous phase q5(r). As represented in (IC), the phase @(t)’ increases or decreases linearly by 7r/2 in the duration Taccording to the symbol stream {CY,,} transmitted in bit rate of f b = 1/T.

From modulated signal s( t ) , a coherent detector should recover a phase-reference carrier r ( t ) represented by the complex form:

iTt

2 7

1, for mark (“m”)

p,

r(t) = Re [ eJZ7~ci]. ( 2 )

There are two approaches to recover the reference carrier as follows.

1) Frequency Shift Keying (FSK) Approach: The MSK signal, as a CP-FSK with modulation index 0.5. takes binary frequencies of fc + f b / 4 , and f, - f b / 4 for “m” and “s.” respectively. Squaring the MSK signal followed by bandpass filtering of center-frequency 2 f c produces a CP-FSK signal with a modulation index of one. known as Sunde’s FSK [6]. The signal has two line spectral components at 2 fc + f b / 2 and

0018-954518511 1OO-0157S01 .OO Z 1986 IEEE

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158 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. VT-34. KO. 4 . NOVEMBER 1985

2 f c - fb/2. The components can be extracted by two phase locked loops (PLL). From the two PLL outputs, the reference carrier and also the timing clock are recovered by utilizing multiplication, division, and filtering techniques [ 11.

2 ) Phase Shift Keying (PSK) Approach: The MSK signal can also be considered as a PSK signal. It is shown later that the MSK signal is obtained by transformations from binary PSK (BPSK) or quadrature PSK (QPSK). Then, carrier recovery techniques for PSK, such as Costas-loop [7], reverse modulation [8], and remodulation [9], [lo], are available for MSK.

The Costas-loop technique is superior to the others in circuit simplicity for the purpose of integrating the demodulator into a single chip.

B. BPSK Costas-loop The conventional Costas-loop is utilized for BPSK coherent

detection. The loop is available for MSK since MSK can be derived from BPSK as follows.

A phase trajectory of MSK for a certain symbol pattern is shown in Fig. 1 by a solid line. The phase changes linearly by k a/2 in each bit duration. The same amount of phase change is observed in d2-shift BPSK. In Fig. 1, the phase trajectory of the ?r/2-shift BPSK signal with differential encoding is shown by a broken line. This figure shows that the MSK signal is obtained from the n/2-shift BPSK signal by transforming the abrupt phase change to the linear phase change.

The a/2-shift BPSK signal Sb(t) is obtained from the differentially encoded BPSK signal with additional abrupt phase change by - at2 in the carrier at t = nT

Sb(t)=Re (Sb(t)ej[2*fcr-~c(1)J}, (34

sb(t) =AeJ6b('), (3b)

where nT 5 t < (n + 1) T. From the above equations, MSK is obtained by transform-

ing the abrupt change, expressed by (3c) and (3d), into the linear phase change of t / T

s(t) = Re [ S '(t)eJ2*Cfc-fb/4)'], ( 4 4

S'(t)=Aej@'('), (4b)

a,+ 1 at 2 T

d)'(t)=- . -++,

$,= a ( a k + 1)/2= n- I [ s: (modulo 2a). (4d)

k = - m

The above equations (4a)-(4d) are equivalent to (la)-(le). The new ones are, however, different in the following aspects: 1) carrier frequency deviates from f c by - fb/4; 2) modulated

, H/Z-SHIFT BPSK H

0

' 77/2

Fig. 1. Phase trajectories of MSK and xi2-shift BPSK

phase d) ' ( t ) linearly changes by a in the duration T for a, = 1 ; and 3) phase $I ' ( t ) remains in two stationary phases 0 and n for a, = - 1. The difference in the signal-space diagram is depicted in Figs. 2(a) and 2(b), where the reference phase of Figs. 2(a) and 2(b) are chosen in accordance with the respective carrier phase. In Fig. 2(a), signal movement is depicted in two modes. The in-phase mode occurs when n is even and quadrature-phase mode occurs when n is odd. Stationary phase characteristics 3) shows that the BPSK Costas-loop, shown in Fig. 3, is applicable for MSK carrier recovery. This technique is known as simplified MSK [ l 13 or deviated-frequency locking [12]. It was shown in [12] that the detection performance of the deviated-frequency locking scheme is inferior to that of the center-frequency locking scheme with respect to phase error in recovered carrier.

C. MSK Costas-loop The BPSK Costas-loop shown in Fig. 3 should be modified

to recover the reference carrier from the MSK signal at the center frequency fc. Instantaneous phase q5(t), given in (IC), can be expressed as

where the first term given in (4c) represents stationary phase characteristic, and the second term represents the phase corresponding to the deviated-frequency - f b/4 in (4a). When the reference carrier for the coherent detection has a phase error of AO, the in-phase and quadrature detected signals, u,(t) and uQ( t ) , become

where Im [ -1 denotes the imaginary part of [ . 1. Therefore, the product u x ( t ) becomes

In the above equation, eJ2@ ' ( I ) is the same that appears in the BPSK Costas-loop in the deviated-frequency locking scheme. The signal-space diagram of eJ2@ ' ( I ) is depicted in Fig. 4(a)


SUZUKI et 01.: MSK COHERENT DEMODULATOR 159

IN-PHASE MODE

QUADRATURE-PHASE MODE

Fig. 2. Signal-space diagram of MSK. (a) Center-frequency representation (la)-(le) (b) Deviated-frequency representation (4a)-(4d).

0 n/2 I

for reference. In (7), there is the additional factor e -j*f'T. The signal-space diagram of factor ej[2$'(t)-*t/Tl is depicted in Fig. 4(b). It should be noted that the signal in the diagram is always moving, and has no stationary phase. The simplest method for making a stationary phase is to eliminate e - J n f / T by multiply- ing a wave cos ( a t / T ) = + e j s t /T ) /2 :

ux(t) x cos (;)

The signal-space diagram of the first and second terms in [ .] are depicted in Fig. 4(c). They have one stationary phase, which is available for carrier recovery. Following the above idea, the MSK Costas-loop shown in Fig. 5 is presented in [3]. The loop-filter output up(t) becomes

1 uF(t) = - - sin (2A8).

8 (9)

This output is used to control the phase of the voltage- controlled oscillator (VCO). Although this carrier recovery circuit intrinsically requires timing-clock recovery PLL in the carrier recovery PLL, it was demonstrated experimentally that phase-tracking performance is sufficient for practical application [3].

D. QPSK Costas-loop

f,=fc-fb/4 8 p q I The MSK signal was considered as a signal transformed from the BPSK signal. On the other hand, the MSK signal is v , ( t )

F I L T E R vco obtained from an offset QPSK (OQPSK) signal sq(t ) :

Fig. 3 . BPSK Costas-loop, sq(t) = Re [Sq(t)ejznfc']. (10)

oo 0

Eye patterns of ur(t) = Re [Sq(t)] and ue(t) = I, [Sq(t)] are shifted by T as shown in Fig. 6(a) [ 1 3 ] . The MSK signal is obtained by transforming the rectangular elemental pulses of uI(t) and uQ(t) in the OQPSK signal into half-sinusoidal

OQPSK is recovered by the QPSK Costas-loop as shown in Fig. 7, the reference carrier of MSK can also be recovered by the QPSK Costas-loop, subject to introduction of a technique to generate the four stationary phases: k a/4, k (3/4)a with

f,=fc-fb/4 pulses, as shown in Fig. 6(b). Since the reference carrier of

(a)

0 which the QPSK Costas-loop is synchronized.

f, = f The simplest method for generating the stationary phases is to smooth the signal waveforms by a predetection bandpass

(b) filter (BPF). For example, application of the optimum Gaussian BPF, whose 3 dB bandwidth B normalized by the bit rate is BT = 0.5887 [ 1 2 ] , results in an eye pattern of u I ( t ) and

Another smoothing technique is to utilize a baseband filter, 0 a signal-space diagram as shown in Fig. 6(c).

f, = f, which narrows the MSK signal bandwidth. . . C)

Fig. 4. Signal-space diagrams. (a) eJ2* ' ( r ) . (b) eJ[2* ' ( r ) - r f 'n , (c) e12$'(r) and E. Applications to Modvied MSK

e J Z ' ( t ) e - J Z z f j T Several narrow-band techniques are being investigated to obtain narrow-band MSK-type signals. One technique utilizing a premodulation low-pass filter (LPF) before the VCO for


160 IEEE TRANSACTIONS ON VEHICULAR TECHNOLOGY, VOL. VT-34, NO. 4, NOVEMBER 1985

SIGNAL f c v,(t)

BPF

0 n f 2

fr=fc

LOOP FILTER

vco Fig. 5 . MSK Costas-loop.

EYE-PATTERN

OQPSK

SIGNAL-SPACE DIAGRAM

0

0

ga

0 Fig. 6. Eye-patterns and signal-space diagrams. (a) OQPSK signal. (b) MSK

signal. (c) MSK signal filtered with the optimum Gaussian predetection BPF. (d) GMSK signal: an MSK-type signal filtered with Gaussian premodulation LPF.

' f r=fc 9 3 4 - 7 - 1 FILTER

Fig. 7 . QPSK Costas-loop.

MSK modulation is a promising approach. For example a Gaussian LPF is applied to obtain GMSK [3], and the class Il partial-response technique with the third Nyquist criteria is applied to obtain TFM [2].

The signal-space diagram of GMSK is shown in Fig. 8(a). Although the signal moves along a circle because of the constant envelope property, signal movement is schematically shown by straight arrows. Angle 8 in the figure decreases as the normalized 3 dB bandwidth B b T of the LPF narrows, e.g.,

For reference, the eye pattern of u I ( t ) for BbT = 0.25 is shown in Fig. 6(d). The signal-space diagram of GMSK shows that GMSK maintains the following MSK properties:

8 + ~ / 2 f o r B b T > 0.5,andO + ~ / 4 f O r B b T = 0.2 - 0.23.

1) signal movements are symmetric with respect to the in- phase and quadrature axes;

2) the diagram is in two modes, and which signals are transmitted is decided along these axes alternately.

Furthermore, there are two kinds of signal movements:

1) the f n/2 phase change like MSK remains; 2) stationary phases: f a/4, * 3nI4, appear especially in

the case Of BbT = 0.2 - 0.23.

The first movement indicates that the BPSK and MSK Costas- loops are also applicable for GMSK carrier recovery, and the latter one allows utilization of the QPSK Costas-loop for it [2].

F. Comparison Three kinds of Costas-loops have been reviewed with

regard to MSK modulation. The relationship is shown in Fig. 9.

Among the three Costas-loops, the MSK Costas-loop was selected for demodulator IC implementation. The main reason is that degradation due to imperfect carrier recovery is relatively small in comparison to the BPSK Costas-loop [ 121. On the other hand, the QPSK Costas-loop is superior to the MSK Costas-loop since the former one intrinsically requires no recovered timing clock for the carrier recovery. However, the QPSK Costas-loop utilizes four reference carriers whose phases are spaced by nI4. This requires a VCO of higher frequency for carrier recovery utilizing digital IC's. This requirement contradicts the low-power and accurate IC design.

111. DIGITAL-IC DESIGN AND DIVIDED-FREQUENCY REFERENCE DETECTION

A . Demodulator Circuit The coherent demodulator consists of several functional

blocks: coherent detector, carrier recovery circuit, data recovery circuit, and timing-clock recovery circuit. To reduce the demodulator dimensions, monolithic IC implementation is indispensable.

In order to fabricate the demodulator by monolithic IC technology, it is better to construct the circuit of digital devices, i.e., building blocks of logic devices. This is because these devices operate stably when integrated into a chip. The only exception is an analog lag-lead loop filter for carrier recovery. This filter is attached externally in order to


SUZUKl et a/.: MSK COHERENT DEMODULATOR 161

Fig. 8 . Schematic signal-space diagram (a) GMSK. QJ) GMSK for BbT = 0.2 - 0.23, or TFM.

MODULATION SCHEME

r

CARRIER RECOVERY COSTAS-LOOP

L

(min. mod. index) ( l i n e a r phase (sinuioidal shaping )

-

I ' I I I -

MSK COSTAS-LOOP BPSK COSTAS-LOOP OPSK COSTAS-LOOP

Fig. 9. Modulation schemes relating to MSK, and Costas-loop applications,

accurately control the VCO phase. Since current drain for mobile radio system equipment should be as low as possible, CMOS-IC is the best choice for integration.

The digital demodulator circuit is shown in Fig. 10 [3]. This circuit has the following features.

1) The phase detector circuit is composed of D-type flip- flop circuits DFFl and DFF2. The sampling technique with D- type flip-flop circuits is adopted to remove the postdetection LPF that is always required in the analog detector to eliminate the harmonics.

2) The in-phase and quadrature reference carriers are generated with digital devices. For this purpose, a multivibrator having an oscillation frequency of 4 fc is used for the VCO. The oscillator output is divided four times with D-type flip- flop circuits, DFF3 and DFF4, and the in-phase and quadrature reference carriers are obtained from these circuits, respectively.

3) Two baseband multipliers in the MSK Costas-loop are realized by exclusive-or circuits EXORl and EXOR2.

4) For timing-clock recovery, a digital PLL (DPLL) is constructed with both a random-wark filter and a phase- selection type digital VCO. The filter, composed of a binary counter, integrates the phase error of the recovered timing clock, and controls the digital VCO phase.

IN -PHASE CAR!& f, QUADRATURE-PHASE 1 C i K

C I R C U I T CLOCK RECOVERY

D F F 3 OFF4 I I

4fr '

In addition to the above circuit features, the demodulator adopts a new divided-frequency locking scheme in which the reference carrier frequency is equal to the input signal carrier frequency divided by an odd number. Then, frequency fi decreases, which reduces the power consumption and design tolerance of the VCO. This scheme is realized by the sample- and-hold operation of the D-type flip-flop circuits in the phase detector circuit.



SIGNAL 7 3 - O REFERENCE

(a)

REFERENCE m SIGNAL RETARD

ADVANCED SIGNAL

-2n rn -I 0 I 2I 4

(C) Fig. 1 1 . Phase comparison characteristic of D-type flip-flop circuit. (a) D-

type flip-flop circuit. (b) Reference and signal. (c) Phase comparison characteristic.

B. D-Type Flip-Flop Detector Circuit The modulated signal and the reference carrier are applied

to D and T terminals, respectively, as shown in Fig. 1 l(a). The respective signals are assumed to be logic-level rectangular waveform as shown in Fig. 1 l(b):

s(t) = rect (2afct + d), (1 1)

r(t) = rect (2~fr t ) , (12)

fr = f c (13)

where rect (x ) = 1 or - 1 for cos (x ) 0 or <O, respectively. This circuit operates as follows.

I ) Phase Comparison Characteristics: The input signal is transferred to the output only on the positive-going edge of the reference carrier. The output becomes - 1 for the retard signal ( - a C 4 c 0), and 1 for the advanced signal (0 C 9 C a). The phase comparison characteristic, therefore, becomes rectangular having a period of 2 a in 6, as shown in Fig. 1 l(c). In this figure, the region of 1 denoted by q50 is ?r because the duty factor 6 of the signal s( t ) is 0.5. In general, do becomes

do = 2x6. (1 4)

2) Low-Pass Filtering Characteristic: The D-type flip-flop circuit holds the output until the next positive-going edge of the reference camer. The in-phase detected signal ul(t) is,

therefore, represented as

rect, ( t ) = 1, for 0 t<r 0, for t c O , t 2 r (17)

where 6(t) and the asterisk denote the Dirac’s delta function and the convolutional integral. The Fourier integral transformation of (15) gives

VT.) = F{ UT<t)} 3 ( 19a)

scn = F{ s(t)I, ( 19b)

where F{ - } denotes the Fourier integral of { } . In the above derivation, the following equation

is used. Equation (1 8) shows that the output of D-type flip-flop circuit is the linearly detected signal of S( f ), and that the output is filtered with low-pass transmission function of sin (?rf r)/(a f r ) on the following conditions.

1) Frequency deviation A f due to modulation is small:

f = f c + Af , A f / f r 1. (2 1 a)

2) The reference carrier frequency f , is f , /N where N is an positive integer:

r = N / f c . (21b) An example of this detection is illustrated in Fig. 12. In this figure, the frequency of S(t) is constantly deviated by Af = 0.04 f c , and the spectrum of odd harmonics corresponding to the rectangular waveform is shown in Fig. 12(a). The spectrum of the detected signal VI( f ) in the case of N = 1 is shown in Fig. 12(b). The spectral components nearflf, = 0 are the linearly detected signal. The other components are jitter noise in the detected signal (Appendix I). Dependence of signal to jitter noise ratio upon A f / f c is calculated, and shown in Fig. 13, This figure indicates that the ratio A f / f should be smaller than 1/50 for good signal to jitter-noise ratio over 20 dB. Supplementary circuits such as a transversal filter using majority logic gate are effective for reducing the noise further.

C. Divided-Frequency Carrier Reference Detection In the same manner described above, the quadrature-phase

detector output ue( t ) and its Fourier integral Ve(t) are given


SUZUKl et ai.: MSK COHERENT DEMODULATOR 163

Fig. 12. An operation example of the D-type flip-flop detector. (a) Spectrum of the input carrier signal. (b) Spectrum of the detected signal.

40

- - 30 m U

0

- w v)

4 20 ci

L w I-

7

0 I-

I .......................................................... 1 \ I

0 I

10-2 10-1

NORMALIZED DEVIATED FREQUENCY : Af/f,

Fig. 13. Signal-to-jitter noise ratio of the detected signal.

and

Considering the detection conditions, (21a) and (21b), the fundamental components of detected signals, V,(Af) and VQ(A f ), become

V Q ( A ~ ) = eJa V,(Af) (24)

where a is the phase of VQ(A f ) relative to that of V,( A f ). When N = 1, CY becomes 7d2, then, V,(Af) and VQ(A f ) represent the outputs of quadrature coherent detection. It should be noted that the odd numbers of N also satisfy the condition of quadrature coherent detection:

a= r/2, for N= 1, 5 , -r/2, for N=3, 7, . (26)

Therefore, the present coherent detector also operates by a reference carrier whose frequency f r is f c divided by an odd number. The detector output VQ( f ), however, should be inverted for N = 3 , 7, * 9 because a = - a12.

The divided-frequency locking scheme, however, causes little degradation in detection performance. This is because the normalized deviated-frequency increases by a factor of N . Therefore, the signal-to-jitter noise ratio, shown in Fig. 13, decreases. It is experimentally confirmed in Section IV that the degradation in BER performance is smaller than 1 dB for A f/ fr < 0.05.

IV. A SINGLE-CHIP MSK COHERENT DEMODULATOR

A . Characteristics of the IC Following the above design, a single-chip MSK coherent

demodulator was fabricated [ 141. I ) CMOS Process: Analog CMOS process was utilized

because the multivibrator for the VCO in carrier recovery loop can be accurately designed. Building blocks of CMOS logic devices, using approximately 800 FET’s, are integrated on a 3.7 x 2.8 m m 2 silicon-chip area. The chip is mounted in a dual-in package with 16 pins. Microscope photograph of the chip is shown in Fig. 14. The dc power supply is 5 V single. The power consumption is no more than 0.75 mW.

2) VCO Circuit: The multivibrator is designed as follows. a) The oscillation frequency linearly changes with

multivibrator input voltage by a current control circuit which charge or discharge the capacitor for determining the oscillation frequency.

b) The output waveform of 50 percent duty is obtained by the following capacitor configuration; one side of the capacitor is grounded and the other side is connected to the current control circuit. This circuit prevents current leak and latch-up at p-i-n junction of the capacitor.

The measured dependence of fr upon the multivibrator input voltage is shown in Fig. 15. At 2 .5 V input, the reference frequency is approximately f J 3 where fc = 455 (kHz).



TIMING-CLOCK R E C O V E R Y CIRCUIT

\ DETECTOR CIRCUIT CARRIER R E C O V E R Y CIRCUIT lI4-DIVIDER CIRCUIT

Fig. 14. Microscope photograph of the demodulator chip.

0 1 2 3 4 5

vco INPUT VOLTAGE (Vol t s )

Fig. 15. Oscillation frequency of the Vco versus input voltage.

3) Clock Recovery Circuit: An internal clock of frequency 32 f b is used for the digital VCO for precise timing-clock recovery. Phase error is integrated by an upldown counter to control the digital VCO. The timing-clock recovery circuit can operate separately from the detector circuit in the IC. The pull- in characteristics near fb = 32 (kHz) as a function of the counter stage is shown in Fig. 16. In practical usage, when counter is above the fifth stage, it shows good detector performance.

B. BER Measurement System The BER performance of the IC demodulator is measured in

a 32 kbitls GMSK experimental transmission system. The GMSK signal is generated by a quadrature modulator. It modulates directly in 800 MHz band using GaAs-FET double- balanced mixers [15]. The baseband signals of in-phase and

32.1

32.0

31.9

31.8

31.7 I b

Fig. 16. Pull-in characteristic of the timing-clock recovery.

quadrature in the modulator are generated by digital signal processing (DSP) [ 161. The baseband filtering parameter BbT is easily changed by replacing a read-only memory (ROM) in the DSP circuit. In this experiment, BbT = 0.25 is used since the value gives the maximum spectrum efficiency in a mobile radio system [ 171, [ 181. The modulator output through a Rayleigh fading simulator is converted in its carrier frequency to 455 kHz in a receiver. Receiver noise is filtered with a predetection Gaussian BPF having 3 dB bandwidth of 25 kHz which is a little wider than the optimum [12]. The filtered signal is amplified to the CMOS logic level.

C. BER Performance in a Static Environment Measured BER performance for MSK and GMSK in a static

environment is shown in Fig. 17 by broken lines, where P,, Eb, and No denote the BER, signal energy per bit duration, and noise power spectral density, respectively. The value of &/No corresponds to carrier-to-noise ratio I' as follows:

where BN is equivalent noise bandwidth. The solid lines in this figure show the performance of the ideal coherent detection calculated theoretically averaging all possible symbol patterns [19]. Impairment of the experiment from the calculated value is as small as 0.5 dB for MSK and 1 dB for GMSK in Eb/N0 at P, = 10 - 3 . Impairment is partly due to the increase in jitter- noise of detected signal in the divided-frequency locking scheme. A comparison is shown in Fig. 18 between measured BER of a discrete-IC demodulator with f r = f , and that of the single-chip demodulator with fi = f J 3 . From this figure, the monolithic IC implementation with divided-frequency reference detection is impaired approximately 0.5 - 1 dB.


SCZUKI ef al.: MSK COHERENT DEMODULATOR

1 0 - 2

01 0.

W

a

E W

t m

1 o - ~

0 5 10 1 5 20

Eb 1 No ( d B )

Fig. 17. BER performance of the IC demodulator in the static environment

GMSK (BbT=0.25)

f, = 4 5 5 ( k H z )

fb = 32 (kbps)

f, = f, / 3

( DISCRETE-IC DEMODULATOR ) DEMODULATOR

( SINGLE-CHIP

165

Considering these facts, it was concluded that the demodulator was well fabricated.

D. Detection Performance in a Fading Environment

The VHF or UHF mobile radio channel is usually character- ized by fading in propagation [20], [21]. The detection performance in a fading environment is measured from two aspects: BER performance and burst error characteristics.

1) BER Performance: The measured average BER performance in the Rayleigh fading environment with a parameter of maximum Doppler frequency f D is shown in Fig. 19 by broken lines. In this figure. the ideal average BER performance of MSK for average Eb/No under the quasistatic condition, f D + 0, P, = B,.T/2(Eb/N,), is shown by a solid line. The measured average BER disagrees with the calculated curve for large average Eb/No, and becomes constant. This is because impairment due to random FM noise becomes dominant over that due to envelope fade. From this figure, constant average BER is approximately proportional to fi similar to the differential detection [20].

The average BER performance is dependent upon carrier recovery characteristics. In Fig. 19. this dependence upon loop noise bandwidth of the carrier recovery BL = 460, 920 (Hz) is shown. For large BL, the recovered carrier becomes noisy. and BER performance in the static environment is impaired. However, average BER in the random FM noise region is reduced since the pull-in range of the carrier recovery is enlarged. Thus, appropriate loop bandwidth BL should be designed to meet the fading condition for actual application.

2) Burst-Error Characteristic: In the fading environment. burst error occurs. One of the practical characterization of the burst error length L is a period which is terminated by error free regions as shown in Fig. 20 [22]. The error free region is defined by a period in which no error is detected successively for more than 50 bits. Measured probability of the burst length p ( L ) is shown in Fig. 21. This is measured under the following conditions. 1) P, = 10 - 3 and BL = 460 (Hz), 2) in static environment Fig. 21(a)? and 3) in fading environment with the parameter f D = 2.5 (Hz) for Fig. 21(b), and 40 (Hz) for Fig. 21(c). The cumulative distributions of L for BL = 460, and 920 (Hz) are shown in Figs. 22(a) and (b) by solid lines, respectively. In Fig. 22(a), measured distribution in the static environment is shown for reference. In the static case, distribution of the Poisson process by computer simulation, whose probability of occurrence is 10 -3, is shown by a broken line. This figure indicates that measured error in the static case occurs almost at random as expected. The two figures, Figs. 21 and 22, show the following burst error characteristics.

The double error ( L = 2) occurs most frequently because 1 o - ~ the detector employs differential decoding. Burst length L

0 5 1 0 1 5 20 becomes longer as f D becomes smaller. The average burst length (L) was calculated, and listed in Table I. The average is

E b 1 No ( d B ) approximately equal to the estimated value given from the fade Fig. 18. Comparison of BER performance: a discrete IC demodulator with duration [221:

f, = fc and the single-chip demodulator with f, = f J 3 .

( L ) e = B v T

(Eb/No)fDTd2.rr In (1 +B.bzT/(Eb/No)} . (28)



1

h

-1

P Y

- : THEORY

E 10‘1 W J

Li CL

m a

k -1

m 4

I

P

3

NORMALIZED BURST-LENGTH : L

(a)

1

0 10 20 30 40 50 60 -1

P v

AVERAGE Eb / No (dB) I RAYLEIGH FADING

g 10-1 Fig. 19. BER performance of the IC demodulator in the Rayleigh fading Y

environment. 2 c v)

m = 3

100000. . . . 000001011011100010100110100000 . . . . 000001 0 U 10-2

L E % ; T E A BU:;f:;RoR +;;;Fj s 8 I m t; 2 a

10-3

1 : ERROR B I T 0 : CORRECT B I T

L : BURST LENGTH F : ERROR FREE LENGTH (b)

NORMALIZED BURST-LENGTH : L

F > 50 1

Fig. 20. Definition of the error burst. v -I

The value of is also listed in Table I. The values (L) and agree well when fD 2 10. The same agreement was

obtained in differential detection [22]. Therefore, distributions of burst length are similar in both coherent and differential detections. The difference between (L) and (15)~ for fo = 2.5 (Hz) is caused by the .approximation in (28).

E L 10’’ W -1

z a m U 0 > t;

s P

-1

m V. CONCLUSION -

A single-chip MSK coherent demodulator was implemented by CMOS-IC technology. The design of the IC demodulator and its performance were described in detail.

Various carrier recovery techniques for MSK were reviewed focusing on Costas-loop applications. Operation of the Costas-loop was described from the viewpoint of stationary phases in a received signal. It was pointed out that there are Fig. 21. Measured probability ofthe burst lengthp(l1. (a) Static condition.

(b) Fading condition, f~ = 2.5 (Hz). (c) Fading condition, f~ = 40 (HZ). three types of Costas-loops, and that these three loops are

L

1 o - ~

1 10 102 1 o3 NORMALIZED BURST-LENGTH : L

(C)

derived from the view of the MSK modulation format relating


SUZUKI et al. : MSK COHERENT DEMODULATOR 167

99.9

99

> U z w a B 2 50 w z s 2

2 10

a

1

0.1

99.9

99

2

x a a 10

40 (Hz) i BL = 460 (Hz)

TABLE I AVERAGE BURST LENGTH MEASURED (L) AND ESTIMATED

\ F A ! k ! E I T I ~ ~ 1 MAXIMUM DOPPLER FREQUENCY : fo

2.5 Hz 40 Hz 10 Hz

B = L 460 Hz

70.8 61.6 14.5 EXPERIMENT

NORMALIZED BURST-LENGTH : L reducing power consumption in the IC and for simplifying the (a) adjustment of the IC.

Finally, IC demodulator performance was described experimentally. Its performance in the static environment showed that the IC demodulator was well-fabricated. Measurements in the Rayleigh fading environment showed that the performance in the envelope fading region is the same as that of differential detection, while in the random FM region coherent detection is a little inferior to differential detection. The impairment was improved by appropriate design of the PLL for carrier recovery.

It was concluded, therefore, that the single-chip coherent demodulator is effective for digital mobile radio application.

40 (Hz) i i BL = 920 (Hz) .......................... * .................... .’....+ ..........................

APPENDIX I

AN EXAMPLE OF D-TYPE FLIP-FLOP DETECTION 1 , , , , , , , , , , , ,,,,; , , , ,,] The D-type flip-flop detection of a carrier s( t ) whose 1 .......................... i ...................................................... frequency is deviated by A f from f , is considered.

0.1 s(t)=rect [2n(fc+Af) t ] . (29) 1 10 102 1 o3 The Fourier integral transformation of s(t) becomes

NORMALIZED BURST-LENGTH : L (b)

Fig. 22. Cumulative distribution of L. (a) BL = 460 (Hz). (b) BL = 920 (Hz).

Intensities of the fundamental ( k = 1) and harmonics ( k = 3 , with d2-shift BPSK, and OQPSK. Among these loops, the 5 ) components are shown in Fig. 12(a). Substituting (30) into MSK Costas-loop was selected as the most suitable for IC (18) on the condition 7 = 1/ f , , VI( f ) becomes implementation. This loop is simple and has good performance because the center-frequency locking scheme is adopted. This j 1 - ( - l ) k sin ( r f / f , ) scheme impairs little due to phase error of the carrier recovery, and does not require high frequency VCO for the

Vim= ; .f I f , S c f - k A f )

k = - m

loop. The digital IC design of the demodulator is next described.

Utilizing the sample-and-hold operation of D-type flip-flop i f 0 k r - a Tf / f c

circuit for detector circuit, a divided-frequency locking scheme was proposed. This scheme allows lower frequency

- S c f - kAf - i fc) . (31)

operation of the VCO for carrier recovery, and is effective for The spectrum of VI( f ) is depicted in Fig. 12(b). The first

j 1 - (- l ) k sin (rf/ f , ) k



term of (31) is similar to the one in (30), and indicates the rectangular wave of linearly detected signal. The second term represents the jitter noise. The signal-to-jitter noise ratio is shown in Fig. 13.

ACKNOWLEDGMENT

Thanks are expressed to Mr. M. Murotani, Mr. Y. Okamoto, Dr. S . Seki, Dr. K. Daikoku, Dr. K. Hirade, and Mr. A. Iwata for their encouragement and guidance. The authors are also grateful to Mr. K. Murota for his advice on circuit design, and Mr. K. Momma, Mr. K. Uchimura, Mr. K. Ohtani, and Mr. H. Kudo for their many helpful sugges- tions and assistance.

REFERENCES

R. de Buda, “Coherent demodulation of frequency-shift keying with low deviation ratio,” IEEE Trans. Commun., vol. COM-20, pp. 429-435, June 1972. F. de Jager and C. B. Dekker, “Tamed frequency modulation, a novel method to achieve spectrum economy in digital transmission,” IEEE Trans. Commun., vol. COM-26, pp. 534-542, May 1978. K. Murota and K. Hirade, “GMSK modulation for digital mobile radio telephony,” IEEE Trans. Commun., vol. COM-29, pp. 1044-1505, July 1981. S. Pasupathy, “Minimum shift keying: A spectrally efficient modulation,” IEEE Commun. SOC. Mag., vol. 19, pp. 4-22, July 1979. G. D. Fomey, “The Viterbi algorithm,” Proc. IEEE, vol. 61, pp.

E. D. Sunde, “Ideal binary pulse transmission by AM and FM,” Bell

J. P. Costas, “Synchronous communications,” Proc. IRE, vol. 44, pp. 1713-1718, Dec. 1956. H. Yamarnoto, K. Hirade, and Y. Watanabe, “Carrier synchronizer for coherent detection of high-speed four-phase-shift-keyed signals,” IEEE Trans. Commun., vol. COM-20, pp. 803-808, Aug. 1972. H. Ishio, S. Seki, and Y. Fujii, “Design and performance of carrier recovery circuit for 806 Mbis 4-phase PSK transmission system,” Trans. IECE Japan, vol. 57-B, pp. 625-632, Oct. 1974 (in Japanese). Y. Morihiro, S. Nakajima, and N. Furuya, “A 100 hfbit/s prototype MSK modem for satellite communications,” IEEE Trans. Commun.,

F. Amoroso and J. Kivett, “Simplified MSK signalling technique,” IEEE Trans. Commun., vol. COM-25, pp. 433-441, Apr. 1977. M. Ishizuka and K. Hirade, “Optimum Gaussian filter and deviated- frequency-locking scheme for coherent detection of MSK,” IEEE Trans. Commun., vol. COM-28, pp. 850-857, June 1980. R . K. Kwan, “The effects of filtering and limiting a double-binary PSK signal,” IEEE Trans. Aerospace Electron. Syst., vol. AES-5, pp.

H. Suzuki, Y. Yamao, and H. Kikuchi, “Single-chip CMOS MSK coherent demodulator,” Electron. Lett., vol. 18, pp. 581-582, June 1982. Y. Yamao and H. Suzuki, “Design of a quadrature modulator consisting of GaAs-FET double-balanced mixers,” Trans. IECE Japan, vol. J65-B, pp. 1140-1147, Sep. 1982 (in Japanese). H. Suzuki and Y. Yamao, “Design of quadrature modulator for digital FM signaling with digital signal processing,” Trans. IECE Japan, vol. J65-B, pp. 1148-1155, Sep. 1982 (in Japanese). H. Suzuki and K. Hirade, “System considerations of M-ary PSK land mobile radio for efficient spectrum utilization,” Trans. IECE of Japan, vol. E-65, pp. 159-165, Mar. 1982.

268-278, Mar. 1973.

S y ~ t . Tech. J., V O ~ . 38, pp. 1357-1426, NOV. 1959.

v01. COM-27, pp. 1512-1518, Oct. 1979.

589-594, July 1969.

K. Murota, K. Kinoshita, and K. Hirade, “Spectrum efficiency of GMSK land mobile radio,” 1981 Int. Conf. Commun., Denver, 23.8, June 1981. K. Murota and K. Hirade, “Transmission performance of GMSK modulation,” Trans. IECE Japan, vol. J64-B, pp. 1123-1130, Oct. 1981 (in Japanese). W. C. Jakes, Jr., Ed., Microwave Mobile Communications. New York: Wiley, 1974. W. C. Y. Lee, Mobile Communications Engineering. New York: McGraw-Hill, 1982. K. Otani, K. Daikoku, and H. Omori, “Burst error performance encountered in digital land mobile radio channel,” IEEE Trans. Veh. Technol., vol. VT-30, pp. 156-160, Nov. 1981.

Yasushi Yamao received the B.S. and M S . degrees in electronics engineering from Kyoto Univer- sity, Kyoto, Japan, in 1977 and 1979, respectively.

Since 1979, he has been with the Yokosuka Electrical Communication Laboratory, Nippon Tel- egraph and Telephone Public Corporation (NTT). From 1979 to 1982, he was engaged in research on digital FM modems. He is now engaged in develop- mental research on portable mobile radio communication systems.

Mr. Yamao is a member of the Institute of Electronics and Communication Engineers of Japan.

Hiroyuki Kikuchi was born in Tokyo, Japan, on April 28, 1948. He received the B.S. and M S . degrees in electronic engineering from Keio Univer- sity, Kanagawa, Japan, in 1972 and 1974, respectively.

In 1974 he joined the Electrical Communication Laboratory, Nippon Telegraph and Telephone Pub- lic Corporation (NTT), Tokyo, Japana. He has been engaged in the research and design of CMOS digital and analog LSI’s for communication use. He is now a Staff Engineer in the Linear Integrated Circuit

Section, Integrated Electronics Development Division, NTT Electrical Com- munication Laboratories. His recent work is in the area of high frequency analog IC’s.

Mr. Kikuchi is a member of the Institute of Electronics and Communication Engineers of Japan.



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