Electronically Tuned UHF Power Amplifier

Frederick H. Raab

Green Mountain Radio Research Company Colchester, Vermont 05446 USA

Abstract -- This electronically tunable UHF power amplifier (PA) is based upon a gallium-nitride (GaN) HEMT operated in class C. The gate input is tuned by two arrays of varactor diodes. The drain output is tuned by a three-stub tuner whose stub lengths are controlled by pin diodes. From 325 to 800 MHz (factor of 2.5:1), the PA delivers an output of 34 to 50 W with an overall efficiency of 47 to 65 percent. Index Terms -- Power amplifier, electronic tuning, UHF, varactor, pin diode, reconfigurable

I. INTRODUCTION Achieving high efficiency and good power output from power amplifiers (PAs) operating at UHF and microwave frequencies generally requires tuning and matching both the input and output for the specific frequency of operation, load impedance, and power level. Electronically tuning the power amplifier offers a means of achieving optimum performance over a range of frequencies, load impedances, and amplitudes [1],[2],[3],[4],[5]. Dynamic load modulation via electronic tuning can also be used to achieve high-efficiency, wideband amplitude modulation [6],[7]. Broadband tuning networks can also be used to extend the operating bandwidth of high-efficiency PAs In these networks, the action of one component approximately counters the action of another over some range of frequencies. In broadband matching, there is an inherent trade-off between the bandwidth and the quality of the match. Nonetheless, bandwidths of 1.3 to 1.67 have been demonstrated [8],[9],[10]. Electronically tuned PAs overcome the limitations of broadband networks by readjusting the input and output matching as necessary to maintain performance when frequency or load impedance changes. This electronically tunable UHF power amplifier (Figure 1) is based upon

• Gallium-nitride (GaN) HEMT, • Varactor tuner for gate input, and • Three-stub output tuner controlled by pin diodes.

The PA is operable from 325 to 800 MHz and delivers an output of about 34 to 50 W with an overall efficiency of 47 to 65 percent.

Figure 1. Simplified circuit of electronically tuned PA.

II. RF-POWER TRANSISTOR The RF-power transistor is a Nitronex NPTP00050 GaN HEMT. It's gate is biased at -1.6 V, which produces a quiescent drain current of 4 mA at VDD = 10 V and class-C

operation. In previous tests in a class-F PA, this device has produced in excess of 50 W with efficiencies of 70 percent or better at frequencies from 500 to 1200 MHz.

III. INPUT TUNING Varactors provide continuously adjustable capacitances, hence perfect impedance matching. The power level of available varactor diodes is adequate for the 2.5-W maximum drive power of this PA. The input tuner is a classic capacitor-line-capacitor-line ladder network (Figure 1). It is implemented by placing two varactor arrays along the 35-Ω line leading to the gate. The GaN HEMT is initially placed in a test fixture and tuned by external tuners in 50-MHz steps across the operating band. The conjugates of the impedances of the input tuner are then transformed to find the gate impedances at each frequency. Design of the input tuner is then accomplished iteratively by a program that places two capacitors at selected points along the input line. For each frequency, the capacitances are stepped through their ranges to determine whether a satisfactory match to 50 Ω can be achieved. Micrometrics abrupt-junction diodes were selected for use in the input tuners based upon breakdown voltage, capacitances, and availability in a chip package. The selected devices have breakdown voltages of 30 V and specified small-signal Q > 140 for 500 MHz. The variations of capacitance with bias voltage are shown in Figure 2. In large-signal operation, part of the voltage range must be allocated to excursions of the signal voltage above and below the bias voltage. The effective tuning range is

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therefore about 3:1. The large-signal performance is evaluated by placing the diodes in a simple series-resonant circuit and driving power through them at various biases and frequencies. These tests show that the losses are no higher than 10 percent.

Figure 2. C(V) characteristic of varactor diodes.

The final design places a variable-capacitance array A3 of 11 - 31 pF 3 cm from the gate and a second variable-capacitance array A2 of 9 - 18.5 pF 3 cm ahead of A3. These capacitances are obtained by 4x4 arrays (Figure 3) of Micrometrics MTV4030-22 and MTV4030-24 varactor diodes. These arrays use anti-series arrangement of the diodes to minimize harmonic distortion. Stacking four diodes in series multiplies their voltage-handling capability by four. Placing four such stacks in parallel then restores the original values of capacitance.

Figure 3. Varactor array for input tuning. The input varactors are easily adjusted for the minimum input SWR. A small amount of trial-and-error variation is needed to find approximate settings, after which the two biases are iteratively adjusted to minimize the SWR. This

adjustment must be made with the gate bias on. Some readjustment is necessary as the supply voltage and drive power are increased. The variation of the bias voltages with frequency is shown in Figure 4. The series resonance of the arrays varies from 600 to 1400 MHz and is always higher than the operating frequency. At some of the lower frequencies, the biases are lower than the anticipated 5-V minimum. It appears that the voltages on two diodes in the stack flattens out, while most of the variable RF voltage appears across the other two. This plus the ability to tune to frequencies significantly below the low end of the design range (450 MHz) suggest that the varactors are useable over a greater range than anticipated.

Figure 4. Variation of bias voltages with frequency.

IV. OUTPUT TUNING Varactors with suitable voltage ratings were not available when this PA was built. Arrays at least as large as 8 x 8 would be needed to use the same diodes that are used for input tuning. Output tuning is therefore accomplished by a three-stub tuner (Figure 1). The lengths of the stubs are controlled by pin diodes. While this arrangement has power-handling capability, it has the disadvantage of tuning in steps and therefore achieves only approximately optimum operation. The pin diode selected for this design is the Micrometrics MMP7091. This diode is available in a convenient chip package, has a breakdown voltage of 500 V, off-state capacitance of 1 pF, and on-state resistance of 0.25 Ω. In the present application, it is biased at -100 V to produce the off state and at 100 mA to produce the on state. Large-signal tests demonstrated efficiencies of 96 to 97 percent when transmitting 25 to 44 W to a 50-Ω load. A simplified circuit of the variable-length transmission line is shown in Figure 5. Ten pin diodes are placed at equal distances along the line. The section of the output transmission line to which the stubs are connected is isolated by blocking capacitors. Bias is fed to the diode through an RF choke and flows through the main transmission line to ground through another choke.

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Figure 5. Simplified variable-length transmission line. The line impedances of the stubs are 50•Ω to allow a good range of stub input impedances. The main-line impedance of 33.85 Ω is intermediate between the 50-Ω load and the lower drain-terminal impedances that must be produced. The stub length of 8.5 cm and spacing of 7 cm are midway between the quarter-wavelengths of 11 cm at 400 MHz and 4.3 cm at 1 GHz. This design achieves a match to the target impedances with a maximum SWR of 1.2 over the band of operation. Tune-up begins by setting the drive power to about 1 W and adjusting the varactor biases to minimize the input SWR. The supply voltage is then set to 5 V and the pin diodes are toggled on and off to find one or more "hot spots". The best operation is obtained within one or two diode positions of one of the hot spots. The supply voltage is then increased to 20 V and the tuning and drive level are readjusted for optimum operation. The pin diodes selected for optimum performance of the PA are shown in Figure 6 as a function of frequency. In general, the trend is for longer stubs (larger N) at the lower frequencies and shorter stubs (smaller N) at the higher frequencies. At frequencies below the lowest design frequency (400 MHz), one or more stubs are at the maximum length (ND = 10). It appears that the need for

longer stubs limits operation with good performance to frequencies above 300 MHz.

Figure 6. Variation of stub lengths with frequency.

V. PERFORMANCE The performance is evaluated in terms of output power, gain, and overall efficiency. Overall efficiency is defined by ηO = Po / (Pi + PDR) ,

where Po is the output power, Pi is the dc-input power, and

PDR is the RF-drive power.

The relationship between supply and output voltages is fairly linear for 400 and 800 MHz, but there is some droop at 600 MHz. The efficiency is somewhat constant over a 15-dB range of output. The performance over frequency is shown in Figure 7 for operation at full output. An output of 30 to 50 W is produced from 325 MHz to 850 MHz. The overall efficiency is over 60 percent for 325 and 350 MHz, and then varies between 50 to 55 percent at frequencies up to 800 MHz. The performance degrades at frequencies of 300 MHz and lower and 850 MHz and higher. The drive power is in the range of 0.5 to 2 W and generally increases with frequency. The resultant gain varies from about 20 dB at the lower frequencies to 10 dB at the higher frequencies. At the lower frequencies, the onset of gate-voltage clamping occurs when the drive is near the optimum level. At the higher frequencies (e.g., 750 MHz and upward), gate clamping does not occur. This suggests that the upper limit on the frequency of operation is due to insufficient drive reaching the gate. For comparison, Figure 7 also includes the performance of the FET in the test fixture operating in class C (red) and class F (green). The output and efficiency of the test-fixture PA are generally somewhat higher than those of the tunable PA because of the more precise tuning and control of the harmonic impedances. The harmonics in the output are generally 24 dB or more below the desired signal.

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Figure 7. Performance over frequency.

VI. CONCLUSIONS The electronically tuned UHF power amplifier produces 40 to 45 W with an efficiency of 50 - 55 percent from 325 to 800 MHz. The prototype PA is operable over a 2.46:1 range (325 - 800 MHz) with good performance and over a 2.83:1 range (300 - 850 MHz) with useful performance. It can be tuned over a 2.67:1 range (300 - 800 MHz) without additional tuning components. It is difficult to achieve optimum power and efficiency with this output tuner for two reasons. First, the tuning becomes somewhat coarse at the high end of the band, resulting in impedances that are only approximately optimum. Second, adjustment of the tuner varies both the fundamental-frequency and harmonic impedances. The best performance is achieved with an available combination of these impedances, rather than impedances that are truly optimum. In general, the varactor-diode tuner is easier to adjust and more precise, but the pin-diode tuner has a greater frequency range. It should be possible to improve performance and extend the frequency range by using a combination of varactors and pin diodes. At the input, the pin diodes could switch varactor arrays or stacks of diodes in and out as needed for the frequency of operation. At the output, a stack of varactor diodes connected to each stub could be used for fine tuning. An additional stub might also

allow independent optimization of both the fundamental and second-harmonic impedances.

REFERENCES [1] F. H. Raab, "Electronically tunable class-E power amplifier," Int. Microwave Symp. Digest, Phoenix, AZ, vol. 3, pp. 1513 - 1516, May 20 - 25, 2001. [2] F. H. Raab, "Electronically tuned power amplifier," U.S. Patent 7,202,734, April 10, 2007. [3] W. C. E. Neo et al., "Adaptive multi-band multi-mode power amplifier using integrated varactor-based tunable matching networks," IEEE J. Solid-State Circuits, vol. 41, no. 9, pp. 2166 - 2176, Sept. 2006. [4] L. Sankey and Z. Popovic, "Adaptive tuning for handheld transmitters," IEEE MTT-S Int. Microwave Symp. Digest, paper TU4E-4, pp. 225 - 228, Boston, MA, June 7 - 12, 2009. [5] H. Kim et al., "A fully integrated CMOS RF power amplifier with tunable matching network for GSM/EDGE dual-mode application," IEEE MTT-S Int. Microwave Symp. Digest, paper WE3F-2, pp. 800 - 803, Anaheim, CA, May 23 - 28, 2010. [6] F. H. Raab, "High-efficiency linear amplification by dynamic load modulation," Int. Microwave Symp. Digest, vol. 3, pp. 1717 - 1720, Philadelphia, PA, June 8 - 13, 2003. [7] H. M. Nemati, C. Fager, U. Gustavsson, R. Jos, and H. Zirath, "Design of varactor-based tunable matching networks for dynamic load modulation of high power amplifiers," IEEE Trans. Microwave Theory Tech., vol. 57, no. 5, pp. 1110 - 1118, May 2009. [8] J. K. A. Everard and A. J. King, "Broadband power efficient Class E amplifiers with a non-linear CAD model of the active MOS device," J. IERE, vol. 57, no. 2, pp. 52 - 58, March/April 1987. [9] K. Narendra, A. Mediano, L. Anand, and C. Prakash, "Second harmonic reduction in broadband HF/VHF/UHF class E RF power amplifiers," IEEE MTT-S Int. Microwave Symp. Digest, paper TUPC-2, pp. 328 - 331, Anaheim, CA, May 23 - 28, 2010. [10] P. Wright, J. Lees, P. J. Tasker, J. Benedikt, and S. C. Cripps, "An efficient, linear, broadband class-J-mode PA realised using RF waveform engineering," IEEE MTT-S Int. Microwave Symp. Digest, paper WE3A-1, pp. 653 - 656, Boston, MA, June 7 - 12, 2009.

ACKNOWLEDGEMENT The work reported here was performed under contract W909MY-05-C-0005 from the U.S. Army Communication and Electronics Command (CECOM). Peter Kaunzinger and David Ruppe were the program managers. Michael Gladu assisted in construction and testing.

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