Microsoft Word - Ahmad_Saad_AlAdsani_Final_PhD_Thesis_Septemper2011_Manchester_UK.docELECTRIC VEHICLES
A thesis submitted to the University of Manchester for the degree of
Doctor of Philosophy
September 2011
2
1.4 Thesis Organization 38
1.5.1 PM Synchronous Machine with Claw Pole Field Excitation
(PSCPF)
40
(SynPM)
48
3
Machine (CPPM)
1.5.7 Dual-Rotor Machine 56
1.5.9 Series Double Excited Synchronous Machine (SDESM) 62
1.5.10 Switch Reluctance Machine with Stator Field Assistance 63
1.5.11 Novel Dual-Stator Hybrid Excited Synchronous Wind
Generator (DSHESG)
CHAPTER 2
70
2.3.1 Machine Back-EMF 78
2.4 Machine Magnetic Lumped Parameter models 82
2.4.1 General 82
2.4.3 WF Rotor Design Study 87
2.4.4 Comparative Analysis of WF Rotor Designs 89
2.4.5 Resistances 92
2.4.7 Rotor PM Demagnetisation 95
4
2.6 HPM Machine Thermal Model 101
2.6.1 General Principle of the Lumped Parameter Method 101
2.6.2 Conduction Heat Transfer 101
2.6.3 Convection Heat Transfer 104
2.6.4 Radiation Heat Transfer 106
2.6.5 HPM Machine Thermal Model 107
2.7 Comparison Between PM and Four HPM Machine
Topologies
111
HYBRID PERMANENT MAGNET GENERATOR
3.3.1 Winding Layout and Normalised Back-EMFs 122
3.3.2 Machine Synchronous Inductance, LS 123
3.3.3 Resistances 130
3.4.2 Simulation Model 134
3.5 System Studies 136
3.5.4 Quality of Generated DC Output 147
3.6 Comparison Between 3- and 9-Phase System Losses 151
3.6.1 Introduction 151
3.6.3 Passive and Active Conversion Stage Loss Prediction for
HPM and PM Generator Systems
152
SERIES HYBRID EV POWER-TRAIN
4.2.1 Control Strategy Analysis 162
4.2.2 DC-link Design Options 169
4.2.3 Simplified HPM Model 171
4.3 HPM Generator Output Power Control 174
4.3.1 Introduction 174
4.3.3 Solving Final Choice with Full Simulation Model 184
4.3.4 Thermal Analysis Results of Final HPM Choice 188
6
AND EXPERIMENTAL RESULTS
5.2.1 Stators 193
5.4 HPM Machine Control Cabinet and Test Rig Setup 204
5.5 HPM Machine Test Results 206
5.6 Conclusions 220
6.3 Future Work 225
Av1 Magnetic vector potential of one coil side Wb/m
Av1' Magnetic vector potential of the other coil side Wb/m
Ac Cross sectional area m 2
Ass Half stator slot area µm 2
Ars Half rotor slot area µm 2
Bn Flux density normal to the surface T
B Flux density T
Bm Peak flux density T
Cρ Coil pitch Degrees
Cø3 Predicted smoothing capacitor via dynamic model
simulation
F
Dc Conductor diameter mm
EMFPk Peak back-EMF per phase V
EMFA Phase A back-EMF V
EMFrms Back-EMF rms value V
ERS Capacitor internal resistance
fe Machine electrical frequency Hz
8
FS Switching frequency Hz
H Flux intensity A.T/m
hst Stator tooth thickness mm
hstt Stator tooth tip thickness mm
hry Rotor yoke thickness (length) mm
hrt Rotor tooth thickness mm
hrtt Rotor tooth tip thickness mm
hry Rotor yoke thickness mm
hgPM Permanent magnet machine air-gap thickness mm
hgWF Wound field machine air-gap thickness mm
If DC excitation current A
Ifmax Rated DC excitation current A
Iin Stator injected current A
IDC-link DC-link current A
Id d-axis current A
Iq q-axis current A
Is Phase current A
Icc Collector current A
Icn Rated Icc A
ICrms DC-link capacitor RMS current A
ICrmsM Data sheet RMS capacitor current A
is Phase current A
9
Jrs Rotor slot current density MA/m 2
Jss Stator slot current density MA/m 2
k Error tuning gain -
kp Slot fill factor -
KP Back-EMF waveform factor -
ke, kh, ka, kd Polynomial coefficient of the iron losses equations -
L Machine axial length mm
Lsc Copper length per stator coil mm
Lfc Copper length per field coil mm
LM Magnet thickness mm
LSelf Self-inductance H
Lm Measured self-inductance H
Ld d-axis inductance H
Lq q-axis inductance H
M Mutual-inductance H
10
MMF Magneto-motive force A.t
µr Relative permeability H/m
µri Relative permeability of iron H/m
µrs Relative permeability of steel H/m
n No. of phases Phase
nsp No. stator slots / rotor poles Slots
ncpp Coils per pole per phase -
nPc No. of conductors per single turn coil Conductor
nl No. of stator winding layers Layer
nstc No. of turns per coil Turns
nsc No. of series coils per phase Coils
nst No. of turns per phase Turn
ncp No. of capacitors in parallel Capacitors
ne Effective turns per-phase Turns
nf Excitation field number of turns Turns
Ns Mechanical speed rpm
11
Piron Iron losses W
PCD Average diode conduction losses W
PSW-on IGBT switch turn on losses W
PIGBT IGBT conduction losses W
PDIODE Power losses in the reverse biased diode W
PSW-off IGBT turn off losses W
PRR Reverse recovery losses W
PDCD Desired DC output power W
PDCM Measured Dc output power W
QRR Reverse recovery charge C
QRRN Rated reverse recovery charge C
Rs Stator phase resistance
Rf Rotor excitation field resistance
RDC DC-link resistor
ℜ Reluctance H -1
stℜ Stator tooth reluctance H -1
12
gℜ Airgap reluctance H -1
rttℜ Rotor tooth tip reluctance H -1
rtℜ Rotor tooth reluctance H -1
ryℜ Rotor yoke reluctance H -1
mℜ Permanent magnet reluctance H -1
S No. of stator slots Slots
Sρ Slot pitch Degrees
T Period s
tfN Rated IGBT turn off fall time s
tr Rise time s
trrN Rated reverse recovery time s
τrewe Rotor mean end-winding radius mm
τst Spacing between stator teeth (at tooth tip) mm
τrt Spacing between rotor teeth (at tooth tip) mm
τPMrp Spacing between rotor poles (at pole tip) mm
U Array of phase slot numbers -
uD0 Maximum forward voltage drop V
Vf DC Excitation field voltage V
VDC-link DC-link voltage V
13
VCO Voltage drop across the voltage P-N junction V
VCE Collector-to-emitter voltage V
VCEO Threshold VCE V
VF Forward voltage V
VA Phase voltage V
Vs Phase rms voltage V
v Volume m 3
Wstt Stator tooth tip width mm
Wrt Rotor tooth width mm
Wrtt Rotor tooth tip width mm
X FEA mesh coordinate in the x-axis direction -
Y FEA mesh coordinate in the y-axis direction -
θr Rotor position angle Degrees
σ Electrical conductivity S/m
θm Main temperature o C
θ1-6 Surface temperatures o C
δ Phase voltage angle with respect to the back-EMF Radians
λe Effective self flux-linkage Wb
λt Total flux per-phase Wb
λel Effective flux-linkage between different coils Wb
λHPM Amplitude of the total excitation flux-linkage Wb
14
-3, -9 Denote three- and nine-phase configurations -
ε Error signal -
CPPM Consequent pole permanent magnet hybrid excitation machine
DSHESG Dual-stator hybrid excitation synchronous wind generator
-D Dimension
HEV Hybrid electric vehicle
HPM Hybrid permanent magnet
HECPSG Hybrid excitation claw-pole synchronous generator
HESM Hybrid excitation synchronous machine
HHESM Homopolar hybrid excitation synchronous machine
ICE Internal combustion engine
IM Induction motor
PM Permanent magnet
SHEV Series hybrid electric vehicle
16
SDESM Series double excitation synchronous machine
SOC State-of-charge
TM Traction motor
Table 1.1 HESPSG advantages and disadvantages. 43
Table 1.2 Advantages and disadvantages of toroidal-stator transverse-flux
machine topologies.
Table 1.3 Advantages and disadvantages of the HESM. 48
Table 1.4 Advantages and disadvantages of the SynPM. 51
Table 1.5 Advantages and disadvantages of CPPM and variants as reported
in [22] to [27].
Table 1.7 SDESM advantages and disadvantages [31]. 63
Table 1.8 PM brushless hybrid generator advantages and disadvantages. 66
Table 1.9 DSHESG advantages and disadvantages. 67
Table 1.10 Main particulars of HPM machines presented in research
publications.
69
Table 2.1 Main design dimensions of benchmark brushless PM machine. 73
Table 2.2 Stator winding details of benchmark brushless PM machine. 74
Table 2.3 Lumped parameter analytic and FEA predicted flux for the PM
machine section.
85
Table 2.4 Lumped parameter analytic and FEA predicted flux for the WF
machine section.
Table 2.5 Dimensions details for WF machine rotor designs. 88
Table 2.6 Case (1) results when Jrs constant at 3.7 MA/m 2 . 90
Table 2.7 Summery of the Case (2) results when nf equals 100 turns. 90
Table 2.8 Predicted and measured phase resistance for HPM machines. 92
Table 2.9 Coefficients of equation (2.33) to (2.36). 99
Table 2.10 Different stator and rotor sections masses and core losses for the
fundamental harmonic only of 3-phase PM and HPM machine at
no-load.
100
18
Table 2.11 Comparison of maximum back-EMF per phase for the selected
HPM topologies and the reference PM machine.
113
Table 2.12 Total volume and mass for the different HPM machines. 113
Table 3.1 FEA predicted values for stator self- and percentage of mutual-
inductances relative to phase A for both HPM machine parts and
winding configurations.
128
Table 3.2 Measured and predicted HPM machine self inductance values. 129
Table 3.3 Mutual inductance calculation based on line-to-line inductance
measurements.
130
Table 3.4 Predicted and measured phase resistance for HPM machines. 130
Table 3.5 Comparison of different HPM machine configurations at rated
Psc when LS = LSelf and without applying excitation current to the
wound field part.
138
Table 3.6 Comparison of 3- and 9-phase HPM generator system data for LS
= LSelf.
142
Table 3.7 Voltage regulation of 3- and 9-phase machine systems for the
first comparison case.
145
Table 3.8 Voltage regulation of the 3- and 9-phase machine systems of the
second comparison case.
Table 3.9 The encountered iterative process for selecting the DC-link
capacitors.
149
Table 3.10 Different stator and rotor sections masses and core losses for the
fundamental harmonic only of 3-phase PM and HPM machine at
no-load.
151
Table 3.11 Losses due to passive rectification stage in HPM generator
system.
152
Table 3.12 Parameter definitions and typical values for semi-conductor loss
calculations.
154
Table 3.13 Calculated loss due to active rectification stage in PM generator
system for several switching frequencies.
155
Table 3.14 Comparison between passive and active rectification stage losses
of 3-phase systems.
Table 3.15 Comparison between passive and active rectification stage losses
of 9-phase systems.
156
Table 4.1 Simulation data for the operational points of Fig. 4.4. 166
19
Table 4.2 HPM machine winding electrical parameters as a function of
turns.
170
Table 4.3 HPM machine rotor excitation options for repetitive NEDC
driving cycles.
Table 4.4 HPM machine rotor excitation options for repetitive ECE-15
driving cycles.
181
Table 4.5 Simulation data for the operational point of the 63-turn HPM
machine.
185
Table 4.6 HPM machine thermal analyses results. 188
Table 5.1 Some RS components ordered parts. 202
Table 5.2 Open circuit measured results at different speeds without the WF
rotor of the 9-phase HPM machine.
208
Table 5.3 Open circuit characteristics of the 14-turn 9-phase HPM machine
at different speeds and WF excitation currents.
211
Table 5.4 Open circuit characteristics of the 14-turn 9-phase HPM machine
at fixed speed and extended range of the WF excitation currents.
213
Table 5.5 No-load characteristics of the 14-turn 9-phase HPM machine
with passive rectification stage at fixed speed and different WF
excitation currents.
Table 5.6 On-load characteristics of the 14-turn 9-phase HPM machine
with passive rectification stage for different WF excitation
currents and almost constant speed and DC-link output power (≈
2 kW).
Table 5.7 On-load characteristics of the 14-turn 9-phase HPM machine
with passive rectification stage for different WF excitation
currents and almost constant speed and DC-link output power (≈
3.2 kW).
20
Fig. 1.1 Different hybrid-electric vehicle power-train configurations [11]. 34
Fig. 1.2 Different ICE/generator machine and power conversion system
implementations for auxiliary power unit in series hybrid electric
vehicles.
35
Fig. 1.3 Variation of generator rectified AC output due to wound
excitation field.
with claw pole field excitation (PSCPF) [14].
40
Fig. 1.5 Flux path of the (PSCPF) machine as reported in [14]. 41
Fig. 1.6 Simplified construction figure of HESG as reported in [15]. 41
Fig. 1.7 A new type hybrid excitation claw-pole synchronous machine
components (HECPSG) [16].
Fig. 1.9 Transverse-flux machine components as reported by Spooner et
al [17].
44
Fig. 1.10 Toroidal transverse-flux machine reported by Spooner et al [17]. 45
Fig. 1.11 Flux paths in the toroidal transverse-flux machine, as reported by
Spooner et al [17].
46
Fig. 1.12 Structure of the HESM with a two-part rotor [20]. 47
Fig. 1.13 Cross-section of the of SynPM machine reported by Xiaogang
and Lipo, showing one phase belt of the stator winding [21].
49
Fig. 1.14 Back-EMF of one coil of the phase belt winding [21]. 49
Fig. 1.15 Flux lines of the six pole SynPM machine presented by Xiaogang
and Lipo [21].
49
Fig. 1.16 Example of resultant coil and phase back-EMF for different field
winding excitation conditions [21].
Fig. 1.17 Consequent pole PM hybrid excitation machine (CPPM) reported
in [22] and [23].
Fig. 1.18 Field controlled Torus-NS machine (FCT-NS) [24, 27]. 53
Fig. 1.19 Field Controlled Tours-NS type (FCT-NS) [24, 25, 27]. 54
Fig. 1.20 Reported combinations of the FCAFPM machines [25]. 55
Fig. 1.21 NN type FCAFPM machine reported in [25, 26]. 55
Fig. 1.22 Dual-rotor machine [27]. 56
Fig. 1.23 Imbricated hybrid excitation machine (IHEM) [28]. 57
Fig. 1.24 Homopolar and bipolar hybrid excitation synchronous machines
[29].
58
Fig. 1.25 HHESM various flux paths due to deferent excitations [30]. 59
Fig. 1.26 BHESM various flux paths created by different excitations [29]. 60
Fig. 1.27 Homopolar paths of fluxes created by PM [29]. 61
Fig. 1.28 SDESM design presented by Fodoren et al [31]. 62
Fig. 1.29 Hardware of SDESM. 63
Fig. 1.30 Cross-section and actual switch reluctance motor with field
assistance.
64
PM brushless hybrid generator [36].
65
Fig. 1.32 Detailed structure and magnetic circuits of DSHESG [37]. 67
Fig. 2.1 Benchmark brushless PM machine. 72
Fig. 2.2 Main dimensions of the benchmark brushless PM machine. 73
Fig. 2.3 Stator lamination and phase A winding scheme for benchmark
brushless PM machine.
Fig. 2.4 Benchmark brushless PM machine 2-D FEA model. 77
Fig. 2.5 One quarter of the benchmark PM machine stator winding layout. 79
Fig. 2.6 Benchmark PM machine open circuit back-EMF waveforms at
3000 rpm.
EMF per coil.
and line to line back EMF’s.
81
Fig. 2.10 Lumped parameter magnetic circuit representation of the PM
machine.
85
Fig. 2.12 Lumped parameter magnetic circuit representation of the WF
machine.
86
Fig. 2.13 FEA results for 3 wound field designs considered for detailed
study.
88
Fig. 2.14 Case (1) results with a vertical line intersection at 35.9 V; 100
turns.
90
Fig. 2.15 Air-gap flux versus slot current density for the three WF rotor
designs.
91
Fig. 2.16 WF machine open circuit back-EMF waveforms at 3000 rpm. 91
Fig. 2.17 FEMM model of PM machine section, highlighting the
circumferential contour line taken for Maxwell stress integration.
93
Fig. 2.18 Phasor relationship of stator currents. 93
Fig. 2.19 Predicted torque versus rotor angle for the HPM machine
sections.
94
Fig. 2.20 Magnetic saturation characteristics of the PM and WF HPM
machine parts.
96
Fig. 2.22 PM pole in FEMM Post-processing mode. 96
Fig. 2.23 Manufacture loss data for United Laminated Steel (0.47 mm)
[62].
97
Fig. 2.24 Curve fitting for HPM machine core loss prediction. 99
Fig. 2.25 Stator sections for iron loss calculations with FEA results at no-
load condition (PM part).
Fig. 2.26 Illustrative lumped parameter thermal networks for three heat
flow directions.
102
23
Fig. 2.27 PM machine steady state thermal analysis results via Motor-
CAD.
108
Fig. 2.28 WF machine steady state thermal analysis results via Motor-
CAD.
109
Fig. 2.29 PM machine thermal transient analysis curves via Motor-CAD. 110
Fig. 2.30 WF machine thermal transient analysis curves via Motor-CAD. 110
Fig. 2.31 WF machine rotor tooth thermal transient analysis curves for
different excitation currents via Motor-CAD.
110
Fig. 2.32 PM and several HPM machine models used in the analysis. 112
Fig. 2.33 Back-EMF waveform for SynPM and the PM machine. 114
Fig. 2.34 Comparison between PM flux density distribution points for the
HESM and DESM designs when If = -4.2A (field weakening) and
at full stator field current.
114
Fig. 3.1 Series HEV power-train schematic with a 3- or 9-phase HPM
generator supplying power to the vehicle DC-link.
119
Fig. 3.2 Three- and nine-phase HPM generator with power circuit
converter.
119
Fig. 3.3 AC L-L voltage and rectified DC output for 3-and 9-phase
machines at no-load.
121
Fig. 3.4 General stator winding connections for 3- and 9-phase HPM
machines.
122
Fig. 3.5 Normalised back-EMF waveforms of the 3- and 9-phase HPM
machines.
123
Fig. 3.6 Single coil representation of the 3- and 9-phase HPM machine. 127
Fig. 3.7 Equivalent steady-state circuits of the HPM machine in the d-q
axes.
132
Fig. 3.8 HPM generator phasor diagram in steady state region for three
operating cases.
dynamic model.
Fig. 3.10 Nine-phase HPM generator with power circuit converter dynamic
model.
136
24
Fig. 3.11 Multi-phase HPM generator without and with power circuit
converters.
137
Fig. 3.12 Phase voltages and current waveforms of the 3-phase HPM
machine configurations with sinusoidal and less trapezoidal back-
EMF waveforms at different loading conditions with and without
passive rectification stage (LS = LSelf).
140
Fig. 3.13 Phase voltages and current waveforms of the 9-phase HPM
machine configurations with more trapezoidal back-EMF
waveforms at different loading conditions with and without
passive rectification stage (LS = LSelf).
141
Fig. 3.14 DC-link voltage waveforms for the 3- and 9-phase HPM
configurations at rated conditions (LS = LSelf).
143
Fig. 3.15 Voltage regulation of HPM machine at different loads. 144
Fig. 3.16 Output power ratio curves for 3-phase machine with sinusoidal
back-EMF versus 14 turns per tooth 9-phase machine.
145
Fig. 3.17 Different terminal and DC-link voltage regulation curves of the
3-phase with sinusoidal back-EMF versus the 14 turns per tooth
9-phase HPM machines.
146
Fig. 3.18 Output power ratio curves for 3-phase machine with trapezoidal
back-EMF versus 18 turns per tooth 9-phase machine.
146
Fig. 3.19 Different terminal and DC-link voltage regulation curves of 3-
phase with trapezoidal back-EMF versus 18 turns per tooth 9-
phase HPM machines.
Fig. 3.20 DC-link voltage waveform with and without smoothing capacitor
effect for the 3-phase trapezoidal back-EMF HPM machine
system at rated conditions.
148
Fig. 3.21 SimPower models for 3-phase HPM generator systems with DC-
link inductance.
150
Fig. 3.22 Impact of DC-link inductance on DC-link voltage ripple in 3-
phase HPM.
150
Fig. 4.1 Schematic of a series hybrid EV power-train and typical DC-link
voltage variation during urban driving.
159
Fig. 4.3 HPM generator open-loop control system block diagram. 164
25
Fig. 4.4 DC-link voltage and peak phase back-EMF with-respect-to the
WF excitation current for a constant DC-link output power of
3 kW.
166
Fig. 4.5 Phase voltage and current waveforms for eight DC-link voltage
levels for the 18-turn, 9-phase HPM machine at PDC = 3 kW.
167
Fig. 4.6 Magnified views of Figs. 4.5(a), (d) and (h) illustrating waveform
phase relationships.
Fig. 4.7 DC-link voltage and peak phase back-EMF with-respect-to the
WF excitation current for varying turns per phase and constant
DC-link output power of 3 kW.
170
Fig. 4.8 HPM generator closed loop control system block diagram. 171
Fig. 4.9 Simplified HPM model for the full battery cycle analysis. 172
Fig. 4.10 Examples of results from the detailed and unadjusted
simplified simulation models.
[4].
175
Fig. 4.12 Examples of battery terminal voltage during repetitive NEDC
driving cycles.
Fig. 4.13 Examples of battery terminal voltage during repetitive ECE-15
driving cycles.
176
Fig. 4.14 Example simulation results of HPM generator variables with the
proposed excitation current control scheme. 177
Fig. 4.15 Five operating scenarios of the HPM generator with respect to
the DC-link voltage levels in volts.
179
Fig. 4.16 Total energy losses for the NEDC and ECE-15 driving cycles at
different DC-link voltages. 181
Fig. 4.17 HPM machine rotor power loss for the five operating scenarios
and two driving cycles.
Fig. 4.18 HPM field excitation current variation for repetitive NEDC
driving cycles.
Fig. 4.19 DC-link voltage and peak phase back-EMF with-respect-to the
WF excitation current for 63-turn case and a constant DC-link
output power of 3 kW.
184
Fig. 4.20 Phase voltage and current waveforms for eight DC-link voltage
levels for the 63-turn, 9-phase HPM machine at PDC = 3 kW.
186
26
Fig. 4.21 Magnified views of Fig. 4.19(a) (b) and (c) illustrating waveform
phase relationships.
Fig. 4.22 Steady-state temperature distribution for PM section of 63-turn
HPM machine.
Fig. 4.23 Steady-state temperature distribution for WF section of 63-turn
HPM machine.
Fig. 5.2 HPM machine stator fabrication with both winding
configurations.
195
Fig. 5.3 HPM machine stator and tooling before and after they placed in
the VPI machine.
196
Fig. 5.4 HPM stators before and after VPI and heating process. 197
Fig. 5.5 Some stages before stators are fitted inside the case through
expanding and shrinking process based on different case
temperatures.
198
Fig. 5.7 PM rotor of the HPM machine. 199
Fig. 5.8 WF rotor iron sheet of the 10 mm HPM machine section. 200
Fig. 5.9 Finalised WF rotor assembly. 200
Fig. 5.10 HPM machine rotor with plain shaft. 201
Fig. 5.11 HPM machine with finalised rotor. 201
Fig. 5.12 Finalised HPM machine assembly. 201
Fig. 5.13 The 3-phase bridge rectifier major characteristic curves [127]. 202
Fig. 5.14 Detailed schematics of the final control cabinet and test rig setup. 203
Fig. 5.15 HPM machine control cabinet parts. 204
Fig. 5.16 Final HPM machine test rig setup. 205
Fig. 5.17 W-L set hardware with open circuit characteristics curve. 205
Fig. 5.18 Measured and predicted phase (A) back-EMFs waveforms at 3
krpm for the 14-turn 9-phase HPM machine.
206
Fig. 5.19 Measured back-EMFs waveforms at Ns ≈ 1.5 krpm for the 14-
turn 9-phase HPM machine.
Fig. 5.20 Projection of Fig. 5.19 measured back-EMFs waveforms on
single plot.
Fig. 5.21 Measured open circuit DC-link voltage waveforms at different
speeds.
208
Fig. 5.22 Measured temperature rise for the 14-turn 9-phase HPM machine
operating at 1.5 and 3 krpm without WF rotor and with no-load.
209
Fig. 5.23 Illustrative diagrams of the open circuit test setups based on the
3-phase HPM machine configuration.
210
Fig. 5.24 Peak back-EMF variation due to the WF part only at different
speeds.
210
Fig. 5.25 Several HPM machine back-EMF waveforms due to different
WF excitation currents at Ns ≈ 1.2 krpm.
212
Fig. 5.26 Predicted and measured peak back-EMF characteristic curves of
the WF part of the 14-turn 9-phase HPM machine at Ns ≈ 3 krpm.
213
Fig. 5.27 Illustrative diagram of the closed circuit test setup based on the 3-
phase HPM machine configuration.
current for almost a constant rectified DC-link output power of ≈
2 kW at Ns ≈ 3 krpm.
215
Fig. 5.29 HPM machine phase voltage and current with the rectified DC-
link voltage and current waveforms for two DC-link voltage
levels at PDC-link ≈ 2 kW and Ns ≈ 3 krpm.
216
Fig. 5.30 HPM machine phase voltage and current with the rectified DC-
link voltage and current waveforms for the second two DC-link
voltage levels at PDC-link ≈ 2 kW and Ns ≈ 3 krpm.
217
Fig. 5.31 Measured and predicted DC-link voltage with-respect-to the WF
excitation current for almost a constant rectified DC-link output
power of ≈ 2 kW at Ns ≈ 3 krpm.
218
Fig. 5.32 Measured and predicted DC-link voltage with-respect-to the WF
excitation current for almost a constant rectified DC-link output
power of ≈ 3.2 kW at Ns ≈ 3 krpm.
219
28
ABSTRACT
In this research study, the feasibility of using one of the Hybrid Permanent Magnet (HPM)
machine topologies acting as a generator with a passive rectification stage is considered.
The primary application area is in the power-train of a series hybrid electric vehicle where
the concept will be considered as an alternative to brushless PM machines interfacing to the
vehicle power-train via active power electronic converters. The electro-magnetic design of
the two main parts in the selected HPM generator topology and their individual system
behaviour at normal and rated conditions will be studied. Prediction of the transient and
steady state temperature in some of the HPM machine parts will be conducted based on
commercial thermal analysis software. Two HPM machine stator winding configurations;
3-phase and 9-phase, with their relevant passive rectification stages will be analysed in
terms of their terminal and DC-link output power along with the quality of the generated
DC output voltage. An investigation of the operational characteristic of the HPM generator
when delivering a fixed power at a fixed speed into a dynamic DC voltage source typical of
a hybrid electric vehicle power-train subject to urban driving regimes will be presented.
The research work will be a mixture of simulation studies using electro-magnetic finite
element analysis (FEA), transient machine and system analysis via SimPower, a
Matlab/Simulink toolbox set, along with test validation via a representative prototype HPM
generator configuration and its interface to an experimental electrical system evaluation
platform.
29
DECLARATION
No portion of the work referred to in this thesis has been submitted in support of an
application for another degree or qualification of this or any other place of learning.
30
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property.pdf), in any relevant Thesis restriction declarations deposited in the
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http://www.manchester.ac.uk/library/aboutus/regulations) and in The University’s
policy on presentation of Theses. from the Head of School or Electrical and
Electronic Engineering (or the Vice-President) and the Dean of the Faculty of Life
Sciences, for Faculty of Life Sciences’ candidates.
31
ACKNOWLEDGEMENTS
First, I wish to express my sincere thanks to my supervisor, Dr. Nigel Schofield. His
knowledge, valuable guidance and unlimited patience inspired the completion of this thesis.
His encouragement, understanding and willingness to spend his precious time with me are
beyond my appreciation. Moreover, especial thanks also go to my supervisor for his
guidance and help in the up-to-date published papers and for the up coming journal
publications.
I would like to express my sincere gratitude towards my parents and all my family members
who have done whatever has been required to support me along the way of my PhD research
study and specially my beloved wife who, without her endless support, encouragement,
understanding, patience and help I would have not been able to complete this essential stage of
my research career.
My deep appreciation also goes to many people whose advice, assistance and
encouragement have been a great help to me: Prof. Sandy Smith for talking the time
attending my first and second year viva committee along with his valuable advice, Dr.
Sinisa Durovic for the many interesting and helpful discussions we had together and
Danny, Paul and John of the School mechanical workshop for their help with experimental
machine manufacture and setup.
I would also like to thank my colleagues who made my stay at the University of Manchester a
pleasant memory, including Ali Jarushi for working with him on several published IEEE
papers.
Finally, I acknowledge the Power Conversion Group in the Electrical and Electronic
Engineering School at the University of Manchester for providing me with excellent
academic circumstances that are essential to the accomplishment of my post graduate
research study. I am really fortunate to have known so many great people so far in my life
that my limited memory cannot accommodate. To those who are missing in this
acknowledgement, and were supposed to be here, I sincerely apologize.
32
Worldwide concerns over increasing energy consumption and pollution due to
transportation systems is a primary motivation for alternatively fuelled or more-electric
vehicles. Electric vehicles are set to play an increasingly predominant role in the
automotive market since they address the energy and environmental impact of an
expanding road transport population by offering a more energy efficient and less polluting
power-train alternative to conventional internal combustion engine (ICE) vehicles [1].
However, when compared with conventional gasoline or diesel fuelled vehicles, all-electric
vehicles are disadvantaged in driving range because of the relatively low energy storage
capacity of the on-board batteries, excessive battery mass and the refuelling, thereof or
recharging time, associated with existing electro-chemical battery technologies [1, 2]. For
example, 2.5 tonne ICE urban vehicle will have a typical range of 400 km [3] compared to
126 km for the equivalent all-electric vehicle [4]. Hybrid electric vehicles address the
vehicle range issue while offering improved fuel economy and emissions reduction when
compared to conventional ICE vehicles, but only when the vehicle has intermittent or
transient duty operation. Such vehicles are now being commercialized in high volumes by
major automotive companies, for example Toyota [5]. There are a number of larger fleet
type vehicles now in operation, in the UK for example, Stagecoach in Manchester are
trialling 11.8 tonne, double-decker hybrid city buses which are claimed to produce 30% less
33
carbon emissions than standard vehicles [6]. Those two vehicles have a parallel hybrid-
electric power-train configuration, as illustrated schematically in Fig. 1.1(a). The parallel
hybrid-electric vehicle format is generally the favoured scheme for most vehicle suppliers
to-date since it allows them to continue to use their existing power-train components while
accommodating an additional power input, thus requiring lower investment and minimising
system risk. However, the series hybrid-electric vehicle, as illustrated in Fig 1.1(b), can
offer more flexibility and lead the way to future vehicle systems where the ICE energy
source is replaced by alternatively fuelled combustion engines (using petroleum products,
methanol, hydrogen) or fuel cell systems [7]. Here, the ICE is mechanically disconnected
from the power-train and is hence independent of vehicle road wheel speed. Energy
conversion is then from the on-board fuels chemical energy to kinetic energy via the ICE,
and then from kinetic to electrical energy via an engine mounted or coupled generator, Fig.
1.1(b) [8]. Fig 1.1 also shows one other possible power-train implementation scheme (c)
although many topologies have been reported. There is, therefore, an interest in on-board
auxiliary power units that would serve as an energy input to the vehicle power-train, for
sub-urban or highway driving.
Such schemes tend to combine the engine start and generator functionalities into one
machine, a philosophy of reduced component count. However, the start and generation
operational requirements are quite different. Here, the machine has to provide high peak
power transients during starting and engine braking. More importantly, the
starter/alternator requires a voltage source inverter for active power conversion between
starter/alternator and the ICE, hence the power converter needs to have active switching
and high silicon VA rating. However, most of the machine operation is in the steady power
generation mode which is usually over a limited speed range and at lower power levels than
for the system starting transient. Thus, if the starting and steady power generation are
realised by two machines, one a traditional starter and the second a lower power generator
operating over a constrained speed range, the installed machines could be simpler in form
and the power electronic conversion requirements reduced.
This thesis therefore considers the design of a generator system that is capable of supplying
the average energy requirements for a road vehicle, but not the peak powers required for
acceleration or recovery of regenerative braking energy. For an urban vehicle the disparity
between peak and average power (energy) can be between 4:1 and 10:1 depending upon the
specified vehicle road duty cycle [9]. For HEV topologies, a significant saving in energy
can be realised by levelling of the vehicle energy demand via the inclusion of a peak power
34
buffer in the vehicle drive-train, downsizing the ICE power capability and operating the
ICE over the optimum region of the engine power-speed characteristic in terms of specific
fuel consumption and emissions [10]. As mentioned, the most flexible implementation of
this scheme is the series hybrid (SHEV) drive-train configuration, as shown in Fig. 1.1(b).
Here, the ICE acts as the prime mover to an electrical generator specified to satisfy the
vehicle net energy or average power demand.
(a) Parallel hybrid
(b) Series hybrid
Fig. 1.1 Different hybrid-electric vehicle power-train configurations [11].
The use of dual power and energy sources requires a detailed analysis of the vehicle drive-
train in order to optimise component specifications and energy management strategies [9].
In-particular, the vehicle power-train DC supply may typically have a 30% voltage
variation during acceleration and deceleration due to the vehicle power transients and
energy source regulation. Hence, some form of voltage management is required for the
electrical generator system.
Brushless Permanent Magnet (PM) machines are an established candidate machine
technology for engine mounted generators since they have lower losses and higher power
and torque densities when compared to competing technologies [12, 13]. However, they do
have some drawbacks, for example the constant permanent magnet flux-linkage
necessitating some form output voltage management.
There are a number of ways in which the output voltage of the PM (or other machine
technology) generator may be facilitated, as illustrated schematically in Fig. 1.2. The PM
generator output (3-phase in this case) can be passively rectified and the output voltage
magnitude regulated by speed control of the ICE prime mover, Fig 1.2(a). However,
voltage transients are usually much faster than the control dynamics of ICEs and hence an
active rectification scheme is usually required and the ICE speed effectively fixed, as
shown in Fig. 1.2(b). This adds power electronic complexity, but gives additional control
functionality that will be discussed later in the thesis. Alternatively, the ICE can be used as
prime mover to a Hybrid Permanent Magnet (HPM) machine the output of which is
connected to a simple passive rectification stage, as illustrated in Fig 1.2(c) showing the
HPM machine (again 3-phase) in the power-train system schematic. The hybrid permanent
magnet machine is predominantly a permanent magnetic machine, but with some facility to
control its output open circuit voltage via a secondary field excitation source. Hence, the
hybrid terminology refers to the hybridisation of permanent magnet and wound field
excitation in one machine package.
(a) ICE/PM machine and passive power converter
(b) ICE/PM machine and active power
converter
converter
PM ICE
magnetic coupling
36
The HPM machine terminal voltage is converted to DC via a passive rectification stage and
the additional wound field excitation moderates that due to the permanent magnets. Fig.
1.3, shows the variation in rectified generator output voltage (VDC-link) due to the back
electromotive force (back-EMF) of both the permanent magnet (EMFPM) and wound field
(WF) excitation (±EMFWF) parts.
Fig. 1.3 Variation of generator rectified AC output due to wound excitation field.
HPM machines are a promising electric machine topology that have attracted some research
effort in recent years and are described in many technical papers and patent applications [14
- 38]. It is worthwhile mentioning that these machines are in the early stages of
development with no commercial product to-date [39]. There are many different topologies
presented in literature, however there is a need for more research into these types of
machine since their applications, and thus performance requirements, are varied. Further,
there is no detailed discussion into the design and related control issues for those machines
in hybrid EV power trains. This thesis aims to address these issues. The remainder of this
Chapter describes different HPM topologies listing their reported and unreported
advantages and disadvantages and sets out the scope of this research study.
EMFPM + EMFWF If > 0
0
37
1.2 Research Objectives
In this research study, the main aim is to establish the feasibility of using one of the HPM
machine topologies acting as a generator with a passive rectification stage. The primary
application area is in the power-train of a series hybrid electric vehicle where the concept
will be considered as an alternative to brushless PM machines interfacing to the vehicle
power-train via active power electronic converters. The electro-magnetic design of the two
main parts in the selected HPM generator topology and their individual system behaviour at
normal and rated conditions will be studied. Prediction of the transient and steady state
temperature in some of the HPM machine parts will be conducted based on commercial
thermal analysis software. Two HPM machine stator winding configurations, i.e. 3- and 9-
phase, with their relevant passive rectification stages will be analysed in terms of their
terminal and DC-link output power along with the quality of the generated DC output
voltage. As an extended investigation, detailed cycle-by-cycle analysis based on a vehicle
power-train load demand will be studied along with the design of the auxiliary generation
system in a HEV. The latter will be optimised to take account of excitation variation and
terminal constraints in the form of a passive rectification stage. If feasibility of the HPM
generator can be established, using the methods developed in this thesis, a major benefit of
the HPM generator with a passive rectification stage can be gained in terms of system costs,
since the active inversion stage accounts for 2/3 of the vehicle power-train cost [10].
The research work will be a mixture of simulation studies using electro-magnetic finite
element analysis (FEA), transient machine and system analysis via SimPower, a
MATLAB/SIMULINK toolbox set, along with test validation via a representative prototype
HPM generator configuration and its interface to an experimental electrical system
evaluation platform.
This research study has significance both fundamentally and practically since, as far as the
author is aware, there has been no clear investigation of the performance requirements for
HPM machines operating in a hybrid electric vehicle power-train. The author has
published four papers on the design, operation, and control of the selected HPM machine
topology with a passive rectification stage along with a comparison study between 3- and 9-
phase machine configuration systems. The novel 9-phase HPM machine design with a
38
passive rectification stage is capable of supplying the average energy requirements for a
road vehicle, but not the peak powers required for acceleration or recovery of regenerative
braking energy, with improved vehicle power-train robustness, reliability and reduced
power conversion costs compared to a 3-phase brushless PM generator with an active
rectification stage.
The design of the WF rotor of the HPM machine, including the back-EMF control
requirements has been conducted and analysed via FEA and thermal analysis software’s,
providing a methodology for designing future generator systems. In addition, several radial
flux HPM machine topologies have been compared for the same volume envelope in terms
of their maximum back-EMF when the DC field excitation is varied and permanent magnet
demagnetisation risk due to the total field weakening by the wound field excitation.
A representative 9-phase HPM machine prototype has been designed constructed (starting
from steel lamination sheet until the overall machine) and successfully tested via an
experimental system evaluation platform.
1.4 Thesis Organization
Chapter 1 discusses the HPM generator concept and reviewed published literature on
different HPM machine topologies, indicating briefly their operational philosophy,
advantages, and disadvantages.
Chapter 2, discusses the steps used to design the selected HPM machine topology via 2-D
FEA and Matlab/Simulink, along with the analytical derivation of an equivalent magnetic
circuit model of the chosen HPM machine topology. In addition, a general description of
the thermal network model is presented along with the technique that has been adopted to
obtain the PM and WF parts of the HPM machine thermal model. Finally, the selected
HPM machine with another three HPM machine topologies considered most likely to be
applicable for high volume automotive manufacture will be investigated by quantitatively
comparing their performance within the same volumetric constraints as that of a reference
PM only machine design.
A dynamic simulation model for 3- and 9-phase HPM generators with an associated
rectification scheme is developed in Chapter 3. The model is then used to perform a
39
back-EMF waveforms. Two 9-phase machines with different stator turns are investigated
in terms of their terminal and DC-link output power along with the quality of the generated
DC output voltage. In addition, sensitivity analysis based on machine synchronous
inductance for 3- and 9-phase machine configurations with varying load conditions and
zero DC wound rotor field excitation are conducted. Finally, a loss comparison between
the HPM and PM machines and their rectification systems for 3- and 9-phase machine
configurations is made. The outcome of this analysis was subsequently used to give some
conclusions concerning the viability of utilising an ICE/HPM generator system with a
passive rectification stage that might or might not be an adequate replacement of a more
traditional ICE/PM generator with an active power electronic converter in the power-train
of a hybrid electric vehicle.
The Matlab/Simulink model developed in Chapter 3 is expanded in Chapter 4 to include the
vehicle power-train load demand and thus used for detailed cycle-by-cycle analysis. To
consider the longer simulation times associated with the full battery discharge of a hybrid
electric vehicle, a simplified dynamic model for the HPM generator with voltage regulation
control is derived. The ICE/HPM generator output power control scheme that maintains
ICE efficiency at its optimal region is then modelled and options concluded. Finally,
several operating scenarios for the HPM generator excitation scheme are assessed to
minimise machine DC excitation losses.
Chapter 5 discusses experimental validation of the analysis undertaken in the proceeding
Chapters. HPM machine hardware, test rig components and measurement instrumentation
are presented. The developed test bench allows operating of the HPM machine at varying
load and speeds conditions while connecting to an active DC system typical of a vehicle
power-train. Experiments were carried out focusing on HPM generator system
performance and thermal behavior. Chapter 5 also includes a comparison of measured and
numerically computed results together with a discussion on the possible causes of
discrepancy between the two.
Chapter 6 concludes this research study by summarizing the main results, presenting
concluding observations and suggesting directions for future research.
40
1.5 Review of Published HPM Machine Topologies
For all HPM machines, there are at least two magnetisation sources, a PM field that
provides the air-gap with constant flux-linkage and a DC excited wound field that acts to
regulate or adjust the PM air-gap flux distribution either by strengthening or weakening the
field magnitude. For the machine to operate as a variable voltage generator the range of
air-gap flux density variation has to be designed to match the anticipated application
requirements, this will be discussed in later thesis Chapters. A number of HPM machine
topologies have been reported in literature in recent years. However, none of these
publications attempt to qualify or quantify the benefits of the reported or competing
topologies. The reported HPM machine topologies will be reviewed here and an
assessment made of each topology with a view of arriving at a preferred topology for study
in Chapter 2.
1.5.1 PM Synchronous Machine with Claw Pole Field Excitation (PSCPF)
Zhao and Yan discussed briefly the PSCPF machine components and the associated flux-
linkages paths in 1988 [14], the machine cross-section is illustrated in Fig. 1.4. The PSCPF
is composed of two parts, one called the main part and the other the assistant part. Both
parts of the machine share one common stator. Referring to Fig. 1.4, the assistant part is
composed of components 2-5; these represent the Claw-pole structure. The field winding is
placed on the stator, therefore slip rings and brushes are not required.
Fig. 1.4 Cross-sections of the permanent magnet synchronous machine with claw pole
field excitation (PSCPF) [14].
When current flows through the field winding (component 5), the magnetic path of the DC
flux is through the inner cylinder of component 3(axial); the bottom of component
1- shaft 2- N pole 3- magnetizer bracket 4- S pole 5- field winding 6- stator iron core 7- magnet 8- non-magnet 9- pole boot
δ1, δ2 adjunctive gap
1
δ2
3
7
10
8
41
3(radial); through the outer cylinder of component 3(axial); the air-gap δ1 (radial); plane
magnet pole (axial); the main air-gap δ (radial); stator iron core (radial); air gap δ (radial);
claw pole magnet pole 2(radial); magnetic shaft (axial); air-gap δ2 ( radial); inner cylinder
of component 3, as discussed in further detail in [14]. The magnetic path of the PM is
through the claw pole magnet pole; air-gap δ (radial); stator iron core; air-gap δ (radial);
claw-plane magnetic pole; PM (N pole); rotor iron core and PM (S pole), as illustrated in
Fig. 1.5.
Fig. 1.5 Flux path of the (PSCPF) machine as reported in [14].
Zhao and Yan discussed an improved PSCPF machine that they referred to as the hybrid
excitation synchronous generator (HESG) [14], as illustrated in Fig. 1.6. It is basically a
similar structure to that of the PSCPF the dissimilarity being that the latter has clapboard
inserts that are made of non-magnetic material. The clapboard introduces an air-gap and
thus reduces the coupling between the PM and wound field excitation making the two fields
independent of each other [14]. For both the PSCPF and HESG designs the PM and wound
field excitations act independently, i.e. they are magnetically in parallel.
(a) Axial section view (b) Radial section view
Fig. 1.6 Simplified construction figure of HESG as reported in [15].
1- rotating shaft 2- non-magnetic clapboard 3-winding holder 4- electrical magnetic winding 5- N side magnetic pole 6- magnetic shaft sleeve
7- S side magnetic pole 8- stator core
9- non-magnetic clapboard 10- permanent magnet poles
11- rotor core
Flux path due
to PM
42
In 2007, Chao-hui et al presented a study of a new HPM machine based on the HESG
topology called the hybrid excitation claw-pole synchronous generator (HECPSG) [16].
The structure of the HECPSG is shown in Figs. 1.7 and 1.8.
Fig. 1.7 A new type hybrid excitation claw-pole synchronous machine
(HECPSG) components [16].
Fig. 1.8 HECPSG machine assembly [16].
The stator of the HECPSG consists of polyphase windings. The claw poles of the rotor are
magnetised by a cylindrical wound coil and a cylinder shaped permanent magnet which is
axially magnetised. The flux under one pole pair consists of two parts, one is produced by
the permanent magnets and the other produced by the coil exciting current [16]. The
magnetic field from one claw-pole, passes through the air-gap and stator core and back to
another claw pole. No detailed discussion is given for the interaction between the PM and
winding fields, i.e. potential for demagnetisation, heating effects, reaction effects. Further,
(a) Magnet (b) Field coil (c) PM+field coil (d) Claw pole
(e) Rotor (f) Stator (g) Claw pole machine
Shaft
43
the contribution from each field source to the stator induced back-EMF is not discussed.
Table 1.1 summarises the advantages and disadvantages of the HECPSG topology.
Advantages Disadvantages
(2) The structure of claw pole is helpful to
arrange more magnet poles when the
rotor diameter is relatively small [16]
(3) Slip rings and brushes are not required
[16]
Note : Deduced by the author (*)
Reported in literature [16]
Table 1.1 HESPSG advantages and disadvantages.
1.5.2 Toroidal-Stator Transverse Flux Machine (TSTFM)
Spooner et al discussed hybrid excitation of AC and DC machines for rail traction and
engine mounted generators in 1989 [17]. Transverse-flux AC synchronous machines are
excited by means of a simple DC coil mounted on the stator, as shown in Fig. 1.9(a).
Consequently, they are naturally brushless, they are reported to have low rotor losses (since
the rotor has no permanent magnet poles) and they are mechanically suited to very high
speed. However, the authors do not consider high frequency losses that may occur in the
solid rotor poles.
The basic machine cross-section schematic is illustrated in Fig. 1.9(a), consisting of two
stator sections joined by a soft-magnetic outer casing and separated by the field coil. The
rotor has two similar sections, one in each stator section and mutually displaced in space, in
this case by 180 0 mechanical. Each rotor section has a salient structure, Fig. 1.9(b). The
field coil DC current establishes a set of north poles on rotor section 1 and a set of south
poles on the rotor section 2, as illustrated in Fig. 1.9(b). Each stator coil encloses both stator
core sections and experiences alternate north and south rotor poles as the rotor turns. The
flux-linking a stator coil is equivalent to that in a conventional radial field machine design
of half the total core length [17] since there are empty spaces between the rotor soft-
magnetic iron poles. A major problem for designers is the provision of sufficient magnetic
44
material to carry flux between the two rotor sections. Further, there is a substantial leakage
flux when the stator sections are faced by the large effective air-gap of the “empty” or high
reluctance rotor sections.
(c) Machine rotor with saliency and permanent magnets
Fig. 1.9 Transverse-flux machine components as reported by Spooner et al [17].
Fixing magnets in the empty spaces of each rotor section, as shown in Fig. 1.9(c), provides
a pole opposite to those established by the field winding and enhances the mechanical
rotational symmetry (balance). The flux that passes through the machine shaft due to the
permanent magnets is subtracted from that due to the excitation field current and so makes
possible a greater flux-per-pole for each rotor section. The required field current can thus
be reduced from the design of Fig. 1.9(b), and leakage flux is also reduced [17]. Thus,
transverse-flux machine arrangements appear to be an attractive option for small and
medium size generators [17].
Field coil
Stator winding
45
Fig. 1.10 Toroidal transverse-flux machine reported by Spooner et al [17].
Spooner et al presented a rotary toroidal version of the transverse-flux hybrid excitation
machine, based on the work of Evans and Eastham transverse-flux AC machine topology
[40]. The machine construction is illustrated in Fig. 1.10 showing a toroidally wound stator
core of multi-phase windings, DC field winding located inside the toroidal core and 2
rotating discs with alternate permanent magnet and soft-magnetic poles. The flux-linkage
paths throughout the machine parts due to both the PM's and stationary field coil are
illustrated in Fig. 1.11. If the two rotor poles are only provided by PM's the flux path can
be traced from one rotor plate containing North pole magnets, crossing the air-gap into the
toroidal stator, then travelling circumferentially across the second air-gap into the South
magnet pole on the opposite plate; through the plate into the shaft and back to the first plate
to close the loop at the North pole [18], as shown in Fig. 1.11(a). A modification to the
design of Fig. 1.11(a) has soft magnetic poles between the respective North and South PM
poles, as illustrated in Fig. 1.11(b) [17] and resulting in additional flux paths. Thus, flux
from the North pole on the right hand side plate crosses to the stator but then comes back to
the same rotor disc via the soft iron pole [18], as shown in Fig. 1.11(b). In this case flux
does not generally pass through the rotor shaft. However, during operation of the machine,
flux travels through both paths subject to reluctance variation in the shaft. Finally, there is
a third flux path due to the field excitation coil that drives flux through the rotor shaft, rotor
plate, iron poles, air-gap, stator and the second iron poles on the opposite disc [18], as
illustrated in Fig 1.11(b) and (c) for both strengthening and weakening modes respectively.
The toroidal transverse-flux machine configurations are brushless machines generating an
AC output that is modified by the DC field winding excitation current [19]. For both
transverse-flux topologies illustrated in Figs. 1.9 to 1.11, the main PM field and moderating
wound field are magnetically in parallel, their advantages and disadvantages being noted in
Table 1.2.
46
(a) Flux paths due to PM's alone; without rotor iron poles
(b) Flux paths due to both PM's and DC field excitation in strengthening mode; with
rotor iron poles.
(c) Flux paths due to both PM's and DC field excitation in weakening mode; with rotor
iron poles.
Fig. 1.11 Flux paths in the toroidal transverse-flux machine, as reported by Spooner et al
[17].
(2) The short axial length makes this
machine suitable for directly mounting to
an engine shaft replacing, in part, the
flywheel [17]
[17]
large, which necessitates relatively
toroidal is restricted by the machine
diameter [17]
Reported in literature [17]
topologies.
1.5.3 Hybrid Excitation Synchronous Machine (HESM)
Naoe and Fukami discussed the structure of a hybrid excitation synchronous machine
(HESM) in [20]. The machine has a conventional AC stator and a two part rotor
construction where each part is separated by an air-gap. One rotor part has PM excitation
and the other part wound field excitation. Each rotor part is independent of the other and,
in the case reported, is of radial field design. The HESM is illustrated schematically in Fig.
1.12. The flux produced by the field winding is designed not to pass through the PM's
because of their large reluctance, thus keeping the field winding MMF low [20]. Hence, the
machine air-gap flux can be modified by the field winding current direction and magnitude.
The PM and rotor wound field excitation sources are magnetically in parallel. Table 1.3
summarises the advantages and disadvantages of the HESM topology.
Fig. 1.12 Structure of the HESM with a two-part rotor [20].
Wound
(2) Short magnetic path [20]
(3) The air-gap flux can be easily
controlled by the field current [20]
(1) Slip rings and brushes exist, which
increases complexity and maintenance
Reported in literature [20]
Table 1.3 Advantages and disadvantages of the HESM.
1.5.4 Synchronous Permanent Magnet Hybrid AC Machine (SynPM)
The synchronous permanent magnet hybrid AC machine (SynPM) design was presented by
Xiaogang and Lipo in 2000 [21]. The machine is a combination of four PM poles and two
wound field excitation poles on the same rotor, as illustrated in Fig. 1.13. The PM poles
provide the major part of air-gap flux, while the wound field excitation poles act as a flux
regulator to adjust the air-gap flux distribution. By appropriate connection of the stator
coils, and rotor winding excitation, the net phase flux-linkage and hence back-EMF may be
weakened or strengthened. Considering one of the stator phase belt coils; the coil back-
EMFs for the three excitation modes are as shown in Fig. 1.14, while Fig. 1.15 illustrates
the corresponding open circuit flux lines due to positive, zero and negative DC field
currents. A phase belt is formed by connecting three coils of the same phase in series, as
shown in Fig. 1.13, thus the resulting phase back-EMF's for the cases of positive, zero, and
negative field winding current are as shown in Fig. 1.16. Slip-rings and brushes are
required for this machine topology. For the machine discussed, excitation produces around
67% of the total air-gap flux [21]. The flow of the flux is radial for both PM and DC field
windings, which are magnetically acting in parallel. Table 1.4 summarises the advantages
and disadvantages of the SynPM topology.
49
Fig. 1.13 Cross-section of the of SynPM machine reported by Xiaogang and Lipo,
showing one phase belt of the stator winding [21].
Fig. 1.14 Back-EMF of one coil of the phase belt winding [21].
Fig. 1.15 Flux lines of the six pole SynPM machine presented by Xiaogang and Lipo
[21].
current
Rotor
Stator
(b) Back-EMF of one circuit with zero excitation
(c) Back-EMF of one phase with full negative excitation
Fig. 1.16 Example of resultant coil and phase back-EMF for different field winding
excitation conditions [21].
Structure *
short magnetic paths. A high power
density is suggested but no data quoted *
(1) Slip rings and brushes exist [21]
(2) The combination of 4-pole or 2-pole
field flux in field weakening, with the
6-poles stator flux, will result in a
number of space and time harmonic
components and undesirable torque
pulsations and vibration [21]
might appear *
Reported in literature [21]
1.5.5 Consequent Pole Permanent Magnet Hybrid Excitation Machine (CPPM)
Tapia et al discussed a consequent pole permanent magnet hybrid excitation machine in
2001 [22]. The machine combines fixed PM excitation with variable flux via a field
winding fixed in the stator. The machine is similar to the Transverse-flux machine reported
by Spooner et al [17]. However, Tapia et al discussed a greater number of design options
and discussed the design in greater depth. The machine consists of a rotor divided into two
sections, each section having radially magnetised surface mounted permanent magnets
interleaved with laminated iron poles, as illustrated in Fig. 1.17(a). The magnetisation of
each rotor section is shifted 1-pole-pitch with respect with the other section.
The stator is composed of two laminated tooth sections inside a solid outer soft magnet
yoke. A conventional three-phase AC winding is located in slots around the periphery of
the inner stator diameter and a circumferential field winding is placed between the two
stator sections, as illustrated in Fig. 1.17(a). The field winding is excited by DC current.
For no field current the machine excitation is due to the rotor PM's alone and is essentially
radial, each PM linking with a consequent soft iron pole on the same machine half. When
excited with positive current flux generated by the field winding flows in a direction such
that it adds to the PM flux and the flux closes its path in the same half stator, as illustrated
52
in Fig. 1.17(b). If the field current is negative, the direction of the air-gap flux is as shown
in Fig. 1.17(c). Fig. 1.17(d) shows further views of the CPPM components.
The stator and rotor yokes provide a low reluctance path for the axial flux, which is
considered an important feature of the machine operation. The current of the field winding
is externally controlled in order to provide variable excitation.
(a) Magnetic structure of the CPPM machine
[23]
(d) 3 kW CPPM
(c) Demagnetising effect of the field flux
Fig. 1.17 Consequent pole PM hybrid excitation machine (CPPM) reported in [22] and
[23].
Laminated
1.5.6 Field Controlled Torus-NS (FCT-NS) Machine
Aydin et al discussed an axial flux machine in 2002, designed to improve the flux
weakening operation of the previously reported axial flux, toroidal PM machines [24]. The
machine is essentially an axial field version of the CPPM and was referred to as the Field
Controlled Torus-NS (FCT-NS) machine. The machine construction consists of 2 outer
rotor discs carrying axially magnetised permanent magnets alternatively placed with slotted
magnetic iron pole pieces. There are two slotted stator cores, an inner and outer core,
realized by tape wound laminations inserted with polyphase AC windings and a DC field
winding between the stator inner and outer cores, as illustrated schematically in Fig. 1.18.
Variations on the FCT-NS design were presented by Lipo and Aydin in 2004 [25] and [26].
(a) Machine components (b) Stator assembly (c) Rotor assembly
Fig. 1.18 Field controlled Torus-NS machine (FCT-NS) [24, 27].
Fig. 1.19 shows the main flux direction of a two-pole portion of the FCT machine at the
average diameter [24] (a); rotor flux directions (b); air-gap flux directions (c); and operating
principle of the FCT machine (d) for zero (i), positive (ii) and negative (iii) field current.
Fig. 1.19 (e) shows the FCT stator and rotor components.
Fig. 1.20 illustrates schematics of the single-rotor-single-stator topology (a) the NN and NS
types, double-rotor-single-stator (b) and (c); double-stator-single-rotor (d) and multi-stage
(e) concepts.
Fig. 1.21 illustrates hardware of the NN type FCAFPM machine as reported in the literature.
The CPPM and variants are all parallel permanent magnet and wound field magnetic
Magnets
54
designs. Table 1.5 summarises the advantages and disadvantages of the CPPM and variants
as reported in [22] to [27].
(a) Main flux direction of the FCT
machine [24]
(b) Rotor flux directions [24]
(c) Air-gap flux directions [24]
(d) Operating principle [27] (e) FCT rotor and stator components [25]
Fig. 1.19 Field Controlled Tours-NS type (FCT-NS) [24, 25, 27].
(i) No field current
(ii) Positive field current
(iii) Negative field current
(i)
single-stator
Fig. 1.20 Reported combinations of the FCAFPM machines [25].
(a) Stator view pre-
Fig. 1.21 NN type FCAFPM machine reported in [25, 26].
56
without affecting the magnetization
characteristics of the PM's.
can be obtained with a low DC
excitation field Ampere-turn
(1) Additional DC winding in the stator
reduces the power density, such that the
additional air-gap surface associated to
this winding dose not participate in the
energy conversion process.
Table 1.5 Advantages and disadvantages of CPPM and variants as reported in [22] to [27].
1.5.7 Dual-Rotor Machine
Amara et al proposed a dual-rotor machine in 2001 that is comprised of two rotors placed
together (one wound and the other with PM's) inside the same stator assembly, as shown in
Fig. 1.22. The design employs juxtaposed magnetic circuits that, according to the authors,
avoids the risk of PM demagnetisation [28]. Flux weakening is achieved via excitation of
the wound rotor to create a flux opposite to that created by the rotor PM's [28]. The design
is similar in form to the HESM presented in Section 1.5.3 [20], but having slightly different
rotor topologies.
Stator
winding
1.5.8 Imbricated Hybrid Excitation Machine (IHEM)
Amara et al also proposed an imbricated hybrid excitation machine (IHEM), as illustrated
in Fig. 1.23. The rotor is composed of two magnetically isolated parts, one containing the
PM excitation, and the other is used to direct flux created by an excitation coil that is
located on either the rotor or the stator, the latter case avoiding all sliding contacts. The
stator is composed of two identical parts linked by a yoke, as shown in Fig. 1.23(a). The
main goal of this design was to ensure that the flux created by the excitation winding does
not pass through the PM hence the possibility of demagnetisation is greatly reduced [28].
(a) Machine cross-section
(b) Rotor structure
Fig. 1.23 Imbricated hybrid excitation machine (IHEM) [28].
Furthermore, Vido et al proposed two improved versions of the IHEM in 2005 [29], the (i)
Homopolar and (ii) Bipolar hybrid excitation synchronous machines, HHESM and BHESM
respectively, as illustrated in Fig. 1.24. Cross-sectional schematics of both prototypes are
shown in Fig. 1.24. The rotors consist of three parts, one a solid core, one part laminated
core and a set of permanent magnets. The schematics show an axial cut of the stator and
rotor for both prototypes which are 6 pole-pair. The two machine rotors have the same
dimensions. By comparing the two topologies, it can be observed that the lateral permanent
magnets are not present in the BHESM prototype [29].
Flux path due
(a) First prototype machine (HHESM) (b) Second prototype machine (BHESM)
Fig. 1.24 Homopolar and bipolar hybrid excitation synchronous machines [29].
(iii) Prototype stator and rotor
(ii) Prototype rotor details
stator core
Solid end
Solid rotor core
59
The various flux paths created by excitation coils, lateral PMs (side magnets) and azimuth
PMs, may be divided into two categories, homopolar and bipolar flux paths. The
homopolar flux path represents a flow of flux through machine parts in axial and radial
directions. The bipolar design has flux paths in either radial or axial direction. Therefore,
the flux generated by the field DC coils has only one path, which is homopolar in nature, as
shown in Fig. 1.25(a). Moreover, the homopolar path for the lateral PMs, can be observed
in Fig. 1.25(a). The flux generated by the PMs has two distinct paths, one of which is
bipolar, as shown in Fig. 1.25(b) and (c), which creates north and south poles under the
active parts [30]. The flux path generated by the azimuth PM's is primarily oriented
perpendicular to axial direction of the machine [30]. A portion of the flux generated by the
lateral PM's is oriented in the axial direction of the machine via the rotor flux collector, as
shown in Fig. 1.25(c).
(a) Homopolar flux path due to DC coils (b) Bipolar flux path due to Azimuthal
magnets
(c) Bipolar flux path due to PM's (d) Homopolar flux path due to PM's
Fig. 1.25 HHESM various flux paths due to deferent excitations [30].
External yoke
End shields
Side magnet
collector
60
In other words, the fluxes created by either the PM's or the wound field excitation that
exhibit a homopolar path only gives rise to one type of pole (either north or south),
depending on the direction in which the magnets are magnetised and the polarity of current
in the DC field coils [30]. Flux only passes once through the air-gap under the active part,
then it returns first via the stator end shields and then via the rotor flux path, as illustrated in
Fig. 1.25(d) [30].
created by a rotor PM (c) Leakage flux path
created by PM
Fig. 1.26 BHESM various flux paths created by different excitations [29].
Fig. 1.26(a) shows flux paths created by the DC field coils for the BHESM design. This
bipolar configuration passes through two annular excitation coils. Each coil acts in one
polarity of pole [29]. The flux created by an excitation coil goes through active parts and an
air-gap (homopolar path). Fig. 1.26(b) shows the bipolar flux path created by PMs, where
this bipolar flux passes through active parts and air-gap, creating north and south poles.
Fig. 1.26(c) shows the PM leakage flux path, which is not through the active parts and
hence does not contribute to torque production [29].
Fig. 1.27 shows homopolar flux paths created by PMs, as reported in [29]. For homopolar
hybrid excitation machines the total flux passing through the stator windings exhibit a DC
component, while for bipolar hybrid excitation machines the total flux passing through the
armature windings does not have a DC component [29].
Excitation coils
Flux paths
Flux path
Permanent magnets
Permanent
magnets
61
Thus, although air-gap flux control is effective for both the HHESM and BHESM
machines, the DC current excitation efficiency is better for the HHESM because of the
solid rotor core parts [29]. For the HHESM operating with enhanced excitation flux,
magnetic saturation occurs when the magnetic pole in which the DC excitation is acting is
saturated, even if the other pole is still not saturated [30]. However, for the BHESM,
magnetic saturation occurs only when both magnetic poles are saturated, from which the
authors conclude that the BHESM has a wider excitation flux variation [30]. The efficiency
of the hybrid excitation is better for the HHESM than it is for the BHESM design because
of the leakage flux path, shown in Fig. 1.26(c) [29], which does not contribute to torque
production. Table 1.6 summarises the advantages and disadvantages of the IHEM topology.
Advantages Disadvantages
[28]
reduced [28]
[28]
Leakage *
Reported in literature [28]
(a) First homoplar path (b) Second homoplar path
Fig. 1.27 Homopolar paths of fluxes created by PM [29].
Homopolar flux
1.5.9 Series Double Excited Synchronous Machine (SDESM)
Fodoren et al presents the series double excited synchronous machine (SDESM) that has
series excitation circuits [31, 32]. The parallel excitation circuit reported in some of the
previously presented topologies suffer from the drawback of construction complexity [31].
The main advantage of the SDESM appears in applications where the electric drive
operates under partial loads for most of the time [31]. Fodoren et al presented a design
procedure, prototype and test results for a SDESM design in [31].
(a) SDESM cross section basic principal (b) SDESM cross section actual design
Fig. 1.28 SDESM design presented by Fodoren et al [31].
The proposed SDESM design has the field excitation winding fixed on the rotor in a series
magnetic configuration with the surface PMs, as shown in Fig. 1.28. The stator is that of a
commercial induction motor, while the rotor was constructed as shown in Fig. 1.29(a). The
3-phase stator winding is single-layer, with three slots-per-pole-per phase [31]. Table 1.7
summarises the advantages and disadvantages of the SDESM topology as reported in [31].
Des
Dis
Wrs
demagnetization risk
operating region
excitation field
(a) SDESM rotor [33] (b) Test bench [31]
Fig. 1.29 Hardware of SDESM.
1.5.10 Switch Reluctance Machine with Stator Field Assistance
In 2006, Afjei et al presented a new configuration of switch reluctance machine with stator
field assistance that represents a hybrid generator topology, albeit with no PM's, as
illustrated in Fig. 1.30 [34, 35]. This machine design was intended to be utilised in hybrid
vehicle motor/generator unit. The proposed hybrid machine consists of two stator and two
rotor sections placed on both sides of the field coil assembly, as illustrated in Fig. 1.30(b)
[34]. Here, the magnetic flux produced by the field coil travels through the guide and shaft
to the rotor poles and then to the stator poles, finally closing through the motor housing [34].
Feeding
ring
PM
pieces
Excitation
coils
DC
machine
DESM
64
[35]
Fig. 1.30 Cross-section and actual switch reluctance motor with field assistance.
A variant of the Afjei design that incorporate PMs was presented by Chau et al for small
wind applications [36]. This hybrid design has a unique structure which, which it is
claimed, contributes to simplified mechanical manufacturing and magnetic fixing. The
machine design is illustrated in Fig. 1.31 [36]. Due to the extra air-bridge that is in parallel
with each PM, an amplification of the effect of the flux in the reinforcing mode is achieved,
as illustrated in Fig. 1.31(b), where if the field winding MMF is opposing the PM MMF.
The PM flux leakage will increase causing an amplification of the effect of the flux
weakening, as shown in Fig. 1.31(b) [36]. Thus, as with a proper design of the air-bridge
width, a wide flux regulating range can be obtained by virtue of a small DC field excitation
[36]. The PM and rotor wound field excitation sources are magnetically in series. Table
1.8 summarises the advantages and disadvantages of the PM brushless hybrid generator
topology.
Motor
housing
Field
coils
Stator
Rotor
(c) Prototype generator
Fig. 1.31 Cross-section, magnetic field distributions and actual prototype PM brushless
hybrid generator [36].
Stator Rotor Permanent
small DC field excitation [36]
(2) The rotor has neither PM's nor field
windings which offers high mechanical
integrity [36]
[36]
Demagnetisation *
Saturation *
Leakage *
Reported in literature [36]
Table 1.8 PM brushless hybrid generator advantages and disadvantages.
1.5.11 Novel Dual-Stator Hybrid Excited Synchronous Wind Generator (DSHESG)
In 2009, Liu et al presented a novel dual-stator hybrid excited synchronous wind generator
(DSHESG), as illustrated in Fig. 1.32 [37]. The proposed generator stator is composed of
outer stator, inner stator and field winding. The rotor consists of the PMs, claw poles, rotor
yoke, and cup rotor [37]. There are two independent parallel magnetic circuits in the
DSHESG due to PMs and the DC excitation coil. Here, the series PM magnetic circuit
consist of PMs, air-gap, cup rotor and laminated stator core. The rotor series magnetic
circuit consist of claw poles, air-gap, laminated core of the outer stator and bracket of the
field winding. It is claimed that this topology overcomes some of the previous hybrid
machine topologies weakness, for example, the benefit of the two independent magnetic
circuits which reduces the leakage flux problems, and there is a reduced risk of PMs
demagnetisation. The PM and rotor wound field excitation sources are magnetically in
parallel. Table 1.9 summarises the advantages and disadvantages of the DSHESG
topology.
67
(a) Structural 3-D FEA model (b) Structure cross section
(c) Magnetic circuit of PM (d) Magnetic circuit of DC field winding
(e) Actual prototype
Fig. 1.32 Detailed structure and magnetic circuits of DSHESG [37].
Advantages Disadvantages
[37]
Note : Deduced by the author (*)
Reported in literature [37]
Bracket of
Field winding
Field winding
1.6 Summary of Literature Review
Published work in the field of HPM machines started around 1989. For all of the reviewed
HPM machine topologies, there are at least two excitation field sources that provide the net
machine excitation. In general, a permanent magnet (PM) source provides the main
excitation and a wound field component acts to regulate the machine flux distribution either
by boosting or weakening the PM field depending on the direction of the wound field DC
excitation current. The DC field winding may be placed on the same part of the machine as
the PM’s [20, 21, 31, 38] which necessitates slip rings and brushes or an exciter, or on the
stator [15 - 20, 22, 24, 27, 29, 35 - 37].
The review of previous publications has highlighted a number of notable issues. The
published HPM machine designs are, in general, complicated and have particular
weaknesses in their design for example, excessive flux paths that lead to high leakage
(magneto-motive force loss) [19, 24, 31] and PM demagnetisation [31]. Table 1.10 collects
data for the different hybrid excitation machines reported in the literature review. Note that
Table 1.10 is not complete due to the lack of some information in the published papers.
The open circuit back-EMF regulation capability that has been reported for some of the
HPM machines designs are generally based on experimental or finite element analysis
(FEA) results. The variation of open circuit back-EMF due to the machine DC field current
of the reported HPM machine topologies varies from 42 to 175% relative to the machine
back-EMF with zero DC field excitation. However, Table 1.10 does not give a clear
picture regarding the best machine topology for vehicle application, thus in order to get an
adequate and fair comparison the machine mass, volume and thermal limits should be
considered along with their performance and back-EMF regulation capabilities.
Furthermore, the power rating of some of the actual HPM machine prototypes discussed in
the literature ranged from 0.65 to 10 kW, as detailed in Table 1.10. Some of the
publications highlighted the area of interest for HPM machine topologies for example,
traction, wind power and vehicle systems. However, as far as the author is aware, there has
been no clear investigation of the performance requirements for HPM machines operating
in a hybrid electric vehicle power-train prior to this thesis study.
69
Topology
Novinschi
[18]
+ 30
+ 100
+ 3 -
Vido
[29]
[30]
Fodoren
[31]
Switch
reluctance
motor
Vehicles
Afjei
[34]
[35]
PM
brushless
hybrid
generator
- -300
FEA (*)
Table 1.10 Main particulars of HPM machines presented in research publications.
70
2.1 Introduction
The multiplicity of machine designs reviewed in Chapter 1 create a certain level of
difficulty when selecting the most appropriate HPM machine design topology since each
author declared some degree of novelty, but did not reference their design against any
benchmark solution. Throughout this PhD research study there has been no fixed specific
application targeted for the HPM generator design. However, prior research [1, 10] had
identified basic performance specification requirements for the electrical and
interconnection attributes of the machine. It was also anticipated from the previous
experiences of the author’s supervisor, that many future HPM machine applications would
be direct engine-mounted having a large outer diameter to active axial length aspect ratio
and possible facility for some through-shaft element (for flywheel or multiple geared
outputs) necessitating an essentially “donut” shaped volume envelope constraint [41 - 44].
Consequently, an available 3 kW, surface magnet mounted, brushless permanent magnet
(PM) machine having the above volumetric attributes was chosen as a benchmark PM
design from which to develop a suitable HPM generator for this thesis study since the
average power demand of 3 kW is typical of small 1.0 to 1.2 tonne urban vehicles [10].
This Chapter discusses the analysis of the benchmark brushless PM machine and testing
thereof to validate the calculation models and procedures. The developed tools were then
used to analyse and design wound field (WF) rotors within the same stator constraints of
71
the brushless PM machine, from which a preferred rotor design solution was chosen. Thus,
a machine that combined the brushless PM and chosen WF design features was proposed.
This HESM structure, as discussed in Chapter 1 (Section 1.5.3) was compared along with
three other HPM machine topologies (all designed around the benchmark brushless PM
machine) to attempt a comparative analysis of the four competing topologies. Conclusions
are presented and the chosen HPM machine design solution subsequently adopted for
further study in Chapters 3 to 5.
The benchmark brushless PM machine is illustrated in Fig. 2.1, showing the stator and rotor
end view (a), cross-sections (b) and a end view photograph of the complete machine
assembly. Fig. 2.2 defines the major dimensions of the machine the parameter values of
which are detailed in Table 2.1. Fig. 2.3 illustrates the benchmark PM machine stator and
phase A winding scheme, details of which are given in Table 2.2.
In this Chapter, one of the previously discussed HPM machine topologies has been
selected, and an analysis process for some of its parts have been undertaken, as will be
explained in the following sections. The selected HPM machine topology consist of two
different synchronous machines, PM and WF machines coupled on the same rotor shaft,
which will be analysed in terms of their machine geometry, excitation field technique,
back-EMF per coil, back-EMF per phase and developed electromagnetic torque (Td) for
both no-load and on-load characteristic cases.
2.2 Magnetic