High Speed Opamp Feedback

7/29/2019 High Speed Opamp Feedback

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SECTION 1

HIGH SP EED OP ER ATIONAL AMP LIFIER S

Wa l t Kes ter

INTRODUCTION

High speed a na log signa l processing applicat ions, su ch as video and

communications, require op amps which have wide bandwidth, fast settling time,

low distort ion a nd n oise, high out put curr ent , good DC perform an ce, an d opera te a t

low su pply voltages. Th ese devices a re widely used as gain blocks, cable dr ivers,

ADC pr e-amps, cur ren t-to-voltage converter s, etc. Achieving higher ban dwidths for

less power is extremely critical in today's porta ble and bat ter y-operat ed

communications equipment. The rapid progress made over the last few years in

high-speed linea r circuits h as hinged n ot only on th e developmen t of IC processes

but also on innovative circuit topologies.

The evolut ion of high speed pr ocesses by using am plifier ba ndwidth as a function of

supply curr ent as a figure of mer it is shown in Figur e 1.1. (In th e case of dua ls,

tr iples, an d qua ds, the cur ren t per am plifier is used). Ana log Devices BiFET pr ocess,

which produced the AD712 and OP 249 (3MHz ban dwidth, 3mA cur ren t), yields

about 1MHz per mA. The CB (Complemen ta ry Bipolar ) process (AD817, AD847,

AD811, etc.) yields about 10MHz/mA of supply cur ren t. F t's of the CB pr ocess PNP

tra nsistors are a bout 700MHz, an d the NP N's about 900MHz.

The lat est gener at ion complemen ta ry bipolar process from Ana log Devices is a

high speed dielectr ically isolated process called XFCB (eXtr a Fa st Complemen ta ry

Bipolar). This process (2-4 GHz Ft m at ching PNP a nd NP N tr an sistors), coupled

with inn ovat ive circuit topologies allow op a mps to achieve n ew levels of cost-effective perform an ce at ast onishin g low quiescent curr ent s. The appr oximat e figure

of merit for th is process is t ypically 100MHz/mA, alth ough t he AD8011 op a mp is

capable of 300MHz ban dwidth on 1mA of supply curr ent due t o its un ique two-sta ge

current-feedback architecture.


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CB:1

0MHz

PERm

A

AMPLIFIER BANDWIDTH VERSUS SUPPLY CURRENTFOR ANALOG DEVICES' PROCESSES

SUPPLY CURRENT (PER AMPLIFIER), mA

BANDWIDTH(MHz)

1000

300

100

30

10

3

1

0.3 1 3 10 30

AD8001

AD8011

AD811

AD847AD817

OP482 AD712OP249

741

1.1

XFCB

:100M

HzPERmA

BiFET

:1MH

zPER

mA

a

In order to select in telligent ly the corr ect op am p for a given a pplicat ion, a n

un derst an ding of the var ious op am p topologies as well as t he t ra deoffs between

th em is r equired. Th e two most widely used t opologies ar e voltage feedback (VFB)

an d curr ent feedback (CFB). The following discussion tr eat s each in det ail an d

discusses t he similar ities and differen ces.

VOLTAGE F EEDBACK (VF B) O P AMP S

A voltage feedback (VFB) op am p is dist inguished from a curr ent feedback (CFB) op

am p by circuit topology. The VFB op am p is certa inly the m ost popular in low

frequency applicat ions, but th e CFB op amp h as some advan ta ges at h igh

frequencies. We will discuss CFB in det ail later , but first t he m ore t ra ditiona l VFB

architecture.

Ea rly IC voltage feedback op a mps wer e ma de on "all NP N" processes. These

processes were optimized for NPN tr ansistors, and the "latera l" PNP tra nsistors h ad

relatively poor performance. Lateral PNPs were generally only used as current

sources, level shifters, or for other non-critical functions. A simplified diagram of a

typical VFB op am p ma nu factu red on such a pr ocess is shown in Figur e 1.2.


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1.2

VOLTAGE FEEDBACK (VFB) OP AMPDESIGNED ON AN "ALL NPN" IC PROCESS

a

vOUT

VBIAS

IT

-VS

IT

2

CP

RT

IT

2

Q2

Q3

Q1

+VS

"LATERAL" PNP

IT

Q4"gm"STAGE

A

i = vgm

v

+

-

IT

2

gm = Ic q

kTvOUT = = v @ HF

i

jCP

gm

jCP

The inpu t st age is a differen tial pa ir consistin g of eith er a bipolar pair (Q1, Q2) or a

FE T pair. This "gm" (tra nscondu ctan ce) stage convert s th e sma ll-signal differen tial

input voltage, v, into a curr ent , i, i.e., it's tr an sfer fun ction is mea sur ed in un its of

condu ctan ce, 1/, (or m hos). The sma ll signal emit ter resista nce, re, i sappr oximat ely equa l to the r eciprocal of th e sma ll-signal gm. The formu la for t he

sma ll-signal gm of a single bipolar tr an sistor is given by th e following equa tion:

( )gmre

q

kTIC

q

kT

I T= = =

1

2, or

gmmV

I T

1

26 2.

IT is the differential pair tail current, IC is th e collector bias curr ent (IC = IT/2), q is

the electr on char ge, k is Boltzmann 's consta nt, a nd T is absolute tem perat ur e. At

+25C, VT = kT/q= 26mV (often called th e Ther ma l Volta ge, VT).

As we will see short ly, th e am plifier un ity gain-bandwidth product, fu , is equa l to

gm/2Cp, where th e capa cita nce Cp is used t o set t he domina nt pole frequ ency. Forthis reason, the ta il cur rent , IT,is ma de proportional to absolute tem perat ure

(PTAT). This cur rent tr acks the variation in r e with temperatur e thereby making

gm independent of tempera tur e. It is relatively easy to make Cp r easonably

constant over temperature.

The outpu t of one side of th e gm stage drives the emitt er of a later al PNP tra nsistor

(Q3). It is importa nt to note th at Q3 is not u sed to am plify the signa l, only to level


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shift, i.e., the signa l cur ren t va riat ion in th e collector of Q2 app ear s a t t he collector

of Q3. The outpu t collector cur ren t of Q3 develops a voltage a cross high impeda nce

node A. Cp sets t he domina nt pole of th e frequency response. Emit ter follower Q4

provides a low impeda nce out put .

The effective load a t t he h igh impedan ce node A can be repr esent ed by a resista nce,

RT, in pa ra llel with th e domina nt pole capa cita nce, Cp. The small-signal outpu t

voltage, vout , is equa l to the sma ll-signal curr ent , i, mu ltiplied by the impeda nce ofth e par allel combina tion of RT and Cp.

Figur e 1.3 shows a simple model for th e single-sta ge amplifier a nd t he

corr espondin g Bode plot. Th e Bode plot is cons tr ucted on a log-log scale for

convenience.

1.3

MODEL AND BODE PLOT FOR A VFB OP AMP

a

6dB/OCTAVE

f = UNITY GAIN FREQUENCYu

f

f = CLOSED LOOP BANDWIDTHCL

1 +

1

R2

R1

NOISE GAIN = G

= 1 +R2

R1

i = v gm

X1

R1 R2

v

vin

+

-

gm

+

- CP RT

OAOf

vOUT

f1

=2R CO PT

fu

fCL

fu

=

=

gm

2C P

fuG

R1

R2=

1 +

The low frequ ency breakp oint , fo,is given by:

foR TCp

=1

2.

Note tha t th e high frequency response is deter mined solely by gm and Cp:

vout vg m

j Cp=

.

The unity gain-bandwidth frequency, fu , occurs where | vout | =| v| . Solving t he

above equat ion for fu ,assuming | vout | =| v| :


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fug m

Cp=

2.

We can use feedback t heory to derive th e closed-loop r elationship bet ween t he

circuit's signal inp ut voltage, vin ,and it 's out put voltage, vout :

vout

v in

R

R

j Cp

gm

R

R

=+

+ +

12

1

1 12

1

.

At th e op am p 3dB closed-loop bandwidt h frequ ency, fcl, the following is true:

21

2

11

fclCpgm

R

R+

= , and hen ce

fclgm

Cp R

R

=+

2

1

12

1

, or

fclfu

R

R

=+1

2

1

.

This demonstr at es a funda menta l property of VFB op amps: T he closed-loop

band width m ultiplied by the closed-loop gain is a constan t, i.e., the VFB op am pexhibits a constant gain-bandwidth product over most of the usable frequency range.

Some VFB op amps (called de-compensa ted) are un stable at u nity gain and a re

designed to be opera ted a t some m inimum am ount of closed-loop gain. F or th ese op

am ps, the gain-ban dwidth pr oduct is still relat ively const an t over t he r egion of

allowable gain .

Now, cons ider t he following typical example: IT = 100A, Cp = 2pF. We find t ha t:

gmI T

VT

A

mV

= = =/ 2 50

26

1

520

fugm

CpMH z= =

=

2

1

2 520 2 10 12153

( )( ).

Now, we mu st consider th e lar ge-signal resp onse of the circuit. The s lew-ra te, SR, is

simply the total available cha rging curr ent, IT/2, divided by the dom in ant pole

capacitance, Cp. For the example under consideration,


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I Cdv

dt= ,

dv

dtSR= , SR

I

C=

SRI T

Cp

A

pFV s= = =

//

2 50

225

.

The full-power ba ndwidth (FPBW) of the op am p can n ow be calculat ed from t he

formula:

FPBWSR

A

V s

VMH z= =

=

2

25

2 14

/

, ,

wher e A is the pea k am plitude of th e out put signal. If we assu me a 2V peak-to-peak

out put sinewave (cert ainly a r easona ble assum ption for high speed applicat ions),

then we obtain a FPBW of only 4MHz, even though the small-signal unity gain-

ban dwidth product is 153MHz! For a 2V p-p outpu t sin ewave, distortion will begin

to occur much lower th an th e actua l FPBW frequency. We mu st increase th e SR by

a factor of about 40 in order for t he F PBW to equa l 153MHz. The only way to do th is

is to increase th e ta il cur rent , IT,of th e input differen tial pa ir by the sa me factor.This implies a bias curr ent of 4mA in order to achieve a FP BW of 160MHz. We ar e

assu ming th at Cp is a fixed valu e of 2pF a nd can not be lowered by design.


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VFB OP AMP BANDWIDTH AND SLEW RATE CALCULATION

nn Assume that IT = 100A, Cp = 2pF

nn gmIc

VT

A

mV== == ==

50

26

1

520

nn

fugm

Cp

MHz== ==

2

153

nn Slew Rate = SR =

ITCp

V s/

/2

25==

BUT FOR 2V PEAK-PEAK OUTPUT (A = 1V)

nnFPBW

SR

AMHz== ==

24

nn Must increase IT to 4mA to get FPBW = 160MHz!!

nn Reduce gm by adding emitter degeneration resistors

a 1.4

In pr actice, th e FP BW of th e op am p should be appr oximat ely 5 to 10 times t he

ma ximum outp ut frequency in order t o achieve acceptable distort ion per form an ce

(typically 55-80dBc @ 5 t o 20MHz, but actu al s ystem requ iremen ts var y widely).

Notice, however, that increasing the tail current causes a proportional increase in

gm an d hence fu . In order t o prevent possible insta bility due to th e large increase in

fu , gm can be reduced by inserting resistors in series with the emitters of Q1 and Q2

(this t echn ique, called emitt er degeneration , also serves t o linear ize the gm transfer

function an d lower distort ion).

A major inefficiency of conventional bipolar voltage feedback op amps is their

inability to achieve high slew rat es without pr oport iona l increa ses in quiescent

curr ent (assum ing tha t Cp is fixed, an d ha s a reasonable minimum value of 2 or

3pF). This of cour se is not m ean t t o say th at high speed op amps designed using th is

ar chitectur e a re deficient, it 's just tha t ther e ar e circuit design techniques available

which allow equivalent per form an ce at lower quiescent curr ent s. This is extrem ely

importa nt in portable batt ery operat ed equipment wh ere every milliwatt of power

dissipation is critical.

VF B O p A m p s D e s ig n e d o n C o m p l e m e n t a r y B i p o la r P r o c e s s e s

With th e advent of complemen ta ry bipolar (CB) processes having high qua lity PNP

transistors as well as NPNs, VFB op amp configurations such as the one shown in

th e simplified diagram (Figur e 1.5) becam e popula r.


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1.5

VFB OP AMP USING TWO GAIN STAGES

a

IT

CP

Q3

Q2Q1

+VS

Q4

X1

-VS

+

D1

-

OUTPUT

BUFFER

Notice th at th e input differen tial pa ir (Q1, Q2) is loaded by a curr ent mirr or (Q3 and

D1). We show D1 as a diode for simplicity, but it is actually a diode-connected PNP

tr an sistor (mat ched to Q3) with t he ba se an d collector conn ected to each oth er. This

simplification will be used in many of the circuit diagrams to follow in this section.

The common emitter transistor, Q4, provides a secondvolta ge gain st age. Since the

PNP tr ansistors ar e fabricated on a complementa ry bipolar pr ocess, they ar e highquality and m atched to th e NPN s an d suita ble for voltage gain. The dominan t pole

of the a mplifier is set by Cp, and th e combinat ion of th e gain st age,Q4, and Cp is

often referr ed to as a Miller Integra tor. The unit y-gain outpu t buffer is usu ally a

complemen ta ry em itter follower.

The model for t his two-sta ge VFB op am p is shown in F igure 1.6. Notice tha t t he

unity gain-bandwidth frequency, fu , is still determ ined by the gm of the input st age

and the dominant pole capacitance, Cp. The second gain stage increases the DC

open-loop gain, but the maximum slew rate is still limited by the input stage tail

curr ent: SR = IT/Cp .


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1.6

MODEL FOR TWO STAGE VFB OP AMP

a

X1

R1

v

vin

+

-gm

+vout

VREF

R2

CP

-

IT

i = vgm

a

+

-

fu = fCL =gm

2CP

fu

R2

R1

SR =IT

CP

1+

The t wo-stage topology is widely used t hr oughout th e IC indu str y in VFB op am ps,

both precision a nd h igh speed.

Another popular VFB op am p ar chitectur e is the folded cascode as shown in Figure

1.7. An in dust ry-stan dar d video am plifier family (th e AD847) is based on t his

ar chitectu re. This circuit t akes a dvanta ge of the fast P NPs available on a CBprocess. The differen tial signa l cur ren ts in th e collectors of Q1 and Q2 a re fed to th e

emitters of a P NP cascode tra nsistor pa ir (hence the t erm folded cascode). The

collectors of Q3 and Q4 a re loaded with th e cur ren t m irr or, D1 an d Q5, an d Q4

provides voltage gain. This single-sta ge architectur e uses t he junction capa cita nce at

th e high-impeda nce node for compen sat ion (an d some variat ions of th e design bring

this node to an extern al pin so that additiona l extern al capacita nce can be added).


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1.7

AD847-FAMILY FOLDED CASCODE SIMPLIFIED CIRCUIT

a

Q2Q1

+VS

X1

-VS

+

-

Q5

Q4Q3

2IT 2IT

2IT

IT

IT

VBIAS

CCOMP

CSTRAY

D1

AC GROUND

With no emitter degenerat ion r esistors in Q1 an d Q2, and n o additiona l extern al

compensating capacitance, this circuit is only stable for high closed-loop gains.

However, un ity-gain compen sat ed versions of th is fam ily ar e available which ha ve

the a ppropriat e amount of emitter degenerat ion.

The a vailability of J FE Ts on a CB process allows not only low input bias curr ent but

also improvement s in t he t ra deoff which must be made between gm and IT foun d in

bipolar inpu t st ages. Figur e 1.8 shows a simplified diagra m of the AD845 16MHz op

amp. J FE Ts have a much lower gm per mA of ta il curr ent t ha n a bipolar tr an sistor.

This allows the input tail current (hence the slew rate) to be increased without

ha ving to increase Cp to mainta in sta bility. The un usua l thing about t his seemingly

poor per forma nce of the J FE T is tha t it is exactly wha t is needed on the inpu t st age.

For a typical J FE T, the value of gm is appr oximat ely Is/1V (Is is the source current),

ra ther than Ic/26mV for a bipolar t r ansis tor , i.e ., abou t 40 t im es lower . Th is a llows

much higher t ail cur rent s (and higher slew rat es) for a given gm when J FETs are

used as the input sta ge.


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a

AD845 BiFET 16MHz OP AMP SIMPLIFIED CIRCUIT

1.8

-VS

CP

X1

Q2

Q3

Q1

+VS

Q4

Q6

Q5

D1

VBIAS

+

-

A New VFB Op Am p Arch i t ec t u r e fo r "Cur re n t -on -Dem a nd " P e r form an ce ,

L o w e r P o w e r , a n d I m p r o v e d Sle w R a t e

Unt il now, op am p designers ha d to mak e th e above tr adeoffs between th e input gmstage quiescent cur rent an d th e slew-ra te a nd distortion per forma nce. Ana log

Devices' has pat ent ed a n ew circuit core wh ich su pplies current-on-demand to charge

an d discha rge th e domina nt pole capa citor, Cp, while allowing the quiescent curr entto be sma ll. The ad ditiona l cur ren t is proport iona l to th e fast slewing input signal

an d a dds to th e quiescent curr ent . A simplified diagra m of th e basic core cell is

shown in Figure 1.9.


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1.9

"QUAD-CORE" VFB gm STAGE FOR CURRENT-ON-DEMAND

a

Q2Q1

+VS

-VS

+-

Q5

Q4Q3

CP1

Q6

Q7

Q8

X1

CP2

Th e quad-core (gm sta ge) consist s of tr an sistors Q1, Q2, Q3, and Q4 with th eir

emitt ers conn ected t ogeth er a s shown. Consider a positive step volta ge on t he

invertin g input. This volta ge produces a pr oportiona l cur ren t in Q1 which is

mirr ored int o Cp1 by Q5. The curr ent th rough Q1 also flows th rough Q4 an d Cp2.

At t he dyna mic ran ge limit, Q2 and Q3 ar e correspondingly tur ned off. Notice th at

the char ging an d discha rging curr ent for Cp1 a nd Cp2 is not limited by the qua dcore bias current. In practice, however, small current-limiting resistors are required

form ing an "H" resist or n etwork a s shown. Q7 and Q8 form th e second gain sta ge

(driven differen tially from t he collectors of Q5 and Q6), an d t he out put is buffered by

a u nity-gain complement ar y emitter follower.

The qua d core configura tion is paten ted (Roy Gosser, U.S. Pa ten t 5,150,074 an d

others pending), as well as the circuits which establish the quiescent bias currents

(not shown in the diagram). A number of new VFB op amps using this proprietary

configura tion have been released an d h ave un surpa ssed high frequency low

distort ion performa nce, bandwidth, a nd slew ra te a t the indicated quiescent curr ent

levels (see F igure 1.10). The AD9631, AD8036, and AD8047 ar e optimized for a gain

of +1, while th e AD9632, AD8037, an d AD8048 for a gain of +2. The sa me qua d-corear chitectur e is used as th e second stage of the AD8041 rail-to-rail output, zero-volt

input single-supply op am p. The input st age is a differentia l PNP pair wh ich a llows

th e input comm on-mode signa l to go about 200mV below th e negat ive supply rail.

The AD8042 an d AD8044 are du al a nd qu ad ver sions of th e AD8041.

"QUAD-CORE" TWO STAGE XFCB VFB OP AMPSAC CHARACTERISTICS VERSUS SUPPLY CURRENT


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PART # ISY / AMP BANDWIDTH SLEW RATE DISTORTION

AD9631/32 17mA 320MHz 1300V/s 72dBc@20MHz

AD8036/37 Clamped 20mA 240MHz 1200V/s 72dBc@20MHz

AD8047/48 5.8mA 250MHz 750V/s 66dBc@5MHz

AD8041 (1) 5.2mA 160MHz 160V/s 69dBc@10MHz





Number in ( ) indicates single, dual, or quad

a 1.10

CUR R E NT F EEDBACK (CFB ) OP AMP S

We will now exam ine th e curr ent feedback (CFB) op am p t opology which h as

recently become popular in h igh speed op am ps. The circuit concepts wer e

intr oduced ma ny year s ago, however m odern high speed complemen ta ry bipolar

processes are required to ta ke full advan tage of the ar chitectur e.

It h as long been kn own t hat in bipolar tra nsistor circuits, current s can be switched

faster than voltages, other things being equal. This forms the basis of non-

sat ur at ing emitter -coupled logic (ECL) an d devices such a s curr ent -out put DACs.

Maintaining low impedances at the current switching nodes helps to minimize the

effects of str ay capacitan ce, one of th e largest detr iment s to high speed operat ion.

The curr ent mirr or is a good example of how cur ren ts can be switched with a

minimum amount of delay.

The curr ent feedback op amp t opology is simply an a pplicat ion of th ese fun dam ent al

principles of curr ent steer ing. A simplified CFB op amp is shown in F igure 1.11. The

non-invert ing inpu t is high impedan ce an d is buffered directly to th e inverting input

thr ough t he complementa ry emitter follower buffers Q1 a nd Q2. Note tha t the

invertin g input impedan ce is very low (typically 10 to 100), becau se of the low

emitter resistan ce. In t he ideal case, it would be zero. This is a funda ment aldifference between a CFB a nd a VFB op amp, an d also a feat ur e which gives the

CFB op am p some un ique advanta ges.


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1.11

SIMPLIFIED CURRENT FEEDBACK (CFB) OP AMP

a

+VS

-VS

+ -

Q3

Q1

Q2

R1 R2

Q4

RTCP

i

i

i

X1

The collectors of Q1 an d Q2 drive cur ren t m irrors wh ich m irror t he invert ing input

curr ent to the h igh impedan ce node, modeled by RT an d Cp. The high impedan ce

node is buffered by a complementa ry un ity gain emitt er follower. Feedba ck from th e

outpu t to the inverting input acts to force the inverting input currentto zero, hence

the term Current Feedback. (In t he ideal case, for zer o invertin g input impedan ce, no

sma ll signal voltage can exist at th is node, only sma ll signal cur ren t).

Consider a p ositive step voltage a pplied to the n on-invertin g input of th e CFB op

amp. Q1 immediately sources a proportional current into the external feedback

resistors creat ing an errorcurr ent which is mirr ored to th e high impedan ce node by

Q3. The voltage developed at th e high impeda nce node is equal to this curr ent

multiplied by the equivalent impedance. This is where th e term transim pedan ce op

am p origina ted, since the tr an sfer function is an impedance, rat her t han a u nitless

voltage ra tio as in a tra ditiona l VFB op amp.

Note th at the er ror current is not limited by the inpu t sta ge bias curr ent, i.e., there

is no slew-rate lim itation in an ideal CFB op am p. The curr ent m irrors supply

current-on-demand from th e power su pplies. The nega tive feedback loop t hen forcesthe outpu t voltage to a value which r educes th e inverting input error curr ent t o zero.

The model for a CF B op am p is shown in Figur e 1.12 along with t he corr esponding

Bode plot. Th e Bode plot is p lott ed on a log-log scale, an d t he open -loop gain is

expressed as a transimpedance, T(s), with units of ohms.


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15

( )

a 1.12

R2R1

CFB OP AMP MODEL AND BODE PLOT

CPRTi

|T(s)|

()

fO

fCL

1

1

2R2CPRO RO

R2 R1

X1

X1

RO

VOUT

VIN

RT

R2

RO

6dB/OCTAVE

12dB/OCTAVE

fCL =

2R2CP

FORRO


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16

Examina tion of this equa tion quickly reveals tha t the closed-loop band wid th of a

CFB op am p is determ ined by th e internal d om inan t pole capacitor, Cp, and th e

external feedback resistor R2, an d is in dependent of the gain-settin g resistor, R1. This

ability to maint ain consta nt bandwidth independent of gain m akes CFB op amps

ideally suited for wideban d program ma ble gain a mplifiers.

Becau se t he closed-loop ba ndwidth is inversely proport iona l to th e extern al feedback

resistor, R2, a CFB op a mp is u sua lly optimized for a specific R2. Increasing R2 fromit's optimu m value lowers t he ban dwidth, an d decrea sing it may lead t o oscillat ion

an d inst ability becau se of high frequen cy par asitic poles.

The frequen cy response of the AD8011 CFB op am p is shown in Figur e 1.13 for

var ious closed-loop valu es of gain (+1, +2, and +10). Note th at even a t a gain of +10,

th e closed loop ban dwidth is st ill great er t ha n 100MHz. The peak ing which occurs a t

a gain of +1 is typical of wideband CF B op am ps when used in t he n on-invert ing

mode and is due primar ily to stra y capacitance at t he inverting input . The peaking

can be redu ced by sacrificing band width a nd u sing a slightly larger feedback

resistor. The AD8011 CFB op amp represents state-of-the-art performance, and key

specifications ar e shown in F igure 1.14.

a

+4

1.13

+3

AD8011 FREQUENCY RESPONSEG = +1, +2, +10

+2

+1

0

2

3

4

1 10 100 500

FREQUENCY - MHz

1

+5

5

G = +1

RF = 1k

G = +10

RF = 500

G = +2

RF = 1k

VS= +5V OR 5V

VOUT= 200mV p-p

NORMALIZEDGAIN-dB

AD8011 CFB OP AMP KEY SPECIFICATIONS

nn 1mA Power Supply Current (+5V or 5V)

nn 300MHz Bandwidth (G = +1)

nn 2000 V/s Slew Rate

nn 29ns Settling Time to 0.1%

nn Video Specifications (G = +2)

Differential Gain Error 0.02%


17/37

17

Differential Phase Error 0.06

25MHz 0.1dB Bandwidth

nn Distortion

70dBc @ 5MHz

62dBc @ 20MHz

nn Fully Specified for 5V or +5V Operation

a 1.14

Traditional current feedback op amps have been limited to a single gain stage, using

curr ent -mirrors a s previously described. The AD8011 (an d also oth ers in t his family:

AD8001, AD8002, AD8004, AD8005, AD8009, AD8013, AD8072, AD8073), un like

tra ditiona l CFB op amps uses a two-stage gain configurat ion as shown in Figure

1.15. Un til now, fully complement ar y two-gain st age CFB op am ps ha ve been

impra ctical because of their high power dissipat ion. Th e AD8011 employs a second

gain st age consist ing of a pair of complement ar y amplifiers (Q3 and Q4). Note t ha t

they ar e not connected as curr ent mirrors but as grounded-emitters. The deta iled

design of curr ent sources (I1 a nd I2), an d t heir respective bias circuits (Roy Gosser,pat ent -applied-for) ar e the key to the success of th e two-sta ge CFB circuit; th ey

keep t he a mplifier's quiescent power low, yet ar e capable of supplying current-on-

demandfor wide curr ent excur sions r equired du ring fast slewing.

a

SIMPLIFIED TWO-STAGE CFB OP AMP

1.15

-VS

CP

X1

Q2

Q3

Q1

+VS

Q4

I1

+ -

I2

CC/2

CC/2

A furt her adva nt age of th e two-sta ge amplifier is the h igher overa ll ban dwidth (for

th e sam e power), which mean s lower signal distortion an d th e ability to drive

heavier externa l loads.


18/37

18

Thus far, we ha ve learn ed severa l key featu res of CFB op amps. The most import an t

is tha t for a given complemen ta ry bipolar I C process, CFB generally always yields

higher FPBW (hence lower distortion) than VFB for the same amount of quiescent

supply current. This is becau se th ere is pra ctically no slew-ra te limiting in CFB.

Because of this, the full power ban dwidth an d th e small signa l bandwidth a re

approxima tely the same.

The second importa nt featur e is tha t th e inverting input im pedance of a CFB op am p

is very low . This can be advant ageous when using the op amp in t he inverting mode

as a n I/V convert er, becau se th ere is much less sensitivity to invert ing input

capa citance tha n with VFB.

The third featu re is tha t the closed-loop bandwidth of a CFB op amp is determined

by the valu e of the int ernal Cp capacitor and the external feedback resistor R2 and is

relatively independent of the gain-settin g resistor R1 . We will now examine some

typical a pplications issues an d ma ke furt her compar isons between CFBs a nd VFBs.

CURRENT FEEDBACK OP AMP FAMILY

PART ISY/AMP BANDWIDTH SLEW RATE DISTORTION


AD8002 (2) 5.0mA 600MHz 1200 V/s 65dBc@5MHz



AD8009 (1) 11mA 1000MHz 7000 V/s 80dBc@5MHz



AD8013 (3) 4mA 140MHz 1000 V/s G=0.02%, =0.06

AD8072 (2) 5mA 100MHz 500 V/s G=0.05%, =0.1

AD8073 (3) 5mA 100MHz 500 V/s G=0.05%, =0.1

Number in ( ) Indicates Single, Dual, Triple, or Quad

a 1.16

SUMMARY: CURRENT FEEDBACK OP AMPS

nn CFB yields higher FPBW and lower distortion than

VFB for the same process and power dissipation


19/37

19

nn Inverting input impedance of a CFB op amp is low,

non-inverting input impedance is high

nn Closed-loop bandwidth of a CFB op amp is determined

by the internal dominant-pole capacitance and the

external feedback resistor, independent of the gain-

setting resistor

a 1.17

E FF E C T S OF F EEDBACK CAP ACITANCE IN OP AMP S

At t his point, the t erm noise gain needs some clarification. Noise gain is th e am oun t

by which a sma ll am plitude noise voltage source in ser ies with an input ter mina l of

an op am p is amplified when measu red a t t he output . The input voltage noise of an

op am p is modeled in th is way. It sh ould be noted th at th e DC noise gain can a lso be

used t o reflect t he inpu t offset voltage (and oth er op am p input err or sour ces) to the

output.

N oise ga in must be distinguished from signal gain . Figure 1.18 shows an op amp in

the inverting and non-inverting mode. In the non-inverting mode, notice that noise

gain is equa l to signal gain. H owever, in th e invertin g mode, th e noise gain d oesn't

chan ge, but th e signa l gain is now R2/R1. Resistors a re sh own a s feedback

elements, however, the networks may also be reactive.

FOR VFB OP AMP:

CLOSED-LOOP BW =

fCL =

a 1.18

+

-

R1R1

NOISE GAIN AND SIGNAL GAIN COMPARISON

VOUT

VIN

fu

G

+

-

VIN

VOUT

R2R2

VN VN

NON-INVERTING INVERTING

SIGNAL GAIN = 1 +

NOISE GAIN = = 1 +

SIGNAL GAIN =

NOISE GAIN = 1 +

UNITY GAIN BANDWIDTH FREQUENCYNOISE GAIN

R2

R1

R2R1

- R2

R1

R2R1


20/37

20

Two oth er configur at ions a re sh own in F igure 1.19 wher e th e noise gain ha s been

increased indepen dent of signal gain by th e addit ion of R3 across the inpu t

ter mina ls of the op amp. This t echnique can be u sed to sta bilize de-compen sat ed op

am ps which ar e un sta ble for low values of noise gain. H owever, the sen sitivity to

input noise and offset voltage is correspondingly increa sed.

R2

R1||R3

a 1.19

+

-

R1R1

INCREASING THE NOISE GAINWITHOUT AFFECTING SIGNAL GAIN

VOUT

VIN

+

-

VIN

VOUT

R2R2

VN VN

NON-INVERTING INVERTING

SIGNAL GAIN = 1 +

NOISE GAIN = = 1 +

SIGNAL GAIN =

NOISE GAIN = 1 +

R2

R1

- R2

R1

R2

R1||R3

R3R3

Noise gain is often plott ed as a fun ction of frequ ency on a Bode plot t o determin e th e

op am p sta bility. If th e feedback is pur ely resist ive, the n oise gain is consta nt with

frequency. However, rea ctive element s in th e feedback loop will cau se it t o cha nge

with frequency. Using a log-log scale for the Bode plot allows the noise gain to be

easily drawn by simply calculatin g the brea kpoints deter mined by the frequen cies of

th e var ious poles an d zeros. The point of int ersection of the n oise gain with th e open-

loop gain n ot only determ ines th e op am p closed-loop band wid th , but a lso can be

used to analyze stability.

An excellent explana tion of how to m ak e simplifying appr oximat ions using Bodeplots t o an alyze gain a nd ph ase per form an ce of a feedback net work s is given in

Reference 4.

J ust as signal gain an d noise gain can be different, so can the signal band width and

the closed-loop band wid th . The op amp closed-loop bandwidth, fcl, is always

deter mined by t he int ersection of the n oise gain with th e open-loop frequ ency

response. The signa l bandwidt h is equ al to th e closed-loop ban dwidth only if th e

feedback net work is pur ely resistive.


21/37

21

It is qu ite comm on t o use a capacitor in th e feedback loop of a VFB op am p to sha pe

th e frequency response a s in a simple single-pole lowpass filter (see F igure 1.20a).

The resu lting noise gain is plott ed on a Bode plot t o an alyze stability and ph ase

ma rgin. Sta bility of th e system is deter mined by th e net slope of th e noise gain an d

th e open loop gain wh ere t hey inter sect. For un conditional st ability, th e noise gain

plot m ust inter sect th e open loop response with a n et slope of less th an 12dB/octave.

In t his case, the net slope wher e they int ersect is 6dB/octa ve, indicat ing a sta ble

condition. Notice for t he case dr awn in F igure 1.20a, th e second pole in th efrequency response occurs a t a considerably higher frequen cy th an fu .

a 1.20

R2R1

NOISE GAIN STABILITY ANALYSIS FOR VFB AND CFB

OP AMPS WITH FEEDBACK CAPACITOR

fp=

1

2R2C2

RO

1 +

+

-

R2

A B

VFB OP AMP CFB OP AMP

fp=

1

2R2C2

fCL

f1 f

fu

UNSTABLE

|A(s)| |T(s)|

()

R1

R2

C2

In the case of the CFB op amp (Figure 1.20b), the same analysis is used, except that

th e open-loop tr an simpeda nce gain, T(s), is used to const ru ct the Bode plot. The

definition ofnoise gain (for th e pur poses of stability ana lysis) for a CF B op amp,

however, mu st be r edefined in t erm s of a currentnoise source at tached to th e

invertin g inpu t (see Figure 1.21). This cur ren t is reflected t o th e out put by an

impedan ce which we define t o be th e "curr ent noise gain" of a CFB op a mp:

" "CURRENT NOISE GAIN

Ro ZRo

Z + +

2 11

.


22/37

22

a 1.21

RO

CURRENT "NOISE GAIN" DEFINITION

FOR CFB OP AMP FOR USE IN STABILITY ANALYSIS

i

Z1

X1

X1

i

RO

VOUT

VOUT

Z1

Z2

Z2

T(s)

CURRENT"NOISE GAIN" =

= RO+Z2(1+ )

VOUTi

ROZ1

Now, retu rn to Figure 1.20b, and observe the CFB current n oise gain plot. At low

frequencies, the CF B current noise gain is simply R2 (making the assum ption tha t

Ro is mu ch less t ha n Z1 or Z2. The first pole is deter mined by R2 and C2. As th e

frequency continu es to increa se, C2 becomes a short circuit, and a ll th e invertn g

input curr ent flows th rough Ro (refer ba ck to Figur e 1.21).

The CF B op am p is n orm ally optimized for best per form an ce for a fixed feedback

resistor, R2. Additional poles in th e tr an simpedan ce gain, T(s), occur at frequenciesabove the closed loop ba ndwidth , fcl, (set by R2). Note th at th e inter section of th e

CFB cur ren t n oise gain with th e open-loop T(s) occur s wher e th e slope of th e T(s)

function is 12dB/octave. This indicates instability and possible oscillation.

It is for t his reason th at CFB op amps are not suitable in configurations which

require capacitance in the feedback loop , such as sim ple active integra tors or lowpass

filters. They can, h owever, be used in certa in active filters su ch as t he Sa llen-Key

configur at ion sh own in F igure 1.22 which do not requ ire capacitan ce in th e feedback

network.


23/37

23

a

EITHER CFB OR VFB OP AMPS CAN BE USED IN

THE SALLEN-KEY FILTER CONFIGURATION

1.22

R2R1

+

-

R2 FIXED FOR CFB OP AMP

VFB op am ps, on t he oth er h an d, ma ke very flexible active filters. A multiple

feedback 20MHz lowpass filter using t he AD8048 is shown in Figur e 1.23.

a 1.23

MULTIPLE FEEDBACK 20MHz LOWPASS FILTERUSING THE AD8048 VFB OP AMP

1VIN

R4154

C150pF

C2100pF

R1154

AD8048

R378.7

+5V

0.1F

3

2

1006 VOUT

10F

5 0.1F

5V

10F

4

7


24/37

24

In genera l, the a mplifier should ha ve a bandwidth which is at least t en times th e

ban dwidth of th e filter if problems due to pha se shift of the am plifier ar e to be

avoided. (The AD8048 ha s a ban dwidth of over 200MHz in th is configur at ion). The

filter is designed as follows:

Choose:

Fo = Cut off Frequ ency = 20MHz = Dam ping Ra tio = 1/Q = 2H = Absolute Value of Circuit Gain

= | R4/R1| = 1

k = 2FoC1

CC H

pF24 1 1

2100=

+=

( )

, for C1 = 50pF

RHk

12

159 2= =

. , use 154

Rk H

32 1

79 6

= + =

( ).

, use 78.7

R4 = H R1 = 159.2, use 154

H IGH SP E E D CUR R E NT -TO-VOLTAGE CONVERTERS , AN D

TH E E FF E C T S OF INVERTING IN P U T CAPACITANCE

Fast op amps are useful as current-to-voltage converters in such applications as high

speed ph otodiode pr eam plifiers a nd curr ent -out put DAC buffers. A typical

application using a VFB op am p as a n I/V convert er is shown in Figur e 1.24.


25/37

25

a

COMPENSATING FOR INPUT CAPACITANCE IN ACURRENT-TO-VOLTAGE CONVERTER USING VFB OP AMP

1.24

+

-

C2

R2

C1 VFBi

|A(s)|

1

fp

fx

fu

f

COMPENSATED

UNCOMPENSATED

fp=

fx =

fx = fp fu

C2 =

FOR 45 PHASE MARGIN

1

1

2R2C1

2R2C2

C1

2R2 fu

NOISEGAIN

The net input capacitan ce, C1, form s a pole at a frequen cy fp in th e noise gain

tr an sfer fun ction a s shown in t he Bode plot, an d is given by:

fpR C

=1

2 2 1.

If left uncompen sat ed, the ph ase sh ift a t t he frequen cy of inter section, fx

, will cause

insta bility an d oscillat ion. In tr oducing a zero at fx by adding feedback capacitor C2

sta bilizes the circuit a nd yields a ph ase m ar gin of about 45 degrees. The location of

th e zero is given by:

fxR C

=1

2 2 2.

Although th e a ddition of C2 actu ally decrea ses t he pole frequen cy slightly, this effect

is negligible if C2


26/37

26

This value of C2 will yield a ph ase m ar gin of about 45 degrees. Increasin g the

capacitor by a factor of 2 increa ses th e pha se ma rgin to about 65 degrees (see

Referen ces 4 a nd 5).

In pr actice, th e optimu m valu e of C2 ma y be optimized experimen ta lly by var ying it

slight ly to optimize th e out put pulse response.

A similar a na lysis can be applied to a CFB op am p as sh own in F igure 1.25. In t hiscase, however, the low invert ing input impeda nce, Ro, great ly reduces the sen sitivity

to inpu t capa cita nce. In fact, an ideal CFB with zero input im pedan ce would be

totally insensitive to any amount of input capacitance!

a 1.25

RO

R2

COMPENSATING FOR INPUT CAPACITANCE IN A

CURRENT-TO-VOLTAGE CONVERTER USING CFB OP AMP

+ 1

fp

UNCOMPENSATED2R2C2

R2 ffx

|T(s)|

-

iC1

C2

R2

RO

fp=

fCL

COMPENSATED

FOR 45 PHASE MARGIN

2ROC1

1

1

2RO||R2C1

fx=

fx= fp

fCL

C2= C1

2R2fCL

The pole cau sed by C1 occur s a t a frequency fp:

fpRo R C RoC

= 1

2 2 1

1

2 1 ( || ).

This pole frequ ency will be genera lly be mu ch higher t ha n t he case for a VFB op

am p, and t he pole can be ignored completely if it occur s at a frequen cy great er t ha n

th e closed-loop band width of the op am p.

We next introduce a compensating zero at the frequency fx by insert ing the

capacitor C2:

fxR C

=1

2 2 2.


27/37

27

As in the case for VFB, fx is th e geomet ric mean of fp and f cl:

fx fp fu= .

Solving the equations for C2 and rearranging it yields:

CRo

R

C

R fcl2

2

1

2 2=

.

There is a significant advant age in u sing a CF B op amp in th is configurat ion as can

be seen by compar ing th e similar equ at ion for C2 r equired for a VFB op am p. If th e

un ity-gain ba ndwidth product of th e VFB is equal t o the closed-loop ban dwidth of

th e CFB (at t he optimum R2), then th e size of th e CFB compen sat ion capa citor, C2,

is reduced by a factor of R Ro2 / .

A compa rison in an actu al a pplicat ion is sh own in F igure 1.26. The full scale outpu t

curr ent of the DAC is 4mA, the n et capacita nce at t he inverting input of the op ampis 20pF, and t he feedback resistor is 500. In th e case of the VFB op amp, th e poledue t o C1 occur s a t 16MH z. A compen sat ing capacitor of 5.6pF is r equired for 45

degrees of phase m ar gin, an d th e signa l bandwidth is 57MHz.

a 1.26

LOW INVERTING INPUT IMPEDANCE OF CFBOP AMP MAKES IT RELATIVELY INSENSITIVE TO INPUT

CAPACITANCE WHEN USED AS A

CURRENT-TO-VOLTAGE CONVERTER

+

1

fu = 200MHz

500-4mA

C1

C2

R2

VFB

fp=

2ROC1

+

-

C1

C2

R2

CFB

5004mA

20pF20pF

fCL = 200MHz

RO = 50

= 160MHz

C2 = 1.8pF

fx = 176MHz

1fp

=2R2C1

= 16MHz

C2 = 5.6pF

fx = 57MHz

For t he CF B op am p, however, becau se of th e low invertin g input impedan ce (Ro =

50 ), th e pole occurs a t 160Mhz, th e requ ired compen sat ion capa citor is a bout1.8pF, and the corresponding signal bandwidth is 176MHz. In actual practice, the


28/37

28

pole frequency is so close to the closed-loop bandwidth of the op amp that it could

probably be left u ncompen sat ed.

It sh ould be noted th at a CFB op amp's r elative insensitivity to inverting input

capacitan ce is when it is used in th e inverting mode. In th e non-invert ing mode,

even a few picofara ds of str ay capacitan ce on the invert ing input can cause

significant gain-peaking and potential instability.

Anoth er a dvan ta ge of th e low inverting inpu t impeda nce of th e CFB op amp is when

it is used as a n I/V convert er t o buffer t he outpu t of a h igh speed cur ren t outpu t

DAC. When a step fun ction curr ent (or DAC switching glitch) is applied to th e

inverting input of a VFB op am p, it can produce a large voltage tra nsient u ntil the

signa l can propagate th rough the op am p to its out put an d negative feedback is

regain ed. Back-to-back Schottky diodes are often used t o limit t his volta ge swing as

shown in F igure 1.27. These diodes mu st be low capa cita nce, small geomet ry devices

becau se th eir capacitan ce adds to the total input capa citance.

A CFB op am p, on t he other ha nd, pr esent s a low impeda nce (Ro) to fast switching

curr ent s even before th e feedback loop is closed, th ereby limiting th e voltage

excursion without the requirement of the external diodes. This greatly improves thesett ling time of th e I/V convert er.

1.27

LOW INVERTING INPUT IMPEDANCE OF CFB OP AMPHELPS REDUCE AMPLITUDE OF FAST DAC TRANSIENTS

a

R2

* SCHOTTKYCATCHDIODES

CURRENT-OUTPUT

DAC

I

* NOT REQUIRED FOR CFB OP AMPBECAUSE OF LOW INVERTING INPUT IMPEDANCE

-

+VFB

NOISE COMP ARISONS BE TWEEN VF B AN D CFB OP AMP S


29/37

29

Op am p n oise ha s t wo component s: low frequency noise whose spectra l density is

inversely proport iona l to th e squa re r oot of th e frequ ency an d white noise at

medium an d h igh frequencies. The low-frequency noise is kn own a s 1/f noise (th e

noise powerobeys a 1/f law - th e noise voltage or n oise curr ent is proport iona l to

1/f). The frequency at which th e 1/f noise spectr al den sity equa ls th e white n oise isknown a s th e "1/f Corn er F requ ency" an d is a figur e of merit for t he op am p, with

th e low values ind icat ing better perform an ce. Values of 1/f corner frequency vary

from a few Hz for th e most m odern low noise low frequency am plifiers t o severa lhu ndr eds, or even t housa nds of Hz for h igh-speed op am ps.

In m ost applicat ions of high speed op amps, it is the tota l out put rm s noise th at is

genera lly of inter est. Becau se of th e high ban dwidths, th e chief cont ribut or t o th e

out put rm s noise is th e white n oise, and th at of th e 1/f noise is negligible.

In order t o bett er u nder sta nd t he effects of noise in high speed op amps, we use th e

classical noise model shown in Figur e 1.28. This diagram identifies a ll possible white

noise sour ces, including th e extern al noise in t he source an d th e feedback resistors.

The equa tion allows you t o calculat e th e tota l out put rm s noise over t he closed-loop

ban dwidth of th e amp lifier. This formu la work s quite well when t he frequen cy

response of th e op am p is relat ively flat. If ther e is more tha n a few dB of highfrequency peaking, however, the actual noise will be greater than the predicted

becau se t he contr ibution over th e last octa ve before th e 3db cutoff frequ ency will

dominate. In most applications, the op amp feedback network is designed so that the

bandwidth is relatively flat, and the formula provides a good estimate. Note that

BW in t he equ at ion is th e equivalent n oise bandwidt h wh ich, for a single-pole

system, is obta ined by m ultiplying th e closed-loop ba ndwidth by 1.57.

1.28

OP AMP NOISE MODEL FOR A

FIRST-ORDER CIRCUIT WITH RESISTIVE FEEDBACK

R1

VON

Rp

In-

In+

VON =2

R2

fcl = CLOSED LOOP BANDWIDTH

In-2 R 2

2 + In+2 RP

2 1 +R1

R2+ Vn

2 + 4kTR 2 + 4kTR1 + 4kTR P1 +

21 +

2

Vn

VR2J

VR1J

VRPJ

BW = 1.57fcl

BWR2

R1

2 R2

R1

R2

R1

+

a


30/37

30

Figur e 1.29 shows a ta ble which indicat es how the individual noise cont ribut ors a re

referred t o th e out put . After calculat ing the in dividua l noise spectra l densities in

this ta ble, they can be squared, added, and then the squ ar e root of the su m of the

squa res yields th e RSS value of th e out put noise spectra l density since all the

sources are u ncorr elated. This value is mu ltiplied by the squ ar e root of th e noise

ban dwidth (noise band width = closed-loop bandwidt h mu ltiplied by a corr ection

factor of 1.57) to obta in t he final valu e for t he out put rm s noise.

REFERRING ALL NOISE SOURCES TO THE OUTPUT

NOISE SOURCE EXPRESSED AS

A VOLTAGE

MULTIPLY BY THIS FACTOR TO REFER

TO THE OP AMP OUTPUT

Johnson Noise in Rp:

4kTRpNoise Gain

R

R== ++1

2

1

Non-Inverting Input Current Noise

Flowing in Rp:

In+Rp

Noise GainR

R== ++1

2

1

Input Voltage Noise:

VnNoise Gain

R

R== ++1

2

1

Johnson Noise in R1:

4 1kTR

R2/R1 (Gain from input of R1 to Output)

Johnson Noise in R2:

4 2kTR 1

Inverting Input Current Noise

Flowing in R2:

In-R2

1

a 1.29

Typical high speed op am ps with ban dwidths greater tha n 150MHz or so, and

bipolar input stages have input voltage noises ranging from about 2 to 20nV/Hz. Toput voltage n oise in perspective, let's look at th e J ohn son noise spectr al den sity of a

resistor:

vn

kTR BW= 4 ,

where k is Boltzman n's consta nt, T is th e absolute t emperat ure, R is th e resistor

value, an d BW is th e equivalent n oise ban dwidth of inter est. (The equivalent noise

ban dwidth of a single-pole system is 1.57 times th e 3dB frequency). Using th e

formula, a 100 resistor h as a noise density of 1.3nV/Hz, and a 1000 resistorabout 4n V/Hz (values are at room temperature: 27C, or 300K).

The base-emitter in a bipolar t ra nsistor ha s a n equivalent n oise voltage sour ce

which is due t o the "shot noise" of th e collector curr ent flowing in t he t ra nsistor's


31/37

31

(noiseless) incremen ta l emitter r esistan ce, re. The curr ent noise is proport iona l to

th e squa re r oot of th e collector curr ent , Ic. The emitter resist an ce, on the other

ha nd, is inversely proport iona l to the collector cur ren t, so the shot-noise voltage is

inversely proportional to the square root of the collector current. (Reference 5, Section

9).

Volta ge noise in F ET-input op am ps ten ds to be lar ger th an for bipolar ones, but

current noise is extremely low (generally only a few tens of fA/Hz) becau se of th elow input bias currents. However, FET-inputs are not generally required for op ampapplicat ions r equiring bandwidths grea ter t han 100MHz.

Op amps also have input curr ent noise on each input. F or high-speed FET-input op

amps, th e gate current s ar e so low tha t input cur rent noise is almost a lways

negligible (measured in fA/Hz).

For a VFB op am p, the inverting a nd n on-inverting input cur rent noise ar e typically

equa l, an d almost a lways un corr elated. Typical values for wideban d VFB op amps

ra nge from 0.5pA/Hz t o 5pA/Hz. The input cur rent noise of a bipolar input stageis increased wh en inpu t bias-cur rent can cellation generat ors ar e added, becau se

their curr ent n oise is not corr elated, and t herefore a dds (in an RSS man ner) to the

intr insic curr ent noise of the bipolar st age.

The inpu t volta ge noise in CF B op am ps ten ds to be lower t ha n for VFB op amps

ha ving the sa me appr oximat e bandwidth. This is because t he input stage in a CFB

op am p is usually operat ed at a h igher curr ent, th ereby reducing the emitter

resista nce an d hen ce the voltage n oise. Typical values for CF B op amps r an ge from

about 1 t o 5nV/Hz.

The input cur rent noise of CFB op am ps tends t o be larger t han for VFB op amps

because of the generally higher bias current levels. The inverting and non-inverting

current noise of a CFB is usually different because of the unique input architecture,and ar e specified separat ely. In m ost cases, the inverting input curr ent noise is the

larger of th e two. Typical inpu t cur ren t n oise for CF B op am ps ra nges from 5 t o

40pA/Hz.

The gener al pr inciple of noise calculation is th at un corr elated n oise sources add in a

root-sum-squar es man ner, which m eans tha t if a noise sour ce ha s a contr ibution t o

th e out put noise of a system wh ich is less th an 20% of th e am plitude of th e noise

from other noise sour ce in th e system, th en its cont ribut ion t o th e total system n oise

will be less tha n 2% of th e total, an d th at noise sour ce can a lmost invar iably be

ignored - in ma ny cases, noise sources sma ller t ha n 33% of the lar gest can be

ignored. This can simplify th e calculations usin g the form ula, a ssum ing the corr ect

decisions a re m ade r egard ing the sour ces to be included a nd t hose to be neglected.

The sources which dominat e the outp ut noise ar e highly dependen t on th e closed-

loop gain of the op amp. Notice that for high values of closed loop gain, the op amp

voltage n oise will tend be th e chief cont ribut or t o the outpu t n oise. At low gains , the

effects of th e input curr ent noise mu st a lso be consider ed, and m ay dominat e,

especially in th e case of a CF B op amp.

Feedforwa rd/feedback resistors in h igh speed op am p circuits m ay r an ge from less

than 100 to more th an 1k, so it is difficult to genera lize a bout th eir cont ribut ion


32/37

32

to th e tota l out put noise with out kn owing th e specific values an d th e closed loop

gain. The best way to make the calculations is to write a simple computer program

which perform s th e calculat ions a ut omatically an d include a ll noise sour ces. In m ost

high speed applicat ions, t he sour ce impeda nce noise can be neglected for source

impedances of 100 or less.

Figur e 1.30 shows an example calcula tion of tota l out put noise for t he AD8011

(300MHz, 1mA) CFB op amp. All six possible sources are included in the calculation.The appropriate multiplying factors which reflect the sources to the output are also

shown on t he diagra m. F or G=2, the close-loop ba ndwidt h of the AD8011 is

180MHz. The corr ection factor of 1.57 in th e final calculation convert s t his sin gle-

pole bandwidth into the circuits equivalent noise bandwidth.

a 1.30

AD8011 OUTPUT NOISE ANALYSIS

(G)

+

-

RS

50

fCL = 180MHz

1.8nV/Hz

0.5nV/Hz

4nV/Hz

4nV/Hz

5nV/Hz

4nV/Hz4nV/Hz

5pA/Hz4nV/Hz

AD8011

5pA/Hz

0.9nV/Hz

2nV/Hz

R11k

R2

1k

OUTPUT NOISE SPECTRAL DENSITY = 8.7nV/Hz

TOTAL NOISE = 8.71.57 X 180 X 106 = 146V rms

(G RS)

(G)

(1)

(R2)

(-R2/R1)

G = 1 +R2R1

In comm un icat ions applications, it is comm on to specify the noise figure (NF) of an

am plifier. Figure 1.31 shows th e definition. NF is the r at io of th e total int egrat ed

out put noise from a ll sour ces to the t otal out put noise which would resu lt if th e op

am p were "noiseless" (this noise would be th at of the sour ce resistan ce multiplied by

th e gain of th e op am p using t he closed-loop bandwidth of th e op am p to ma ke th ecalculat ion). Noise figure is expressed in dB. The value of the source resist an ce must

be specified, an d in m ost RF systems, it is 50. Noise figure is use ful incomm un icat ions r eceiver design, since it can be used t o mea sur e th e decrease in

signal-to-noise ra tio. For inst an ce, a n am plifier with a noise figure of 10dB following

a st age with a s ignal-to-noise ra tio of 50dB redu ces the signa l-to-noise rat io to 40dB.


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33

a 1.31

NOISE FIGURE OF AN OP AMP

+

-RS

50

NOISE FIGURE = 20log = 13.7dB

AD80110.9nV/Hz

R2

1k

R1

1k

8.7

1.8

TOTAL = 8.7nV/Hz

NOISE FIGURE = 20logTOTAL OUTPUT NOISE

OUTPUT NOISE DUE TO RS

1.8nV/HzG

The ra tio is comm only expressed in dB an d is useful in signal chain a na lysis. In th e

previous exam ple, the tota l out put voltage noise was 8.7nV/Hz. Integrat ed over theclosed loop ba ndwidth of th e op am p (180MHz), this yielded an out put noise of

146V rm s. The noise of th e 50 source resista nce is 0.9nV/Hz. If the op am p werenoiseless (with noiseless feedback r esistors), th is noise would appear at th e outpu t

mu ltiplied by th e noise gain (G=2) of th e op am p, or 1.8nV/Hz. The total outputrms noise just due t o the sour ce resistor int egrated over t he sa me ban dwidth is

30.3V rm s. The n oise figure is calculat ed as:

NF dB=

=20 10146

30 313 7log

.. .

The same result can be obtained by working with spectral densities, since the

bandwidths used for th e integration ar e the sam e and cancel each other in the

equation.

NF dB=

=20 108 7

1 813 7log

.

.. .

HIGH SPEED OP AMP NOISE SUMMARY

nn Voltage Feedback Op Amps:

uu Voltage Noise: 2 to 20nV/Hzuu Current Noise: 0.5 to 5pA/Hz

nn Current Feedback Op Amps:


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34

uu Voltage Noise: 1 to 5nV/Hzuu Current Noise: 5 to 40pA/Hz

nn Noise Contribution from Source Negligible if < 100

nn Voltage Noise Usually Dominates at High Gains

nn Reflect Noise Sources to Output and Combine (RSS)

nn Errors Will Result if there is Significant

High Frequency Peaking

a 1.32

DC CHARACTERISTICS OF H IGH SP E E D OP AMP S

High speed op amps a re optimized for ba ndwidth an d sett ling time, not for pr ecisionDC characteristics as found in lower frequency op amps such as the industry

standard OP27. In spite of this, however, high speed op amps do have reasonably

good DC per form an ce. The m odel shown in Figur e 1.33 shows how to reflect th e

input offset voltage and the offset currents to the output.

a 1.33

VO = VOS 1 + + Ib+R3 1 + - I b-R2

IF Ib+ = Ib- AND R3 = R1||R2

V O = VOS 1 +

MODEL FOR CALCULATING TOTAL

OP AMP OUTPUT VOLTAGE OFFSET

Ib-

VOS

+

-

R3

R2R1

R2

R1

Ib+

R2

R1

R2

R1

VO


35/37

35

Inpu t offset voltages of high speed bipolar in put op am ps ar e ra rely tr immed, since

offset voltage ma tching of th e inpu t st age is excellent, typically ra nging from 1 to

3mV, with offset temperature coefficients of 5 to 15V/C.

Input bias curr ents on VFB op am ps (with n o input bias curr ent compensat ion

circuits) ar e appr oximat ely equa l for (+) an d () input s, an d can r an ge from 1 to 5A.

The output offset voltage due to the input bias curr ents can be nulled by mak ing the

effective source resista nce, R3, equa l to th e pa ra llel combina tion of R1 an d R2.

This scheme will not work, however, with bias-curr ent compen sat ed VFB op am ps

which h ave additional curren t genera tors on t heir inputs. In t his case, the net input

bias curr ent s ar e not necessar ily equa l or of th e sam e polar ity. Op am ps designed for

rail-to-rail input operation (parallel PNP and NPN differential stages as described

later in th is section) ha ve bias cur ren ts wh ich a re a lso a function of the comm on-

mode inpu t voltage. Exter na l bias cur ren t cancellation schemes a re ineffective with

th ese op am ps also. It sh ould be noted, however, th at it is often desira ble to ma tch

th e source impedan ce seen by the (+) an d () input s of VFB op am ps to minimize

distortion.

CFB op amps generally ha ve unequal an d un corr elated input bias curren ts becau seth e (+) an d () input s ha ve completely differen t a rchitectur es. For this r eason,

extern al bias curr ent can cellation schemes ar e also ineffective. CFB input bias

curr ents ra nge from 5 to 15A, being generally higher at the invert ing input

OUTPUT OFFSET VOLTAGE SUMMARY

nn High Speed Bipolar Op Amp Input Offset Voltage:

uu Ranges from 1 to 3mV for VFB and CFB

uu Offset TC Ranges from 5 to 15V/C

nn High Speed Bipolar Op Amp Input Bias Current:uu For VFB Ranges from 1 to 5A

uu For CFB Ranges from 5 to 15A

nn Bias Current Cancellation Doesn't Work for:

uu Bias Current Compensated Op Amps

uu Current Feedback Op Amps

a 1.34

P S R R C HARACTERISTICS OF H IGH SP E E D OP AMP S

As with most op am ps, th e power supp ly rejection ra tio (PSRR) of high speed op

am ps falls off ra pidly at higher frequencies. Figur e 1.35 shows the P SRR for t he

AD8011 CFB 300MHz CFB op amp. N otice th at at DC, the P SRR is near ly 60dB,

however, a t 10MH z, it falls to only 20dB, indicatin g th e need for excellent ext ern al

LF a nd H F decoupling. These nu mber s ar e fairly typical of most h igh speed VFB or

CFB op amps, alth ough t he DC PSRR m ay ra nge from 55 to 80dB depending on t he

op am p.


36/37

36

a 1.35

AD8011 POWER SUPPLY REJECTION RATIO

+10

0

-10

-20

-30

-50

-60

-70

100k 1M 10M 100M 500M

FREQUENCY - Hz

-40

-80

-90

+PSRR

-PSRR

PSRR-dB

VS = +5V OR 5V

G = +2

RF = 1k

The power pins of op am ps mu st be decoupled directly to a la rge-area ground pla ne

with capa citors wh ich h ave minimal lead length. It is genera lly recommended tha t a

low-indu ctan ce cera mic surface mount capacitor (0.01F to 0.1F) be used for t he

high frequency noise. The lower frequency noise can be decoupled with low-

inducta nce ta nt alum electr olytic capacitors (1 to 10F).


37/37

R E F E R E N C E S

1. Thoma s M. Freder iksen , In t u i t i ve Oper a t i ona l Am p l if ie r s ,

McGraw-Hill, 1988.

2. Sergio Franco, Current Feedback Amplifiers, E DN , Ja n.5, 1989.

3. Roy Gosser , U .S Pa t en t 5,150,074.

4. J a m es L. Melsa a nd Don ald G. Sch ult z, L in e a r C o n t r o l S ys t e m s ,

McGraw-Hill, 1969, pp. 196-220.

5. Am p l ifi e r App l ica t i on s Gu i de , Analog Devices, Inc., 1992,

Section 3.

6. Walter G. J ung, I C O p a m p C o o k b o o k , T h i r d E d i t i o n ,

Howar d S am s & Co., 1986, ISBN: 0-672-22453-4.

7. Pa ul R. Gray an d Robert G. Meyer, Ana l ys i s an d Des i gn o f Ana l ogI n t e g r a t e d C i r c u i t s , T h i r d E d i t i o n , J ohn Wiley, 1993.

8. J . K. Roberge, O p e r a t i o n a l Am p li fi e r s -T h e o r y a n d P r a c t i ce ,

J ohn Wiley, 1975.

9. Henry W. Ot t , N o is e R e d u c t i on T e c h n i q u e s i n E l e ct r o n i c S ys t e m s ,

S e c o n d E d i t i on , J ohn Wiley, Inc., 1988.

10. Lewis Sm it h an d Da n Sh ein gold ,N oise and Opera tional Am plifier Circu it s,

Ana l og Di a logue 25t h Ann i ve r s a r y I s s ue , pp. 19-31, 1991.

11. D. St ou t, M. Ka ufm an , H a n d b o o k o f O p e r a t i o n a l Am p l ifi e r C i r c u i t

Des ign , New York , McGra w-Hill, 1976.

12. J oe Buxton , Careful Design Tam es High-Sp eed Op Am ps, E l e c t r o n i c

Des ign , April 11, 1991.

13. J . Dosta l, Oper a t i on a l Am p l if ie r s , Elsevier Scientific Publishing,

New York, 1981.

14. Bar rie Gilbert , Cont emporary Feedback Am plifier Design ,

15. Sergio Franco, Des i gn w i t h Oper a t i on a l Am p l ifi e r s an d Ana l og ICs ,McGraw-Hill Book Company, 1988.

16. J era ld Graeme, P ho t od i ode Am p l ifi e r s -Op Am p So l u t i on s ,

Gain Technology Corporation, 2700 W. Broadway Blvd., Tucson,

AZ 85745, 1996.

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High Speed Opamp Feedback