Final Report

Design of a Compact Vivaldi Antenna ArrayChapter 1

2011

Introduction

1.1 IntroductionThe objective of this project has been to design a compact tapered slot Vivaldi antenna array for UWB see through wall radar. Vivaldi antennas have received considerable attention due to their high gain, relatively wide band, simple structure, easy fabrication, and wide use in UWB applications. Their small lateral dimensions and simple integration make them excellent candidates for array development [1.1]. Federal Communication Commission (FCC) approved a 1.99 GHz to 11.6 GHz frequency band for use in UWB through-wall imaging systems [1.2]. Yang et.al. [1.3] designed a Vivaldi antenna array around 10 GHz for UWB see through wall radar utilizing antipodal Vivaldi antennas with Wilkinson power divider for the binary feed. However, the size of this 16 element array is relatively too large if the antenna array is duplicated for the lower band UWB applications; i.e. close to 3 GHz. Therefore, we used here only a 4- element array and optimized its performance to sustain similar almost constant gain over its operating band. Similar concepts to that utilized by Abbosh et al. [1.4] to design a compact UWB antipodal Vivaldi antenna have been utilized here. In this project, we have developed a Vivaldi antenna array for see through wall UWB applications. The configuration of the array element was optimized to have a compact size. Then, a 1 2 Vivaldi antenna array is developed using tapered slot antennas (TSA) and a 3 dB power divider. After that, a 1 4 Vivaldi antenna array is developed using tapered slot antennas (TSA) and Wilkinson power divider. Details of the developed Vivaldi antenna, Wilkinson power divider, Vivaldi antenna arrays, its simulation and experimental results are presented in this project report.

Department of Electronics, CUSAT

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1.2 Antenna ParametersThe performance of an antenna can be gauged from a number of parameters. Cetain r critical parameters are discussed below [1.5]. 1.2.1 Gain Gain is a parameter which measures the degree of directi it of the antenna's radiation pattern. An antenna with a low gain emits radiation with about the same power in all directions, whereas a high-gain antenna will preferentiall radiate in particular direc tions. Specificall , the ant nna gain, directive gain, or power gain of an antenna is defined as the ratio of the intensit (power per unit surface) radiated by the antenna in the direction of its maximum output, at an arbitrary distance, di ided by the intensity radiated at the same distance by a hypothetical isotropic antenna. The gain of an antenna is a passi e phenomenon - power is not added by the antenna, but simply redistributed to provide more radiated power in a certain direction than would be transmitted by an isotropic antenna. An antenna designer must take into account the application for the antenna when determining the gain. High -gain antennas have the advantage of longer range and better signal quality, but must be aimed carefully in a particular direction. Low-gain antennas have shorter range, but the orientation of the ante nna is relatively inconsequential. For example, a dish antenna on a spacecraft is a highgain device that must be pointed at the planet to be effective, whereas a typicalWi-Fi antenna in a laptop computer is low-gain, and as long as the base station is within range, the antenna can be in any orientation in space. It makes sense to improve hori ontal range at the expense of reception above or below the antenna. Thus most antennas labeled "omnidirectional" really have some gain. Power gain is a unit less measure that combines an antenna's efficiency and directivity figures: (1.1)

If the radiation intensity U in the desired solid angle is known, then power gain for that solid angle can be calculated:

Depart ent of E ectronics CUSAT

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Design of a Compact Vi aldi Antenna Array 2011

(1.2) 1.2.2 Radiation pattern The radiation pattern of an antenna is a plot of the relative field strength of the radio waves emitted by the antenna at different angles. It is typically represented by a three dimensional graph, or polar plots of the hori ontal and vertical cross sections. The pattern of an ideal isotropic antenna, which radiates equally in all directions, would look like a sphere. Many non directional antennas, such as monopoles and dipoles, emit equal power in all hori ontal directions, with the power dropping off at higher and lower angles; this is called an omnidirectional pattern and when plotted looks like a torus or donut. The radiation of many antennas shows a pattern of maxima or "obes" at various l angles, separated by "nulls", angles where the radiation falls to zero. This is because the radio waves emitted by different parts of the antenna typically interfere, causing maxima at angles where the radio waves arrive at distant points in phase, and zero radiation at other angles where the radio waves arrive out of phase. In a directional antenna designed to project radio waves in a particular direction, the lobe in that direction is designed larger than the others and is called the "main lobe". The other lobes usually represent unwanted radiation and are called "sidelobes". The axis through the main lobe is called the "principle axis" or "boresight axis".

Fig. 1.1 Radiati n pattern of a directional antenna

Department of Electronics CUSAT

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Design of a Compact Vivaldi Antenna Array

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1.2.3 Impedance As an electro-magnetic wave travels through the different parts of the antenna system (radio, feed line, antenna, free space) it may encounter differences in impedance (E/H, V/I, etc.). At each interface, depending on the impedance match, some fraction of the wave's energy will reflect back to the source, forming a standing wave in the feed line. The ratio of maximum power to minimum power in the wave can be measured and is called the standing wave ratio (SWR). A SWR of 1:1 is ideal. A SWR of 2.5:1 is considered to be marginally acceptable in low power applications where power loss is more critical, although an SWR as high as 6:1 may still be usable with the right equipment. Minimizing impedance differences at each interface (impedance matching) will reduce SWR and maximize power transfer through each part of the antenna system. Complex impedance of an antenna is related to the electrical length of the antenna at the wavelength in use. The impedance of an antenna can be matched to the feed line and radio by adjusting the impedance of the feed line, using the feed line as an impedance transformer. More commonly, the impedance is adjusted at the load with an antenna tuner, a balun, a matching transformer, matching networks composed of inductors and capacitors, or matching sections such as the gamma match. 1.2.4 Efficiency Effi iency is the ratio of power actually radiated to the power put into the antenna terminals. A dummy load may have an SWR of 1:1 but an efficiency of 0, as it absorbs all power and radiates heat but very little RF energy, showing that SWR alone is not an effective measure of an antenna's efficiency. Radiation in an antenna is caused by radiation resistance which can only be measured as part of total resistance including loss resistance. Loss resistance usually results in heat generation rather than radiation, and reduces efficiency. Mathematically, efficiency is calculated as radiation resistance divided by total resistance. 1.2.5 Polarization The polarization of an antenna is the orientation of the electric field (E-plane) of the radio wave with respect to the Earth's surface and is determined by the physical structure of the antenna and by its orientation. It has nothing in common with antenna directionality terms: "horizontal", "vertical" and "circular". Thus, a simple straight wire antenna will haveDepartment of Electronics, CUSAT Page 4


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one polarization when mounted vertically, and a different polarization when mounted horizontally. "Electromagnetic wave polarization filters are structures which can be employed to act directly on the electromagnetic wave to filter out wave energy of an undesired polarization and to pass wave energy of a desired polarization. Reflections generally affect polarization. For radio waves the most important reflector is the ionosphere - signals which reflect from it will have their polarization changed unpredictably. For signals which are reflected by the ionosphere, polarization cannot be relied upon. For line-of-sight communications for which polarization can be relied upon, it can make a large difference in signal quality to have the transmitter and receiver using the same polarization; many tens of dB differences are commonly seen and this is more than enough to make the difference between reasonable communication and a broken link. Polarization is largely predictable from antenna construction but, especially in directional antennas, the polarization of side lobes can be quite different from that of the main propagation lobe. For radio antennas, polarization corresponds to the orientation of the radiating element in an antenna. A vertical omnidirectional WiFi antenna will have vertical polarization (the most common type). An exception is a class of elongated waveguide antennas in which vertically placed antennas are horizontally polarized. Many commercial antennas are marked as to the polarization of their emitted signals. Polarization is the sum of the E-plane orientations over time projected onto an imaginary plane perpendicular to the direction of motion of the radio wave. In the most general case, polarization is elliptical, meaning that the polarization of the radio waves varies over time. Two special cases are linear polarization (the ellipse collapses into a line) and circular polarization (in which the two axes of the ellipse are equal). In linear polarization the antenna compels the electric field of the emitted radio wave to a particular orientation. Depending on the orientation of the antenna mounting, the usual linear cases are horizontal and vertical polarization. In circular polarization, the antenna continuously varies the electric field of the radio wave through all possible values of its orientation with regard to the Earth's surface. Circular polarizations, like elliptical ones, are classified as right-hand polarized or left-hand polarized using a "thumb in the direction of the propagation" rule. Optical researchers use the same rule of thumb, but pointing it in the direction of the emitter, not in the direction of propagation, and so are opposite to radio engineers use.


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In practice, regardless of confusing terminology, it is important that linearly polarized antennas be matched, lest the received signal strength be greatly reduced. So horizontal should be used with horizontal and vertical with vertical. Intermediate matchings will lose some signal strength, but not as much as a complete mismatch. Transmitters mounted on vehicles with large motional freedom commonly use circularly polarized antennasso that there will never be a complete mismatch with signals from other sources. 1.2.6 Return loss Return loss or reflection loss is the loss of signal power resulting from the reflection caused at a discontinuity in a transmission line or optical fiber. This discontinuity can be a mismatch with the terminating load or with a devic inserted in the line. It is usually e expressed as a ratio in decibels (dB);

Where RL (dB) is the return loss in dB, Pi is the incident power and Pr is the reflected power. Properly, loss quantities, when expressed in decibels, should be positive numbers However, return loss has historically been expressed as a negative number, and this convention is still widely found in the literature. Taking the ratio of reflected to incident power results in a negative sign for return loss

Where RL'(dB) is the negative of RL(dB). Caution is required when discussing increasing or decreasing return loss since these terms strictly have the opposite meaning when return loss is defined as a negative quantit . y

1.3 Project Report OrganizationThis repot has been organized in six chapters. The second chapter gives an insight into the tapered slot antenna radiation characteristics and design.Department of Electronics CUSAT Page 6


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The third chapter gives a qualitative description of the full wave analysis technique, Finite Element Method (FEM). Ansofts HFSS software based on FEM is used for antenna simulation in this project. A brief description of HFSS is also given in this chapter. The fourth chapter gives the design of Tapered Slot Antenna, its simulation and measurement results. The fifth chapter gives the design and fabrication of 12 Vivaldi antenna array, its simulation and measurement results. The sixth chapter discusses the 14 Vivaldi antenna array. A comparison between the results of 2-element and 4-element array is also done. The seventh chapter is conclusion.

1.4 References

[1.1]

Sng-Gyu Kim, Kai Chang, A low cross polarized antipodal Vi aldi antenna array for wideband operation, Antennas and Propagation Society International Symposium, vol.3, June 2004, pp. 2269 -2272.

[1.2]

Federal Communications Commission, Revision of part 15 of the Commissions R les Regarding Ultra Wideband Transmission Systems, March 12, 2003

[1.3]

Yunqiang Yang, Cemin Zhang, Song Lin and Aly E. Fathy, Development of an ultra wideband antipodal antenna, Antenna and Propagation Society International Symposium, vol. 1A, July 2005, pp. 606 -609.

[1.4]

A.M. Abbosh, H.K. Kan and M.E. Bialkowski, Design of compact directive ultra wideband antipodal antenna, Microwave and Optical Technology Letters, vol. 48, no.12, Dec. 2006, pp. 2448-2451.

[1.5]

www.wikipedia.org


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Design of a Compact Vivaldi Antenna ArrayChapter 2

2011

Tapered Slot Antennas

2.1 IntroductionThe objective of this chapter is to give an insight into the radiation mechanism and various topologies possible while designing a Tapered Slot Antenna (TSA). Further, comparison of their performances is also given based on the papers published by the other authors. The terms, notch antenna, tapered notch, flared slot antennas are used synonymously to represent TSAs. Tapered slot antennas belong to the class of travelling wave antennas. They are end fire radiators. Tapered slot is formed by gradual widening of a slotline. In 1979 Gibson [2.1] demonstrated an exponentially tapered slot antenna demonstrating a bandwidth of 8-40 GHz and he called it Vivaldi antenna. In the same year Prasad and Mahapatra [2.2] first introduced the linearly tapered slot antenna (LTSA). However, the tapered slot antenna was introduced as an array element by Lewis et al. [2.3] in 1974. The conventional resonant microstrip antenna size becomes very small as the operating frequency shifts to millimeter wave frequency band. This increases the cost because fabrication tolerance level decreases. In addition, the skin effect conductor losses in the microstrip feed network tends to become excessive at higher frequencies thus lowering antenna efficiency. Tapered slot antennas can circumvent these problems. The dimensions of tapered slot antennas are several times the free space wavelength at the frequency of operation which eases the fabrication tolerance. Many variations of these antennas were fabricated for frequency of operation of up to about 800 GHz [2.4] and higher within the required fabrication tolerances using standard printed circuit fabrication techniques. Furthermore, active circuits like mixers and amplifiers can be integrated with the antenna using stripline, slotline, microstrip line, finline and coplanar waveguide. In addition to this, other advantages are:y y y

Multi-octave bandwidth. Moderate gain. Symmetric E and H plane radiation patterns.Page 8



Some of the disadvantages as compared to the conventional microstrip patch are: y y y Cannot be designed for dual-frequency operation. Dual polarization cannot be obtained without structure complexity. Loses its planar architecture when used in 2D array.

2.2 Taper ProfilesAccording to the different taper profile they are generally classified into: y y y Linear tapered (LTSA) Non-linear tapered (exponential, parabolic) Constant width (CWSA)

(a)

(b)

(c)

(d)

(e)

(f)

Fig 2.1. Various Tapered slot Antenna Profiles: a) Step constant b) Exponential c) Linear d) Parabolic e) Linear constant f) Broken linear Various tapered slot profiles are illustrated in Fig 2.1. A variant of the conventional TSA is the antipodal tapered slot antenna, see Fig. 2.2. In practice, the conventional planar TSA is fed by a balanced slotline. One serious drawback of the conventional TSA is in the fabrication and impedance matching of the slotline. Slotline fabricated on a low dielectric constant substrate has relatively high impedance which makes matching to a low impedance microstrip feed very difficult. The antipodal TSA replaces the band limiting microstrip/ slotline transition by a tapered balun section which gives a very wide bandwidth. However, it has exhibited very poor cross polarization characteristics. The antenna is formed by graduallyDepartment of Electronics CUSAT Page 9


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flaring the strip conductors of the balanced microstrip on opposite sides of the dielectic substrate with respect to the antenna axis, thus allowing the antenna to be directly fed by a microstrip feed [2.5]. The antenna suffers from the high cross polarization due to skewing of the E-field components within the antenna with respect to the physical axis of the antenna. At the low frequency end of the band this skew is small because the ratio of slot width to dielectric thickness is large. However as we move to the high frequency end the angle of skew increases and ultimately tends to 90o. Therefore the antenna has poor cross polarization (of the order of 5 dB) and also there is severe polarization tilt as the frequency of operation increases [2.6].

Fi 2.2. Antipodal Vivaldi Antenna

To overcome the high cross polarization problem, Langley et al. [2.6] introduced a new technique. The idea was to negate the effect of skew by applying another dielectric layer and metallization layer. This new antenna, known as balanced antipodal Vivaldi which is fed by stripline, has demonstrated -15 dB lower crosspolarizations across an 18:1 band as compared to the conventional Vivaldi antenna.

2.3 Radiation mechanismThe Vivaldi antenna is essentially frequency independent, since at a given wavelength only a section of the exponential curve actually radiates efficiently. As the wavelength varies, radiation occurs from a different section which is scaled in size in proportion to the wavelength, and has the same relative shape. This translates into antenna with a large bandwidth. The main lobe of the antenna is linearly polarized with the electric field parallel to the aperture.


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(a) Fi 2.4. Vivaldi Antenna

(b)

A TSA can be divided into two regions: a non radiating feed region and radiating slot region [2.7]. With respect to Fi 2.4 these regions can be identified as:y y

Propagating area defined by Ws < W < Wa Radiating area defined by Wa < W < Wo

Where Ws- input slot width Wa- slot width at radiating area Wo- output width The main, non resonant, travelling wave mechanism of radiation is produced by higher order Hankel function (H o(n)) mode generated by waves travelling down a curved path along the antenna [2.3]. The energy in the travelling wave is tightly bound to the conductors when the separation is very small compared to the free space wavelength and that becomes progressively weaker and more coupled to the radiation field as the separation is increased. The guide wavelength and characteristic impedance of a slotline increase with the increase in width of the slot. For a given dielectric substrate of certain thickness at a specified frequency, a 20% increase in slot width leads to 1% increase in / and 6% increase in Zo. If is less than about 40% of the fields will be adequately contained and the slotline behaves like transmission line. Departure from this condition would result in radiation [2.2].


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2.3.1 Antipodal Vivaldi Antenna:

The electric field lines at different cross section along the feed and the antenna are illustrated in Fi . 2.5. The electric field lines which are spread out in the conventional microstrip structure, concentrate between the metal strip of the balanced microstrip, and finally rotate while travelling along the TSA [2.8]. This is the reason for higher crosspolarizations in these types of antennas. At the higher frequencies (narrower slot width) the skew angle is almost 90o due to which there can be a severe polarization tilt also.

(a) Fi

(b)

(c)

2.5. Electric field distribution at cross sections a) Conventional microstrip b)

Balanced microstrip c) Radiatin ed e

2.4 Desi n ConsiderationsDesign of TSA has been primarily based on empirical approach and as a starting point, one can use the following guidelines [2.8]:y y y y

The aperture width of slot: W 2.05 c/v 2.2.

o.

The taper angle, 2 , is typically 5 to 12o The length of TSA, L, is typically 2 to 12o o

Where c is the velocity of light in free space; v is guided wave velocity along the slot; the operating wavelength in free space. In general, the design of TSA involves two major tasks:y

is

The design of a broadband transition and feed structure with very wide frequency range and low return loss, and Determining the dimensions and shape of the antenna in accordance with the required half power (3 dB) beam width, side lobe, and back lobe etc. over the operating

y


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frequency range. The geometric parameters such as length, width, dielectric thickness, ground plane size, and taper- profile have direct impact on the impedance, directivity, bandwidth, and radiation pattern of the antenna.

2.5 Feedin MechanismThis is the most critical part of Vivaldi antenna design. The impedance matching of the low impedance stripline/ microstrip line to the high impedance slotline is crucial. For a wide bandwidth operation one should design the feed transition with a wide bandwidth. Some of the feed mechanisms are explained below. 2.5.1 Coaxial- Slotline Transition: Coaxial line is useful as a feed structure [2.9] because of its compatibility with the slotline, coplanar microstrip, or balanced microstrip to form wideband transitions; thus, it can be used to excite all the variants of planar TSA described earlier.

Fi 2.6. Coaxial-Slotline Transition The coaxial feed can be directly used to excite a TSA by extending the center conductor over the slotline section of the TSA and anchor the coaxial feed with solder connection to the ground plane as shown in Fi 2.6. The disadvantage is that it is not planar and has high losses at higher frequencies. Further, coaxial line is an unbalanced feed line. All of the currents flow inside the line, i.e. the inner connector and the inside of the shield. Feeding a balanced antenna with unbalanced coaxial feed may cause currents to flow on the outside of the shield, which could results in significant power loss and serious distortion in radiation pattern [2.8].


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2.5.2. Microstrip- Slotline Transition Microstrip line is an unbalanced line and slotline is a balanced line. So feeding a TSA with a microstrip line requires a balanced-to-unbalanced transition (balun). This is essential for broadband antenna performance.

Fi . 2.7 4th order Printed Marchand Balun The most common microstrip/slot transition, the Marchand balun [2.10, 2.11], has demonstrated a VSWR of 2:1 over an octave bandwidth with an integrated wideband Vivaldi antenna (DETSA) [2.12]. The balun consists of four quarter-wave sections with the end opencircuited section extended past the center of the slotline by about one quarter of a guided wavelength ( m). The Fi .2.7 shows the fourth order Marchand balun. Another practical microstrip to slot transition consists of a slot, etched on one side of the substrate, crossing an open circuited microstrip line, located on the opposite side, at a right angle [2.13]. The slot extends to one quarter of a wavelength ( s) beyond the microstrip and the microstrip extends one quarter of a wavelength ( m) beyond the slot [2.12]. Further broadening of the bandwidth can be achieved when the microstrip is terminated by a radial stub and the slot line is terminated by an elliptical shaped cavity [2.15] as shown in Fi 2.8.


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Fi 2.8 Microstrip-Slotline Transition A microstrip- slotline transition with radial stubs illustrated in Fi 2.8 provides a very wide bandwidth [2.16-2.17]. This feeding is used in this project. 2.5.3 CPW Slotline Transition Another way of exciting the slotline in a Vivaldi antenna is to use a coplanar waveguide (CPW) feed. Any transmission line with coplanar conductors can be considered a coplanar waveguide line. The signal line and ground plane are on the same side of a printed circuit board in a coplanar waveguide (CPW). The normal propagating mode on this transmission line is the quasi-TEM mode with the electric fields in the two slots oriented in opposite directions. Fi 2.9 (a) depicts a simple CPW slotline transition. CPW may be used to feed a TSA as shown in Fi 2.9 (b): one half of the CPW transitions into a slotline and feeds a TSA while the other half is terminated in a short circuit [2.18]. This technique may sometimes yield a bandwidth greater than that obtained with a conventional microstrip feed [2.19]. The advantage of using a CPW feed is that it can be used in applications where high circuit density is important as in microwave integrated circuit (MIC) applications. Another benefit is that it has low radiation loss and the center conductor width can be chosen independently for the line impedance, which leads to low dispersion and conductive losses [2.19].


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(a)

(b)

Fi 2.9 Examples of CPW- Slotline transition

2.6 Factors Affectin the RadiationSchaubert et al. [2.20 - 2.26] has done an extensive parametric study of the Vivaldi antenna. The inferences drawn in his work have set major thought process during this project. The major parameters affecting the performance of the Vivaldi antenna are:y y y y y

Dielectric constant of the substrate Diameter of the slotline cavity and the stripline stub Input stripline and slotline width Taper profile Aperture height

The following paragraphs describes briefly about the effect of these parameters. 2.6.1 Effect of Dielectric The performance of a tapered slot antenna (TSA) is sensitive to the thickness and dielectric constant of the dielectric substrate. The presence of a dielectric substrate has the primary effect of narrowing the main beam of the antenna. The substrate thickness primarily affects H plane beam width. As the thickness increases the H-plane pattern becomes narrower. The increase in substrate thickness causes increase in cross polarization level. Increasing the dielectric thickness generally results in increased gain, but with higher side lobes. For good performance, a TSA should have an effective substrate thickness in the range of 1.0025o

teff 1.028 o, where teff = t(

r

- 1), is the effective thickness of the substrate.o,

For substrate thickness above the upper bound of 1.028

unwanted substrate modes

develop which degrade the antenna performance resulting in low efficiency and narrowDepartment of Electronics, CUSAT Page 16


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bandwidth, particularly for dielectrics with high dielectric constants. Further, for millimeterwave operations, the upper bound on the effective thickness, constrains to using mechanically fragile substrates with thickness of only a few hundreds of microns. The methods to overcome these problems by increasing the effective substrate thickness and suppressing the excitation of the surface modes are explained in [2.27 2.28]. Kasturi and Schaubert [2.23] studied the effect of dielectric permittivity on infinite array of single polarized Vivaldi antennas. Fi 2.10 illustrates the result from their study.

Fi 2.10 Effect of Dielectric Permittivity on Infinite Vivaldi Array 2.6.2 Effect of length, width and taper profiles Tapered slot antennas radiate in the end fire direction with symmetric radiation patterns, and have cross polarization level of -20 dB or lower. But it has significant cross polarized radiation in the diagonal plane. The lowest cross polarization level in the diagonal plane is about 10 to 15 dB higher than that of the principle planes which are typically better than -15 dB. In general, the cross polarization characteristics of planar TSA are superior to those corresponding to their antipodal counter part.Department of Electronics, CUSAT Page 17


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Being a travelling wave antenna, the phase velocity and guide wavelength

varies

with any change in the geometrical and material parameters of the antenna, which in turn impacts the radiation characteristics of the antenna. The gain of a TSA increases with the length L of the antenna, typically from a few dB to over 10 dB as L increases from 2-5 reported for long TSA with L greater than 6g

[2.29]. Maximum measured gain of 16-17 dB with radiation efficiency of 80% has beeno

[2.30]. The beam widths, decrease rapidly asg.

the length is increased which is obvious. The H-plane beam width varies more slowly in comparison to the E-plane beam width particularly for L less than 5 polarized Vivaldi antenna array [2.21]. Fi 2.11 illustrates the variation in SWR with variation in tapered slot length at broad side scan for dual

Fi 2.11 Effect of Tapered Slot len th at Broadside Scan onVSWR The taper profile has been found to have strong effects on both the beam width and side lobe level (SLL) of the antenna. In general, the beam widths are narrower for CWSA, followed by the LTSA, and then Vivaldi for antennas with the same length, same aperture size, and on the same substrate. And the side lobes are highest for the CWSA, followed by the LTSA, and the Vivaldi. Being a travelling wave antenna, the H-plane beam width follows1/L dependence, while the E-plane beam width depends more on the aperture width or tapered angle [2.31]. Varying the tapered angle will change the phase velocity and henceg,

which will in turn change the E-plane beam width. Thus, constant beam width in both E

and H plane can be achieved with proper choices of L and tapered angles.


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Fi 2.12 Effect of Openin Rate at Broadside Scan on VSWR Fi 2.12 demonstrates the effect of taper profile on VSWR at broadside scan for a dual polarized Vivaldi antenna array [2.21] Schaubert et.al [2.32] have demonstrated a balanced antipodal antenna with elliptical radiating taper with a constant E and H plane beam width was observed for a LTSA with tapered angles in the range of 15 to 20 degrees [2.33]. 2.6.3 Effect of slot line cavity and stripline stub The slotline in a Vivaldi antenna is usually terminated with a cavity as can be seen in Fi 2.4. The stripline extending past slotline on opposite sides of the substrate is terminated with a stub. The cavity minimizes any reflections at the point of termination of the slotline by acting as an open circuit. At lower frequencies, the slotline cavity act as a poor open circuit, so is the large reflections in the VSWR at low frequencies [2.20]. The cavity can be made either circular or rectangular. The stripline feed is terminated either using a via or a virtual short using a radial stub.


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Fi 2.13 Effect of Slotline Cavity Diameter at Broadside Scan on VSWR Fi 2.14 illustrates the effect of slotline cavity diameter on the VSWR for a dual polarized Vivaldi antenna array from Chio and Schubert [2.21].

2.7 SummaryIn this chapter Tapered Slot Antennas have been studied and explained in detail. This chapter discusses the work carried out worldwide on this structure and explains the various electrical traits of the structure and its variants. Various feeding techniques to obtain a good impedance match over a wide bandwidth have been explained.

2.8 References[2.1] P.J.Gibson, The Vivaldi Aerial, Proc.,9th European Conf., Brighton, U.K. 1979, pp. 101-105. [2.2] S.N.Prasad and S. Mahapatra, A Novel MIC Slotline Aerial, Proc. 9th European Microwave Conf., Brighton, U.K. 1979, pp. 120 -124 [2.3] L.R.Lewis, M.Fassett and J.Hunt, A Broadband Stripline Array Element, IEEE APS International Symposium, Atlanta, GA, pp-335-337, 1974


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[2.4]

Pranay Acharya, Hans Ekstrom, Steven S Gearhart, Stellan Jacobson, Jaokim F Johansson, Erik L Kollberg, Gabriel M Rebeiz, Tapered Slot Antenna at 802 GHz, IEE Transactions on Microwave Theory and Techniques, vol.41, no.10, October 1993, pp. 1715-1719

[2.5]

E.Gazit, Improved design of the Vivaldi antenna, IEEE Proc., Part H, vol. 135, No.2, 1988, pp. 89-92.

[2.6]

J D S Langely, P S Hall, and P.Newham, Balanced antipodal Vivaldi antenna for wide bandwidth phased arrays, IEEE Proc. Antenna Proag., vol. 143, no. 2, April 1996, pp. 97-102.

[2.7]

D.H. Schaubert, Endfire Tapered Slot Antenna Characteristics, ICAP89, 4-7 April 1989, vol.1, pp.432-436

[2.8] [2.9]

Richard Q Lee, Notch Antennas, http://gltrs.grc.nasa.gov P.Knott and A.Bell, Coaxially fed Tapered Slot Antenna, Electronics Letters, vol.37, no.18, Aug. 2001, pp. 1103-1104.

[2.10] N. Marchand, Transmission Line Conversion, Electronics, vol.17, 1944, pp.142145 [2.11] Velimir Trifunovic, Branka Jokanovic, Review of printed Marchand and Double Y Baluns: Characteristics and application, IEEE Transactions on Microwave theory and techniques, vol.42, No.8, August 1994, pp. 1454-1462 [2.12] A.B. Smolders and M.J. Arts, Wideband Antenna Element with Integrated Balun, 1998 IEEE AP-S Int. Symposiums Digest, Atlanta, USA, June 1998 [2.13] Knorr J.B, Slotline Transitions, IEEE Trans., vol. MTT-22, 1974, pp. 48-554. [2.14] Schuppert, Microstrip/ Slotline Transitions: Modeling and experimental

investigations,IEEE Trans. On Microwave theory and techniques, vol. MTT-36, 1988, pp. 1272-1282. [2.15] Oraisi and Jam, Optimum Design of TSA profile, IEEE Trans. On antennas and propagation, vol.51, no.8, August 2003, pp. 1987-1995.


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[2.16] M. M. Zinieris, R. Sloan and L.E. Davis, A broadband microstrip to slotline transition, Microwave and Optical technology letters, vol.18, no.5, August 5 1988 [2.17] A.H. Atwater, The design of the radial line stub: A useful microstrip circuit element, Microwave J., vol. 28, 1985, pp. 149 -156. [2.18] A.Nesic, Endfire slotline antennas excited by a coplanar waveguide, 1991 IEEE AP-S International Symposium, vol.2, Ontario, pp.700-702, 1992. [2.19] R.Q. Lee and R. N. Simons, chapter 9 in Advances in microstrip and printed antennas, John Wiley and Sons, 1997. [2.20] J. shin and D. H. Schaubert, A parameter study of stripline fed Vivaldi notch antenna arrays, IEEE Trans. On Antennas and Propagation, vol. 47, no.5, May 1999, pp.879-886. [2.21] D.H. Schaubert and T.H. Chio, Parameter study and design of wideband, widescan dual polarized tapered slot antenna arrays, IEEE Transactions on antennas and propagation, vol. 48, no. 6, June 2000,pp. 879-886 [2.22] S.Kasturi, A.O. Boryssenko and D.H. Schaubert, Infinite arrays of tapered slot antennas with and without dielectric substrate, proceedings of the 2002 Antenna Applications Symposium, Monticello, IL., Sept. 2002, pp. 372-390 [2.23] S.Kasturi and D.H.Schaubert, Effect of dielectric substrate on infinite arrays of single polarized Vivaldi antennas, proceedings of the 2003 Antenna Applications Symposium, 2003. [2.24] D.H. Schaubert, A.O. Boryssenko and T.H Chio, Analysis of finite arrays of wideband tapered slot antennas, Proceedings of the 2002 URSI General Assembly, Maastricht, The Netherlands, 2002. [2.25] D.H. Schaubert and T.H. Chio, wideband Vivaldi arrays for large aperture antennas, NFRA International Conference on Perspectives in radio astronomy: Technologies for large antenna arrays, Dwindeloo, Netherlands, pp. 49-57, Apr 1999. [2.26] D.H Schaubert, S.Kasturi, A.O Borryssenko and W.M Elsallal, Vivaldi antenna arrays for wide bandwidth and electronic scanning.


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[2.27] J.S. Colbum and Y. Rahmat Samii, Printed antenna pattern improvement through substrate perforation for high dielectric constant material: An FDTD Evaluation, Microwave and optical technology letters, vol. 18, no.1, May 1998, pp. 27-32. [2.28] Thomas J. Ellis and Gabriel M. Rebiz, MM-wave tapered slot antennas on micromachined Photonic Band gap dielectrics, 1996 IEEE MTT-S International Symposium Digest, pp. 1157-1161. [2.29] Kai Fong Lee and Wei Chen, Advances in microstrip and Printed antennas, Wiley Interscience, New York 1997. Chapter 9, p. 443. [2.30] K. Sigfrid Yngvesson, T.L. Korzeniowski, Young-Sik Kim, Erik L. Kollberg, and Jaokim F. Johansson, The tapered slot antenna- A new integrated element for millimeter wave application, IEEE Trans. Microwave and Techniques, vol. 37, no. 2, Feb. 1989, pp. 365-374. [2.31] T.Thungren, E.L. Kollberg and K.S. Yngvesson, Vivaldi antennas for single beam integrated receiver, Proceedings of the 12th European Microwave Conference, 1982, pp. 474 - 481. [2.32] T.L. Korzeniowski, D.M. Pozar, D.H. Schaubert, and K.S. Yngvesson, Imaging system at 94 GHz using tapered slot antenna elements, Proceedings of the 8th International Conference on Infrared and Millimeter waves, 1983. [2.33] P.S.Kooi, T.S.Yeo, and M.S.Leong, Parametric studies of the Linearly Tapered Slot Antenna (LTSA), Microwave and Optical Tech. Lett., vol.4, no.5, Apr 1991, pp. 200-206


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Chapter 3

Analysis Methods

3.1 IntroductionThe numerical electromagnetic simulators solve the Maxwells equations. The Maxwells equations are:

(3.1) (3.2) (3.3) (3.4)

With associated consecutive equations (3.5) (3.6) The actual solution of the Maxwell equations is complex, and for realistic problems, approximations are usually required. The numerical approximation of Maxwells equations is known as Computational Electromagnetics (CEM)

3.2 Numerical MethodsThe differences between various numerical techniques reside essentially in the following aspects [3.1]:y y y

The electromagnetic quantity that is being approximated; The expansion functions that are used to approximate the unknown solution; The strategy employed to determine the coefficients of expansion functions.


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The solution of an electromagnetic problem may require finding the electric or magnetic field, a potential function, or a distribution of charges and/or currents. While these quantities are related, they have different properties; hence, problem formulations for field, potential, and charge or current solutions are different. Finding fields or potentials will require expansion functions in the field space (domain methods), while unknown charge or current distributions are expanded into functions defined mostly on boundaries (boundary methods). Finally, there exists a variety of strategies for computing the unknown coefficients, which involve the inversion of large matrices, implicit and explicit iteration schemes, evolutionary algorithms etc. The various existing numerical methods employ different combinations of these aspects. The widely used full-wave techniques are:y y y

Method of Moments (MoM) Finite Difference Time Domain Method (FDTD) Finite Element Method (FEM) The electromagnetic simulation tool HFSS which is used in this project, is based on

FEM. In the following sections, a brief description of FEM and HFSS are given.

3.3 Finite Element Method (FEM)The Finite Element Method (FEM) is one of the best-known methods for the solution of partial differential equations. It is a method for solving a differential equation subject to certain boundary values. The FEM may be derived on two view points: one uses variational analysis, the other weighted residuals. Both start with the partial differential equation (PDE) form of Maxwells equations. The former finds a variational functional whose minimum corresponds with the solution of the PDE, subject to certain boundary conditions. The latter also starts with the PDE form of Maxwells equations, and then introduces a weighted residual (error); using Greens theorem, one of the differentials in the PDE is shifted to the weighing functions [3.2, 3.3-3.5]. For most applications, these procedures result in identical equations. In both cases, the unknown field is discretized using a finite element mesh; typically, triangular elements are used for surface meshes and tetrahedrons for volumetric meshes, although many other types of elements are available.Department of Electronics, CUSAT Page 25


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This represents a very large class of electromagnetic engineering applications of the FEM, including antenna, radar cross-section, microwave circuit and periodic structure analysis. As with the FDTD method, the FEM does not include the radiation condition. For closed regions (e.g. waveguide devices or cavities) this is of no concern. However, for open regions (e.g. radiation or scattering problems), this requires special treatment, and this must be incorporated using either an artificial absorbing region within the mesh (the numerical analogy of an anecholic chamber) or using a hybridization with the MoM to terminate the mesh. Traditionally, the FEM has been formulated in the frequency domain, although time domain formulations have also been used for specialized applications. Ansofts HFSS package is widely regarded as the market leader among the commercially available packages based on FEM. A fairly recent entry, FEMLAB, has also attracted users. The strong points of the FEM are the following [3.2]:y y

Very straightforward treatment of complex geometries and material in-homogeneities. Very simple handling of dispersive materials (i.e. materials with frequency dependant properties). Ability to handle eigen problems. Potentially better frequency scaling than the MoM although the requirement to mesh a volume rather than a surface means that the number of unknowns in the problem is usually much larger. Straightforward extension to higher-order basis functions. It is also possible to use conformal elements to better approximate curved geometries. The weak points of the FEM include the following [3.2]:

y y

y

y

Inefficient treatment of highly conducting radiators when compared to the MoM (due to the requirement to have some mesh between the radiator and the absorber). The FEM meshes can become very complex to implement than the FDTD method. In conclusion, the FEM is the preferred method for microwave device simulation and

y

eigen problem analysis. Using FEM / MoM hybrids, scattering problems involving


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electromagnetically penetrable media and specialized antenna problems can be accurately and efficiently solved.

3.4 Hi h Frequency Structure Simulator (HFSS)The basic approach of FEM is to divide a complex structure into smaller sections of finite dimensions known as elements. These elements are connected to each other via joints called nodes. Each unique element is then solved independently of the others thereby drastically reducing the solution complexity. The final solution is then computed by reconnecting all the elements and combining their solutions. These processes are named assembly and solution respectively in the FEM [3.3].

Fi 3.1 Tetrahedral Element FEM is the basis of the simulation in HFSS [3.6]. HFSS divides the geometric model into a large number of tetrahedral elements, see Fi 3.1. Each tetrahedron is composed of four equilateral triangles and the collection of tetrahedra forms what is known as the finite element mesh. At each vertex of the tetrahedron, components of the field tangential to the three edges meeting at that vertex are stored. The other stored component is the vector field at the midpoint of selected edges, which is also tangential to a face and normal to the edge. Using these stored values, the vector field quantity such as the H-field or the E-field inside each tetrahedron is estimated. A first-order tangential element basis function is used for performing the interpolation. Maxwells equations are then formulated from the field


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quantities and are later transformed into matrix equations that can be solved using traditional numerical techniques. Fi 3.2 illustrates the meshing of a Vivaldi antenna in HFSS.

Fi 3.2 Meshin of Vivaldi antenna in HFSS 3.4.1 Size of Mesh Vs. Accuracy There is a trade-off among the size of the mesh, the desired level of accuracy, and the amount of available computing resources. The accuracy of the solution depends on the size of each of the individual elements (tetrhedron). Generally speaking, solutions based on meshes using thousands of elements are more accurate than solutions based on course meshes using relatively few elements. To generate a precise description of a field quantity, each element must occupy a region that is small enough for the field to be adequately interpolated from the nodal values. However, generating a field solution involves inverting a matrix with approximately as many elements as there are tetrahedral nodes. For meshes with a large number of elements, such an inversion requires a significant amount of computing power and memory. Therefore, it is desirable to use a mesh fine enough to obtain an accurate field solution but not so fine that it overwhelms the available computer memory and processing power. To produce the optimal mesh, HFSS uses an iterative process, called an adaptive analysis, in which the mesh is automatically refined in critical regions. First, it generates a solution based on a course initial mesh. Then, it refines the mesh in areas of h error igh density and generates a new solution. When selected parameters converge within a desired limit, HFSS breaks out of the loop.Department of Electronics, CUSAT Page 28


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3.4.2 The HFSS Solution Process To calculate the S-matrix associated with a structure with ports, HFSS does the following:y y

Divides the structure into a finite element mesh. Computes the modes on each port of the structure that are supported by a transmission line having the same cross-section as the port. Computes the full electromagnetic field pattern inside the structure, assuming that one mode is excited at a time. Computes the generalized S-matrix from the amount of reflection and transmission that occurs. The resulting S-matrix allows the magnitude of transmitted and reflected

y

y

signals to be computed directly from a given set of input signals, reducing the full 3D electromagnetic behavior of a structure to a set of high frequency circuit parameters. 3.4.3 HFSS Antenna Design Kit HFSS Antenna Design Kit is a guide that creates parametric HFSS models for a variety of common antenna types [3.5]. It can be used to easily generate antenna models and assist in learning proper usage of HFSS for antenna design. The parameters of the initial model can be easily modified. The taper profile of the antenna is drawn with the help of Antenna Design kit.

3.5 SummaryIn this chapter, at first, the various full wave techniques widely used for computational electromagnetics is specified. The FEM method is explained briefly. The simulation tool used in this project, which is Ansofts HFSS, has been explained. Next chapter will describe about the design of Tapered slot antenna, and its simulated results.

3.6 References[3.1] Daniel G. Swanson Jr., Wolfgang J. R. Hoefer, Microwave Circuit Modeling Using Electromagnetic Field Simulation, Artech house, Inc., 2003


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[3.2]

David B. Davidson, Computational Electromagnetics for RF and Microwave Engineering, Cambridge University Press, 2005

[3.3]

Anastasis C. Polycarpou, Introduction to the Finite Element Method in Electromagnetics, Morgan & Claypool Publishers, 2006

[3.4]

Susanne C. Brenner, L. Ridgway Scott, The Mathematical Theory of Finite Element Methods, Third Edition, Springer, 2008

[3.5]

John L.Volakis, Arindam Chatterjee, Leo C. Kempal, Finite Element Method for Electromagnetics, IEEE Press, 1998

[3.6]

http://www.ansoft.com


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Chapter 4

Desi n of Tapered Slot Vivaldi Antenna

4.1 IntroductionComputer Aided Design (CAD) of the Vivaldi antenna can be described in the following hierarchical manner: 1. Choice of substrate material 2. Choice of primitive antenna dimensions to start with [4.1]. 3. Modeling and Optimization of antenna using CAD software. In this project, the Finite Element Method (FEM) based software by Ansoft Corporation; HFSS v12 [4.2] has been used.

4.2 Choice of substratePerformance of tapered slot antennas is very sensitive to thickness and dielectric constant of the substrate. The acceptable range of dielectric thickness for good antenna operation was found to be 0.00250

teff 0.028 o, where teff = t( r - 1)

Bandwidth and efficiency are generally having an inverse relationship with the substrate dielectric constant. The line width becomes smaller with increase in dielectric constant. So, it becomes difficult to realize microstrip lines especially at higher frequencies. Substrate thickness and dielectric constant are chosen such that microstrip / stripline trace is realizable. Line width increases with substrate thickness. Hence, an optimum thickness for the substrate has to be chosen for the correct design. FR4 epoxy substrate with dielectric constant,r

= 4.4, thickness of 1.6mm and dielectric loss tangent of 0.02 has been chosen in

this project.


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4.3 Tapered Slot Antenna Confi urationsThe design parameters of the proposed TSA are shown in Fi . 4.1. The top layer shows the microstrip line and the series radial stub used for feeding the tapered slot antenna. The bottom layer indicates the exponential taper profile [4.3], which is drawn with the help of an HFSS tool called Antenna Design Kit [4.2].

Fi . 4.1 Confi urations of the proposed tapered slot antenna Given the highest frequency of operation (fH), the width, Wtaper of the tapered slot antenna should satisfy equation given below to circumvent the grating lobes of Vivaldi array.

Wtaper

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