EMC Characteristics of

Lighting Systems

Zoran Zegarac

This thesis is presented for the Degree of Master of Engineering

of The University of Western Australia

School of Electrical, Electronic and Computer Engineering

The University of Western Australia

Crawley, WA 6009, Australia

2013

Statement

I, Zoran Zegarac from 25 Westminster Road, Leeming, WA, hereby declare that my thesis

“EMC Characteristics of Lighting Systems” is my original work and :

• all sources are acknowledges.

• the thesis has not previously been accepted for any other degree in this or another

institution.

• the thesis has been substantially accomplished during enrolment in the degree.

• the thesis is wholly my own composition.

• a full electronic copy of the thesis is available on request.

____________________________

Zoran Zegarac, May 2013

Abstract

This thesis examines Electromagnetic Compatibility (EMC) characteristics of lighting

systems. The issues examined have become more significant with the rapid

development of both lighting systems and general electronics devices such as Blue

Tooth and RFID tags.

Every commercial and industrial building has lighting systems consisting of wiring,

electronic ballasts, dimmers, luminaires and the like. Except for the occasional need

to change a lamp, typically each lighting system remains unaltered during the life of

the building which can be several decades.

Computers and related electronic products which are operated within such buildings

are continually being updated or replaced as they become obsolete and new

products become available. Computer systems are expected to perform flawlessly

at all times and yet Electro Magnetic Interference (EMI) emanating from lighting

systems can cause such systems to experience disruption. Such disruption causes

delay and cost to commercial enterprise. A luminaire must necessarily be located

very close to a work area. Data transmission can be adversely affected by coupling

of radiated electromagnetic field from a luminaire. Even if there are no radiation

emissions, conducted emissions are almost always present, transmitted via power

lines and thereby exposing nearby computers to the risk of disruption.

Where EMI causes disruption, it may become necessary to modify the entire lighting

system in the work area or building. Often such remedial work will incur substantial

costs by way of labour and replacement parts, as well as the business costs

associated with computer down time in the work environment.

The concepts and results presented in this thesis will provide a means to reduce the

impact of certain EMI from lighting systems in computer work environments by

ensuring that luminaires are properly characterized to screen out those which

exhibit unpredictable behaviour. The scope of this thesis has been limited to

conducted emissions from linear fluorescent lamps in the high frequency range

below 30 MHz. A novel approach that uses a combination of measurements and

computer simulation for the luminaires was adopted and validated. Its main

advantage lies in replacing a significant volume of measurements in the EMC

laboratory with computer simulations. This method could be extended for use with

other types of luminaires or even other devices.

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Table of Contents

1 Introduction ............................................................................................................. 5

1.1 Background and Motivation ............................................................................. 5

1.2 Research Methodology .................................................................................... 6

1.3 Thesis Outline ................................................................................................ 10

1.4 Thesis Contributions ...................................................................................... 14

2 Measurements ...................................................................................................... 15

2.1 Preliminary Time and Frequency domain measurements ............................. 15

2.2 Measurements on a CISPR TR 30 like test bench ........................................ 19

2.3 Measurements for CISPR 15 like model ........................................................ 25

2.4 Measurements Summary ............................................................................... 30

3 Numerical Model development ............................................................................. 31

3.1 Modelling in general ....................................................................................... 31

3.2 Numerical Model ............................................................................................ 32

3.3 CISPR TR30 like model ................................................................................. 33

3.3 CISPR15 like Model ....................................................................................... 36

3.4 Numerical Model Summary ............................................................................ 38

4 The N-Port Model Development ........................................................................... 39

4.1 The N-Port Model Description........................................................................ 39

4.2 Estimating Port’s Termination Impedances ................................................... 40

4.3 Excitation scenarios ....................................................................................... 46

4.4 The N-Port Model summary ........................................................................... 48

5 Simulation ............................................................................................................. 50

5.1 Motives for simulation .................................................................................... 50

5.2 Model similar to CISPR TR30 Variations ....................................................... 50

5.2 CISPR 15 like Model with variable gear tray width ........................................ 53

5.3 Testing of emissions over 30 MHz ................................................................. 58

5.4 Simulation Summary ...................................................................................... 60

6 Conclusion and future work .................................................................................. 61

6.1 Conclusions ................................................................................................... 61

6.2 Future work .................................................................................................... 61

References .............................................................................................................. 63

Appendix A .............................................................................................................. 65

Current probes and Equipment used ................................................................... 65

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Laboratory equipment used ................................................................................. 66

Ballast test samples ............................................................................................. 66

Appendix B .............................................................................................................. 67

Formulas for time domain measurements (Chapter 2.1) ..................................... 67

Formulas for frequency domain measurement (Chapter 2.1 and 2.2) ................. 67

Conversion from linear to logarithmic values and back ....................................... 67

EMI Receiver sweep time calculations ................................................................ 68

Additional time domain results ............................................................................. 68

Appendix C .............................................................................................................. 70

Wires and surfaces modelling .............................................................................. 70

Appendix D .............................................................................................................. 74

Post-processing of preliminary testing ................................................................. 74

Appendix E .............................................................................................................. 82

Emergency lighting measurements...................................................................... 82

Appendix F .............................................................................................................. 96

Some of Matlab Code used in Thesis .................................................................. 96

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1 Introduction

1.1 Background and Motivation

The Western Australian Telecommunication Research Institute (WATRI) had been established in October 2002 to provide a supporting environment for a range of cutting edge research and development projects. Until its closure in 2009, WATRI was a pinnacle of innovation work in WA. One of its core functions was to collaborate with Australian industry and government on a range of projects. WATRI was approached by the Lighting Council of Australia (LCA), the peak body for Australia’s lighting industry, for help in developing a new method for verification and testing of lighting products.

Lighting systems must be well characterised from the Electro Magnetic Compatibility (EMC) standpoint, as they are an important contributor to electromagnetic noise. Many parts, like dimmers, ballasts and lamps are getting more and more sophisticated. In conjunction with wiring, these are complex electromagnetic systems that are expected to function flawlessly for at least a couple of decades, while computer and mobile technology becomes obsolete and is renewed every 3-5 years.

Lighting systems and their electromagnetic emissions pose a serious EMC issue. This has been recognized by regulatory bodies worldwide. Emission limits and methods for their measurement have been regulated and prescribed. In Australia and New Zealand, the guiding document is AS/NZ CISPR 15 [1]. This standard specifies a test set-up and way to measure disturbance voltages. The noise level for luminaires is specified and defined in term of mains terminal disturbance voltage.

No luminaire can be sold in Australia without passing these tests. If a device or a group of devices fail, consequences for a manufacturer and/or reseller could be the banning of sales or withdrawal of type approval.

Unfortunately, compliance testing is time consuming, expensive and does not provide many answers. Further to this, AS/NZ CISPR 15 only looks at a luminaire as a whole, not considering the contribution of individual parts to the emissions.

These problems prompted the investigation presented in this thesis. There is a clear need for a research project into EMC characterisation of lighting systems. Cheaper, quicker and more flexible tools are needed to facilitate approval process for luminaires.

Fundamental objectives and questions that this thesis aims to answer are:

• What are the typical emissions and what are the limit values, for example frequency range, voltages and currents?

• What are the inherent source characteristics, i.e. what kind of internal equivalent circuit describes the electromagnetic behaviour or the sources best, and what does that mean in respect to their ability to drive currents through attached cables?

• What is the influence of cabling layout and gear tray shape on emissions level?

The expected results characterise EMC in lighting systems in respect to:

• Noise sources • Coupling mechanism • Impedance characteristics

Experience gained in preparing this thesis should not only facilitate EMC characterisation, but also help in developing guidelines for design and installation of lighting systems.

1.2 Research Methodology

Figure 1-1 illustrates the major components of a linear fluorescent luminaire. This type of luminaire is most frequently used in commercial and residential buildings. It consists of a fluorescent lamp, a gear tray, wiring and electronic ballast.

Disturbance noise is the result of coupling between the mains power wires and wires connecting the ballast output with the lamp ends. It is interesting to note that none of the documents regarding emission limits and their measurement provide any information on coupling between the wires. The coupling process is also dependent on frequency [2] and there is evidence showing that, at least at some frequencies, emissions increase with increased wire separation [3].

Regarding disturbance emissions (voltages and currents), the most important part of a luminaire is the electronic ballast. Ballast is used to keep the lamp current constant by regulating lamp voltage and deactivating the lamp if there is any problem [4]. Regulation of the current is crucial to avoid variations is the amount of light emitted by a fluorescent lamp. As the ballast is an active circuitry part, it is considered to be the main noise source.

Figure 1-1 A linear fluorescent luminaire with a lid removed

Following the AZ/NZ CISPR 15 standard, a complete luminaire is set-up on a test bed and its conducted emissions measured in the prescribed manner. The problem with this approach is that each combination of these four parts represents a specific luminaire configuration that needs to be tested.

For example, the gear tray in Figure 1-1 could be replaced with a gear tray produced by a different manufacturer. Even if the dimensions were the same, the standard testing procedure requires that luminaire be tested again. Further, with 2 different ballasts, 3 different lamps, 2 methods of wiring and 2 different gear trays, it

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is possible to make 2x3x2x2=24 different luminaire types. This represents a significant cost and possible time-to market delay.

This project aims to use a combination of electronic ballast measurements, a numerical model with estimated parameters, and develop an N-Port model which characterises linear fluorescent luminaries reasonably well from a disturbance emissions point of view [5] [6].

Unlike standard conducted emissions testing, which treat every combination of a ballast, wiring and gear tray as a single luminaire, the approach taken in this thesis examines ballast and the rest of the luminaire as two separate issues.

An impedance characteristic of the noise source determines the coupling in the different scenarios. In this way, the coupling and subsequent emission level could be predicted from the model, without testing a luminaire. In other words, instead of assembling and testing every type of luminaire, it is only necessary to measure and characterise the noise source, thus reducing the need for a very large number of tests. So instead of 24 tests in the previous example, it is only necessary to conduct 2 tests.

The N-Port model developed in this thesis is not intended to provide absolute prediction for disturbance voltages on lamp terminals or to remove the need to test luminaires completely, but to examine the effect of ballast impedances, wiring layout and gear tray shape on emissions. These effects are not mentioned in AS/NZ CISPR 15.

Referring to Figure 1-1, on the ballast input there is a standard mains power 220-240 VAC/ 50 Hz voltage. Voltage on the ballast output has a typical value of 690 Volts/45 kHz and a rich harmonic content, due to its trapezoidal nature [6] [7]. Part of the wiring (pink area in Figure 1-1), represents the coupling mechanism whereby the noise spreads from the wires connected to the lamp ends to the wires on the ballast input, and then further down the power lines.

A schematic diagram based on Figure 1-1 is shown in Figure 1-2. It shows 2 ports on the input side of ballast, and 4 ports at the output. The active and neutral voltages, and , measured across 50 Ω ports of a Line Impedance Stabilization Network (LISN), are the disturbance signals of interest. It is the value of disturbance voltages measured on these terminals that is compared with the AS/NZ CISPR 15 limits. That is why these voltages are also computed using the N-Port model developed in this thesis.

Although there might be some coupling in the ballast, its inputs and outputs should be unrelated, and the ballast should inject signals only into 4 wires on its output side, to the lamp Far and Near Ends. But, because of coupling, the electronic ballast injects signals on both its input and output, in all of its 6 terminals into the wiring network, either due to direct propagation from input terminals or due to capacitive and inductive coupling between the lamp and power wires.

Figure 1-2 Lamp schematic diagram

There are two ways for noise to spread:

• Noise signals injected into power side terminals (Ballast In), are directly transmitted to the observation points, but still may experience a distortion. This is because the length of the connection, at the highest frequency of interest (30 MHz in this case), can be a significant portion of the wavelength. The same electronic ballast can therefore generate different disturbance voltages, depending on the characteristics of the transmission line between ballast and LISN, the grounding arrangements of the ballast, and the capacitive coupling between the body of the luminaire and reference potential (this capacitance is not shown in the diagram).

• Noise signals injected into the lamp terminals (Ballast Output) can couple capacitively and inductively into the power wires and thus also generate a disturbance voltage at the LISN. Again, the same ballast must be expected to produce a different result when the wiring is modified, and thus capacitances and inductances are changed.

The results from a conducted emission measurement, obtained for electronic ballast in one particular luminaire can therefore not be used, without further analysis, to predict the emission when the same ballast is used in another luminaire with different wiring or with a different shape of the gear tray.

To illustrate the model from an N-Port network point of view, the schematic in Figure 1-2 is re-drawn in Figure 1-3.

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Figure 1-3 Luminaire as N-Port model

The gear tray and wiring, including 50 Ω terminations as representation for the LISN, form an N-Port network [4]. In this case the model has 6 ports; each port is excited by a source. If the characteristics of these sources are known (internal impedance and open circuit voltage, for instance, i.e. its Thevenin equivalent), and the impedance characteristics of the N-Port network are also known, then the currents injected into the ports can be predicted. More details are given in chapter 3.

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∑ ∑

For linear, passive systems, port voltages and currents are related by the

(2)

In this case the elements of the impedance matrix can be obtained by numerical

,

Furthermore, if the transfer functions between the disturbance voltage at the LISN and currents injected into each port are known then the disturbance voltages and

can be calculated.

The following equation expresses the previous discussion in mathematical form:

(1)

where is the number of ports , 6 in this case and are the actual currents injected in each port, and are the weighting coefficients for these port currents. The disturbance voltages are linear combinations of all port currents, and in order to use (1) port currents, , and the coefficients and must be obtained.

impedance matrix [Z]:

simulation using the Method of Moments. Exciting the network at one port, i, with a current , and determining the open circuit voltages at all other ports provides the elements for one column, i, of the impedance matrix (1):

(3)

). (Vj: Voltage at port j while feeding port i with current Ii

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llows the computation of the weighting coefficients and :

,

Recording the disturbance voltages and as well a

, (4)

When sources and loads are also connected to this network the following equation

ll source voltages, and [ ] is a diagonal matrix containing the impedances, res ect ly. Po t curre

3) a

] and [ ] are the hevenin equivalents of the ballast terminals.

cteris the n in Chapter 2. Short circuit currents

can be measured based on CE02, a conducted emission measurement procedure

particular se values may be

unreliable or not available.

-linear effects. The electronic ballast is a highly non-linear element but this is of no concern, as this component is not part of the N-

e terminal does not influence the open circuit voltage or internal impedance at the other terminals.

the organisation of the thesis is as follows:

• Chapter 1 gives an introduction on the motivation for the project, ethodology. The

main standard is presented and its limitations exposed. It outlines the main

must be satisfied:

(5)

[ ] contains aload impedances and internal source p ive r nts [I] and port voltages [v] in the equations (1) and ( re identical, and by combining both equations the port currents can be expressed in terms of the N-Port-impedance matrix [Z], the load impedances, and the source voltages:

(6)

Connected to the ports is the electronic ballast, and [T

Measurements can be made to find the circuit chara tics of electronic ballast. More details on Measurement are give

in Mil-Std 461 C. In this case a suitable capacitor is used to short-circuit the terminal for the frequency range of interest and the current through this capacitor is measured. The open circuit voltage can be obtained by multiplying measured currents with the internal impedance of the source.

There are also some possible drawbacks in using this approach:

• Impedance and short circuit currents are subject to an uncertainty; in the measurement of their complex values is difficult, and pha

• The model assumes a linear system. The fluorescent lamp is part of the network, and will introduce some non

Port network.

• The terminals of the ballast are assumed to be independent of each other, i.e. the load on on

1.3 Thesis Outline

Referring to Figure 1-4,

emphasizes the project objectives and proposed research m

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rned with measurements, whereby the luminaires were haracterised by conducting a series of measurements to get voltages and

• odel consists of surfaces

nd wires, and is a part of a numerical model. Surface meshing was a

cent lamps” [8].

2.

Usicomsta to calculate weighting coefficients. These

• ighting coefficients and measured

urrents, detailed analysis of a model was performed and results verified.

focus of this thesis which seeks to develop the N-Port model of a luminaire, and to perform a set of simulations to determine its behaviour. The terminology and definitions that are used throughout the thesis are introduced.

• Chapters 2 and 3 are undertaken independently, as per Figure 1-4. Chapter 2 is conceccurrents. Although the measurement procedure is similar to the one required by CISPR conducted emissions measurement standards [1] [8] or similar [9] [10], the aim was not to see if the device would fail or pass this test, but to extract values that are later used for luminaire characterisation. Currents on wires were measured with different types of current probes; voltages on the LISN were also measured. Measured values are analysed and used for N-Port Model excitation. Problems encountered were a relatively small signal strength and wide frequency range, from 9 kHz to 30 MHz. To overcome this, a set of current probes was made and tested. Chapter 3 develops the numerical model, on a basis of a simplified representation of the luminaire. The 3D geometry maseparate issue and it is explained in the Appendix C. It should be noted that the numerical model is used only to produce input coefficients for the N-Port model, and not for computing predicted disturbance LISN voltages. Two types of numerical models have been developed in order to make the characterisation of lighting systems simple and repeatable. 1. A model labelled “TR 30 like model” relates to CISPR TECHNICAL

REPORT 30 (TR30) “Test method on electromagnetic emissions from electronic ballasts for single- and double-capped fluoresThe ballast is mounted on a reference luminaire, which is practically a test bench, as is further explained in chapter 2.2. A model referred to as “CISPR 15 like model” is based on AS/NZ CISPR 15 “Limits and methods of measurement of radio disturbance characteristics of electrical lighting and similar equipment” [1]. This method refers to testing of a luminaire “as is”, whereby ballast and wiring are enclosed in a gear tray. ng the Method of Moments [11], currents, voltages and impedances were puted and used as raw data for the N-Port model. In the post-processing

ge, these results are used coefficients are a form of impedances and they determine the extent to which currents at each port contribute to disturbance voltages. Numerical problem solution must satisfy some basic criteria, like smooth current distribution, boundary conditions met etc. Chapter 4 discusses in detail the main part of the thesis, the N-Port model. Using arbitrary impedances, computed wec

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the changes in utput values observed. The main objective is to provide an insight into

chapters and summarises the work. Future work and commendations are identified.

These input values for an N-Port model are critical, so this is a possible Garbage In Garbage Out (GIGO) step. It should be noted that it is difficult to extract or calculate port impedances. In this thesis they are estimated using a combination of knowledge of particular circuitry and trial - and- error. For example, from the wiring and functional position, it can be estimated that a particular port has the value between 10 Ω and 1 k Ω. Disturbance voltages for a particular combination of ballast and a luminaire are calculated and compared with the measured LISN voltages. If the differences are too large, it can be argued that the termination impedance value was incorrect and the process needs to be repeated with a different impedance.

• Chapter 5 contains a discussion of the N-Port model implementation in practice. Several “What if“ scenarios were modelled andocoupling mechanisms and suggest that the narrowest gear tray is the worst case scenario. Some prediction for the higher frequency range was also proposed.

• Chapter 6 concludes the thesis by consolidating the results from the previous re

While simplified, the flowchart in Figure 1-4 logically depicts the thesis as a group of interrelated steps. Chapter headings in the thesis coincide with the respective blocks in the diagram.

Figure 1-4 Thesis outline

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1.4 Thesis Contributions

The key contributions of this thesis are:

• An N-Port model has been developed and tested. • Some of the main ballast parameters, input and output impedances, have

been examined and their effect on conducted emissions characterised. • Confirmation that the EMC characteristics of lighting systems can be

reasonably and accurately predicted. • Some practical case studies for different luminiares were performed.

Using a combination of electronic ballast measurements, a numerical model and estimated parameters, it is possible to develop an N-Port model which characterises a linear fluorescent luminaire very well from the EMC stand point.

The model developed in this thesis was never intended to provide a 100 % accurate prediction for disturbance voltages on lamp terminals. Its main purpose was to characterise the coupling process between the wires and the gear tray.

In summary:

Firstly, currents on the ballast output were measured. A numerical model was used to compute the impedance matrix. Together with estimated impedances, this matrix is a basis for a simulated transfer function (N-Port) model. The N- Port model is excited by injecting measured ballast output currents. Results from The N-port model are calculated LISN voltages which reasonably well represent LISN voltages measured in the laboratory.

2 Measurements

2.1 Preliminary Time and Frequency domain measurements

While the model development required frequency domain data, the first measurements were done in time domain. The motivation for these preliminary measurements was not to extract luminaire parameters, but to get a better insight into the nature of disturbance emissions and compare it with a luminaire with a non-fluorescent lamp.

The basic set-up was quite simple, as per Figure 2-1, where the upper picture shows the set- up with a linear fluorescent lamp and the lower with a table lamp with an incandescent bulb.

Figure 2-1 Measurement set-up with a fluorescent lamp (top) and an incandescent light bulb (bottom)

More details on measurements, equipment used and most of the results are set out in Appendix D. Only one of the results is shown in Figure 2-2. A current probe was positioned on one of the wires at the ballast output, as shown in Figure 2-1. In this way, differential mode currents can be detected. In case that the probe were to enclose both wires, any differential mode current would be largely cancelled by the probe.

The signal was sampled with 2000 points, from -50 to +50 µs. Time domain data are presented in the upper part of Figure 2-2. As is common with many oscilloscopes today, the one used for the time domain measurements also had a built-in FFT (Fast Fourier Transformation) function. This was used to compute a frequency domain representation of the measured signal [12] [6]. The normalized frequency spectrum is shown in the lower part of Figure 2-2. It shows the main harmonic at around 47 kHz, corresponding to a typical switching frequency of the electronic ballast.

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Figure 2-2 Ballast output current, measured in time domain and its computed frequency domain

As the next step, a series of measurements was taken at different points, at the input and output of the ballast (Appendix C). It is clear that the noise level is much higher for the output from the ballast than for the input.

From these measurements, another observation was made; the attenuation of the signal in the current probe. The commercial current probe that was available in the laboratory showed low sensitivity for the frequencies below 10 MHz. This is a significant problem, as the core idea of the thesis was to use measured current to excite the N-Port model. Inadequate measurement accuracy could seriously compromise the model accuracy. This problem was solved by developing and testing a set of current probes, some of which were used through the project, [13] [14] [15] [16], see Appendix A .The best one, labelled Hand Made 1 (HM-1) was used for all the measurements presented in this chapter. By observing the measured results comparison in Figure 2-3, it is clear that this in-house made probe has much better sensitivity. It is of note that both measurements have been conducted with the vertical resolution of 10 mA per division.

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he harmonics of the nominal oscillator frequency of a standard electronic ballast [4].

prominent. After the 3rd harmonic, there is a big drop in the measured signal level.

Figure 2-3 Signal measured with the HM-1 (left) and a commercial probe (right)

In the frequency domain, the voltage on the LISN neutral line was measured, using two pre-set frequency ranges as per the standards. A screen snapshot for the first measurement band, covering frequencies from 9 kHz to 150 kHz, is shown in Figure 2-4. Emissions at around 45, 90 and 135 KHz can be seen, which correspond to t

Figure 2-4 Frequency domain measurements 9 kHz to 150 kHz

For the frequency range from 150 kHz to 30 MHz, Figure 2-4, peaks are not very

Figure 2-5 Frequency measurements 0.15 MHz to 30 MHz

The results showed that the harmonics of the disturbance emissions measured on LISN correspond with the electronic ballast oscillator frequency. This confirms the electronic ballast as the main source of the excessive emissions. To investigate this point further, a simple experiment was devised. An ordinary (non-fluorescent) lamp was connected and measurements with the current probe conducted.

This was just an ordinary table lamp with an incandescent light bulb, Figure 2-1. Unlike the measurements with the fluorescent lamp, in this case the entire power cable was enclosed. Finally, a measurement without lamp or ballast was taken, as shown in Figure 2-6 below.

Figure 2-6 Signal spectra for 3 different measurements configuration

While there is minimal difference between the standard lamp and no lamp at all, it is obvious that the emissions level increases significantly when the fluorescent lamp is

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used. Harmonics are only visible in the case of the fluorescent lamp. Also, these harmonics apply only on to fluorescent lamps with electronic ballast, as the older style passive ballast fluorescent lamps do not produce harmonics in their emission spectrum.

2.2 Measurements on a CISPR TR 30 like test bench

CISPR has created a Technical Report (TR) 30 with the purpose of regulating the way an electronic ballast is to be tested in order to measure its radio disturbance characteristics [8]. A body of a luminaire (gear tray) has been replaced with a test bed, which is essentially a metal plate on an insulating base. This enables easy access to the ballast and wiring, and makes it possible to quickly replace the ballast and repeat the measurements.

The CISPR TR 30 like test bench shown in Figure 2-7 looks similar to the CISPR TR 30 test bed but it is not identical. It is intended for exploring impedance characteristics of the ballast and extracting currents or voltages to be used for exciting the N-Port Mode.

The CISPR TR 30 test bed is symmetrical, with two lamps. This test board has only one lamp and the set-up is highly asymmetrical. The reason for the change in the layout was to get maximum coupling in the LISN active wire (red wire in Figure 2-7), by bringing it very close to the ballast output wire. On the other hand, the LISN neutral wire ( blue wire in Figure 2-7) is positioned in such a way that coupling from the ballast is minimal.

Of particular interest are the grooves in the board. They were very useful in preventing the wires from moving and kept them straight. This also kept the distance between the wires constant. It was possible to quickly replace the ballast without affecting the wiring. A set of measurements on standard ballasts have been conducted in order to enable model development and verification.

Three different ballast types were used:

1. Osram QTP8 2. Tridonic ATCO T 1/36 3. Tridonic ATCO T 1/14

The set-up can be seen in Figure 2-7, together with the schematic for the Tridonic T 1/36 ballast.

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Figure 2-7 Ballast measurement set- up

A set of measurements were conducted for 2 frequency ranges:

• From 0.1 to 5 MHz • From 5 to 30 MHz

Due to the large dynamic range of the measured values, occasionally it was necessary to use an attenuator (12 dB) or an amplifier (25 dB) [16]. These adjustments were later added or subtracted from the measured results. Empty cells indicate that no attenuator of amplifier needed to be used (0 dB gain). A detailed summary of measurements is shown in Table 2-1. Again, red indicates that attenuator was used, green the amplifier.

Table 2-1 CISPR TR 30 like measurement summary

In general, the currents on the ballast input correspond to the LISN voltages.

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Difficulty arises in measurements when using the Ballast’s terminal labels. The difference in terminal labeling for Osram and Tridonic ballast should be noted. The ballast active input port is labelled 3 for Osram and 4 for Tridonic ballast. The ballast output ports are also differently labeled. The terminals at the lamp input are labelled in opposite order for these 2 ballast types, so Port 21 for Osram corresponds to Port 12 for Tridonic and 22 Osram to 11 Tridonic.

To prevent confusion, port labelling throughout the thesis was unified by using port numbering for the model presented in chapter 3 and 4, and corresponding port numbers for specific ballast can be found in Table 2-2.

Table 2-2 Port notation

Model Ports Osram QTP8 Tridonic 1/36 Tridonic 1/14 Port 1 T 1 T 2 T 2 Port 2 T 3 T 4 T 4 Port 3 N/A N/A N/A Port 4 T 21 T 12 T 12 Port 5 T 22 T 11 T 11 Port 6 T 21_22 T 11_12 T 11_12

Considering the results for all the frequency ranges, the highest level of emissions of all the ballasts was below 1 MHz. This has been confirmed in later measurements. For this reason, only the results up to 2 MHz are shown in Figures 2-6 to 2-12, which was also the frequency range used for model verification in Chapter 4.

LISN voltages were measured using the EMC receiver in QP average mode and are shown in Figure 2-6. For this frequency range, it appears that the Tridonic 1/36 generates the highest emission; much higher than the other two ballasts. Osram QTP8 and Tridonic 1/14 perform similarly, except for the significant first peak of over 80 dBµV for Osram. Oscillator frequency is around 50 kHz for all ballasts.

Figure 2-8 depicts currents measured at the ballast input Port 2 and output Port 6. In the first graph, there is a significant difference between the Tridonic ballast 1/36 and other two. This is presumably due to the difference in the ballast’s internal circuitry. Except for the first couple of harmonics, the measured values represent a noise floor for Osram QTP8 and Tridonic 1/36, while emissions from Tridonic 1/14 are still reasonably high.

The lower graph in Figure 2-7 illustrates measured currents on ballast output (Port 6). They are very similar for all 3 ballast and there is no clear indication that any of the ballasts is better or worse than the other.

Figure 2-8 Measurement Results - 0.1 to 2 MHz

Figure 2-9 Measurement Results - 0.1 to 2 MHz

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Measured currents on ballast output Ports 4 and 5 are of particular interest, as these are the values used to excite the N – Port model developed in later sections. Again, Tridonic 1/36 has the highest values. Unlike the other two ballasts, Tridonic 1/14 shows different behavior for Ports 4 and 5 and its emissions level decreases very rapidly with the increase in frequency.

Figure 2-10 Measurement Results – 0.1 to 2 MHz

Results in Figures 2-8 to 2-10 show that the coupling process is different for different ballasts. Figure 2-7 shows that the wiring stayed the same, so the coupling between the wires on the ballast output side and LISNA wire stayed the same. But this coupling is not the primary source of emissions. There is also the coupling from the currents on the ballast input terminals.

To get insight into this coupling, it was necessary to determine how LISN voltages correspond to current on the ballast input terminals. This ratio depends on impedance of the ballast input terminals.

The following group of graphs, shown in Figures 2-11 to 2-13, shows ballast impedances calculated as the difference (in dB) between LISN Active line voltages and current at the ballast input, which is terminal 3, for Osram ballast, and terminal 4, for Clevertronics ballasts. The ratio is different for all different ballasts. The Y axis of these graphs are in dBµΩ and are mostly negative, as the numbers are very small. For example QTP8 minimum impedance is near -55 dBµΩ.

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Figure 2-11 QTP8 ballast impedance

Figure 2-12 Clevertronics T1/36 ballast impedance

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Figure 2-13 Clevertronics T1/14 ballast impedance

2.3 Measurements for CISPR 15 like model

Unlike the previous model, the CISPR 15 like model refers to disturbance emissions measurement on a complete luminaire, with ballast and lamp enclosed in a gear tray. The luminaires with single- and double- width gear trays were tested and a photograph of them without a lid and lamp is shown in Figure 2-14.

Particular attention was paid to keep the wires in a fixed position. Noticeably there is not much similarity with the wiring in the typical luminaire, presented in the introduction (see Figure 1-1). For these measurements, wires were kept straight, with almost no curved sections. Lengthwise, wiring was identical in both trays, but due to the larger width of a gear tray with two luminaires, the wires to the lamp Far End (blue and brown) were 60 mm longer. Setting the wiring in this way has made it possible to insert different ballasts with the minimum change in wiring position (and coupling). It also made it easier to create a 3D computer model, which is further explained in section 4.2.

All 3 ballasts were inserted in both trays and LISN voltages measured.

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Figure 2-14 Test luminaries without lid

Four frequency ranges, tabulated in Table 2-3 were used, similar to CISPR 15. There are no details on the attenuator or amplifier used, as in Table 2-1. This was because the EMC receiver’s (Appendix A) built-in signal magnitude control was enabled. This has no effect on the accuracy of the measurement, but has made post processing simpler.

Table 2-3 Measurement settings

Frequency 9-150 kHz 0.1-5 MHz 5-30 MHz 30-100 MHz Resolution BW 200 Hz 9 kHz 9 kHz 9 kHz Sweep time 10.3 s 7.5 s 2.08 s 10 s

Ease and repeatability of measurements presented an opportunity to expand the frequency range measured beyond the requirements for model development. Frequency range above 30 MHz was beyond the scope of current work, but it was included in the measurement set-up to extend knowledge about the disturbance voltages in this region.

These measurements required a different set-up compared to those referred to in the previous section. This was necessary due to the CISPR 15 specifications, whereby the Device Under Test (DUT) has to be placed 400 mm above a ground plane. The length of the power cable was 800 mm (Figure 2-15).

Figure 2-15 Measurement set-up

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In Figure 2-16 it is clear that LISN voltages for all 3 ballasts have similar amplitude, with a small frequency shift. Osram QTP8 appears to be the best, with the maximum value at about 80 dBµV. Again, Tridonic T36 was the noisiest ballast, peaking at 92 dBµV, around 35 kHz. The frequency range from 9 kHz to 100 kHz is not included in current models. On closer examination it can be seen that the oscillator frequency of the Tridonic T 1/14 ballast is slightly higher than the other two ballasts.

What all 3 ballasts have in common is a decrease in the LISN level from their corresponding peak values to below 60 dBµV when the frequency reaches 0.1 MHz.

Figure 2-16 Measurement results 9 to 150 kHz

Results for the frequencies from 0.1 to 5 MHz are depicted in Figure 2-17 and, as in the case of measurements for TR 30 like model, both LISN voltages drop rapidly with the frequency increase. Above approximately 2 MHz, the difference between a single and double gear tray begins to appear, but not for T14 ballast.

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Figure 2-17 Measurement results 0.1 to 5 MHz

The results for the frequency range from 5 to 30 MHz, depicted in Figure 2-18, show that, regardless of the ballast used, maximum values are higher for the single lamp luminaire.

This frequency range is also of interest for a different reason. As noted in the introduction to this chapter, almost all the wires in the models with a single and double gear tray width have the same length. However, there is a 60 mm difference in the length of wires from ballast to the lamp Far End, shown previously in Figure 2-13. Different wire length results in different resonant frequency for a single and double gear tray width model, which can be seen in Figure 2-18.

In the resonant mode range, between 22 and 30 MHz, emissions increase significantly. This increase raised the question of emissions level above 30 MHz. This was, however, outside of the project scope. As the LISN device used (Appendix A) had an upper frequency limit of 100 MHz, it was possible to conduct additional measurements for the region between 30 and 100 MHz. The results are shown in Figure 2-19.

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Figure 2-18 Measurement results 5 to 30 MHz

Currently, there are now emission limit values for the frequency range above 30 MHz. But the magnitude of the harmonics would possibly be a cause for concern, in particular for the Tridonic T36 ballast. More details on these resonances are given in section 5-3.

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Figure 2-19 Measurement results 30 to 100 MHz

2.4 Measurements Summary

The measurements presented in this section were conducted for the following reasons:

• To gain insight into the nature and behaviour of disturbance emissions • To record current on the ballast output terminals (only for the lamp Far End

wires). These values were later used to feed N-Port model, section 4. 1. • To measure disturbance voltages on LISN and compare them with the LISN

values calculated in section 4. In this way it was possible to calibrate the model and confirm the validity of the modelling approach.

• To emphasize the difference in testing set up and results between CISPR T30 and CISPR 15

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3 Numerical Model development

3.1 Modelling in general

In general, modelling is a process of expressing the behaviour of a system and its components by a set of mathematical equations. This mathematical approximation of a physical system can then be used to calculate different properties, like forces, voltages or fields.

Figure 3-1 General description of modelling process

It is imperative to develop a model to solve a particular problem, not to model the system. While there are only a small number of physical systems, there are many models and simulation scenarios, and it is the same for the results.

Modelling is primarily based on understanding the nature of a specific system. The work in this thesis is focused primarily on creating a model of a luminaire from the EMC perspective. EMC problems are challenging because of dynamic complexity that arises from interactions, not just the number of parts. EM coupling plays a significant role.

While the usefulness of modelling is indisputable in many engineering and scientific applications, there is no unified agreement about the difference between modelling and simulation. Although the Institute of Electrical and Electronics Engineers (IEEE) contends that modelling and simulation could be used interchangeably [19] [20], this is not the view in wider modelling community [21] [19].

This thesis has adopted the view that the simulation is done after modelling. The modelling process is divided into Chapter 3 (numerical model) and Chapter 4 (N-Port model).

During the simulation process, some of the model’s input parameters are changed and the change in the output results observed. This makes simulation a separate process; a further development of a model. It is presented in Chapter 5.

Generally speaking, models are limited, imprecise and a simplified representation of the real world and it is important to decide:

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These results represent the first column of the impedance matrix. The same process was repeated five more times to calculate transfer impedances (weighting

• What is the problem? • What are the key variables and boundaries?

In the case of this project, the main problem is the disturbance voltages measured on LISN. Key variables and boundaries are wiring layout, gear tray shape and electronic ballasts impedance characteristics.

3.2 Numerical Model

Generally speaking, a numerical model can be described as a set of computer instructions that performs numerical computation, e.g. computing currents, voltages or fields values for every single parameter of the model, like a wire segment or surface patch. Different methods can be used for solving these equations. In this project, the Method of Moments was applied.This method is widely used for solving problems stated in the form of an electric field integration equation (EFIE) or a magnetic field integration equation (MFIE). The full description of MoM is beyond the scope of this thesis and it is given in the references [11] [20] [21]. The software used was Concept [25]. The program creates and solves the differential-algebraic equations of a circuit model. Running a complex model is computationally difficult and time consuming, but, providing it is properly done, there is no need to do it more than once.

In the numerical model developed in this thesis, each ballasts’ terminals are modelled as ports. The model is used to calculate voltages and currents on the ports, as illustrated in Figure 3-2. The blue picture in its centre is just a screenshot of a model which is further explained in section 3.3, Figure 3-3. In the numerical model used the number of ports was six. As per CISPR 15 standard, the frequency range was 0.15 to 30 MHz, but any frequency range can be applied.

This section explains the first part of its practical implementation. In Figure 1-4, one of the inputs into the N-Port model is the transfer impedance matrix. Calculating this matrix was undertaken using a numerical model.

Thevenin’s theorem states that any two-terminal linear network may be replaced by a voltage source equal to the open circuit voltage between the terminals in series with the output impedance as seen at this port.

As a first step, Port 1 has been excited with 1V, do its transducer factor is 1mV/Amp=1mΩ. Note that this port shunt impedance is not related to the Thevenin’s theorem mentioned above. It is just used to measure the current . All other ports have high impedance and are considered to be open. These voltages are calculated using the Method of Moments.

Using these voltages and currents, the impedances can be calculated as shown described in the equation (3.1):

= / ; = / ; = / ; = / ; = / ; (3.1)

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• Feeding current on - Port 1, I

Figure 3-2 Numerical model schematic

Applying the same model, but exciting Port 2, the second column of the impedance is repeated for all ports, until all six columns are

el are:

Voltages

is model was that it provided an opportunity to generate , currents on different wires or on different sections of a

CISPR TR30 like model

s few details as possible. For this reason, a simple odel, without any surfaces, was introduced first. When this model was determined

coefficients) for open circuit voltages on all other open ports. These are usually the voltages on LISN connectors, but they can be computed on any wire in the model. After the first run, the following values are known:

• 5 open circuit voltages, V to V

• Input impedance on - Port 1, Z

matrix is computed. The processcomputed.

The outputs from the numerical model that are needed as input parameters for the N-Port mod

• Open Circuit Voltages • Disturbance• Feeding currents • Input impedances

One of the advantages of thadditional data. For examplesingle wire could be calculated, providing a better insight into the coupling process.

Two basic numerical (and N-Port, Chapter 4) models were developed, corresponding to the two basic testing set-ups and are shown in the followingsections.

3.3

The initial model should include am

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this model, which was based on the CISPR TR 30 test bench set-up, presented in chapter 2.2 (Figure 2-5). In other words, the physical model of

lamp has been replaced by two 5 Ω resistors representing filaments on lamp ends.

ith wire ports.

the LISN port impedances, was set on each of

e set-up is above a finite ground

nd 20 mm. This is due to the layout presented in Figure 2-7 in the previous section. Some

ut the curvature of the lines was not

Figure 3-3 TR 30 like model

to be appropriate, the second model was developed with the surfaces added and the wires modified.

Figure 3-3 illustrates

a luminaire test set-up was transferred into a computer model. As usual in modelling, some simplifications were made, resulting in the omission of number of elements:

• The

• Ballast as such is not included, but all ballast terminals have been represented w

• Similar wire ports were used for LISN active and neutral terminals. A 50 Ω resistance, corresponding to these two ports and used as observation points. In reality, this 50 Ω is just an approximation, as explained in chapter 4.2.

• The metal plate used in the CISPR TR 30 like measurement was not necessary, as this standard requires that thplane that extends beyond all the wiring. This finite ground plane has been replaced with the infinite MoM ground. By having no plane, it was computationally easier to run the model, speeding up the calculations.

Wires have been placed at two different heights above the ground plane: 2 a

of the wires were placed on a wooden base 20 mm thick. Others, like the LISN neutral wire and the lamp Near End wires, were laid directly on the ground; hence, the height of only 2 mm. Difference in height is not relevant, as this element has no meshing surfaces (ground was not modelled).

It was imperative to get the topology correct, breplicated exactly, as it should not affect the accuracy of the simulation.

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In the post-processing phase, previously calculated voltages and currents were

The reason for this particular topology was to create a highly non symmetrical layout, thus increasing coupling in the LISN active wire and keeping the coupling into the LISN neutral wire to a minimum.

The model in Figure 3-3 is a practical application of a 6 port network schematic presented in section 3.2. It is a six port network, with 3 ports on the input and 3 ports on the output of the ballast. The model port structure is quite different to the terminals used in chapter 2. This was explained in Table 2.2 in the previous section. All models developed follow the labelling presented in Table 2.2.

By using the same ports numbers throughout, it was easier to keep track of voltages and currents, both measured and calculated. It should be noted that Port 3 and Port 6 did not appear in Chapter 2.

It is important to clarify the difference between the observation point and a port. The observation point is a wire on which a voltage or current was calculated using the Method of Moments. In theory, every wire in the model could be an observation point. A port is an observation point (wire) that is used to excite the model. While every port is an observation point, not every observation point is a port. While Port termination impedance can be (and were) changed later in the N-Port model, the observation point impedance is fixed.

For example, in Figure 3-3, a middle section of wire 1 with an impedance of 1MΩ is Port 1. A middle section of wire 17 with an impedance of 5Ω is an observation point, not used for model excitation. In the case of the TR 30 like model, 6 wires are designated ports and voltage is calculated on a total of 8 wires, 6 ports and 2 additional wires, number 9 and 10. These two wires were used for calculating voltages on virtual LISN ports.

As explained in the previous section, the model was firstly excited by applying 1 V voltage on Port 1. Open port voltages v , v ... to v are then calculated. In this way, the first row of the Voltage matrix is created. Current i generates voltage drop over 1 mΩ resistance.

The process was repeated by exciting Port 2. Port 2 voltage v was 1V and open port voltages from v , v ... to v were computed. Current i was again 1 mA. The process was repeated for Ports 3 to 6. Finally, all 36 voltages and 6 currents were identified.

The voltages on LISN, labelled as wires 9 and 10 in Figure 3-3, are calculated in the same way. For example, v is the voltage calculated on LISNA when Port 3 was excited. In this way all the voltages and currents needed were calculated and stored as ASCI files.

used to calculate transfer impedance and weighting coefficients shown in Figure 3-4. For example, impedance on open Port 5 when Port 2 was excited is calculated as open circuit voltage v divided by current i .

Weighting coefficients were computed in the same way, as shown in Figure 3-4. They are correction factors used to ensure that the voltages and currents calculated using Method of Moments are consistent with the previously calculated impedance matrix. A Matlab file, CalculateWeightingCoefficients.M (set out in Appendix F) was one of the files used to calculate transfer impedances and weighting coefficients.

While typical, this modelling scenario was just one of many that could be easily implemented and run with any of the models developed. The same model can be used subsequently for the frequency range over 30 MHz.

Figure 3-4 Impedances and weighting coefficients calculations

It is important to check results obtained for inconsistency or unexpected values [17] [23] [18] [24]. In this case it was only necessary to plot all calculated voltages and currents to confirm that magnitudes were plausible and there was no strange resonances or behaviour.

3.3 CISPR15 like Model

The CISPR 15 standard specifies the limits and method of measurements of a luminaire as a standard enclosed device. Unlike the previous model that consisted of wires only, the model for this standard requires a complete model of the wiring and the gear tray. The gear tray was modelled as a box, as was the ballast. Ballast terminals were modelled as ports, like the previous model.

Modelling a complete luminaire creates a new set of problems. Ideally, surfaces should represent the gear tray in sufficient detail and patches should be small in the area of interest near the wires. On the other hand, the higher the number of patches, the longer the time needed to solve the necessary equations. While the model based on CISPR TR 30 took several minutes to run, this one took significantly more time. Different meshing techniques were used to find the optimal solution between accuracy and computing time. Details are given in Appendix C, together with the pictures of a complete luminaire model, with the lid and sides. Basically:

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• The gear tray was simplified, without reproducing its exact curvature and some of the small openings. It was modelled as a box with all sides as perfect conductors.

• The ballast was modelled as a hollow box with openings at both ends.

The picture of the model with gear tray sides, lid and ballast removed is shown in Figure 3-4.

When one compares this model with the CISPR TR 30 like model, it can be seen that the wiring is quite different, reflecting the different luminaire layout. This is most obvious in a smaller wire loop, smaller general wire length and different positions of LISN observation points. Only the ballast ports were set-up the same way as in the previous model.

Figure 3-5 CISPR 15 like model

Special attention was paid to surface meshing and calculation of current propagation. All wire segments have a length of less than 20 mm and patch sizes vary from 5 mm by 20 mm. In order to allow for a different power cable position (LISN wires), the hole for the power was cut out. The body of this luminaire was located 800 mm above the infinite ground plane.

Verifying this model involved checking the results for a smooth current distribution and input impedance magnitude, example of which is shown in Figure C-8 in Appendix C.

The results were consistent with the numerical method applied. If the numerical model results had not been satisfactory, it would have been necessary to refine the model, e.g. change meshing, or to begin again and to develop a new model. Using the meshing approach as explained in Appendix C adequate mesh was utilised and proved unproblematic.

Despite increased complexity, this remains essentially a simple 6 port model as presented in Section 3.2. It was excited in the same way as CISPR TR 30 like model. Results were stored as rows of calculated voltages and the same Matlab

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38

script was utilised to post-process these results and produce impedance matrix and weighting coefficients that were used as an input for the N-Port model.

3.4 Numerical Model Summary

Two quite different numerical models were developed, corresponding to two different standards applied in the lighting industry. What they have in common is the same port configuration. This is important as the N-Port model to be described in the following section does not relate to a topology of any particular luminaire, only to ports and observation voltages.

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4 The N­Port Model Development

4.1 The N­Port Model Description

This is the main model and the core of this thesis. It is essentially a practical implementation of the equations in Section 1.2. A schematic presentation of the N- Port model is shown in Figure 4-1.

Following the main diagram 1-4 given in introduction, at this stage almost all input values for the N-Port model have been determined. For excitation, the currents measured on ballast output ports (Port 4, 5 and 6) were used. This is because the longest wires, the main source of emissions, are connected to these ports. Weighting coefficients and impedance matrix have been obtained from the numerical model presented in the previous chapter.

Port termination values are still unknown and it is necessary to ascertain this values. These values are specific for each ballast and they are difficult to determine without detailed knowledge of the ballast’s internal circuitry, like the IRS2166D data sheet [3].

However, the method presented in this thesis stipulates treating each ballast as a “Black Box” and that no ballast internal circuitry data is available. Without this specification, it is necessary that an iteration process is applied to ascertain appropriate impedances. This general approach to characterising any ballast is further explained in section 4.2.

Next, the impedance matrix is created by adding port termination impedances to transfer impedance matrix calculated previously using the numerical model. This impedance matrix is then inverted to get admittance matrix, . Each port current is then calculated by multiplying excitation voltage with the admittance, corresponding to equation (6) in section 1.2.

With known N-Port currents, it is now possible to ascertain the voltages, as per equation (2). Port currents are multiplied with weighting coefficients, previously calculated with the numerical model, and the output is a predicted disturbance voltage. These are the voltages LISN active and LISN neutral . This was the starting equation (1) in the Chapter 1.2.

Figure 4-1 has been implemented in a Matlab script N-Port.m, which is set out in Appendix F. It is a simple script that can easily be modified for use in “For” loops, thus allowing LISN voltages to be calculated for different scenarios.

Figure 4-1 N-Port model schematic

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The N-Port model has been configured to utilize voltage rather than current as excitation. As a first step, excitation voltages are computed by multiplying the measured current with the port termination impedances. So instead of a current source for the ballast terminals, their Thevenin equivalents, voltages and

are used.

Model parameters, like port terminations, can be optimised until measured and predicted emissions substantially agree. It can also be used for a parameter study to isolate worst case scenarios.

4.2 Estimating Port’s Termination Impedances

Although the same procedure for selecting termination impedances and excitation scenarios presented in this and the following section was applied for all 3 ballasts, ultimately only the Osram QTP8 case is explained in detail.

The aim is to find values that will assure the lowest possible difference between measured and simulated data. This required some experimentation to determine not only impedance values, but also their nature (real or complex). Also an excitation

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46656

strategy had to be developed, as it was not exactly clear as to how to excite the model. Initially, Ports 4 and 5 were excited using measured currents.

As noted in the chapter on measurements, the highest level of emissions is in the region up to 2 MHz, so only this frequency range was evaluated.

It must be noted that the impedance presented by the LISN to the load is not 50 Ω as assumed in the model. The CISPR specifies that LISN impedance increases from 25 Ω at 0.1 MHz to 49 Ω at 0.8 MHz. From that it is flat up to 30 MHz, with the +/- 20% tolerance. This means that the 50 Ω LISN impedance assumption is only valid of approximately half the frequency range of interest (up to 2 MHz).

At first, all six impedances were set to 10Ω and they are all real impedances. The complex impedances are used later in this chapter.

Using the N-Port model, LISN voltages were computed and then subtracted from the measured voltages, ascertained in chapter 2. One of the purposes of measurement was in fact to ascertain these voltages to enable evaluation of the model. The lower the remainder of this subtraction, the better the correlation between measured and calculated data. The difference was stored as a numerical value. This process was repeated, with the Zt1 increased to 100 Ω, while keeping other impedances the same. To obtain approximate values, the range of possible impedances used was very wide, from 10 Ω to 10 GΩ. This method was labour intensive and proved too cumbersome due to the large number of possible combinations.

As an alternative, an algorithm was developed to speed up the process. Firstly, a Matlab script was used to provide a string with many different combinations of impedance values. Each of the six impedances could have any of the following six values:

(10Ω 1KΩ 100KΩ 1MΩ 10MΩ 1GΩ)

In this case there were 6 combinations. For each of these combinations LISN voltages were computed and subtracted from the measured LISN Voltages. In the next step, the sum of absolute differences was calculated, which indicated the best fit. Initial results for all 3 ballasts are tabulated in Table 4-1. However, this method is certainly not error free and there are impedance combinations that differ quite substantially from those shown below, but still result in the same level of accuracy.

Table 4-1 Best values after first iteration

Impedance Zt1 Zt2 Zt3 Zt4 Zt5 Zt6 Osram QTP8 100 Ω 100 Ω 10 Ω 10 MΩ 100 MΩ 10 kΩ Tridonic T 1/36 10 kΩ 10 kΩ 10 Ω 100 MΩ 100 MΩ 1 MΩ Tridonic T 1/14 10 kΩ 10 kΩ 10 MΩ 10 MΩ 10 MΩ 1 MΩ

In the previous step the impedance changed values by one order of magnitude, so it is quite possible that some of the values were rounded up or down too much. In the next step, the values from the Table 4-1 were varied +/- 50 %.

For example, if the best value for Zt4 was 10 MΩ (Osram QTP8), in the second iteration values for Zt4 were: 5, 7, 9, 11, 13 and 15 MΩ.

The results for the second step are tabulated in the Table 4-2. No more test runs were conducted, as the algorithm in the current form was not capable of finding the most accurate values for both LISNA and LISNN voltages at the same time.

Table 4-2 Best values after second iteration

Impedance Zt1 Zt2 Zt3 Zt4 Zt5 Zt6 Osram QTP8 50 Ω 50 Ω 9 Ω 9 MΩ 150 MΩ 13 Ω Tridonic T 1/36 10 kΩ 10 kΩ 10 Ω 50 MΩ 50 MΩ 1 MΩ Tridonic T 1/14 10 kΩ 10 kΩ 10 Ω 25 MΩ 25 MΩ 1 MΩ

The results after the second iteration were slightly better, as illustrated in Figure 4-2. There is a slight difference in magnitude, but not identical for all harmonics. Graphs for other two ballasts appear to be similar. Due to the frequency dependence of coupling, it is difficult to get complete correlation in the whole frequency range.

Figure 4-2 Resulting LISN voltages for impedance’ iterations

There are also the effects of measurement system noise floor on the simulation results for the cases where the measured ballast had low emissions. For example,

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QTP8 Active line near 1 MHz the emission level is less than 10 dB above the measurement system noise floor. These values would be more affected by the measurement system noise than the stronger harmonics below 0.6 MHz.

The previous figures show some of the effect different impedance combinations have on results.

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Another approach was adopted to acquire a better understanding of how individual impedances influence calculated LISN active and neutral voltages. Firstly, all Zt1 to Zt6 were set as per Table 4-2. While keeping all other values fixed, the value for Impedance Zt1 was varied from 5 Ohm to 1G Ohm and the change in and observed.

The process was repeated by changing Zt2 values while keeping Zt1, Zt3 ... Zt6 fixed. For many cases, the resulting voltages would increase or decrease, but not uniformly over the whole frequency range. An interesting point was a change in frequency response at about 0.5 MHz, for some of the impedances. For example, for the ballast Tridonic T14 ballast, changing impedance Zt1 from 5 to 10 Ohm resulted in the LISN active voltage mostly increasing slowly from 0.1 to 0.5 MHz, and more rapidly from 0.5 to 2 MHz. LISN neutral voltages slightly increased up to 0.5 MHz, and decreased afterwards, as depicted in Figure 4-3.

Figure 4-3 An example of individual ballast's impedance influence on results.

In order to record these effects, a colour coded method was developed as shown in Table 4-3. This method was not ideal, as the increase or decrease was both very slow and very abrupt, depending on the particular impedances. Decreasing voltages are labelled red, increasing with green, where the darker colour represents a slower rate of change. If there was no change, the colour used was white. The example in Figure 4-3 has been coded as follows:

• LISN Active voltage increasing slightly up to around 0.5 MHz (F1) – dark green.

• LISN Active voltage increasing faster above 0.5 MHz (F2) – light green • LISN Neutral voltage increasing slightly up to around 0.5 MHz (F1) – dark

red • LISN Neutral voltage decreasing slightly up to above 0.5 MHz (F2) – light red

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Although not highly accurate, this method shows different behaviour for each of the 3 ballasts tested. Even if the voltages and were decreasing or increasing at the same time, the rate of change was usually different. This simply confirms that each ballast has different impedance characteristics.

Table 4-3 Individual impedances effect on LISN voltages

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1 1

10 1 1

A similar approach was adopted to determine the influence of complex impedances on calculated LISN Voltages. After all, there is always a possibility that the impedances might be complex values. To determine if this is indeed the case, the real part of the impedance was kept constant, and inductances were added ranging from to . The first column in the Table 4-4 represents imaginary impedances added to real impedance.

In other words, all impedances had firstly fixed values as per Table 4-2. LISN active and neutral voltages were computed and stored as a reference value. Then, the real Impedance Zt1=10 kΩ (Tridonic T1/36) was replaced with a complex impedance

, LISN voltages were then calculated and subtracted from the reference value.

Table 4-4 Individual complex impedances effect on LISN voltages

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1 1

As with Table 4-3, each ballast shows different impedance characteristics. For example, adding an inductance of 10 kΩ to the Zt3 resulted in an increase of computed LISN voltage for QTP8 ballast, a decrease in case of Tridonic T14 and no difference for Tridonic T36.

The results for adding capacitances (e.g. 10 ) are similar and are not shown here. The general conclusion is that only adding a significant inductive or capacitive component to the termination impedances would result in visible changes in computed LISN voltages. Including complex impedances into the model and estimating the required values would increase complexity of the project without obvious advantages. It was therefore concluded that using only real impedances is enough to produce acceptable results.

4.3 Excitation scenarios

In the previous section it was noted that the N-Port model was excited using measured currents. Both measured currents for Port 4 and 5 were used. It was decided to try exciting the model with one of the currents negative (180 degrees phase difference) to ascertain if this would increase the accuracy of the model. The resulting voltages confirmed that the phase difference plays no rolle. The red and green traces in the Figure 4.3 are almost identical. There was no difference for the frequency range up to 30 MHz either.

Figure 4-4 LISN voltages when feed with 180 degrees phase shift

The currents used to feed the model on Port 4 and Port 5 are the currents measured on the ballast terminals 21 and 22 for the Osram ballast, and 11 and 12 for the Tridonic ballasts.

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Due to the nature of the measurement equipment, only the magnitude of the currents was measured, not the phase. Determining the phase from the frequency domain measurements provides information about common or differential mode coupling, which as such is not available from measurements. A phase angle between the currents measured can be calculated using the law of cosines:

,

Again, this variation did not provide any improvement. There are very small differences, of no impact. It was interesting that for the Osram QTP8 and TRIDONIC T 1/36, exciting Port 5 makes very little difference to the resulting LISN voltages.

(7)

Where and are the currents measured on Port 4, Port 5 and between ports 4 and 5.

This phase difference was added to excitation voltages as the imaginary part of the complex voltage. In the example in Figure 4-5, the blue line represents the LISN voltages measured previously. The green line is the LISN voltage calculated by exciting Ports 4 and 5 with the measured currents, which are in scalar (magnitude only) form. The red line shows the calculated LISN voltages when Port 4 is excited as previously, while Port 5 was excited by combination of measured current and phased angle calculated using equation 7.

Figure 4-5 LISN voltages when feed with one of the currents with measured phase

4.4 The N­Port Model summary

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Using the current knowledge described in the forgoing chapters, it was clear that d to have different values for different

equencies. So, as a first step, the frequency range from 0.1 to 0.5 MHz was termination impedances values neefrdivided in 9 frequency bands (Figure 4-6). Port impedances for each of these frequencies were varied using values in Table 4-5.

Table 4-5 Impedance’s values for different graph parts

1 2 3 4 5 6 7 8 9 7 Ω 15 Ω 15 Ω 10 Ω 25 Ω 10 Ω 15 Ω 18 Ω 40 Ω 12 Ω 20 Ω 10 Ω 15 Ω 25 Ω 15 Ω 20 Ω 20 Ω 35 Ω 0.7 Ω 5 Ω 5 Ω 1 Ω 0.6 Ω 0.7 Ω 3 Ω 7 Ω 2 Ω

Us g th roa a y pr d n in ed to igni r ffe b lc a ea

m sion n Figure 4-pproach proposed in this thesis is a valuable alternative to measurements in the

Figure 4-6 Frequency range 0.1 to 0.5 MHz divided into 8 bands

in is app ch thee

ccuraci

of the edicted isturba ce wasd

creasm

so asds ficantly

s, as shown ieduc the d rence

7. It is theretween

efore assertthe ca ulate

ed that thend

modelling sure

ea

is

laboratory. Because of time constraints, the procedure was not extended over the range of 0.5 MHz, but instead the focus was shifted to undertaking this process by way of a mathematical algorithm.

Figure 4-7 Comparison between measured and calculated data

From the graphs it can be seen that:

• The source model for the ballast is reasonably accurate and the magnitude of the calculated LISN voltages does in fact follow the measured values well, as seen in Figure 4-7.

• All 3 ballasts show different behaviour when their individual port impedances are increased in a uniform manner. This is shown in Tables 4-3 and 4-4. Each ballast has a specific combination of components, like inductors and conductors, so different frequency dependence and impedance characteristics are to be expected.

• The phase of the impedances is not critical. Only when the complex value is bigger than the imaginary part do emission levels increase. Magnitude of impedances is more important.

• The requirement to use a frequency dependent set of load impedances adds some complexity to the proposed method, gut id does not necessary reduces its usefulness in terms of applying a generic set of ballast results to arbitrary luminaries.

49

5 Simulation

5.1 Motives for simulation

In combination with measurements, modelling and simulation are the best tools to determine the optimum tradeoffs between complex variables influencing system design and enhancements for any given EMC situation. With the N-Port model’s accuracy verified, it is now possible to move to simulation, change some of the model’s parameters and examine changes in the resulting disturbance voltages.

Both models from Chapter 4 were extended to four different configurations with different distances between the wires and gear tray dimensions. Note again that the models described below are used to characterise a coupling process and not to predict LISN voltages from a particular luminaire on the market.

5.2 Model similar to CISPR TR30 Variations

The motivation for this particular configuration was to determine the influence of distances between the wires on the coupling process. Different coupling would result in different disturbance voltages. A CISPR TR 30 like model, introduced in chapter 4, was slightly modified. Modifications are illustrated in Figure 5-1. First of all, the coupling observed was between the LISN active wire and the wires connecting the ballast and the lamp Near End. The connection to the lamp Far End was therefore not included.

Figure 5-1 Schematic of model variations

50

The key parameters of the model were the same as in the TR 30 like model with Osram QTP8 ballast, shown previously in Figure 3-3, with impedance values given in Table 4-2,chapter 4. Also the same as in Chapter 4, the model is not connected to the power source, just excited at the ports.

The critical factor was d1; being the distance between the LISN active wire (red) and a lamp Far End wire connected to the Port 5 (purple). This distance was increased from 10 to 70 mm. Variations to the CISPR 30 TR 30 like model are set out in the models A to C in Table 5-1. Model D is a special case where the LISNA wire (red) was lifted 200 mm from its original position. The reason for this layout was to determine if a large increase in wire separation would cause a corresponding decrease in coupling.

For all models, the expectation was that level of calculated LISN voltages would drop with the increase in distance, but by how much needed to be explored.

Table 5-1 Distances from Figure 5-1 used for models A-D

Model d 1 (mm) d 2 (mm) d 3 (mm) Height (mm) A 10 70 10mm 20 B 40 40 10mm 20 C 70 10 10mm 20 D 10 70 10mm 220

Screen shots of computer models A-D are shown in Figure 5-2, with the LISNA line highlighted in red.

Figure 5-2 Geometry of the models A - D

The same excitation and loads settings as in the basic CISPR TR 30 like model

51

were applied. The LISN active and neutral voltages were computed using the same procedure which was explained in Chapters 4 and 5. Results for all the models are shown in Figure 5-3.

Figure 5-3 Calculated LISN voltages fors the models A - D

Both LISN active and neutral voltages decreased with the increased distance, but this difference between the models B and C proved to be minimal, despite the 30 cm increase in separation.

For model D, the distance between the coupled conductors had been increased by about a factor of 20 compared with model A. Mutual inductances or coupling capacitance are determined mostly by the natural logarithm of distances. Thus the decrease from the Model A to Model D of about 10 dB was what could be expected.

In summary, the coupling does not reduce dramatically with an increase in wire separation. This is explained by referring Figure 5-1. The coupling process between the LISN active wire and lamp is also affected by the distance d3. While distance d1 was increasing, the distance d3 was left unchanged, enabling constant and significant coupling between the wires.

It was also observed that similar values were recorded for LISN A and LISN N although the LISN N is not subject to direct coupling. The same phenomena was observed during the measurements, referred to in Chapter 2. This can be explained by signals coupled into one conductor leading to a differential mode current in Active and Neutral conductors, and this was the reason why Port 3 was introduced.

52

53

5.2 CISPR 15 like Model with variable gear tray width

Linear luminaires come in different sizes, depending on the required level of light intensity. While one or two lamps are most common, up to 6 lamps can be installed within a single gear tray.

The intention here was to test the hypothesis that coupling between lamp and power wires (Ballast Out and Ballast In) decreases in larger gear trays because of larger distances. In that case, a luminaire with the narrowest gear tray would to be the worst possible case, meaning the highest emissions level.

As mentioned in the introduction, there is very little published material about the coupling process within a luminaire. This model may provide some insight.

To develop a plausible model for the coupling, the width of the model based on CISPR 15 was gradually increased so the gear tray could accommodate 2, 3 and 4 lamps. Regardless of the gear tray width, only one lamp was connected to the ballast.

This is a computer model only, and it was not based on any actual luminaire available in the market. Installing a wide gear tray with only a single lamp serves no practical purpose. Also, for performance reasons, placing lamps very close to each other is not a recommended practice.

Four different models were simulated, with gear tray width gradually extending as per Table 5-1.

Table 5-1 Gear tray dimensions

Model Length Width Height Distance - d Single 1220 50 45 10 Double 1220 110 45 70 Triple 1220 160 45 120 Quadruple 1220 210 45 170

The quadruple lamp is shown in Figure 5-4, with and without a lid. Ports labelling and excitation ports were the same as for the CISPR 30 like model, see Section 3.3. The wiring for all the models is the same, except for the conductors positioned widthwise in the gear tray, as shown in the upper part of Figure 5-4.

The lower part of Figure 5-4 shows all the lamps. This shows how an existing model or even parts of a model can easily be modified and re-used. In this case, it was only necessary to make a gear tray lid and bottom wider and increase the conductor length for 4 wires, as marked in Figure 5-4. The distance between the ballast output terminals and wires to the lamp Far End is labelled with the letter “d”. This distance is an important factor in the level of coupling.

Figure 5-4 Quadruple lamp and its sections

Unlike the models presented in Chapter 4, this one was excited with a constant 1V voltage source, not with the measured currents on Ports 4 and 5. The reason was that there was no physical luminaire of this type available that could be properly characterized in the laboratory.

At any time, only one of the 6 ports was excited and the calculated LISN voltages observed. Port impedances were 1 µΩ for the excitation Port and 1 MΩ for the other ports.

The results for the first two cases, when Port 1 and Port 2 were excited, are presented in Figures 5-5 and 5-6. Most noticeable is the symmetry in the results, e.g. values for LISN_A with excitation on Port 1 are the same as values for LISN_N when Port 2 was excited and vice versa. This comes as no surprise as the wiring for these ports is symmetrical, which can be seen in Figure 3.4, Chapter 3. The length and the shape of the wires connecting Port 1 and the LISN_N are the same as the wiring between Port 2 and the LISN_A.

There is a difference in potential between the conductors connected to Ports 1 and 2. While one conductor is high, the other one is on floating ground and has a gear tray on its end. The gear tray acts as a capacitance, the values of which change with the increase in gear tray width. This in turn causes differences in resonant frequencies.

54

Figure 5-5 Calculated LISN voltages, Port 1 excite

Figure 5-6 Calculated LISN voltages, Port 2 excited

55

When Port 3 is excited, there is very little difference in LISN voltages, regardless of the gear tray’s width. This is shown in Figure 5-7. This observation was also made during the measurements. Both LISN wires are connected to the gear tray, which is on zero potential and is not affected by the change in capacitance (gear tray width). There is no common mode current, only the differential mode current.

Figure 5-7 Calculated LISN voltages, Port 3 excited

Irrespective of whether Port 4 or Port 5 is excited, the results presented in Figures 5-8 and 5-9 are similar. The emissions level is decreasing with the increase in gear tray width. It was noted that there is a significant difference between the Single and other models. This is explained by examining the values for distance “d”, previously shown in Table 5-1. In the Double model, this difference increases from 10 mm to 70 mm, which is a sevenfold increase. But from Double to Triple, the distance barely doubles. As the coupling is inversely proportional to the square of distance, it decreases very quickly. Consequently, the LISN voltages are highest for the Single model, drop significantly for the Double model and then decrease further for the Triple and Quadruple models.

As previously mentioned, the resonances appear for the frequency over around 22 MHz.

56

Figure 5-8 Calculated LISN voltages, Port 4 excited

Figure 5-9 Calculated LISN voltages, Port 5 excited

57

Results for the case when Port 6 is excited are shown in Figure 5-10. LISN voltages are generally lower in comparison with the case when Port 4 and Port 5 were excited. Again the highest emissions are for the Single model. This confirms that disturbance emissions resulting from using the narrowest gear tray represented the worst case scenario for EMC.

Figure 5-10 Calculated LISN voltages, Port 6 excited

The following conclusions can be drawn from all graphs in this Section:

• The smallest gear tray is the worst case scenario, at least for the frequencies

below 22 MHz.

• At resonances, which are at different frequencies for different models, peak

values are almost the same.

• To accommodate for resonances, there would need to be a safety margin of

about 20 dB for the frequencies above 22 MHz.

5.3 Testing of emissions over 30 MHz

Currently there is no requirement for conducted emissions testing over 30 MHz. This is likely to change in the future and the upper limit is likely to be extended up to

58

300 MHz. While the scope of the work presented in this thesis was limited to 30 MHz, it is possible that the model is accurate over that limit.

It is only necessary to measure ballast output currents and LISN voltages up to 300 MHz. Unfortunately, the LISN device used in the measurements (Chapter 3) has a limit of only 100 MHz, so there was no opportunity to verify the model accuracy over that range.

Nevertheless, in order to get some insight on conducted emissions over 30 MHz, the TR 30 like model was used to calculate LISN voltages for the frequency ranges from 30 to 300 MHz. As there was no measured ballast output emissions available, this N – Port model was excited using a constant 1V source.

For comparison purposes, the same excitation was used for the previously characterised range from 0.1 to 30 MHz. The results for both frequency regions are shown in Figure 5-12. Below 30 MHz, the graphs are relatively flat, and do not correlate in any significant way the same graphs shown in the Chapter 4, which is obviously the result of different excitation. It is of note that, due to the wire length, there are many resonances in the region between 30 and 300 MHz. There are many spikes, with voltage values higher than anything in the region below 30 MHz.

In respect to luminaires causing interference, this is potentially a major problem.

Figure 5-11 Simulated disturbance voltages up to 300 MHz

59

60

5.4 Simulation Summary

In this chapter, 3 different simulation scenarios were presented.

Changing the distance between the wires in the CISPR TR 30 like model did reduce the coupling, but significant coupling through the ballast remained.

Varying the gear tray width in this CISPR 15 like model showed that the narrowest gear tray is the worst case scenario regarding the disturbance emissions level.

Simulating the model up to 300 MHz has indicated there are many resonances and corresponding voltage spikes in the region over 30 MHz.

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6 Conclusion and future work

6.1 Conclusions

• From both the test results and numerical model it can be concluded that the main source for noise signals are the lamp terminals of the electronic ballast. The current on the lamp terminals is higher than at the power terminals, as shown in the measurement results in Figure 2-8, section 2.3.

• The modified CISPR TR 30 test set-up, introduced in Chapter 2, can be utilised to gain useful data and characterise the ballast reasonably well. Its open configuration with wiring widely spread makes it easy to conduct measurements. Measured values are the ballast lamp terminal currents, as well as any other current or voltage of interest. The data gained can be used to create a test report for each individual ballast. This test report can then be used in estimating the disturbance voltages.

• Measurement results in Chapter 2 have confirmed that the gear tray with the smallest width is indeed a worst case scenario as far as disturbance emissions are concerned, at least at frequencies below first resonance, at about 22 MHz. This was later confirmed by simulation, see Chapter 5.

• A model of a single lamp luminaire without a gear tray has been developed and verified in Chapters 4 and 5. This model can be further developed and applied to other types of the luminaires.

• A weak point in the procedure is the estimation of ballast termination impedances. This can be significantly simplified if the impedance characteristics were provided by the ballast manufacturers. Another option to obtain values ballast impedances would be by measurement.

• More work needs to be done to find better parameter values. The results in Figure 4-7 do not show calculated values above 0.5 MHz, because the process of estimating the values manually is very time consuming. However, using an automated optimisation algorithm would speed up the process significantly. More work is needed to improve the robustness of the script.

• Using the models developed it is possible to obtain insight in the luminaire behaviour for a frequency range up to 300 MHz. In general, the highest emissions occur for the frequencies below 2 MHz (Chapter 2.2), and then they diminish with the increase in frequency. From about 22 MHz, resonances occur, causing a significant rise in emissions. This is particularly the case for the frequency range above 30 MHz, as shown in Chapter 5.3.

6.2 Future work

• The proposed method of using the N-Port model for luminaries EMC characterisation can be applied to a compact fluorescent lamp (CFL). It

consists of built-in ballast and a non-linear fluorescent lamp. The only requirement would be to change the geometry of the model, but the principal is the same. As this type of lamp is more and more used, the modelling of this type of luminaire would be the next logical step.

• Comprehensive measurements were undertaken to enable the characterisation of emergency lighting. From the measurement results it can be concluded that the level of emissions falls significantly when the inverter and/or battery are installed. More details are provided in the Appendix E. However, no emergency lighting model was developed due to the time constraints.

• The results can be put to practical use by the industry by developing a Graphical User Interface (GUI), like the one shown in Figure 6-1. It would enable a quick estimate to assess if a luminaire would fail or pass conducted emissions testing. The user would use dialog boxes to choose the main components of a luminaire, as well as the frequency range. On the basis of these parameters, the program would retrieve measured ballast output currents, weighting coefficients for the appropriate gear tray and wiring combination and calculate LISN voltages using the N-Port model. In addition to the calculated LISN voltages, the required safety margin, e.g. 20 dB would be added. If this sum was higher than given CISPR limit, the user would be warned that the particular luminaire would need to be tested in an EMC laboratory.

Figure 6-1 A sample GUI

• This thesis did not incorporate measurement of ballast parameters as this was out of the scope of work and would require significant time. Ballast

62

63

characterisation would be greatly facilitated by including ballast manufacturer’s data and equivalent circuit models.

References

[1] Standards Australia/ Standards New Zealand, “Limits and methods of measurements of radio disturbance characteristics of electrical lighting and similar equpment,” AS/NZS CISPR 15:2006, Sydney, 2006.

[2] S. Ramo, Field and waves in communication electronics, New York: John Willey & Sons, 1984.

[3] G. Daniel, “EMI Between wires above a ground plane,” Lighting Council Australia, Sydney, 2009.

[4] International Rectifier, “IRS2166 Data Sheet,” Internatinal Rectifier, El Seguendo , 2006.

[5] K. F. A. C. F. Schlagenhaufer, “Using N-Port Models for the Analysis of Radiation Structures,” in International Symposum on EMC, Minneapolis, 2002.

[6] D. K. Y. S. Junhong Den, “Characterization of RF Noise Source Impedance for Switched Mode Power Supply,” in International Zurich Symposium on Electromagnetic Compatibility, Zurich, 2006.

[7] R. Bracewell, The Fourier Transform and Its Application, Sydnay: McGraw-Hill, 1965.

[8] T. William, EMC for Product Designers, Oxford: Newnes, 1998.

[9] Standards Australia/Standards New Zealand, “Test Method on Electromagnetic Emissions from Electronic Ballast for Single- and Double Capped Clourescent Lamps,” AS/NZS CISPR TR30, Sydnay, 2006.

[10] C. Marshman, The Guide to the EMC Directive 89/336/EEC, Wendens Press: EPA Press, 1995.

[11] T. Williams, The Circuit Designer's Companion, Oxford: Newnes, 2010.

[12] R. Harrington, Field Computation by Momento Method, New York: The McMillan Company, 1968.

[13] C. R. Paul, Introduction to electrical engineering, Sydney: McGraw - Hill, 1986.

[14] J. Nijenhus, “Characterization fo RF Noise Source Impedances with the two

64

current probe method.,” WATRI, 2008.

[15] M. Montrose, Testing for EMC Compliance, New York: IEEE Press, 2004.

[16] K. Amstrong, “EMC Testing Part - Conducted Emissions,” EMC Compliance Journal, 2001.

[17] D. Morgan, A Handbook for EMC Testing and Measurement, London: The Institute of Engineering and Technology, 1994.

[18] HP, EMC Receiver - HP 85462A user manual.

[19] IEEE, “Standard for Validation of Comutational Electromagnetics Computer Modeling and Simulations,” IEEE , New York , 2009.

[20] IEEE, “A Method to Validate Computational Eelectromagnetic Computer Modeling and Simulation,” IEEE Standards, 2008.

[21] J. Sterman, Business Dynamics, System Thinking and Modeling for a Complex World, New York: McGraw-Hill., 2000.

[22] N. Boccara, Modellling complex systems, New York: Springer- Verlag, 2004.

[23] N. V. Kantartzis, Modern EMC Analysis Techniques, New York: Morgan & Claypool , 2008.

[24] P. Argus, “New Basis Functions for Connections between Wire and Surface Structures for Time Domain MoM,” 2002.

[25] C. Software, “http://www.tet.tu-harburg.de/concept/index.de.html,” [Online].

[26] H. D. Bruens, “Validation of MoM Simulation Results,” 2003.

[27] IEEE, “IEEE Std 1597.2-2011 Recommended Practice for IEEE Std 1597.1 ™-2008, Examples and Problem SETS,” IEEE Standards, 2011.

Appendix A

Current probes and Equipment used

As seen in Figure 1-4 and explained in previous sections, one of three necessary inputs into the model were disturbance emissions on ballast Output ports (lamp side). Measuring these values was not a simple exercise. Any use of voltage probes in this case was deemed unsatisfactory due to the large loop area between the probe and ground. Current probes, on the other side, also have some shortcomings, like frequency dependent sensitivity.

The only commercially manufactured current probe available in the lab, FCC 2000, had low sensitivity for the frequencies up to 10 MHz, which proved problematic. To overcome this issue, 4 current probes were developed in the laboratory, as shown in Figure A-1.

Figure A-1 Current probes used for measurements

Hand-made probes 1 to 3 consisted of a wire coiled over a magnetic core; however, the last probe developed was based on the concept of two currents current probe (2CCP). The basic theory of two currents probes is to use one probe to inject a voltage over impedance to be measured, and then use a second probe to measure the current induced by the injected voltage. This probe should be able to detect much smaller currents than all the other probes. The relatively large size of this probe also proved to be problematic and consequently this probe was not used for measurements. However, this concept may have some applications in future work.

The frequency range of probes did not present a major problem as the frequency range for the model is only 30 MHz, in which case the length of wires in the probes and ferrites is definitely less than λ/4 (2.5 m).

A manufacturer’s calibration data for FCC probe were available. This probe has been wrapped around a short wire connected to the tracking signal generator and terminated with 50 Ohm impedance. The measured signal (9Khz to 30 MHz) was very closed to the manufacturers data. So using this known FCC probe parameters as a guide, the current probe factor for the other 4 probes was measured and shown in Figure A-2, together with the voltages measured.

This approach was deemed sufficient, as there was no opportunity to conduct proper current probe calibration as recommended in some standards.

65

Figure A-2 Current probes results

The best results were for the 2CCP and HM-1, and only HM-1 was used.

Laboratory equipment used

• Oscilloscope - Tectronix TDS 744A • EMC Receiver - HP 85462A • LISN - EMCO 3825/2 • Current probes

o FCC F-2000 VHF/UHF o HM1

Ballast test samples • Type

o Osram QTP8 Quicktronic professional o Tridonic ATCO 1/36 o Tridonic ATCO 1/14

66

67

∆ 1

Appendix B

Formulas for time domain measurements (Chapter 2.1)

Time domain data and its computed frequency spectrum were calculated using the following general equations:

∆ 1

2 ∆

∆ ;

In case in Figure 2_2, the values were:

100 10 6 100

1100

;

10

∆100 10 6

20005 10 8 50

12 ∆

1 10 7 10

Formulas for frequency domain measurement (Chapter 2.1 and 2.2)

the first two chapters, all the results measured with the EMC receiver were in

Conversion from linear to

hroughout the investigation, it was necessary to use both linear and logarithmic

IndBm and had to be converted in dBµV, which was easily done by adding 107.

107 dBµV(RMS)=107+Vmeasured (dBm)

logarithmic values and back

Tvalues. For example, the ballast port currents were measured in the logarithmic scale. In practice, the voltages were measured and converted into currents using

68

10 /

20 log

known impedances. As the N-Port model needs linear input data, it was necessary to convert these currents into a linear scale.

On the other hand, calculated LISN voltages were linear values and they had to be converted into logarithmic values in order to be compared with the measured LISN voltages.

.

EMI Receiver sweep time calculations

The minimum sweep time can be calculated using the following equations:

[15]

;

RB Resolution Bandwidth ;

However, sweep time used in measurements (see Chapter 2, Table 2-3) was

Additional time domain results

In Chapter 2, a set current probe was used to measure currents on 6 points on both

somewhat longer. This was because the signal measure was not constant, butmodulated with 100 Hz.

sides of the ballast. The position of the measuring points is seen in Figure B-1, and results are shown in Figure B-2.

Figure B-1 Time domain measurement set-up

The difference in the currents on the wires entering ballast, points 1 to 3, and those exiting the ballast, points 4 to 6, is obvious. This result shows that the lamp terminals rather than power terminals are the source of noise currents.

Figure B-2 Voltage measured at points 1 - 6

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Appendix C

Wires and surfaces modelling

Creating a simplified Computer Aided Design (CAD) model is always a challenge. A computer model needs to represent a physical model sufficiently to gain accurate results while omitting as many features as possible. Regardless of the electrical device being modelled, the steps involved are very similar:

• Draw wires • Construct Surfaces • Define Excitations • Set-up Analysis

These steps have been applied to a typical linear fluorescent lamp gear tray with wiring. The gear tray was Thorn Cadet one with dimensions 1220 x 50 x 45 mm.

- Wires

Wires (Figure C-1) were modelled first, based on the dimensions obtained by measuring the wire length in the laboratory. As explained in Chapter 3, all the wires have been modelled as straight lines as the curvatures have negligible impact on result. All wires have a diameter of 0.25 mm. Dielectric coating on the wires has been ignored.

Figure C-1 Wires

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- Gear tray

The gear tray is a U shaped surface, formed from a bottom and 2 sides. The bottom is the most important part, as all the wires are located above it. The initial mesh was simple to construct, but needed considerable manual refining. All patches are of the same size and reasonably large.

Figure C-2 Gear tray bottom initial mesh

The areas immediately below and near the wires were of highest interest. Making a mesh element (patches) smaller in the areas where the values of currents or voltages change the most enables more accurate current and field calculation, hence resulting in better models.

Firstly, a hole was created for the wires entering from the outside. Then the mesh was refined at the edges, below the wires, and at the points where the port wires were connected to the gear tray (ground). The modified gear tray bottom is shown in Figure C-3.

Figure C-3 Gear tray bottom mesh refined

- Ballast

For the modelling methodology proposed in this thesis, ballast size and exact dimensions were not significant. Consequently, the gear tray was modelled as a U shaped box with no covers at the ends. Ballast terminals were modelled as wire ports, which have been explained in section 3.4.

Finished gear tray bottom with ballast and wires can be seen in Figure C-4.

Figure C-4 partially finished model

- Lid

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A lid is essentially just a plain sheet of metal. The only wires close to the lid are the wires at the lamp ends, so the lid mesh is very coarse, except at the ends. The lid is somewhat shorter than the rest of gear tray.

The sides of the gear tray were modelled in the same way as the gear tray bottom, only simpler in construction. Mesh refinement was minimal, as shown on a complete model of the luminaire shown in Figure C-5.

Figure C-5 Complete model

The process shown in Figures C-1 to C-5 describes the creation of only one of the models, called a Model B. In fact, three different models were developed, as shown in Figure C-6. After all, modelling is an interactive process and usually there are alternative ways of achieving same or similar results.

Table C-1: Model dimensions and details Dimensions Length Width Height Elements Mesh Ballast Model A 1220 50 45 2774 Medium/Fine 360x30x30Model B 1220 50 80 1489 Coarse/Medium 320x30x30Model C 1220 50 80 2846 Triangulated 320x30x30

Model C has the same dimensions as model B, but uses triangular mesh.

Figure C-6 CISPR 15 like models A, B and C

- Excitation and analysis set-up

Excitation details for all the models are the same, using a constant voltage of 1V, applied on Ports 1 to 6, one port at the time. The numerical model was computed for the frequency range from 0.1 to 30 MHz, with a frequency step of 0.1 MHz. While

72

the CISPR TR 30 like model took several minutes to run, the one with surfaces requires significantly more computation time.

It is possible to reduce the simulation time by using only 21 frequency steps and Concept built-in interpolating tools, but this was left for larger models which may be developed in the future.

On the basis of numerical models A to C, the N – Port models were used to calculate LISN voltages, which are shown in Figure C-7. Model A differs significantly, mainly to the difference in geometry. It is interesting to notice almost identical results for models B and C, although the number of elements and subsequent simulation time is much larger for model C. On the basis of these results, a new model, derived as a combination of models A and B, was created. So this model was subsequently used in the Chapter 3.4. It has the dimensions of Model A and meshing from Model B.

Figure C-7 LISNA voltages calculated for models A, B and C

Figure C-8 Current distribution plot for one of the models

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Appendix D

Post­processing of preliminary testing

A standard fluorescent lamp and its electronic ballast were tested under controlled conditions by the Lighting Council Australia in Sydney [2]. The purpose of these tests was to determine the influence of the conductor’s separation and height over ground on the disturbance emissions level. While the test results and a comprehensive report are available, some additional work on post processing and data interpretation has been done as a part of this thesis and is presented here.

The test setup in Figure D-1 is based on CISPR TR 30 and is the same test bed previously described in section 2.2.

While omitted in Figure D-1, LISN has been inserted in order to prevent external noise contaminating the test and to present defined impedance between the phase conductor and the ground and between the neutral conductor and ground. Noise is generated in the electronic ballast as harmonics of the switching frequency and emitted to the main wires (right hand side in Figure D-1) and the lamp wires (left hand side in Figure D-1). The noise signal measured on the active and neutral conductors is then due to:

1. Direct propagation from the source; 2. Coupling from the lamp wire close to the main wire (blue wire in Figure D-1); 3. Coupling from the lamp wires to the main wires due to sharing a common

reference plane which is not perfectly grounded.

What are the expectations?

• Coupling Path 1: no effect of wire separation and wire height; • Coupling Path 2: coupling decreases with increasing distance and increases

with increased height; • Coupling Path 3: little influence of height and distance between lamp wire

and main wires.

Wire separation distances as well as wire elevation above ground have been varied as per Table D-1 to check and verify these expectations.

Figure D-1Measurement set-up for height and wire separation variation – from [2]

Disturbance voltages in dBµV have been measured for neutral and live conductor, for the frequency range from 9 Hz to 30 MHz, using Peak, Quasi Peak and Average Detector (as specified in CISPR 16). Thus results represent a 3 x 5 x 6 matrix (3 wire heights, 5 wire separations, 6 results) which is shown in Table D -1.

Table D - 1 Measurement results matrix details

Wire Separation Distance 2mm 5 mm 10 mm 15 mm 30 mm

01 m

m 1

9 m

m 3

8mm

02_01 05_01 10_01 15_01 30_01 LPK

LQP

LAV

NPK

NQP

NAV

LPK

LQP

LAV

NPK

NQP

NAV

LPK

LQP

LAV

NPK

NQP

NAV

LPK

LQP

LAV

NPK

NQP

NAV

LPK

LQP

LAV

NPK

NQP

NAV

02_19 05_19 10_19 15_19 30_19 LPK

LQP

LAV

NPK

NQP

NAV

LPK

LQP

LAV

NPK

NQP

NAV

LPK

LQP

LAV

NPK

NQP

NAV

LPK

LQP

LAV

NPK

NQP

NAV

LPK

LQP

LAV

NPK

NQP

NAV

02_38 05_38 10_38 15_38 30_38 LPK

LQP

LAV

NPK

NQP

NAV

LPK

LQP

LAV

NPK

NQP

NAV

LPK

LQP

LAV

NPK

NQP

NAV

LPK

LQP

LAV

NPK

NQP

NAV

LPK

LQP

LAV

NPK

NQP

NAV

All results in this document will be labelled according to this table, e.g.NAV0519 means

• NAV – Neutral Wire, Average power detected • 05 – Wire separation is 5 mm • 19 – Distance from the ground is 19 mm

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Due to the large total number of measurements (90), it is not advisable to represent all graphs at once but it is still possible to make some preliminary conclusions. Figure D-2 shows measurement data using a peak detector, for all separation and elevation combinations.

Although there are no defined limits for peak values, and the test set-up deviates from the description in the standard, emissions are notably high.

In accordance to the CISPR requirements there are two separate regions, with different measurements settings, due to the different bandwidth settings:

• From 9-150 kHz, 200 Hz bandwidth, with frequency steps of 100 Hz; • From 150 kHz -30 MHz, 9 kHz bandwidth, with frequency steps of 4.5 kHz.

Figure D-3 shows quasi peak measurement data and, as expected, the values are lower than the peak values, but they still exceed the limits, by about 10 dB, for some parameters, just above 150 kHz.

Figure D-4 shows measurement results for average values; emissions still exceed limits by a few dB for some parameters at frequencies just above 150 kHz.

All measurements data (Peak, Quasi Peak and Average) for the frequency range from 0.15 to 6 MHz are presented in Figure D-5. Although cluttered, this graph is important as it shows that the majority of excessive emissions occur between 0.15 MHz and 1.2 MHz.

Figure D-2 Peak power measurement data

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Figure D-3 Quasi peak measurement data

Figure D-4 Average measurement data

Some conclusions that can be drawn from looking at these measurements are:

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• Increasing the distance between wire and the ground results in greater EMI. As expected, the increase is significantly more for the increase from 1 to 19 mm from the ground, than for 19 to 38 mm, as seen in Figure D-6. The red line (h=1 mm) is always lower than the blue (19 mm), and the blue one is always lower than the green one (38 mm), although the difference between the 19 mm and the 38 mm results is only marginal.

• Increasing the distance between the wires decreases EMI, most of the time, but not always. In some cases, an increase in wire separation increases EMI, which shows some frequency dependence that we need to investigate. Looking at Figure D-7, one would expect the green line (w=1 mm) to be the highest throughout the frequency range, as this is the smallest separation between the lines. Around 4 MHz, the 15 mm results are more or less equal to the 5 mm results, although the separation between the wires is three times as large, and coupling should therefore be weaker. Around 27 MHz, the 30 mm results, those for the widest separation, are the highest, exceeding even the 1 mm results.

• The oscillator frequency of the ballast under test is around 40 kHz and multiples of this frequency is seen as harmonics in all graphs. These harmonics are easy to identify in the frequency range 9 kHz to 150 kHz, as shown in Figure D-8. The emission is spread over approximately 7 kHz for the fundamental. For higher harmonics, the spectrum of each peak spreads wider, and above the 4th harmonics the individual peaks begin to overlap. The highest frequency belonging to the 4th harmonic (4 x 43 kHz = 172 kHz) is only 8 kHz lower than the lowest frequency belonging to the 5th harmonic (5 x 36 kHz = 180 kHz); this difference is less than the resolution bandwidth in this frequency band which is 9 kHz.

As for all linear fluorescent lamps, around 40 kHz is the fundamental switching frequency, 80 kHz and 120 kHz are the 2nd and 3rd harmonic.

Figure D-9 gives a closer look at the fundamental for some of the measurement data. The second and the third harmonic are shown in Figures D-9 and D-10 respectively. In each of these 3 figures, a family of curves for the Average Power measurements is presented, in order to see how EMI values were changing for the change in elevation from the ground (1 to 38 mm, solid lines) and separation between the wires (2 to 30 mm, dotted lines).

Results for the fundamental (Figure D-8) are independent of wire separation and height, but with the second and third harmonic the signal increases with increasing height and decreasing separation. There is no frequency shift.

0

10

20

30

40

50

60

70

80

90

0.00

90.05

220.09

540.13

860.94

2.02 3.1

4.18

5.26

6.34

7.42 8.5

9.58

10.66

11.74

12.82

13.9

14.98

16.06

17.14

18.22

19.3

20.38

21.46

22.54

23.62

24.7

25.78

26.86

27.94

29.02

Level in dB

µV

Frequency in MHz

Emissions level vs Elevation

Elevation 1 mm Elevation 19 mm Elevation 38 mm

Figure D-5 Emissions levels vs. wire elevation

0

10

20

30

40

50

60

70

80

90

0.00

90.04

960.09

020.13

080.68

1.69

52.71

3.72

54.74

5.75

56.77

7.78

58.8

9.81

510

.83

11.845

12.86

13.875

14.89

15.905

16.92

17.935

18.95

19.965

20.98

21.995

23.01

24.025

25.04

26.055

27.07

28.085

29.1

Level in dB

µv

Frequency in MHz

Emissions level vs Wires separation

1 mm 5 mm 10 mm 15 mm 30 mm

1 mm separation

Figure D-6 Emissions levels vs. wire separation

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Figure D-7 Harmonics

Figure D-8 First harmonic-zoomed in

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Figure D-9 Second harmonic - zoomed in

Figure D-10 Third harmonic – Zoomed in

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Appendix E

Emergency lighting measurements

Emergency lighting is designed to continue functioning even if the mains power is switched off. This is generally achieved by using a battery, which provides light for about 2 hours. Regardless of the number of lamps in the light fixtures, only a single lamp functions in this mode. When mains power is on, it charges the battery through the inverter.

Though regulations vary between the countries, some 5 to 10 % of all lighting

fixtures are emergency lamps.

Emergency lighting has 3 main modes of operation investigated here:

1. Maintained mode – mains power on, battery not used

2. Emergency mode – mains power off, battery used

3. Non Maintained mode – mains power not connected, battery used

There are good reasons for close examination of emergency lighting. As an additional component, the inverter is another possible source of excessive emissions. The combination of compliant electronic ballast and a compliant emergency inverter can also lead to excessive emission due to:

• The wiring and thus the coupling between noisy lamp wires and sensitive mains wires can be different;

• Running lamp wires from the ballast through the inverter can change the load characteristics and thus lead to higher emission.

• This hypothesis is supported by the measurements conducted in the laboratory.

Due to the addition of an inverter and the battery, emergency lighting measurement is quite complex. Not only are there inputs and outputs on the ballast, there are also connections on the inverter and the battery. The measurements conducted at Curtin University between November and December 2009 wer comprehensive. With one exception, all measurements were done in 2 frequency ranges, 0.1 to 2 MHz, and 0.5 to 50 MHz. Only portion of the measured data has been used for model development, with the remainder retained for future references.

Four different groups of measurements were made:

• Basic mode – no inverter/battery connected

• Maintained mode – mains power on, battery not used

• Emergency mode – mains power off, battery used

• Non Maintained mode – mains power not connected, battery used

In addition to this, 2 different types of inverters were used, a Clevertronics PCP36 and a Luxalite BILW2/364EIDC.

Figure E-1 is a pictorial representation of the measurements and requires some explanation. All voltage measurements were done on LISN using a 10dB attenuator in order to prevent damage to the EMC receiver and this value was later added during post processing. The measurement was done using a FCC F-2000 VHF/UHF current probe. As currents were reasonably small, a 25 dB amplifier was used and this value later subtracted from the measurement in the post processing stage. The first measurement, presented in Figure E-2, was made to establish a benchmark, with no inverter or battery, just the ballast. It is has been designated “the basic configuration”.

Figure E-1 Measurements summary

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Figure E-2 Basic configuration measurements schematic

As noted in the measurement summary, not all the current and voltages measured are important, only the wires at the lamp 1 near and Far End, as well as conductors connected to LISN. The shorter pair of wires in diagram was considered Near End, and longer pair Far End, like the measurements in Chapter 2-2. In reality, ballast is usually located in the middle of a tray and all wires are approximately the same length.

The following table explains the labelling standard used for measurements, as different manufacturers use different labelling methods. As shown in the schematic on following pages, the lines at the short end were colored purple, and at Near End, red. Ballast’s neutral input was blue and active brown. The top row represents wire labels for the Numerical model, presented in section 3. B, C and L stands for the basic model, Clevertronics and Luxalite inverter. EM and NM stand for emergency mode and non-maintained mode. Results for Lamp input and output are presented in Figure E-3.

Table E-1. Labelling table for measured and simulated data

Model W11 W12 W3 W16 W15 W14 W13 W8 W23 W35 B T3 T4 T3_4 T11 T12 T13 T14 T13_14 W23 W35 C T3 T4 T3_4 T3 T8 T6 T4 T13_14 W23 W35 CEM T3 T4 T3 T8 T6 T4 CNM N A T4 T5 T6 T7 L T3 T4 T3_4 T2 T4 T1 T3 T13_14 W23 W35 LEM T3 T4 T2 T4 T1 T3 W23 W35 LNM N A T2 T4 T1 T3

Wires 23 and 35 are located at the end of a coupling path, so these are important observation points for all configurations.

The next set of measurements was conducted using the basic configuration as per Figure E-3 with the addition of an inverter, firstly Clevertronics and then Luxalite.

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Finally, the last set of measurements were conducted for a non-maintained mode. The schematic is presented in Figure E-3. In this mode ballast was not used and the wiring is a somewhat different, though the conductors of interest are still the same as the basic model, so the labelling is the same.

Figure E-3 Measurement set-up for non –maintained mode

The following graphs show measured voltages and currents. There were two frequency ranges used for measurements:

• 0.5 MHz to 50 MHz

• 0.1 to 2 MHz

Results for LISN voltages, currents on wires 23 and 35 as well as on the lamp near and Far End for the frequency first frequency range are shown in Figures E-4 to E-7.

The same values for the second frequency range are depicted in Figures E-8 to E-16.

From all the graphs, it is clear the inclusion of inverter causes slight increase in LISN voltages, mostly in the frequency range up to 2 MHz.

The emissions drop in emergency and non maintained mode, regardless of the ballast. Harmonics of the ballast are sharper and have different frequencies compared to the basic model.

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Figure E-4 LISN voltages measured, 0.5 – 50 MHz

86

Figure E-5 Measured currents on wires 23 and 35, 0.5 – 50 MHz

87

Figure E-6 Measured Near End currents, 0.5 – 50 MHz

88

Figure E-7 Measured Far End currents, 0.5 – 50 MHz

89

Figure E-8 Measured LISN voltages 0.1 - 2 MHz for B, C and L models

Figure E-9 Measured LISN voltages 0.1 - 2 MHz for C, CEM and CNM models

90

Figure E-10 Measured LISN voltages 0.1 - 2 MHz for B, C and L models

91

Figure E-11 Measured currents on wires 23 and 35 , 0.1 to 2 MHz

92

Figure E-12 Measured currents on Near End wires for B, C and L models

Figure E-13 Measured currents on Near End wires for C, CEM and CNM models

93

Figure E-14 Measured currents on Near End wires for L, LEM and LNM models

Figure E-14 Measured currents on Far End wires for B, C and L models

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Figure E-15 Measured currents on Far End wires for C, CEM and CNM models

Figure E-16 Measured currents on Far End wires for L, LEM and LNM models

95

Appendix F

Some of Matlab Code used in Thesis

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97

98

99

100

101

102

103

104

105

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