Electro-Mechanical Structures for Channel

Emulation

Satyajeet Shinde #1

, Sen Yang #2

, Nicholas Erickson#3

, David Pommerenke #4

, Chong Ding*1

, Douglas White*1

,

Stephen Scearce*1

, Yaochao Yang*2

# Missouri S&T EMC Laboratory, Missouri University of Science and Technology, Rolla, MO 65409, USA

*Cisco Systems, Inc. 1Research Triangle Park, NC,

2San Jose, CA, USA

Abstract— Channel emulators are used to evaluate

communication system performance either in absence of the real

channel or to test the system’s response for varying channel

characteristics. For high speed differential digital channels

bandwidths in excess of 20 GHz are common making it difficult

to recreate the channel performance by electronic means such as

FIR filters. An alternative solution is using a low loss short

transmission line and having its properties modified by

mechanical means. Passive structures are robust, have a

frequency range only limited by the low loss trace, do not add

noise, cannot be damaged by ESD and are very economical. This

paper describes two electro-mechanical structures for

introducing loss and nulling into the frequency response of a

channel. The first part describes the design of a mechanically

tuned quarter-wavelength stub filter that can be used to emulate

the resonances of a channel. In the second part, an electro-

mechanical structure, consisting of Bragg grating and lossy

materials, is constructed to emulate the loss behaviour and the

resonances of a channel.

I. INTRODUCTION

The performance of high speed digital communication

systems can be measured using the real channel, or by

emulating a channel. Emulation allows testing a broad range

of channel characteristics. The response of an electrical

channel is typically characterized by a smooth roll off with

rising frequency caused by copper and dielectric losses and by

nulling caused by reflections, or possibly resonant radiation

losses. The easiest way to emulate the channel is to have a

fixed channel, such as a long cable, or a fixed filter structure.

However, this is inflexible if it comes to observing the

system’s performance for varying channel parameter.

Wideband channel emulators and driver emphasis devices

such as the Tektronix LE320, are implemented using finite

impulse response (FIR) filters such as the Hittite

HMC6545LP5E [1], [2]. The CLE1000 uses lossy materials in

close proximity to a trace to adjust the loss [3]. The MP1825B

allows adjusting the general loss, however it cannot be used to

create nulls [4]. However, all active emulation is limited by its

added noise, the IC’s frequency response, none linear

distortion and the range of adjustability often given by the tap

delay and the number of taps. Cost and ESD sensitivity maybe

further considerations. An alternative approach is to use a low

loss trace and to vary its frequency response by mechanical

means. The bandwidth is then only determined by the low loss

trace and its connectors and the structure is robust against

overload and ESD. We have designed two systems to emulate

the actual channel’s response. These structures are constructed

using low loss Megtron 6 PCB material. These structures form

individual blocks which can be cascaded to construct a more

complex channel. The first part describes the design of a

mechanically tuned quarter-wavelength stub filter that can be

used to emulate the resonances of a channel [5]. In the second

part, an electro-mechanical structure, consisting of Bragg

grating (periodic disturbances of a trace) and lossy materials,

is constructed to emulate the loss behaviour and the

resonances of a channel.

II. MECHANICALLY TUNED BAND-STOP FILTER

A. Concept

A typical channel response may consist of one or multiple

band stops or nulls. To emulate/introduce these band stops in

the channel emulator, we design a mechanically tuneable

quarter-wavelength open ended stub transformer. The tuning

of the resonance frequency, in the range 1GHz to 20 GHz, can

be achieved by changing the length of the stub by mechanical

means. Nulls at odd multiples of the fundamental frequency

are also present, however they are usually not of great concern.

Nulls in the frequency range below one-half of the data rate

strongly influence the eye-diagram, nulls between one-half

and the data rate have a moderate influence, and nulls at

frequencies greater than the data rate show little influence if

the width of the null is not too large. Thus, the first harmonic

dominates the effect of the null on the eye, while the third and

subsequent odd harmonics, which are unavoidable in this

concept, may already fall into a frequency, at which a null in

the channel’s response has little influence on the eye

parameter. The quarter-wave transformer concept is shown in

Fig. 1. Trace Impedance = 50ohm Trace Impedance = 50ohm

50ohm 50ohm

Transmission Line

Stub impedance = ZL

Zin

RL

(Open)

Fig. 1: The quarter-wave open-ended stub transformer.

978-1-4799-5545-9/14/$31.00 ©2014 IEEE 939

Since for the open ended stub, RL is infinity, Zin transforms to

zero, or a short at the stub resonance frequency.

B. Structure

The movable rod is supported by placing it inside a metal

(brass) tube that is soldered onto and along the length of the

micro-strip trace. The metal rod and the brass tube are chosen

such that the outer diameter of the metal rod matches the inner

diameter of the brass tube. Conductive grease is applied

between the movable rod and the brass tube. This ensures

sufficient contact between the brass tube and the metal rod.

Further, in order to allow for tuning two independent nulls at

two different frequencies, two pairs of metal rods and brass

tubes are used. Two tubes are connected on the two top and

bottom side traces with a via-transition in between.

PORT 1

PORT 2

Metal Tube

(Dia=0.8mm)

Ground

Plane

Ground

Via

Signal Via

Metal Rod – Stub

(Dia=0.5mm)

Low Loss Substrate

Microstrip Trace

Airgap

(0.15mm)

Fig. 2: Diagram describing the structure of the mechanically tuned band-

stop filter. Its main structure comprises two microstrip lines on opposite sides of a PCB that are connected by a via.

C. Simulation Model

The structure is simulated in Ansoft HFSS 15.0 using the

frequency domain solver as shown in Fig. 3.

Fig. 3: Full-wave simulation model of the mechanically tuned band-stop filter

in Ansoft HFSS 15.

Lumped ports terminate the microstrip traces. The structure

requires a metal tube soldered onto the micro-strip trace.

Therefore, characteristic impedance of the tube-over-trace

combination must be matched to 50-ohms. The addition of the

metal tube above the trace, gives rise to additional fringing

capacitance from the tube to the ground plane as described in

Fig 4. The combined characteristic impedance the trace-tube

combination becomes lower than that of a micro-strip having

the same trace width. Thus, the trace width of the micro-strip

under the tube is reduced to compensate for the additional

capacitance due to the tube and match the characteristic

impedance to 50-ohms. The S21 magnitude is simulated for

different lengths of the movable metal rod to obtain different

resonance frequencies for the band-stop filter.

Metal Tube

(Outer Diameter=0.8mm)

Substrate

Thickness=20mils

Modified Microstrip

Trace Width = 0.8mm

Ground Plane

Original Microstrip

Trace Width = 1mm

Fringing fields due to the

metal tube

Fields due to the

microstrip

Fig. 4: Fringing fields due to the metal tube and the micro-strip trace.

D. Design and Construction

The optimized dimensions obtained from the full-wave

simulations are used to construct a 2-layer, printed circuit

board on the low loss substrate – Megtron 6. The layout is

shown in Fig. 5. Brass tubes are soldered onto the microstrip

trace and SMA connectors are mounted. The assembled

structure is shown in Fig. 6 below.

Fig. 5: Printed circuit board layout.

Trace

Width=0.82mm

Trace

Width=1mmSMA

Connector

Fixed tube mounted on this

portion of the micro-strip

trace

Signal-via

transition

Via cage along the signal

lineSMA

Connector

Top ground for movable

rod impedance transformerMovable Rod

(Stub)

Fig. 6: Photo of the structure. The board is mounted on a metal sheet for

mechanical support.

E. Measurement setup and results

The S21 magnitude of the structure is measured using a

Vector Network Analyzer up to 20 GHz, using two ports

connected to the two SMA connectors. The measurement

setup is shown in Fig. 7 below. Only one movable rod is used

940

for the measurements. The measurement results are compared

with the simulation results for two different lengths of the

movable rod. Two band stops corresponding to the two

different lengths can be observed in Fig. 8.

Fig. 7: Measurement setup to measure the S21 of the band-stop filter.

0 5 10 15 20-50

-45

-40

-35

-30

-25

-20

-15

-10

-5

0

Frequency (MHz)

Ma

gn

itu

de

S21 (

dB

)

Meas-Rod-len-0mm

Sim-Rod-len-0mm

Meas-Rod-len-10mm

Sim-Rod-len-10mm

Meas-Rod-len-3.4mm

Sim-Rod-len-3.4mm

Different lengths of the movable rod: 0mm, 2mm,

4mm, 7mm

Fig. 8: Measurement and simulation comparison for different lengths of the

movable rod length.

III. METHODS TO CHANGE THE Q-FACTOR AND DEPTH OF THE

BAND STOPS

The mechanical structure described above can be used to

introduce band stops at different frequencies by changing the

position of the movable rod. For a practical application of

such a filter for channel emulation of a channel, there is a

requirement to be able to change the Q-factor and depth of the

band stops, such that the desired ‘shape’ of the resonance can

be emulated. We investigate some of the methods to change

the Q-factor of the band stops.

A. Effect of lossy materials in close proximity to the stub:

The Q-factor of the band stop can be reduced by placing

lossy materials in close proximity to the movable rod/stub.

The measurement result comparison before and after

introducing lossy material close to the stub is shown in the Fig.

9. It must be noted here that the lossy material also introduces

a shift in the resonant frequency; however the desired

resonance frequency can be obtained by tuning the stub length.

0 5 10 15 20-50

-45

-40

-35

-30

-25

-20

-15

-10

-5

0

Frequency (MHz)

Ma

gn

itud

e S

21 (

dB

)

Without Lossy Material

With Lossy Material

Lossy material

Fig. 9: Simulated S21 magnitude for proximity of lossy material to the

movable rod.

B. Effect of increasing the height of the stub (movable rod)

above the ground plane:

On increasing the height of the stub above the ground plane,

the characteristic impedance of the stub is increased. This

results in the reduction of the null depth. The simulation result

comparison for two different heights 1.05mm and 0.25mm is

shown in Fig. 10. The rod length is kept at 7.3mm and 6.6mm

for the rod height of 0.15mm and 1.05mm respectively. The

length is changed to correct for the slight shift in resonance

frequency caused due to the change in height. The simulation

result shows a reduction in the null depth, caused due to an

increase in the characteristic impedance of the stub.

0 5 10 15 20-40

-35

-30

-25

-20

-15

-10

-5

0

Frequency (GHz)

Mag

nit

ud

e S

21 (

dB

)

Rod-height-0.15mm

Rod-height-1.05mm

Movable Rod: 1.05 mm

above ground plane

Movable Rod: 0.25 mm

above ground plane

Fig. 10: Simulated S21 magnitude for different heights of movable rod above

the ground plane – 1.05mm and 0.25mm.

941

C. Effect of loss of the Rod Material: Brass and Graphite-

composite

The material of the movable rod has an influence on the Q-

factor of the resonance. The comparison between the results of

the measurement carried out using a metal rod and a graphite-

composite rod, in Fig. 11, shows the difference in the null

depths. The loss factors of the rods affect the null depth of the

resonance. The null depth can be adjusted from -10dB to -

35dB using different lossy materials for the rod.

Fig. 11: Measured S21 magnitude for different movable rod materials – Graphite and Brass. The null depth is reduced by about 20dB.

D. Effect of ground plane impedance variation:

The depth of the null can be tuned by changing the

impedance of the current return path under the quarter-wave

stub. In the simulation model, the ground plane structure

under the stub was modified by designing a copper patch and

connecting the patch with a resistive boundary to the

surrounding ground plane as described in Fig. 12. The

resistance of this boundary was parametrically varied to obtain

the S21. The simulated S21 for different resistance values are

shown in Fig. 13. The results show that the null depth reduces

as the resistance value is increased. For practical

implementation, PIN diodes, used as variable resistors, can be

used to change the impedance of the current return path.

Ground

Plane

Ground

Plane

Copper

Patch

Variable

resistance

boundary

Fig. 12: Simulation model with the modified ground plane and copper patch

connected to the ground plane with a variable resistance boundary.

Fig. 13: Simulated S21 magnitude of the modified ground structure for

different resistance values.

E. Effect of Rod diameter:

Changing the diameter of the movable rod influences the

width of the resonance. The simulation result comparison for

rod diameters 0.5mm and 0.1mm is shown in Fig. 14. The rod

length in both cases is kept at 7.3mm. The result shows that a

smaller rod diameter results in a reduction in the width of the

null. This can be explained as result of a smaller rod diameter

and an increase in the characteristic impedance of the stub.

Fig. 14: Simulated S21 magnitude for different diameters of the movable rod –

0.5mm and 0.1mm.

IV. TARGET S-PARAMETERS VS EMULATED S-PARAMETERS

The mechanical band-stop filter is used to emulate the s-

parameters of a measured channel. The S21 magnitude of the

measured channel shows a notch at around 4.4 GHz and

general loss behaviour. Fig. 15 shows the target and the

emulated S21 magnitude. The notch is tuned by tuning the

movable rod of the mechanical band stop filter and the general

loss behaviour is emulated by placing lossy materials on the

trace.

0 5 10 15 20-35

-30

-25

-20

-15

-10

-5

0

Frequency (GHz)

Mag

nit

ud

e S

21 (

dB

)

0-ohm

1-ohm

3-ohm

10-ohm

50-ohm

942

Fig. 15: Target and emulated S21 magnitude by tuning the movable rod and

lossy materials on the microstrip trace.

V. ELECTROMECHANICAL STRUCTURE FOR CHANNEL

EMULATION

The mechanical structure for channel emulation utilizes a

combination of two structures, for emulating a given channel

response. The Bragg introduces band stops, based on the

concept of periodic and non-periodic discontinuities on a

transmission line. The lossy material lifter emulated the

smooth loss function of the channel.

A. Structure and Construction:

The structure mainly consists of two parts, as mentioned

above. The mechanical Bragg sliders are placed on the top of

a differential microstrip pair and the lossy material lifter on

the bottom. The schematic is shown in Fig. 16. The

mechanical Bragg structure, shown in Fig. 17, drives five

identical and equally spaced sliders on top of the trace. These

sliders function as periodic discontinuities for the transmission

line. The sliders over the trace are shown in Fig. 19. By

changing the distance between each slider, the notch

frequency on S21 can be changed. By setting them in non-

periodic distances other perturbations of the channel can be

achieved. The microstrip has a slotted ground. This way,

perturbations of the field can be introduced from the top and

from the bottom. The lossy material lifter, shown in Fig. 18,

lifts the lossy material to the underside of the PCB and

introduces losses by attenuating the fields that pass through

the slotted ground plane. By varying the space between the

lossy material and the slot, the general loss function of the

channel response can be emulated.

A prototype of the structure was built using copper-clad

PCB structure. DC motors are used to control the position of

individual sliders over the differential trace pair. One motor is

used to control the height of the lossy material under the slot.

The control circuit for the DC motors consists of motor

drivers which are controlled by a microcontroller having a

USB interface. The position of the sliders and the height of the

lossy material under the trace is controlled using the motor

drivers.

GND plane

with slot

PCB

Lossy material

Trace

Slot

Bragg on top of the trace

Fig. 16: Diagram describing the structure with the Bragg sliders and the lossy

material.

Slider

Potentiometer

Motor

Fig. 17: The mechanical structure showing the sliders of the Bragg structure.

Lossy

material

Lifter

Fig. 18: The lossy material lifter to lift the loss material towards the slot under

the trace.

B. Measurement Setup and Results:

The S21 magnitude of the structure is measured using a

Vector Network Analyzer up to 20 GHz, using two ports

connected to the two SMA connectors. This is a single ended

measurement; however the structure can support differential

microstrip lines. The measurement setup is shown in Fig. 20.

Two measurements are recorded – in the first case, the

separation between the Bragg grating is changed while

keeping the height of the lossy material at a fixed height. In

the second measurement, the height of the lossy material

under the slot is changed while keeping the Bragg sliders in a

fixed position. The S21 measurement result comparison for the

first case is shown in Fig. 21 which shows the variation of the

0 2 4 6 8 10-50

-45

-40

-35

-30

-25

-20

-15

-10

-5

0

Frequency (GHz)

Ma

gn

itu

de

S21 (

dB

)

Target S-parameter

Emulated S-parameters

943

band stops due to the change in the separation between the

Bragg sliders.

Bragg sliders over trace

Fig 19. Bragg sliders over trace.

VNA

DC Power

Supply

Control

Circuit

USB

Controller

Mechanical

structure

Mechanical

Bragg

Lossy Material Lifter

Fig. 20: Measurement Setup showing the mechanical structure and the control

circuitry.

Fig.21: Measured S21 magnitude for different spacing between the Bragg grating showing different band stops.

Fig. 22 shows the measurement comparison results for the

second case which shows the change in the roll-off function of

the S21 magnitude response with the change in the height of

the lossy material under the slots in the ground plane of the

board.

Fig. 22: Measured S21 magnitude for different distances of proximity of the

lossy material to the slot under the trace.

VI. CONCLUSION

This paper describes two electro-mechanical structures for

emulating different characteristics of the frequency response

of a channel. The structures when constructed using low loss

substrate (eg. Megtron 6) for the printed circuit boards can be

cascaded. The designs described here are relatively low cost

and simple to implement as compared to other methods that

use integrated circuit based channel emulators. The upper

limit of usable frequency range for the mechanically tuned

band-stop filter is determined by the via-transition, edge-

launch connectors, and the diameter of the fixed and movable

tube. Band stops at the higher order odd harmonics of the stub

are also present, which place a lower limit on the resonance

frequency that can be emulated. For the motor controlled

mechanical structure, the accuracy of emulating the frequency

response of a given channel depends on the precision with

which the Bragg grating and lossy material lifter can be

positioned using the control circuit.

VII. ACKNOWLEDGEMENT

This material is based upon work supported by the National

Science Foundation under Grant No. 0855878. We also thank

Cisco Systems Inc., USA for their support towards this work.

VIII. REFERENCES

[1] http://www.tek.com/bit-error-rate-tester/digital-preemphasis.

[2] http://www.hittite.com/products/view.html/view/HMC6545LP5E

[3] http://www.aceunitech.com/products/cle1000.html

[4] http://www.anritsu.com/en-US/Products-

Solutions/Products/MP1825B.aspx.

[5] D. M. Pozar, and D. H. Schaubert, Microstrip Antennas, IEEE

Press 1995.

0 5 10 15 20-30

-25

-20

-15

-10

-5

0

Frequency (GHz)

Ma

gn

itu

de

S21 (

dB

)

No Bragg

Bragg Spacing-1

Bragg Spacing-2

0 5 10 15 20-20

-18

-16

-14

-12

-10

-8

-6

-4

-2

0

Frequency (GHz)

Ma

gn

itu

de

S21 (

dB

)

No Loss

Loss-1

Loss-2

944