Effective mirror-mapping-based intercarrier interference cancellation for OFDM underwater acoustic communications

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Ad Hoc Networks xxx (2014) xxx–xxx

ADHOC 1072 No. of Pages 13, Model 3G

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Contents lists available at ScienceDirect

Ad Hoc Networks

journal homepage: www.elsevier .com/locate /adhoc

Effective mirror-mapping-based intercarrier interferencecancellation for OFDM underwater acoustic communications q

http://dx.doi.org/10.1016/j.adhoc.2014.07.0151570-8705/� 2014 Published by Elsevier B.V.

q Parts of the results in this paper were presented at the International Conference on Underwater Networks and Systems (WUWNet), Kaohsiung2013, and the Asilomar Conference on Signals, Systems and Computers, Pacific Grove, CA, 2013.⇑ Corresponding author.

E-mail addresses: [email protected] (X. Cheng), [email protected] (M. Wen), [email protected] (X. Cheng), dduaedu (D. Duan), [email protected] (L. Yang).

Please cite this article in press as: X. Cheng et al., Effective mirror-mapping-based intercarrier interference cancellation for OFDMwater acoustic communications, Ad Hoc Netw. (2014), http://dx.doi.org/10.1016/j.adhoc.2014.07.015

Xilin Cheng a, Miaowen Wen b, Xiang Cheng c, Dongliang Duan d, Liuqing Yang a,⇑a Department of Electrical and Computer Engineering, Colorado State University, Fort Collins, CO 80523, USAb School of Electronic and Information Engineering, South China University of Technology, Guangzhou 510641, Chinac School of Electronics Engineering and Computer Science, Peking University, Beijing 100871, Chinad Department of Electrical and Computer Engineering, University of Wyoming, Laramie, WY 82071, USA

a r t i c l e i n f o

29303132333435363738394041

Article history:Received 24 March 2014Received in revised form 16 July 2014Accepted 29 July 2014Available online xxxx

Keywords:OFDMUnderwater acoustic communicationsICI cancellation algorithmsMirror-mapping ruleCIR derivationMultipath Rayleigh fading channels

424344454647484950

a b s t r a c t

Orthogonal frequency-division multiplexing (OFDM) techniques are promising for under-water acoustic (UWA) communications due to their robustness against large delay spread.However, OFDM systems in UWA channels suffer from intercarrier interference (ICI) causedby the Doppler effect. In this paper, we propose four low-complexity mirror-mapping-basedICI cancellation schemes in OFDM UWA communications without explicitly estimating ICIcoefficients or carrier frequency offset (CFO), namely the mirror symbol repetition (MSR)scheme, the mirror conjugate symbol repetition (MCSR) scheme, the mirror conversiontransmission (MCVT) scheme, and the mirror conjugate transmission (MCJT) scheme. Wetheoretically evaluate their performance by deriving their carrier-to-interference powerratio (CIR) expressions in multipath Rayleigh fading channels with CFO. The key idea ofour proposed schemes is to employ data repetition within one or two consecutive OFDMsymbols according to the mirror-mapping rules and apply the conjugate or conversion oper-ations to remove ICI. Theoretical analyses and numerical results show that the proposedmirror-mapping-based schemes can effectively improve the CIR in comparison with theplain OFDM and the adjacent-mapping-based schemes. In addition, the effects of the chan-nel length and the CFO deviation between two consecutive OFDM symbols are investigated.Finally, to demonstrate their benefits for OFDM UWA communications, these schemes havealso been tested in a recent sea experiment conducted in Taiwan in May 2013. The decodingresults validate that the proposed mirror-mapping-based schemes significantly improvethe bit error rate (BER) performance of OFDM communications in UWA channels.

� 2014 Published by Elsevier B.V.

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1. Introduction

Orthogonal frequency-division multiplexing (OFDM)has recently attracted great interests from researchers inthe field of underwater acoustic (UWA) communications

due to its capability of combating inter-symbol interfer-ence (ISI) caused by large delay spread, which is quite com-mon in UWA communications [1–15]. The key principle ofOFDM is that the wideband time-invariant channel is con-verted into multiple orthogonal flat fading sub-channels,

, Taiwan,

n@uwyo.

under-

http://dx.doi.org/10.1016/j.adhoc.2014.07.015

mailto:[email protected]







http://www.sciencedirect.com/science/journal/15708705

http://www.elsevier.com/locate/adhoc


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and modulated symbols transmitted over differentsubcarriers do not interfere with each other. However, forUWA channels, the Doppler effect is usually very severedue to the transmitter/receiver motion and ocean waves,which result in rapidly time-varying behaviors. The time-varying features of UWA channels would then destroythe orthogonality among subcarriers and lead to intercarri-er interference (ICI). To ensure satisfactory performance ofOFDM communications in UWA channels, ICI has to besuppressed to an acceptable level.

In the literature, there are mainly three categories of ICIsuppression methods. In the first category, the basisexpansion model (BEM) has been considered for ICI mitiga-tion in some work [10–12]. However, the BEM coefficientestimation introduces additional computational burden atthe receiver. In the second category, the ICI is treatedexplicitly by first estimating the ICI coefficients followedby various ICI cancellation algorithms (see e.g. [1,2,16–18]). This strategy is robust against wide Doppler spread.However, it usually requires significant modifications onthe traditional OFDM transceiver and greatly increasesthe complexity. Therefore, in the third category, anotherlow complexity strategy is taken to treat ICI implicitly. In[3–9], it is believed that ICI can be eliminated by receiveddata resampling and Doppler shift compensation. Thesereceivers do not need to estimate the ICI coefficients andthus facilitate low-complexity channel estimation andsymbol detection. However, additional null subcarrieroverhead, exhaustive search, and/or iterative operationsare often involved. In [19], the adjacent-mapping-basedICI cancellation method is proposed. Each data symbol istransmitted in two adjacent subcarriers. Depending onthe conversion or the conjugate relation between adjacentsubcarrier pairs carrying the same information, there arethe adjacent symbol repetition (ASR) scheme and the adja-cent conjugate symbol repetition (ACSR) scheme. Theseschemes can implicitly cancel ICI by combining receivedsignals on adjacent subcarrier pairs. However, for bothschemes, ICI has not been sufficiently suppressed and therestill exists residual interference from neighbor subcarriers.

In this paper, we also adopt the rationale of the thirdcategory and attempt to bypass the ICI coefficient estima-tion during the ICI cancellation process. Inspecting the ICIcoefficients of the plain OFDM systems, we find that forany particular subcarrier, there is a very interesting rela-tionship between the interference from mirrored subcarri-er around. Based on this observation, we exploit datarepetition within OFDM symbols according to some care-fully designed subcarrier mapping rules and operationssuch that after combining the subcarrier pairs carryingthe same information, the carrier-to-interference powerratio (CIR) can be significantly improved. The resultantICI cancellation schemes feature the simplicity of imple-mentation and the effectiveness of ICI mitigation.

There are two options to implement data repetition andthe desired mirror-mapping operation. One is a self-can-cellation approach, where the data is repeated within oneOFDM symbol by mirror-mapping and the other one is ageneral two-path cancellation approach, where the datais repeated across two consecutive OFDM symbols. Com-pared with the self-cancellation approach, the latter one

Please cite this article in press as: X. Cheng et al., Effective mirror-mappwater acoustic communications, Ad Hoc Netw. (2014), http://dx.doi.org

is conveniently compatible with the traditional OFDMtransceiver design without any modifications. Detailedcomparisons are made analytically and by numericalresults and experiments. After data repetition, due to theproperty of the ICI coefficients, one also has two optionsto process the mirror-mapped data, including the conver-sion and conjugate operations. Combining all theseoptions, we have four schemes in total, namely the mirrorsymbol repetition (MSR) scheme, the mirror conjugatesymbol repetition (MCSR) scheme, the mirror conversiontransmission (MCVT) scheme, and the mirror conjugatetransmission (MCJT) scheme. For comparison purposes,we derive closed-form expressions of their CIRs. The CIRanalyses and numerical results show that significant ICIcancellation can be achieved compared with the plainOFDM and the adjacent-mapping-based schemes, and theCIR expression for flat fading channels is identical to thatfor additive white Gaussian noise (AWGN) channels.Furthermore, we find the underlying relationship betweenthe two options to implement the data repetition,namely the self-cancellation schemes and the two-pathcancellation schemes. In addition, the effects of channellength and CFO deviation between two consecutive OFDMsymbols are investigated. It is found that the two-path can-cellation schemes are sensitive to the CFO deviation, andthe CIR performance of the conversion-based schemesdegrades with large channel length. Therefore, the schemeselection depends on the actual system requirement andthe channel condition. Finally, all four schemes have beentested in a recent sea experiment conducted in Taiwan inMay 2013. Decoding results confirm that all proposedschemes significantly improve the OFDM UWA systemperformance.

The rest of the paper is organized as follows. In Section 2,the properties of the ICI coefficients are investigated. In Sec-tion 3, we describe the proposed mirror-mapping-based ICIcancellation schemes. The ICI cancellation modules at thetransmitter and the receiver are explained. In Section 4, CIRsare derived for all schemes and the CIR comparison is alsoprovided. Section 5 provides decoding results based onthe sea experiment data. Finally, summarizing remarksare given in Section 6.

Notation: Superscripts ð�ÞT and ð�ÞH stand for the trans-pose and the Hermitian transpose, respectively; IN is theN � N identity matrix; 0M�N denotes the all-zero matrixof size M � N; FN represents the N � N discrete Fouriertransform (DFT) matrix with the ðm;nÞ-th entry being

e�j2pN ðm�1Þðn�1Þ; % is the modulo operator.

2. Properties of ICI coefficients

In this section, the ICI coefficients of the plain OFDMsystem are investigated in order to guide our proposedICI cancellation schemes. Assume the multipath Rayleigh

fading channels are given by h ¼ h0;h1; . . . ;hL�1½ �T , whereL is the number of channel taps. Each tap is subject toindependent Rayleigh fading with hl � CN ð0;r2

l Þ for l 2f0;1; . . . ; L� 1g, where Ef hlj j2g ¼ r2

l and Ef hlj j4g ¼ 2r4l .

When L ¼ 1, the multipath Rayleigh fading channel isreduced to the flat fading channel. For an OFDM-based

ing-based intercarrier interference cancellation for OFDM under-/10.1016/j.adhoc.2014.07.015


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communication system, the channel frequency response(CFR) in the frequency domain (FD) is given by

H ¼ffiffiffiffiNp

FNPh, where P ¼ IL;0L�ðN�LÞ� �T is the zero-padding

matrix. The k-th element of H is Hk ¼PL�1

l¼0 e�j2pN klhl, and is

circularly symmetry complex Gaussian distributed, i.e.,Hk � CNð0;gÞ, where g is the total power of all channel

taps with g ¼PL�1

l¼0 Ef hlj j2g ¼PL�1

l¼0 rlj j2. It is worth notingthat the channel at each subcarrier has the same power

level, i.e., Ef Hkj j2g ¼ g for k 2 f0;1; . . . ;N � 1g. We alsoassume that in the rest of the paper, the major Dopplereffect has been removed through received signal resam-pling and only residual CFO exists [7].

In multipath Rayleigh fading channels, the receivedbaseband signals in the time domain (TD) is given by:

yn ¼1N

XN�1

k¼0

HkXkej2pnðkþeÞ=N þxn; n ¼ 0;1; . . . ;N � 1; ð1Þ

where Hk is the CFR at subcarrier k;xn is the noise, and e isthe CFO.

After DFT, the received signals in the FD can beexpressed as:

Ym¼ S0HmXmþXN�1

k¼0;k–m

Sk�mHkXkþWm; m¼0;1; . . . ;N�1;

ð2Þ

where Wm is the noise in the FD, and

Sk ¼sin peð Þ

N sin p kþ eð Þ=Nð Þ ej pe 1�1

Nð Þ�pkNð Þ ð3Þ

is the ICI coefficient [20,19].The ICI coefficient Sk is a periodic function with period N.

The amplitude of Sk is plotted in Fig. 1 with e ¼ 0:1 andN ¼ 1000. It can be observed that for large values of kj j;Skj j goes to 0. This means ICI mainly comes from neighbor-

ing subcarriers. The phase of Sk is \Sk ¼ p e 1� 1N

� �� k% N

N

� �for 0 < e < 1. Thus, for small values of kj j;\Sk � pe and\S�k � pe� p. This implies Sk þ S�k � 0. In addition, forsmall values of e;\Sk � 0 and \S�k � �p. This meansSk þ S��k � 0. It is worth noting that Sk þ S�k � 0 is a more

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accurate approximation than Sk þ S��k � 0. As we will seein the next sections, the mirror-mapping-based schemesare designed on the basis of the aforementioned properties(see Fig. 2).

3. Proposed ICI cancellation schemes

3.1. ICI self-cancellation with mirror-mapping

Fig. 3 depicts the system architecture of the ICI self-can-cellation schemes with mirror-mapping. Compared withthe plain OFDM, the ICI self-cancellation schemes havetwo additional modules, i.e., the ICI self-canceling modula-tion before the inverse fast Fourier transform (IFFT)operation at the transmitter and the ICI self-cancelingdemodulation after the fast Fourier transform (FFT) opera-tion at the receiver.

For the ICI self-canceling modulation, the input modu-lated data symbols are first grouped into transmit blocks.Each block consists of ðN=2� 1Þ modulated data symbolsfXgN=2�1

k¼1 , which are then mapped onto N subcarriers usingthe one-to-two mirror-mapping rule as follows:

eXk ¼0; k ¼ 0;N=2Xk; k ¼ 1;2; . . . ;N=2� 1O XN�kð Þ; k ¼ N=2þ 1; . . . ;N � 1;

8><>: ð4Þ

where feXkgN�1

k¼0 are the actual transmitted data symbols on

the OFDM subcarriers. O xð Þ is defined as the mappingoperation which reflects the relationship between thetwo modulated data symbols with the same information.

It is worth noting that eXN�k ¼ OðeXkÞ; k ¼ 1;2; . . . ;N=2� 1represents mirror-mapping. The conversion operationand the conjugate operation can be represented asO xð Þ ¼ �x and O xð Þ ¼ x�, respectively. The 0-th and N=2-th subcarriers are vacant in order to meet the oppositepolarity condition. Thus, we have the MSR scheme andthe MCSR scheme, corresponding to the conversion opera-tion and the conjugate operation, respectively.

After the ICI self-canceling demodulation, the receivedsignals on subcarrier m;m 2 f1;2; . . . N=2� 1g and its cor-responding mapped subcarrier pair ðm0 ¼ N �mÞwill carry

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Fig. 2. The phase of Sk .



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Tx

Modulation Add CP LowpassFiltering

Rx

Demodulation Discard CP

LowpassFiltering

MirrorMapping

Conversion /Conjugate IFFT

Conversion /Conjugate

MirrorDe-Mapping FFT

Channel Estimation

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X

ICI self-canceling modulation

ICI self-canceling demodulation

Fig. 3. Block diagram of an OFDM transceiver with the ICI self-cancellation modules using mirror-mapping in the baseband.



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the same data information. This signal redundancy rendersit possible to improve the ICI mitigation performancethrough a coherent combining technique:

bXm ¼H�mYm þO H�N�mYN�m

� �Hmj j2þ HN�mj j2

; ð5Þ

where Hm;HN�m;Ym, and YN�m are the CFR and the receivedsignals at subcarrier m and its corresponding subcarrierpair ðm0 ¼ N �mÞ, respectively. Note that Eq. (5) is essen-tially maximum ratio combining (MRC).

3.2. ICI two-path cancellation with mirror-mapping

Fig. 4 depicts the system architecture of the ICI two-path cancellation schemes with mirror-mapping. Thetwo-path cancellation schemes transmit the input modu-lated data symbols in two consecutive OFDM symbols,which are usually referred to as two independent pathsseparated by time division multiplexing (TDM). Evidently,the main additional operations due to the introduction oftwo-path cancellation schemes are integrated inside theprecoding and decoding modules.

In general, for the two-path cancellation schemes withmirror-mapping, at the precoding module, one OFDM sym-

bol input fXkgN�1k¼0 will become two OFDM symbol outputs

Fig. 4. Block diagram of an OFDM transceiver with the ICI two-path


fXð1Þk gN�1

k¼0 and fXð2Þk gN�1

k¼0 , where the first OFDM symbol

fXð1Þk gN�1

k¼0 is identical to the input OFDM symbol, i.e.,

Xð1Þk ¼ Xk, and the second OFDM symbol fXð2Þk gN�1

k¼0 obeysthe subcarrier mirror-mapping rule and can be obtained

as Xð2Þk ¼ O XN�kð Þ; k ¼ f0; . . . ;N � 1g. For ease of exposition,if the conversion operation is utilized for the mappingoperation, i.e., O xð Þ ¼ �x, we refer to it as the MCVTscheme, and if the conjugate operation is used, i.e.,O xð Þ ¼ x�, we call it the MCJT scheme.

After the deliberate design of the transmitted signal inthe precoding module, the received signals at the m-thsubcarrier and the ðN �mÞ-th subcarrier of the first OFDMsymbol and the second OFDM symbol, respectively, willcarry the same data information. Therefore, at the decod-ing module, it is reasonable to use MRC for decoding,yielding:

bXm ¼H�mY 1ð Þ

m þO H�N�mY 2ð ÞN�m

n oHmj j2 þ HN�mj j2

; ð6Þ

where Hm is the CFR at subcarrier m, and Y ið Þm is the received

signals in the FD corresponding to the i-th transmittedOFDM symbol ði 2 f1;2gÞ.

cancellation modules using mirror-mapping in the baseband.



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4. Carrier-to-interference power ratio (CIR) evaluation

CIR is a widely used metric for evaluating the system ICIpower level without considering the noise power. In thissection, we derive the CIRs of the plain OFDM and the pro-posed mirror-mapping-based schemes for performancecomparisons.

4.1. Plain OFDM

Suppose that the transmitted data symbols are mutu-ally independent. According to Eq. (2), the instantaneousCIR of the plain OFDM can be readily derived as:

CIROFDM; Inst ¼S0Hmj j2PN�1

k¼0;k–m Sk�mHkj j2: ð7Þ

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CIRMSRðmÞ ¼E S0ð Hmj j2 þ HN�mj j2Þ � S�2mH�mHN�m � S2mH�N�mHm

�� 2� PN=2�1

k¼1;k–mE Sk�mH�mHk þ Sm�kH�N�mHN�k � S�k�mH�mHN�k � SmþkH�N�mHk

�� 2n o :

The average CIR can be calculated by averaging the aboveinstantaneous CIR expression over the distribution of thechannel gains. However, the calculation of the average

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CIRMSRðmÞ ¼PL�1

l¼0 r4l 2S0 � S�2mej4p

N lm � S2me�j4pN lm

�� 2PN=2�1k¼1;k–m

PL�1l¼0 r4

l Sk�me�j2pN ðk�mÞl þ Sm�kej2p

N ðk�mÞl � S�k�mej2pN ðkþmÞl � Skþme�j2p

N ðkþmÞl�� 2 ; ð10Þ

CIR using multiple integral is overly complicated. Thus,an approximate average CIR expression can be derived bytaking the average of the numerator and the denominatorof Eq. (7) separately. As it is a good predictor of the averageCIR and simple enough to compare different schemes [21],the approximate average CIR is utilized for the CIR deriva-tion in this paper. Then, the CIR of the plain OFDM is:

CIROFDM ¼S0j j2E Hmj j2

n oPN�1

k¼0;k–m Sk�mj j2E Hkj j2n o : ð8Þ

As Ef Hkj j2g ¼ Ef Hmj j2g ¼ g,

CIROFDM ¼S0j j2PN�1

k¼1;k–m Skj j2: ð9Þ

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4.2. MSR

For the MSR scheme, the conversion operationis adopted. The transmitted data symbols in the FD


after the ICI self-canceling modulation areeX1 ¼ �eXN�1 ¼ X1, eX2 ¼�eXN�2 ¼ X2; . . . ; eXN=2�1 ¼�eXN=2þ1 ¼XN=2�1, and eX0 ¼ eXN=2 ¼ 0. According to Eq. (5), the decisionvariable at the m-th subcarrier (m 2 f1; . . . ;N=2� 1g)becomes:

bXm ¼ H�mYm � H�N�mYN�m

¼ ðS0 Hmj j2 þ S0 HN�mj j2 � S�2mH�mHN�m

� S2mH�N�mHmÞXm þXN=2�1

k¼1;k–m

ðSk�mH�mHk

þ Sm�kH�N�mHN�k � S�k�mH�mHN�k � SmþkH�N�mHkÞXk

þ H�mWm � H�N�mWN�m:

The factor 1=ð Hmj j2 þ HN�mj j2Þ is removed from the decisionvariable expression, as it does not affect the CIR value. TheCIR at the m-th subcarrier is expressed as:

For the multipath Rayleigh fading channels, the CIR ofthe MSR scheme at the m-th subcarrier is given by:

and the derivation is given in Appendix A.For the flat fading channel, i.e., L ¼ 1, the CIR becomes:

CIRMSRðmÞ ¼2S0 � S2m � S�2mj j2PN=2�1

k¼1;k–m Sk�m þ S� k�mð Þ � Skþm � S� kþmð Þ�� 2 :

ð11Þ

As Sk þ S�k � 0, the CIR of the MSR scheme can be mark-edly improved.

The average CIR of the MSR scheme is then given by:

CIRMSR ¼2

N � 2

XN=2�1

m¼1

CIRMSRðmÞ: ð12Þ

4.3. MCSR

Similarly, for the MCSR scheme, the conjugate operationis adopted. Accordingly, we have eX1 ¼ eX�N�1 ¼ X1,eX2 ¼ eX �N�2 ¼ X2; . . . ; eXN=2�1 ¼ eX �N=2þ1 ¼ XN=2�1, andeX0 ¼ eXN=2 ¼ 0.



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From Eq. (5), the decision variable at the m-th subcarri-er is given by:bXm ¼ H�mYm þ HN�mY�N�m

¼ ðS0 Hmj j2 þ S�0 HN�mj j2ÞXm þXN=2�1

k¼1;k–m

ðSk�mH�mHk

þ S�m�kHN�mH�N�kÞXk þXN�1

k¼N=2þ1

ðSk�mH�mHk

þ S�m�kHN�mH�N�kÞX�N�k þ H�mWm þ HN�mW�

N�m:

Then, the CIR of the MCSR scheme can be expressed as:

CIRMCSRðmÞ ¼E S0 Hmj j2 þ S�0 HN�mj j2�� 2�

PN�1k¼1;kR m;N=2f gE Sk�mH�mHk þ S�m�kHN�mH�N�k

�� 2n o :ð13Þ

For the multipath Rayleigh fading channels, the CIR ofthe MCSR scheme is:

CIRMCSRðmÞ ¼4R S0f g2PN�1

k¼1;kR N=2�m;N�mf g Sk þ S��k

�� 2 : ð14Þ

See Appendix B for the detailed derivation. As Sk þ S��k � 0,the CIR of the MCSR scheme is improved compared withthat of the plain OFDM. Note that the CIR of the MCSRscheme is not affected by the channel length L.

Accordingly, the average CIR of the MCSR scheme isgiven by:

CIRMCSR ¼2

N � 2

XN=2�1

m¼1

CIRMCSRðmÞ: ð15Þ

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4.4. MCVT

For the MCVT scheme, the conversion operation isadopted. The two consecutive transmitted OFDM symbolsare of the form Xð1Þ ¼ X0;X1; . . . ;XN�1½ � andXð2Þ ¼ �X0;�XN�1; . . . ;�X1½ �. From Eq. (6), the decision var-iable at the m-th subcarrier m ¼ 0; . . . ;N � 1ð Þ is given by:

bXm ¼H�mY ð1Þm � H�N�mY ð2ÞN�m ð16Þ

¼ S0ðeÞ Hmj j2 þ S0ðeþ DeÞ HN�mj j2h i

Xm

þXN�1

k¼0;k–m

Sk�mðeÞH�mHk þ Sm�kðeþ DeÞH�N�mHN�k� �

Xk

þ H�mW ð1Þm � H�N�mW ð2Þ

N�m; ð17Þ

where the CFO of the first symbol is e, the CFO of the sec-ond symbol is eþ De, and W pð Þ

m p ¼ 1;2ð Þ is the noise at them-th subcarrier of the p-th path.

According to Eq. (16), the CIR of the MCVT scheme canbe expressed as:

CIRMCVT ¼E S0ðeÞ Hmj j2 þ S0ðeþ DeÞ HN�mj j2�� 2�

PN�1k¼0;k–mE Sk�mðeÞH�mHk þ Sm�kðeþ DeÞH�N�mHN�k

�� 2n o :ð18Þ


For the multipath Rayleigh fading channels, the CIR of theMCVT scheme is:

CIRMCVT ¼S0ðeÞ þ S0ðeþ DeÞj j2

PL�1l¼0 r4

lPN�1k¼1

PL�1l¼0 r4

l SkðeÞe�j2pN kl þ S�kðeþ DeÞej2pN kl

�� 2 :ð19Þ

The derivation is given in C. Notice that for kl N;

Ske�j2pN kl þ S�kej2p

N kl � Sk þ S�k � 0. Thus, significant CIRimprovement can be expected for a small channel length L.

For the flat fading channel, i.e., L ¼ 1, the CIR of theMCVT scheme becomes:

CIRMCVT ¼S0ðeÞ þ S0ðeþ DeÞj j2PN�1

k¼1 SkðeÞ þ S�kðeþ DeÞj j2: ð20Þ

4.5. MCJT

When the conjugate operation is adopted, we obtain theMCJT scheme. In this case, the two consecutive transmittedOFDM symbols are given by Xð1Þ ¼ X0;X1; . . . ;XN�1½ � andXð2Þ ¼ X�0;X

�N�1; . . . ;X�1

� �. According to Eq. (6), the decision

variable at the m-th (m 2 f0;1; � � � ;N � 1g) subcarrier isgiven by:bXm ¼ H�mY ð1Þm þ HN�mY ð2Þ�N�m

¼ S0ðeÞ Hmj j2 þ S�0ðeþ DeÞ HN�mj j2h i

Xm

þXN�1

k¼0;k–m

Sk�mðeÞH�mHk þ S�m�kðeþ DeÞHN�mH�N�k

� �Xk

þ H�mW ð1Þm þ HN�mW ð2Þ�

N�m:

ð21Þ

According to Eq. (21), the CIR of the MCVT scheme can beexpressed as:

CIRMCJT ¼E S0ðeÞ Hmj j2 þ S�0ðeþ DeÞ HN�mj j2�� 2�

PN�1k¼0;k–mE Sk�mðeÞH�mHk þ S�m�kðeþ DeÞHN�mH�N�k

�� 2n o :ð22Þ

For the multipath Rayleigh fading channels, the CIR ofthe MCJT scheme is given by:

CIRMCJT ¼S0ðeÞ þ S�0ðeþ DeÞ�� 2PN�1

k¼1 SkðeÞ þ S��kðeþ DeÞ�� 2 ; ð23Þ

which is derived in Appendix D. Also, it is worth notingthat the CIR of the MCJT scheme is not affected by thechannel length L.

Similarly to MCSR, we see here that the denominator inthe CIR expression is the summation overSkðeÞ þ S��kðeþ DeÞ, which is approximately zero for k–0and small De. In addition, by comparing with Eq. (14), itcan be found CIRMCJT � CIRMCSR when De ¼ 0.

Finally, it is worth noting that in flat fading channels,the CFRs at all subcarriers are same, and the CFR coeffi-cients are inherently cancelled in the CIR expression. Thus,the CIR of each scheme, namely Eqs. 9, 11, 14, 20, 23, hasexactly the same expression as that in AWGN channels.



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0.02 0.04 0.06 0.08 0.1 0.12 0.14 0.16 0.18 0.2

10

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70

ε

CIR

(dB

)

Normal OFDMMSRMCSRMCVT, Δε = 0

MCJT, Δε = 0

MCVT, Δε = 0.03

MCJT, Δε = 0.03ASRACSR

Fig. 5. CIR comparison among the plain OFDM, the adjacent-mapping-based schemes, and the mirror-mapping-based schemes for differentvalues of e over flat fading channels.

0.02 0.04 0.06 0.08 0.1 0.12 0.14 0.16 0.18 0.2

10

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30

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60

ε

CIR

(dB

)

Normal OFDMMCJTMCVT, L=1MCVT, L=10MCVT, L=50MCVT, L=100

Fig. 7. CIR comparison among the plain OFDM, the MCVT scheme, and theMCJT scheme for different values of e over multipath Rayleigh fadingchannels.



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4.6. CIR comparison

To compare the CIR performance among the plainOFDM, the adjacent-mapping-based schemes, and the mir-ror-mapping-based schemes, an OFDM system withN ¼ 1024 subcarriers is considered. Fig. 5 presents theCIR results of flat fading channels, i.e., L ¼ 1. For the ICItwo-path cancellation schemes, the CIR results withDe ¼ 0 and De ¼ 0:03 are presented. From the figure, thefollowing facts can be observed: (1) All mirror-mapping-based schemes express much better CIR performance thanthe plain OFDM; (2) MSR and MCSR outperform ASR andACSR, respectively. This shows that the mirror-mappingrule has better ICI suppression capability than the adja-cent-mapping rule. The main reason is that for the mir-ror-mapping-based schemes, the interference fromneighbor subcarriers is sufficiently suppressed, while thereis still some residual interference from neighbor subcarri-ers for the adjacent-mapping-based schemes; (3) with

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0.02 0.04 0.06 0.08 0.1 0.12 0.14 0.16 0.18 0.2

10

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30

40

50

60

70

ε

CIR

(dB

)

Normal OFDMMCSRMSR, L=1MSR, L=10MSR, L=50MSR, L=100

Fig. 6. CIR comparison among the plain OFDM, the MSR scheme, and theMCSR scheme for different values of e over multipath Rayleigh fadingchannels.


De ¼ 0, the CIR curves of the MCSR scheme and the MCJTscheme coincide, and the CIR of the MSR scheme is closeto that of the MCVT scheme. This is because the same oper-ation is adopted for both schemes; (4) with De ¼ 0, the CIRperformance of the conversion-based schemes, namelyMSR and MCVT, is better than that of the conjugate-basedschemes, namely MCSR and MCJT. This is because in theirCIR expressions Sk þ S��k � 0 is a rougher approximationthan Sk þ S�k � 0; and (5) with De ¼ 0:03, the CIR perfor-mance of the MCVT scheme and the MCJT scheme isdegraded due to the CFO deviation between the first OFDMsymbol and the second OFDM symbol. Figs. 6 and 7 presentthe CIR results of multipath Rayleigh fading channels.Assume each channel tap has the same power, i.e.,r2

0 ¼ r21 ¼ � � � ¼ r2

L�1. The effect of the channel length L isinvestigated. From the figure, it can be observed that forthe multipath Rayleigh fading channels, the mirror-map-ping-based schemes have better CIR performance com-pared with that of the plain OFDM. The CIR performanceof the plain OFDM and the conjugate-based schemes,namely MCSR and MCJT, is not affected by the channellength L. For the conversion-based schemes, namely MSRand MCVT, the CIR performance is degraded with the chan-nel length L. For the OFDM-based communications, tomaintain the spectral efficiency, N is usually much largerthan the channel length L. Thus, the conversion-basedschemes can effectively reduce ICI for the long OFDM sym-bol design.

In summary, based on the CIR results of flat fadingchannels and multipath Rayleigh fading channels, we con-clude that our proposed mirror-mapping-based schemescan effectively mitigate ICI.

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5. Sea experiment results

Up to now, we have shown theoretically the effective-ness of the mirror-mapping-based schemes in multipathRayleigh fading channels with simple CFO. In this section,we will verify the applicability of our proposed schemesin UWA communications by sea experiments.



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Fig. 8. Geographical locations of transceiver nodes. The GPS coordinates of nodes 4, 5, and 9 are (N22.66038, E120.21450), (N22.64844, E120.21405), and(N22.64169, E120.23450), respectively. Their relative distances are 1549.58 m (from node 4 to node 5), 2187.60 m (from node 5 to node 9), and 3377.72 m(from node 9 to node 4).

Table 1System parameters.

Sampling rate at the transmitter 48 kHzSampling rate at the receiver 48 kHzSignal bandwidth 5.36 kHzCarrier frequency 17 kHzNumber of total subcarriers 1600Number of data subcarriers 1278Number of pilot subcarriers 214Number of null subcarriers 108Subcarrier spacing 3.35 HzOFDM symbol duration 299 msGuard interval 50 msNumber of hydrophones 4



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5.1. Experiment settings

The experiment was conducted in a sea area about 3 kmeast of Gushan, Taiwan, in May 21–22, 2013, which is illus-trated in Fig. 8. Three nodes were deployed, i.e., node 4,node 5, and node 9, each of which consisted of 1 transducerand 4 hydrophones. The sea depth is around 20 m and thenode depth is around 10 m. The transducer and the hydro-phones may drift due to waves. During the sea test, thenodes transmitted with each other and the received datapackets were recorded.

The basic system parameters are provided in Table 1.The bandwidth of the system is 5.36 kHz. The total numberof subcarriers is 1600, within which there are 1278 datasubcarriers, 214 pilot subcarriers, and 108 null subcarriers.The guard interval has a length of 50 ms, which is muchlonger than the maximum channel delay spread. The pilotsubcarriers are used for channel estimation. For the ICItwo-path cancellation schemes, the pilots are uniformlyinserted among the data subcarriers. However, for the ICIself-cancellation schemes, to avoid loss of spectral effi-ciency, the mirror-mapped pilot structure has to beadopted. Thus, the ICI self-cancellation schemes do notfacilitate the OFDM system standardization. In addition,QPSK modulation is adopted for the mirror-mapping-basedschemes.

To demonstrate the performance of the mirror-mapping-based schemes, there are two benchmark candi-dates, both of which have the same spectral efficiency asthe mirror-mapping-based schemes. The first one is theplain OFDM with BPSK modulation (OFDM-B). The secondone is the OFDM scheme with half of total subcarriersoccupied by QPSK data symbols and each of data symbols


surrounded by two null subcarriers (OFDM-QH). ForOFDM-QH, the ICI from direct neighbors is removed.Fig. 9 compares the BER performance of OFDM-B andOFDM-QH under different CFOs. It is found that OFDM-Bhas lower BER than OFDM-QH in low-medium SNR range(SNR < 20, BER > 10�3). This is because of the higher sym-bol detection error for QPSK modulation. Considering highsignal attenuation of underwater acoustic channels andrelatively low SNR at the receive hydrophones, OFDM-B,the one with better BER, is chosen as the benchmark inthe sea experiment.

The frame structure for transmission is given in Fig. 10.The transmitted frame consists of a preamble, transmittedOFDM data symbols, and a postamble. We allocate 5 OFDMsymbols with BPSK modulation for the plain OFDM and 10OFDM symbols with QPSK modulation for either the ICIself-cancellation schemes or the two-path cancellationschemes. Therefore, 17 OFDM symbols are involved in eachframe in the experiment. In addition, Gray code is adopted



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−10 −5 0 5 10 15 20 25 30 35 40

10−4

10−3

10−2

10−1

SNR

BE

R

OFDM−BOFDM−QH

ε = 0.1

ε = 0

ε = 0.2

Fig. 9. BER performance comparison between OFDM-QH and OFDM-B forCFOs e ¼ 0;0:1;0:2. The total subcarrier number is N ¼ 1024, the band-width is 6 kHz, and the channel spread length is 9 ms. For OFDM-QH,Gray code is adopted for QPSK modulation.

Preamble Plain OFDM (BPSK) 5 symbols

ICI Cancellation (QPSK) 10 symbols Postamble

Fig. 10. Frame structure.

Table 2Packet information.

Packet index Scheme EffectiveSNR

Transmit time T–R pair

M0000043.DAT MCJT 12.47 18:16:00 05/21/13 N5–N9M0000044.DAT MCJT 12.55 18:16:25 05/21/13 N5–N9M0000046.DAT MCSR 12.27 18:16:55 05/21/13 N5–N9M0000047.DAT MCSR 12.15 18:17:21 05/21/13 N5–N9M0000049.DAT MSR 12.21 18:17:51 05/21/13 N5–N9M0000050.DAT MSR 12.29 18:18:16 05/21/13 N5–N9M0000052.DAT MCVT 12.31 18:18:46 05/21/13 N5–N9M0000053.DAT MCVT 12.36 18:19:11 05/21/13 N5–N9

Fig. 11. BER performance of the plain OFDM and the mirror-mapping-based schemes.



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for QPSK modulation. Depending on the ICI cancellationstructure at the last 10 OFDM symbols, there are four dif-ferent frames, implementing the MSR scheme, the MCSRscheme, the MCVT scheme, and the MCJT scheme, respec-tively. In the experiment, ignoring preamble and CP over-head, the data rate of mirror-mapping-based schemesand OFDM-B is 4.27 kbps.

5.2. Time synchronization and resampling

For time synchronization, the received signal is corre-lated with the preamble and the postamble to obtain thestarting and ending time of the received data frame. Then,the resampling factor is calculated by comparing thereceived signal length and the transmitted signal length.Finally, to remove the major Doppler effect, the receivedsignal is resampled according to the resampling factor [7].

5.3. Channel estimation

The CFR of each subcarrier can be estimated via thereceived signals on the pilot subcarriers. The CFRs on thepilot subcarriers are estimated first. Then the CFRs on thedata subcarriers can be obtained through the piecewisecubic spline interpolation. To combat the time-varying fea-ture of UWA channels, channel estimation is done for eachOFDM symbol.

5.4. Experiment results

To obtain the bit error rate (BER) results and enable thefair comparison of different schemes, the packages with


indices ‘‘M0000043.DAT’’, ‘‘M0000044.DAT’’, ‘‘M0000046.DAT’’, ‘‘M0000047.DAT’’, ‘‘M0000049.DAT’’, ‘‘M0000050.DAT’’, ‘‘M0000052.DAT’’, and ‘‘M0000053. DAT’’ are utilizedwith their information provided in Table 2. They are trans-mitted consecutively with similar receive SNR levels andthrough the same transmitter–receiver (T–R) pair. Fig. 11illustrates the BER performance of OFDM-B, the mirror-mapping-based ICI cancellation schemes, and the plainOFDM with QPSK modulation (OFDM-Q). The BER ofOFDM-Q is calculated by means of directly decoding themirror-mapping-based schemes without combining datasubcarrier pairs and thus ICI is not suppressed. Asexpected, all mirror-mapping-based schemes have lowerBER than OFDM-B and OFDM-Q for all hydrophones. Thisconfirms that the mirror-mapping-based schemes canachieve superior ICI mitigation in OFDM UWA communica-tions. Different from the CIR results, it is observed that allfour schemes have similar BER performance. This is notsurprising since the CIR performance is not directly relatedto the BER performance [22]. The theoretical analyses ofthe proposed mirror-mapping-based schemes over fre-quency-selective UWA channels in terms of BER will beconsidered as our future work.

6. Conclusions

In this paper, to mitigate the detrimental effect of ICI inOFDM UWA communications, we have proposed foureffective low-complexity mirror-mapping-based ICI can-cellation schemes without explicitly estimating ICI coeffi-cients or CFO and derived their CIRs in multipathRayleigh fading channels. From the theoretical analysesand numerical results, it has been revealed that the mir-



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ror-mapping-based schemes outperform the plain OFDMand the adjacent-mapping-based schemes. By comparingthe ICI self-cancellation schemes and the ICI two-pathcancellation schemes, the ICI self-cancellation schemes,namely MSR and MCSR, are robust against the CFO devia-tion between the first OFDM symbol and the second OFDMsymbol. However, they require the mirror-mapped pilotstructure, which does not facilitate the OFDM systemstandardization. In addition, compared with the conju-gate-based schemes, the conversion-based schemes,namely MSR and MCVT, have better CIR performance.However their CIR performance degraded for multipathRayleigh fading channels with large channel length. Thus,the scheme selection depends on the actual systemrequirement and channel conditions. Finally, all mirror-mapping-based schemes have been tested in a recent seaexperiment conducted in Taiwan in May 2013. Decodingresults have shown that the proposed mirror-mapping-based schemes provide much lower BER than the plainOFDM. This confirms that the mirror-mapping-basedschemes are very effective for ICI mitigation in OFDMUWA communications.

Acknowledgments

This work was jointly supported by the National Sci-ence Foundation (Grant No. CNS-1129043), National Natu-ral Science Foundation of China (Grant No. 61101079), theScience Foundation for the Youth Scholar of Ministry ofEducation of China (Grant No. 20110001120129), the Min-istry of Transport of China (Grant No. 2012-364-X03-104),and the National 863 Project (Grant No. 2014AA01A706).We would like to take this opportunity to thank Dr. ShengliZhou for accommodating our data in the Taiwan sea exper-iment in May 2013.

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Appendix A. Derivation of CIR of MSR

The channel response at subcarrier m is given by

Hm ¼PL�1

l¼0 hle�j2pN ml. Accordingly,

S0ð Hmj j2 þ HN�mj j2Þ � S�2mH�mHN�m � S2mH�N�mHm

¼XL�1

a¼0

XL�1

b¼0

h�ahbF1ða; bÞ;

where F1ða; bÞ ¼ S0e�j2pN ðb�aÞm þ S0ej2p

N ðb�aÞm � S�2mej2pN ðaþbÞm

�S2me�j2pN ðaþbÞm. Then,

S0ð Hmj j2 þ HN�mj j2Þ � S�2mH�mHN�m � S2mH�N�mHm

�� 2¼XL�1

a¼0

XL�1

b¼0

XL�1

c¼0

XL�1

d¼0

h�ahbhch�dF1ða; bÞF�1ðc;dÞ:

The multipath Rayleigh fading channels have the fol-lowing property:

E hah�bhch�d �

¼ E haj j4n o

; a ¼ b ¼ c ¼ d

0; else:

(


Based on this property, we can obtain:

E S0ð Hmj j2 þ HN�mj j2Þ � S�2mH�mHN�m � S2mH�N�mHm

�� 2� ¼XL�1

l¼0

E hlj j4n o

F1ðl; lÞj j2

¼XL�1

l¼0

2r4l 2S0 � S�2mej4pN lm � S2me�j4pN lm�� 2:

In a similar way,

Sk�mH�mHk þ Sm�kH�N�mHN�k � S�k�mH�mHN�k � SmþkH�N�mHk

¼XL�1

a¼0

XL�1

b¼0

h�ahbF2ða; bÞ;

where F2ða; bÞ ¼ Sk�me�j2pN ðkb�maÞ þ Sm�kej2p

N ðkb�maÞ

� S�k�mej2pN ðkbþmaÞ � Skþme�j2p

N ðkbþmaÞ:

Then,

E Sk�mH�mHk þ Sm�kH�N�mHN�k � S�k�mH�mHN�k � SmþkH�N�mHk

�� 2n o¼XL�1

l¼0

E hlj j4n o

F2ðl; lÞj j2

Thus, CIRMSR in Eq. (10) can be obtained.

Appendix B. Derivation of CIR of MCSR

It can be shown that:

S0 Hmj j2 þ S�0 H2N�m

�� ¼XL�1

a¼0

XL�1

b¼0

hah�bðS0e�j2pN ða�bÞm þ S�0ej2pN ða�bÞmÞ

¼XL�1

a¼0

XL�1

b¼0

2hah�bR S0e�j2pN ða�bÞm

n oand


�� 2 ¼XL�1

a¼0

XL�1

b¼0

XL�1

c¼0

XL�1

d¼0

4hah�bh�c hdR S0e�j2pN ða�bÞm

n o�R S0e�j2pN ðc�dÞm

n o:

Thus, for the multipath Rayleigh fading channels, we have:

E�


�� 2�

¼ 4R S0f g2XL�1

l¼0

E hlj j4n o

:

In a similar way,

Sk�mH�mHk þ S�m�kHN�mH�N�k ¼XL�1

a¼0

XL�1

b¼0

e�j2pN ðkb�maÞðSk�mh�ahb

þ S�m�khah�bÞ

and

Sk�mH�mHkþ S�m�kHN�mH�N�k

�� 2 ¼XL�1

a¼0

XL�1

b¼0

XL�1

c¼0

XL�1

d¼0

e�j2pN ðkb�maÞ�ðkd�mcÞ½ �

� ðSk�mh�ahbþS�m�khah�bÞ�ðS�k�mhch�dþ Sm�kh�chdÞ:



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9 August 2014

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For the multipath Rayleigh fading channels, we have:

E Sk�mH�mHk þ S�m�kHN�mH�N�k

�� 2n o¼XL�1

l¼0

E hlj j4n o

ð Sk�mj j2 þ Sk�mSm�k þ S�m�kS�k�m þ Sm�kj j2Þ

¼XL�1

l¼0

E hlj j4n o

Sk�m þ S�m�k

�� 2:Thus, CIRMCVT in Eq. (14) can be obtained.

Appendix C. Derivation of CIR of MCVT

Since

S0ðeÞ Hmj j2 þ S0ðeþ DeÞ HN�mj j2

¼XL�1

a¼0

XL�1

b¼0

hah�b S0ðeÞe�j2pN mða�bÞ þ S0ðeþ DeÞej2pN mða�bÞ

h i;

it can be derived that:

S0ðeÞ Hmj j2 þ S0ðeþ DeÞ HN�mj j2�� 2¼XL�1

a¼0

XL�1

b¼0

XL�1

c¼0

XL�1

d¼0

hah�bh�chd

� S0ðeÞe�j2pN mða�bÞ þ S0ðeþ DeÞej2p

N mða�bÞh i� S�0ðeÞej2p

N mða�bÞ þ S�0ðeþ DeÞe�j2pN mða�bÞ

h i:


E S0ðeÞ Hmj j2 þ S0ðeþ DeÞ HN�mj j2�� 2� ¼XL�1

l¼0

E hlj j4n o

S0ðeÞ þ S0ðeþ DeÞj j2:

In the same way,

Sk�mðeÞH�mHk þ Sm�kðeþ DeÞH�N�mHN�k

¼XL�1

a¼0

XL�1

b¼0

h�ahb Sk�mðeÞe�j2pN ðkb�maÞ þ Sm�kðeþ DeÞej2p

N ðkb�maÞh i

and

Sk�mðeÞH�mHk þ Sm�kðeþ DeÞH�N�mHN�k

�� 2¼XL�1

a¼0

XL�1

b¼0

XL�1

c¼0

XL�1

d¼0

h�ahbhch�d

� Sk�mðeÞe�j2pN ðkb�maÞ þ Sm�kðeþ DeÞej2p

N ðkb�maÞh i� S�k�mðeÞej2p

N ðkd�mcÞ þ S�m�kðeþ DeÞe�j2pN ðkd�mcÞ

h i:


E Sk�mðeÞH�mHk þ Sm�kðeþ DeÞH�N�mHN�k

�� 2n o¼XL�1

l¼0

E hlj j4n o

Sk�mðeÞe�j2pN ðk�mÞl þ Sm�kðeþ DeÞej2p

N ðk�mÞl�� 2:


Thus, CIRMCVT in Eq. (19) can be obtained.

Appendix D. Derivation of CIR of MCJT

It can be readily derived that:

S0ðeÞ Hmj j2 þ S�0ðeþ DeÞ H2N�m

�� ¼XL�1

a¼0

XL�1

b¼0

hah�b S0ðeÞe�j2pN ða�bÞm

hþS�0ðeþ DeÞej2pN ða�bÞm

iand


�� 2¼XL�1

a¼0

XL�1

b¼0

XL�1

c¼0

XL�1

d¼0

hah�bh�chd

� S0ðeÞe�j2pN ða�bÞm þ S�0ðeþ DeÞej2p

N ða�bÞmh i� S�0ðeÞej2p

N ðc�dÞm þ S0ðeþ DeÞe�j2pN ðc�dÞm

h i:

Thus, for the multipath Rayleigh fading channels, wehave:

E�


�� 2� ¼ S0ðeÞ þ S�0ðeþ DeÞ�� 2XL�1

l¼0

E hlj j4n o

:

In a similar way,


¼XL�1

a¼0

XL�1

b¼0

e�j2pN ðkb�maÞðSk�mðeÞh�ahb þ S�m�kðeþ DeÞhah�bÞ

and


�� 2¼XL�1

a¼0

XL�1

b¼0

XL�1

c¼0

XL�1

d¼0

e�j2pN ðkb�maÞ�ðkd�mcÞ½ �!

� ðSk�mðeÞh�ahb þ S�m�kðeþ DeÞhah�bÞ� ðS�k�mðeÞhch�d þ Sm�kðeþ DeÞh�chdÞ:


E Sk�mðeÞH�mHk þ S�m�kðeþ DeÞHN�mH�N�k

�� 2n o¼XL�1

l¼0

E hlj j4n o

Sk�m þ S�m�k

�� 2:Accordingly, CIRMCVT in Eq. (23) can be obtained.

References

[1] Kai Tu, Dario Fertonani, Tolga M. Duman, Milica Stojanovic, John G.Proakis, Paul Hursky, Mitigation of intercarrier interference forOFDM over time-varying underwater acoustic channels, IEEE J.Ocean. Eng. 36 (2) (2011) 156–171.

[2] Jianzhong Huang, Shengli Zhou, Jie Huang, Christian R. Berger, PeterWillett, Progressive inter-carrier interference equalization for OFDMtransmission over time-varying underwater acoustic channels, IEEEJ. Select. Topics Sig. Process. 5 (8) (2011) 1524–1536.



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[3] Taehyuk Kang, Ronald A. Iltis, Iterative carrier frequency offset andchannel estimation for underwater acoustic OFDM systems, IEEE J.Select. Areas Commun. 26 (9) (2008) 1650–1661.

[4] Milica Stojanovic, Low complexity OFDM detector for underwateracoustic channels, in: Proceedings of MTS/IEEE Oceans Conference,Boston, MA, September 18–21, 2006, pp. 1–6.

[5] Milica Stojanovic, OFDM for underwater acoustic communications:Adaptive synchronization and sparse channel estimation, in:Proceeding of International Conference on Acoustics, Speech andSignal Processing, Las Vegas, NV, March 30–April 4, 2008, pp. 5288–5291.

[6] Yunus Emre, Vinod Kandasamy, Tolga M. Duman, Paul Hursky,Subhadeep Roy, Multi-input multi-output OFDM for shallow-waterUWA communications, J. Acoust. Soc. Am. 123 (5) (2008)3891.

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[10] Fengzhong Qu, Liuqing Yang, Basis expansion model for underwateracoustic channels? in: Proceedings of MTS/IEEE Oceans Conference,Quebec City, Canada, September 15–18, 2008, pp. 1–7.

[11] Geert Leus, Paul A. van Walree, Multiband OFDM for covert acousticcommunications, IEEE J. Select. Areas Commun. 26 (9) (2008) 1662–1673.

[12] Sean Frederick Mason, Christian R. Berger, Shengli Zhou, Keenan R.Ball, Lee Freitag, Peter Willett, Receiver comparisons on an OFDMdesign for Doppler spread channels, in: Proceedings of MTS/IEEEOceans Conference, Bremen, May 11–14, 2009, pp. 1–7.

[13] Xiaoka Xu, Zhaohui Wang, Shengli Zhou, Lei Wan,Parameterizing both path amplitude and delay variations ofunderwater acoustic channels for block decoding of orthogonalfrequency division multiplexing, J. Acoust. Soc. Am. 131 (June)(2012) 4672.

[14] Xiaoka Xu, Gang Qiao, Jun Su, Pengtao Hu, Enfang Sang, Study onturbo code for multicarrier underwater acoustic communication, in:Proceedings of 2008 International Conference on WirelessCommunications, Networking and Mobile Computing, Dalian,October 12–14, 2008, pp. 1–4.

[15] Xilin Cheng, Rui Cao, Fengzhong Qu, Liuqing Yang, Relay-aidedcooperative underwater acoustic communications: selectiverelaying, in: Proceedings of MTS/IEEE Oceans Conference, Yeosu,Korea, May 21–24, 2012, pp. 1–7.

[16] Tiejun Wang, John G. Proakis, James R. Zeidler, Techniques forsuppression of intercarrier interference in OFDM systems, in:Proceedings of Wireless Communications and NetworkingConference, vol. 1, March 13–17, 2005, pp. 39–44.

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Xilin Cheng (S’10) received his M.S. degree inDepartment of Electrical and Computer Engi-neering from University of Florida, Gaines-ville, Florida. He is currently pursuing thePh.D. degree with the Department of Electricaland Computer Engineering, Colorado StateUniversity, Fort Collins, Colorado. His currentresearch interests include underwater acous-tic communications, cooperative relay com-munications, and rateless erasure coding.

Miaowen Wen received the B.Eng. degree ininformation engineering from Beijing JiaotongUniversity, Beijing, China, in 2009. He is cur-rently pursuing his Ph.D. degree in PekingUniversity, Beijing, China. Since September2012, he has been a visiting student researchcollaborator in Princeton University, Prince-ton, USA. His recent research interests includemultiple access techniques, MIMO, and coop-erative communication.

Xiang Cheng (S’05-M’10-SM’13) received thePhD degree from Heriot-Watt University andthe University of Edinburgh, Edinburgh, U.K.,in 2009, where he received the PostgraduateResearch Thesis Prize. He has been with Pek-ing University, Bejing, China, since 2010, firstas a Lecturer, and then as an Associate Pro-fessor since 2012. His current research inter-ests include mobile propagation channelmodeling and simulation, next generationmobile cellular systems, intelligent transpor-tation systems, and hardware prototype

development. He has published more than 80 research papers in journalsand conference proceedings. He received the best paper award from theIEEE International Conference on ITS Telecommunications (ITST 2012)
and the IEEE International Conference on Communications in China (ICCC2013). Dr. Cheng received the ‘‘2009 Chinese National Award for Out-standing Overseas Ph.D. Student’’ for his academic excellence and out-standing performance. He has served as Symposium Co-Chair and aMember of the Technical Program Committee for several internationalconferences.
Dongliang Duan received the B.S. degreefrom Huazhong University of Science andTechnology, Wuhan, China in 2006, the M.S.degree from the University of Florida,Gainesville, FL in 2009, and the Ph.D. degreefrom the Colorado State University, Fort Col-lins, CO in 2012, all in electrical engineering.Since graduation from CSU, he has been anassistant professor in the department ofElectrical and Computer Engineering at theUniversity of Wyoming. His research interestsinclude signal processing techniques for

wireless communications and power systems, estimation and detectiontheory, and energy resource management in wireless communicationsystems. Currently, he is particularly interested in statistical signal pro-
cessing and wireless communications and networking for power gridmonitoring and control.


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Liuqing Yang (S’02-M’04-SM’06) received herPh.D. degree from the University of Minne-sota, Minneapolis in 2004. After that, shejoined the University of Florida, Gainesville,and became an Associate Professor in 2009.She is currently an Associate Professor atColorado State University. Her general inter-ests are in areas of communications and signalprocessing. Dr. Yang was the recipient of theONR YIP award in 2007, the NSF CAREERaward in 2009, the IEEE Globecom Outstand-ing Service Award in 2010, George T. Abell

Outstanding Mid-Career Faculty Award at CSU in 2012, and Best PaperAwards at IEEE ICUWB’06 and ICCC’13. She has been actively serving the


technical community, including organization of many IEEE internationalconferences, and the editorial board for a number of journals includingIEEE Transactions on Communications, IEEE Transactions on WirelessCommunications, IEEE Transactions on Intelligent Transportation Sys-tems, and IEEE Intelligent Systems Magazine.



Documents

Effective mirror-mapping-based intercarrier interference cancellation for OFDM underwater acoustic communications