Dual-Phase, Parallelable, Average Current-Mode Controllers · PHASE = VCC 115 120 125 PHASE = unconnected 85 90 95 CLKOUT Phase Shift (at fSW = 125kHz) φCLKOUT PHASE = SGND 55 60

General DescriptionThe MAX5038/MAX5041 dual-phase, PWM controllersprovide high-output-current capability in a compactpackage with a minimum number of external compo-nents. The MAX5038/MAX5041 utilize a dual-phase,average current-mode control that enables optimal useof low RDS(ON) MOSFETs, eliminating the need for exter-nal heatsinks even when delivering high output currents.

Differential sensing enables accurate control of the out-put voltage, while adaptive voltage positioning providesoptimum transient response. An internal regulatorenables operation with input voltage ranges of +4.75V to+5.5V or +8V to +28V. The high switching frequency, upto 500kHz per phase, and dual-phase operation allowthe use of low-output inductor values and input capacitorvalues. This accommodates the use of PC board-embedded planar magnetics achieving superior reliabili-ty, current sharing, thermal management, compact size,and low system cost.

The MAX5038/MAX5041 also feature a clock input(CLKIN) for synchronization to an external clock, and aclock output (CLKOUT) with programmable phase delay(relative to CLKIN) for paralleling multiple phases. TheMAX5038 offers a variety of factory-trimmed preset outputvoltages (see Selector Guide) and the MAX5041 offers anadjustable output voltage from +1.0V to +3.3V.

The MAX5038/MAX5041 operate over the extendedindustrial temperature range (-40°C to +85°C) and areavailable in a 28-pin SSOP package. Refer to theMAX5037 data sheet for a VRM 9.0-compatible, VID-controlled output voltage controller in a 44-pin MQFP orQFN package.

ApplicationsServers and Workstations

Point-Of-Load High-Current/High-DensityTelecom DC-DC Regulators

Networking Systems

Large-Memory Arrays

RAID Systems

High-End Desktop Computers

Features♦ +4.75V to +5.5V or +8V to +28V Input Voltage

Range

♦ Up to 60A Output Current

♦ Internal Voltage Regulator for a +12V or +24VPower Bus

♦ True Differential Remote Output Sensing

♦ Two Out-Of-Phase Controllers Reduce InputCapacitance Requirement and Distribute PowerDissipation

♦ Average Current-Mode ControlSuperior Current Sharing Between IndividualPhases and Paralleled Modules

Accurate Current Limit Eliminates MOSFET andInductor Derating

♦ Integrated 4A Gate Drivers♦ Selectable Fixed Frequency 250kHz or 500kHz Per

Phase (Up to 1MHz for 2 Phases)

♦ Fixed (MAX5038) or Adjustable (MAX5041) OutputVoltages

♦ 0.5% Accurate Reference (MAX5041B)

♦ External Frequency Synchronization from 125kHzto 600kHz

♦ Internal PLL with Clock Output for ParallelingMultiple DC-DC Converters

♦ Thermal Protection

♦ 28-Pin SSOP Package

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Dual-Phase, Parallelable, Average Current-ModeControllers

________________________________________________________________Maxim Integrated Products 1

19-2514; Rev 3; 8/04

For pricing, delivery, and ordering information, please contact Maxim/Dallas Direct! at 1-888-629-4642, or visit Maxim’s website at www.maxim-ic.com.

Ordering Information

PART TEMP RANGEPIN-PACKAGE

OUTPUTVOLTAGE

MAX5038EAI12 -40°C to +85°C 28 SSOP Fixed +1.2V





MAX5041EAI -40°C to +85°C 28 SSOPAdj +1.0V to+3.3V

MAX5041BEAI -40°C to +85°C 28 SSOPAdj +1.0V to+3.3V

Pin Configuration appears at end of data sheet.

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ABSOLUTE MAXIMUM RATINGS

ELECTRICAL CHARACTERISTICS(VCC = +5V, circuit of Figure 1, TA = -40°C to +85°C, unless otherwise noted. Typical specifications are at TA = +25°C.) (Note 1)

Stresses beyond those listed under “Absolute Maximum Ratings” may cause permanent damage to the device. These are stress ratings only, and functionaloperation of the device at these or any other conditions beyond those indicated in the operational sections of the specifications is not implied. Exposure toabsolute maximum rating conditions for extended periods may affect device reliability.

IN to SGND.............................................................-0.3V to +30VBST_ to SGND…………………………………….…-0.3V to +35VDH_ to LX_ ................................-0.3V to [(VBST_ - VLX_) + 0.3V]DL_ to PGND..............................................-0.3V to (VCC + 0.3V)BST_ to LX_ ..............................................................-0.3V to +6VVCC to SGND............................................................-0.3V to +6VVCC to PGND............................................................-0.3V to +6VSGND to PGND .....................................................-0.3V to +0.3V

All Other Pins to SGND...............................-0.3V to (VCC + 0.3V)Continuous Power Dissipation (TA = +70°C)

28-Pin SSOP (derate 9.5mW/°C above +70°C) ............762mWOperating Temperature Range ...........................-40°C to +85°CMaximum Junction Temperature .....................................+150°CStorage Temperature Range .............................-60°C to +150°CLead Temperature (soldering, 10s) .................................+300°C

PARAMETER SYMBOL CONDITIONS MIN TYP MAX UNITS

SYSTEM SPECIFICATIONS

8 28

Input Voltage Range VIN Short IN and VCC together for +5V inputoperation

4.75 5.5V

Quiescent Supply Current IQ EN = VCC or SGND 4 10 mA

Efficiency η ILOAD = 52A (26A per phase) 90 %

OUTPUT VOLTAGE

MAX5038 only, no load -0.8 +0.8Nominal Output VoltageAccuracy

MAX5038 only, no load, VIN = VCC =+4.75V to +5.5V or VIN = +8V to +28V(Note 2)

-1 +1%

MAX5041 only, no load 0.992 1.008

MAX5041 only, no load, VIN = VCC =+4.75V to +5.5V or VIN = +8V to +28V

0.990 1.010

MAX5041B only, no load 0.995 1.005SENSE+ to SENSE- VoltageAccuracy

MAX5041B only, no load,VIN = +8V to +28V

0.995 1.005

V

STARTUP/INTERNAL REGULATOR

VCC Undervoltage Lockout UVLO VCC falling 4.0 4.15 4.5 V

VCC Undervoltage LockoutHysteresis

200 mV

VCC Output Accuracy VIN = +8V to +28V, ISOURCE = 0 to 80mA 4.85 5.1 5.30 V

MOSFET DRIVERS

Output Driver Impedance RON Low or high output 1 3 Ω

Output Driver Source/SinkCurrent

IDH_, IDL_ 4 A

Non-Overlap Time tNO CDH_/DL_ = 5nF 60 ns

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ELECTRICAL CHARACTERISTICS (continued)(VCC = +5V, circuit of Figure 1, TA = -40°C to +85°C, unless otherwise noted. Typical specifications are at TA = +25°C.) (Note 1)


OSCILLATOR AND PLL

CLKIN = SGND 238 250 262Switching Frequency fSW

CLKIN = VCC 475 500 525kHz

PLL Lock Range fPLL 125 600 kHz

PLL Locking Time tPLL 200 µs

PHASE = VCC 115 120 125

PHASE = unconnected 85 90 95CLKOUT Phase Shift(at fSW = 125kHz)

φCLKOUT

PHASE = SGND 55 60 65

degrees

CLKIN Input Pulldown Current ICLKIN 3 5 7 µA

CLKIN High Threshold VCLKINH 2.4 V

CLKIN Low Threshold VCLKINL 0.8 V

CLKIN High Pulse Width tCLKIN 200 ns

PHASE High Threshold VPHASEH 4 V

PHASE Low Threshold VPHASEL 1 V

PHASE Input Bias Current IPHASEBIA -50 +50 µA

CLKOUT Output Low Level VCLKOUTL ISINK = 2mA (Note 2) 100 mV

CLKOUT Output High Level VCLKOUTH ISOURCE = 2mA (Note 2) 4.5 V

CURRENT LIMIT

Average Current-Limit Threshold VCL CSP_ to CSN_ 45 48 51 mV

Cycle-by-Cycle Current Limit VCLPK CSP_ to CSN_ (Note 3) 90 112 130 mV

Cycle-by-Cycle OverloadResponse Time

tR VCSP_ to VCSN_ = +150mV 260 ns

CURRENT-SENSE AMPLIFIER

CSP_ to CSN_ Input Resistance RCS_ 4 kΩCommon-Mode Range VCMR(CS) -0.3 +3.6 V

Input Offset Voltage VOS(CS) -1 +1 mV

Amplifier Gain AV(CS) 18 V/V

3dB Bandwidth f3dB 4 MHz

CURRENT-ERROR AMPLIFIER (TRANSCONDUCTANCE AMPLIFIER)

Transconductance gmca 550 µS

Open-Loop Gain AVOL(CE) No load 50 dB

DIFFERENTIAL VOLTAGE AMPLIFIER (DIFF)

Common-Mode Voltage Range VCMR(DIFF) -0.3 +1.0 V

DIFF Output Voltage VCM VSENSE+ = VSENSE- = 0 0.6 V

Input Offset Voltage VOS(DIFF) -1 +1 mV

MAX5038/MAX5041 (+1.2V, +1.5V, +1.8Voutput versions)

0.997 1 1.003Amplifier Gain AV(DIFF)

MAX5038 (+2.5V and +3.3V output versions) 0.495 0.5 0.505

V/V

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ELECTRICAL CHARACTERISTICS (continued)(VCC = +5V, circuit of Figure 1, TA = -40°C to +85°C, unless otherwise noted. Typical specifications are at TA = +25°C.) (Note 1)


3dB Bandwidth f3dB CDIFF = 20pF 3 MHz

Minimum Output Current Drive IOUT(DIFF) 1.0 mA

SENSE+ to SENSE- InputResistance

RVS_ 50 100 kΩ

VOLTAGE-ERROR AMPLIFIER (EAOUT)

Open-Loop Gain AVOL(EA) 70 dB

Unity-Gain Bandwidth fUGEA 3 MHz

EAN Input Bias Current IB(EA) VEAN = +2.0V -100 +100 nA

Error-Amplifier Output ClampingVoltage

VCLAMP(EA) With respect to VCM 810 918 mV

THERMAL SHUTDOWN

Thermal Shutdown TSHDN 150 °C

Thermal-Shutdown Hysteresis 8 °C

EN INPUT

EN Input Low Voltage VENL 1 V

EN Input High Voltage VENH 3 V

EN Pullup Current IEN 4.5 5 5.5 µA

Note 1: Specifications from -40°C to 0°C are guaranteed by characterization but not production tested.Note 2: Guaranteed by design. Not production tested.Note 3: See Peak-Current Comparator section.

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EFFICIENCY vs. OUTPUT CURRENT ANDINTERNAL OSCILLATOR FREQUENCY

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IOUT (A)

η (%

)

4844403632282420161284

50

60

70

80

90

100

400 52

f = 500kHz

f = 250kHz

VIN = +5VVOUT = +1.8V

EFFICIENCY vs. OUTPUT CURRENTAND INPUT VOLTAGE

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2

IOUT (A)

η (%

)

4844403632282420161284

50

40

30

20

10

60

70

80

90

100

00 52

VIN = +12V

VIN = +5V

VOUT = +1.8VfSW = 250kHz

EFFICIENCY vs. OUTPUT CURRENT

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IOUT (A)

η (%

)

4844403632282420161284

50

40

30

20

10

60

70

80

90

100

00 52

VIN = +24VVOUT = +1.8VfSW = 125kHz

EFFICIENCY vs. OUTPUT CURRENTAND OUTPUT VOLTAGE

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IOUT (A)

η (%

)

4844403632282420161284

50

40

30

20

10

60

70

80

90

100

00 52

VOUT = +1.1V

VOUT = +1.5VVOUT = +1.8V

VIN = +12VfSW = 250kHz

EFFICIENCY vs. OUTPUT CURRENTAND OUTPUT VOLTAGE

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5

IOUT (A)

η (%

)

4844403632282420161284

50

40

30

20

10

60

70

80

90

100

00 52

VOUT = +1.1V

VOUT = +1.5V VOUT = +1.8V


SUPPLY CURRENTvs. FREQUENCY AND INPUT VOLTAGE

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FREQUENCY (kHz)

I CC

(mA)

550500400 450200 250 300 350150

6.57.07.58.08.59.09.5

10.010.511.011.512.0

6.0100 600

VIN = +24V

VIN = +12V

VIN = +5V EXTERNALCLOCKNO DRIVER LOAD

SUPPLY CURRENTvs. TEMPERATURE AND FREQUENCY

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TEMPERATURE (°C)

I CC

(mA)

603510-15

10

20

30

40

50

60

70

80

90

100

0-40 85

250kHz

125kHz

VIN = +12VCDL_ = 22nFCDH_ = 8.2nF

SUPPLY CURRENTvs. TEMPERATURE AND FREQUENCY

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TEMPERATURE (°C)

I CC

(mA)

603510-15

50

75

100

125

150

175

25-40 85

600kHz

500kHz

VIN = +5VCDL_ = 22nFCDH_ = 8.2nF

SUPPLY CURRENTvs. LOAD CAPACITANCE PER DRIVER

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CDRIVER (nF)

I CC

(mA)

13117 953

10

20

30

40

50

60

70

80

90

100

01 15


Typical Operating Characteristics(Circuit of Figure 1. TA = +25°C, unless otherwise noted.)

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Typical Operating Characteristics (continued)(Circuit of Figure 1, TA = +25°C, unless otherwise noted.)

CURRENT-SENSE THRESHOLDvs. OUTPUT VOLTAGE

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0

VOUT (V)

(VCS

P_ -

V CSN

_) (m

V)

1.71.61.4 1.51.2 1.31.1

46

47

48

49

50

51

52

53

54

55

451.0 1.8

PHASE 2

PHASE 1

OUTPUT VOLTAGE vs. OUTPUT CURRENTAND ERROR AMP GAIN (RF / RIN)

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ILOAD (A)

V OUT

(V)

5045403530252015105

1.65

1.70

1.75

1.80

1.85

1.600 55

VIN = +12VVOUT = +1.8V

RF / RIN = 15

RF / RIN = 12.5

RF / RIN = 10

RF / RIN = 7.5

DIFFERENTIAL AMPLIFIER BANDWIDTHMAX5038/41 toc12

FREQUENCY (MHz)

GAIN

(V/V

)

PHAS

E (d

eg)

10.1

0.5

1.0

1.5

2.0

2.5

3.0

3.5

00.01 10

-225

-270

-180

-135

-90

-45

0

45

90

PHASE

GAIN

DIFF OUTPUT ERRORvs. SENSE+ TO SENSE- VOLTAGE

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∆VSENSE (V)

ERRO

R (%

)

1.91.81.1 1.2 1.3 1.5 1.61.4 1.7

0.025

0.050

0.075

0.100

0.125

0.150

0.175

0.200

01.0 2.0

VIN = +12VNO DRIVER

VCC LOAD REGULATIONvs. INPUT VOLTAGE

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ICC (mA)

V CC

(V)

13512015 30 45 75 9060 105

4.85

4.90

4.95

5.00

5.05

5.10

5.15

5.20

4.800 150

VIN = +24V

VIN = +12V

VIN = +8V

DC LOAD

VCC LINE REGULATION

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5

VIN (V)

V CC

(V)

262420 2212 14 16 1810

4.80

4.85

4.90

4.95

5.00

5.05

5.10

5.15

5.20

5.25

4.758 28

ICC = 0

ICC = 40mA

VCC LINE REGULATION

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VIN (V)

V CC

(V)

13.012.09.0 10.0 11.0

4.80

4.85

4.90

4.95

5.00

5.05

5.10

5.15

5.20

5.25

4.758.0

ICC = 80mA

DRIVER RISE TIMEvs. DRIVER LOAD CAPACITANCE

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CDRIVER (nF)

t R (n

s)

312616 21116

102030405060708090

100110120

01 36

DL_

DH_


DRIVER FALL TIMEvs. DRIVER LOAD CAPACITANCE

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8

CDRIVER (nF)

t F (n

s)

312616 21116

102030405060708090

100110120

01 36

DL_

DH_


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100ns/div

HIGH-SIDE DRIVER (DH_)SINK AND SOURCE CURRENT

DH_1.6A/div

MAX5038/41 toc19

VIN = +12VCDH_ = 22nF

100ns/div

LOW-SIDE DRIVER (DL_)SINK AND SOURCE CURRENT

DL_1.6A/div

MAX5038/41 toc20

VIN = +12VCDL_ = 22nF

100µs/div

PLL LOCKING TIME250kHz TO 350kHz AND

350kHz TO 250kHz

CLKOUT5V/div

MAX5038/41 toc21

PLLCMP200mV/div

VIN = +12VNO LOAD

350kHz

250kHz

0

100µs/div


500kHz TO 250kHz

CLKOUT5V/div

MAX5038/41 toc22

PLLCMP200mV/div

0

VIN = +12VNO LOAD

500kHz

250kHz

100µs/div


150kHz TO 250kHz

CLKOUT5V/div

MAX5038/41 toc23

PLLCMP200mV/div

0

VIN = +12VNO LOAD

250kHz

150kHz

40ns/div

HIGH-SIDE DRIVER (DH_)RISE TIME

MAX5038/41 toc24


DH_2V/div

40ns/div

HIGH-SIDE DRIVER (DH_)FALL TIME

MAX5038/41 toc25

DH_2V/div


40ns/div

LOW-SIDE DRIVER (DL_)RISE TIME

MAX5038/41 toc26

DL_2V/div


40ns/div

LOW-SIDE DRIVER (DL_)FALL TIME

MAX5038/41 toc27

DL_2V/div



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500ns/div

OUTPUT RIPPLEMAX5038/41 toc28

VOUT(AC-COUPLED)10mV/div

VIN = +12VVOUT = +1.75VIOUT = 52A

2ms/div

INPUT STARTUP RESPONSEMAX5038/41 toc29

VIN 5V/div

VIN = +12VVOUT = +1.75V

IOUT = 52A

VPGOOD1V/div

VOUT1V/div

1ms/div

ENABLE STARTUP RESPONSEMAX5038/41 toc30

VEN 2V/div

VPGOOD1V/div

VOUT1V/div

VIN = +12VVOUT = +1.75V

IOUT = 52A

40µs/div

LOAD-TRANSIENT RESPONSEMAX5038/41 toc31

VIN = +12VVOUT = +1.75VISTEP = 8A TO 52AtRISE = 1µs

VOUT50mV/div

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Pin DescriptionPIN NAME FUNCTION

1, 13CSP2,CSP1

Current-Sense Differential Amplifier Positive Input. Senses the inductor current. The differential voltagebetween CSP_ and CSN_ is amplified internally by the current-sense amplifier gain of 18.

2, 14CSN2,CSN1

Current-Sense Differential Amplifier Negative Input. Senses the inductor current.

3 PHASEPhase-Shift Setting Input. Connect PHASE to VCC for 120°, leave PHASE unconnected for 90°, or connectPHASE to SGND for 60° of phase shift between the rising edges of CLKOUT and CLKIN/DH1.

4 PLLCMPExternal Loop-Compensation Input. Connect compensation network for the phase lock loop (see Phase-Locked Loop section).

5, 7CLP2,CLP1

Current-Error Amplifier Output. Compensate the current loop by connecting an RC network to ground.

6 SGND Signal Ground. Ground connection for the internal control circuitry.

8 SENSE+Differential Output Voltage-Sensing Positive Input. Used to sense a remote load. Connect SENSE+ toVOUT+ at the load. The MAX5038 regulates the difference between SENSE+ and SENSE- according to thefactory preset output voltage. The MAX5041 regulates the SENSE+ to SENSE- difference to +1.0V.

9 SENSE-Differential Output Voltage-Sensing Negative Input. Used to sense a remote load. Connect SENSE- toVOUT- or PGND at the load.

10 DIFF Differential Remote-Sense Amplifier Output. DIFF is the output of a precision unity-gain amplifier.

11 EANVoltage-Error Amplifier Inverting Input. Receives the output of the differential remote-sense amplifier.Referenced to SGND.

12 EAOUTVoltage-Error Amplifier Output. Connect to the external gain-setting feedback resistor. The external erroramplifier gain-setting resistors determine the amount of adaptive voltage positioning

15 EN Output Enable. A logic low shuts down the power drivers. EN has an internal 5µA pullup current.

16, 26BST1,BST2

Boost Flying-Capacitor Connection. Reservoir capacitor connection for the high-side FET driver supply.Connect 0.47µF ceramic capacitors between BST_ and LX_.

17, 25DH1,DH2

High-Side Gate Driver Output. Drives the gate of the high-side MOSFET.

18, 24 LX1, LX2Inductor Connection. Source connection for the high-side MOSFETs. Also serves as the return terminal forthe high-side driver.

19, 23 DL1, DL2 Low-Side Gate Driver Output. Synchronous MOSFET gate drivers for the two phases.

20 VCCInternal +5V Regulator Output. VCC is derived internally from the IN voltage. Bypass to SGND with 4.7µFand 0.1µF ceramic capacitors.

21 INSupply Voltage Connection. Connect IN to VCC for a +5V system. Connect the VRM input to IN through anRC lowpass filter, a 2.2Ω resistor and a 0.1µF ceramic capacitor.

22 PGNDPower Ground. Connect PGND, low-side synchronous MOSFET’s source, and VCC bypass capacitorreturns together.

27 CLKOUTOscillator Output. CLKOUT is phase-shifted from CLKIN by the amount specified by PHASE. Use CLKOUTto parallel additional MAX5038/MAX5041s.

28 CLKIN

CMOS Logic Clock Input. Drive the internal oscillator with a frequency range between 125kHz and 600kHz.The PWM frequency defaults to the internal oscillator if CLKIN is connected to VCC or SGND. ConnectCLKIN to SGND to set the internal oscillator to 250kHz or connect to VCC to set the internal oscillator to500kHz. CLKIN has an internal 5µA pulldown current.

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MAX5038MAX5041

IN

EN

PHASE 1

CSP1DRV_VCC

RAMP1

GMIN

CLK

CLP1

CSN1

SHDNBST1

DL1

LX1

DH1

VCC

TO INTERNAL CIRCUITSCSP1

CSN1

CLP1

PHASE 2

CSP2

DRV_VCC

GMIN

CLK

CLP2

CSN2

SHDN

BST2

DL2

LX2

DH2

CSP2

CSN2

CLP2

PHASE-LOCKED

LOOP

RAMPGENERATOR

RAMP2

CLKIN

CLKOUT

PLLCMP

DIFFAMP

ERRORAMP

SENSE-

SENSE+

DIFF

EAN

EAOUT

PGND

PGND

PGND

SGND

VREF = VOUT for VOUT ≤ 1.8V (MAX5038)VREF = VOUT/2 for VOUT > 1.8V (MAX5038)VREF = +1.0V (MAX5041)

+5VLDO

REGULATOR

UVLOPOR

TEMP SENSOR

0.6V

Functional Diagram

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Detailed DescriptionThe MAX5038/MAX5041 (Figures 1 and 2) average cur-rent-mode PWM controllers drive two out-of-phasebuck converter channels. Average current-mode con-trol improves current sharing between the channelswhile minimizing component derating and size. Parallelmultiple MAX5038/MAX5041 regulators to increase the

output current capacity. For maximum ripple rejectionat the input, set the phase shift between phases to 90°for two paralleled converters, or 60° for three paralleledconverters. Paralleling the MAX5038/MAX5041simproves design flexibility in applications requiringupgrades (higher load).

CLKIN

PLLCMP

PGND

PHASE

DL2

LX2

DH2

DL1

LX1

DH1

VCC

EAOUT

EAN

DIFF

EN

CSP2

CSN2

CSP1

CSN1

MAX5038

3

R1

C39

VIN = +12V

C1, C2

21

15

IN

C25

C26

R4

R7

R8

R6C29

C30

R5C27

C28

SGND

CLP2

CLP1 Q2

Q1

D2

Q2D1

VIN

Q1

VIN

D4

D3

C32

C12

C31

C3–C7

L2 R3

L1 R2

C13

C8–C11 C14,C15

C16–C24,C33

LOAD

+1.8V AT 60AVOUT

SENSE -

SENSE +

17

18

19

16

20

25

24

23

26

9

8

14

13

1

2

BST1

VCC

28

4

10

11

12

7

5

6

22

BST2

NOTE: SEE TABLE 1 FOR COMPONENT VALUES.

VCCRX

Figure 1. MAX5038 Typical Application Circuit, VIN = +12V

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Dual-phase converters with an out-of-phase lockingarrangement reduce the input and output capacitor ripple current, effectively multiplying the switching fre-quency by the number of phases. Each phase of theMAX5038/MAX5041 consists of an inner average cur-rent loop controlled by a common outer-loop voltage-

error amplifier (VEA) that corrects the output voltageerrors. The MAX5038/MAX5041 utilize a single control-ling VEA and an average current mode to force thephase currents to be equal.

CLKIN

PLLCMP

PGND

PHASE

DL2

LX2

DH2

DL1

LX1

DH1

VCC

EAOUT

EAN

DIFF

EN

CSP2

CSN2

CSP1

CSN1

MAX5041

3

R1

C39

VIN = +12V

C1,C2

21

15

IN

C25

C26

R4

R7

R8

R6C29

C30

R5C27

C28

SGND

CLP2

CLP1 Q2

Q1

D2

Q2D1

VIN

Q1

VIN

D4

D3

C32

C12

C31

C3–C7

L2 R3

L1 R2

C13

C8–C11 C14,C15

C16–C24,C33

LOAD

+1.8V AT 60AVOUT

SENSE -

SENSE +

17

18

19

16

20

25

24

23

26

9

8

14

13

1

2

BST1

VCC

28

4

10

11

12

7

5

6

22

BST2

NOTE: SEE TABLE 1 FOR COMPONENT VALUES.

RH

RL

VCC

RX

Figure 2. MAX5041 Typical Application Circuit, VIN = +12V

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VIN and VCCThe MAX5038/MAX5041 accept a wide input voltagerange of +4.75V to +5.5V or +8V to +28V. All internalcontrol circuitry operates from an internally regulatednominal voltage of +5V (VCC). For input voltages of +8Vor greater, the internal VCC regulator steps the voltagedown to +5V. The VCC output voltage regulates to +5Vwhile sourcing up to 80mA. Bypass VCC to SGND with4.7µF and 0.1µF low-ESR ceramic capacitors for high-frequency noise rejection and stable operation (Figures 1and 2).

Calculate power dissipation in the MAX5038/MAX5041as a product of the input voltage and the total VCC reg-ulator output current (ICC). ICC includes quiescent cur-rent (IQ) and gate drive current (IDD):

PD = VIN x ICC

ICC = IQ + fSW x (QG1 + QG2 + QG3 + QG4)

where, QG1, QG2, QG3, and QG4 are the total gatecharge of the low-side and high-side externalMOSFETs, IQ is 4mA (typ), and fSW is the switching fre-quency of each individual phase.

For applications utilizing a +5V input voltage, disablethe VCC regulator by connecting IN and VCC together.

Undervoltage Lockout (UVLO)/Power-On Reset (POR)/Soft-Start

The MAX5038/MAX5041 include an undervoltage lock-out with hysteresis and a power-on reset circuit for con-verter turn-on and monotonic rise of the output voltage.The UVLO threshold is internally set between +4.0Vand +4.5V with a 200mV hysteresis. Hysteresis atUVLO eliminates “chattering” during startup.

Most of the internal circuitry, including the oscillator,turns on when the input voltage reaches +4V. TheMAX5038/MAX5041 draw up to 4mA of current beforethe input voltage reaches the UVLO threshold.

The compensation network at the current error ampli-fiers (CLP1 and CLP2) provides an inherent soft-start ofthe output voltage. It includes a parallel combination ofcapacitors (C28, C30) and resistors (R5, R6) in serieswith other capacitors (C27, C29) (see Figures 1 and 2).The voltage at CLP_ limits the maximum current avail-able to charge output capacitors. The capacitor onCLP_ in conjunction with the finite output-drive currentof the current-error amplifier yields a finite rise time forthe output current and thus the output voltage.

Internal OscillatorThe internal oscillator generates the 180° out-of-phaseclock signals required by the pulse-width modulation(PWM) circuits. The oscillator also generates the 2VP-Pvoltage ramp signals necessary for the PWM compara-tors. Connect CLKIN to SGND to set the internal oscillatorfrequency to 250kHz or connect CLKIN to VCC to set theinternal oscillator to 500kHz.

CLKIN is a CMOS logic clock input for the phase-locked loop (PLL). When driven externally, the internaloscillator locks to the signal at CLKIN. A rising edge atCLKIN starts the ON cycle of the PWM. Ensure that theexternal clock pulse width is at least 200ns. CLKOUTprovides a phase-shifted output with respect to the ris-ing edge of the signal at CLKIN. PHASE sets theamount of phase shift at CLKOUT. Connect PHASE toVCC for 120° of phase shift, leave PHASE unconnectedfor 90° of phase shift, or connect PHASE to SGND for60° of phase shift with respect to CLKIN.

The MAX5038/MAX5041 require compensation on PLLCMP even when operating from the internal oscillator.The device requires an active PLL in order to generatethe proper clock signal required for PWM operation.

Control LoopThe MAX5038/MAX5041 use an average current-modecontrol scheme to regulate the output voltage (Figures3a and 3b). The main control loop consists of an innercurrent loop and an outer voltage loop. The inner loopcontrols the output currents (IPHASE1 and IPHASE2)while the outer loop controls the output voltage. Theinner current loop absorbs the inductor pole reducingthe order of the outer voltage loop to that of a single-pole system.

The current loop consists of a current-sense resistor(RS), a current-sense amplifier (CA_), a current-erroramplifier (CEA_), an oscillator providing the carrierramp, and a PWM comparator (CPWM_). The precisionCA_ amplifies the sense voltage across RS by a factorof 18. The inverting input to the CEA_ senses the CA_output. The CEA_ output is the difference between thevoltage-error amplifier output (EAOUT) and the gained-up voltage from the CA_. The RC compensation net-work connected to CLP1 and CLP2 provides externalfrequency compensation for the respective CEA_. Thestart of every clock cycle enables the high-side driversand initiates a PWM ON cycle. Comparator CPWM_compares the output voltage from the CEA_ with a 0 to+2V ramp from the oscillator. The PWM ON cycle termi-nates when the ramp voltage exceeds the error voltage.

(1)(2)

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DRIVE 2

DRIVE 1CPWM1

CPWM2

CEA1

CEA2

VEA

DIFFAMP

CA1

CA2

CLP2

CSP2

CSN2

CLP1

CSN1

CSP1

SENSE+

SENSE-

VIN

VIN

LOADCOUT

VOUT

RIN*

RF*

RS

RS

IPHASE1

IPHASE2

RCF

CCFF

CCF

RCF

CCCF

CCF

MAX5041

VREF =+1.0V

*RF AND RIN AREEXTERNAL TO MAX5041(RF = R8, RIN = R7, FIGURE 2)

Figure 3b. MAX5041 Control Loop

DRIVE 2

DRIVE 1CPWM1

CPWM2

CEA1

CEA2

VEA

DIFFAMP

CA1

CA2

VREF

CLP2

CSP2

CSN 2

C LP1

C SN 1

C SP1

SENSE+

SENSE-

VIN

VIN

LOADCOUT

VOUT

RIN*

RF*

RS

RS

IPHASE1

IPHASE2

RCF

CCFF

CCF

RCF

CCCF

CCF

*RF AND RIN ARE EXTERNAL TO MAX5038(RF = R8, RIN = R7, FIGURE 1)

MAX5038

Figure 3a. MAX5038 Control Loop

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The outer voltage control loop consists of the differen-tial amplifier (DIFF AMP), reference voltage, and VEA.The unity-gain differential amplifier provides true differ-ential remote sensing of the output voltage. The differ-ential amplifier output connects to the inverting input(EAN) of the VEA. The noninverting input of the VEA isinternally connected to an internal precision referencevoltage. The MAX5041 reference voltage is set to +1.0Vand the MAX5038 reference is set to the preset outputvoltage. The VEA controls the two inner current loops(Figures 3a and 3b). Use a resistive feedback networkto set the VEA gain as required by the adaptive volt-age-positioning circuit (see the Adaptive VoltagePositioning section).

Current-Sense AmplifierThe differential current-sense amplifier (CA_) provides aDC gain of 18. The maximum input offset voltage of thecurrent-sense amplifier is 1mV and the common-modevoltage range is -0.3V to +3.6V. The current-sense ampli-fier senses the voltage across a current-sense resistor.

Peak-Current ComparatorThe peak-current comparator provides a path for fastcycle-by-cycle current limit during extreme fault condi-tions such as an output inductor malfunction (Figure 4).Note that the average current-limit threshold of 48mVstill limits the output current during short-circuit condi-tions. To prevent inductor saturation, select an outputinductor with a saturation current specification greater

than the average current limit (48mV). Proper inductorselection ensures that only extreme conditions trip thepeak-current comparator, such as a cracked outputinductor. The 112mV voltage threshold for triggeringthe peak-current limit is twice the full-scale averagecurrent-limit voltage threshold. The peak-current com-parator has a delay of only 260ns.

Current-Error AmplifierEach phase of the MAX5038/MAX5041 has a dedicatedtransconductance current-error amplifier (CEA_) with atypical gm of 550µS and 320µA output sink and sourcecurrent capability. The current-error amplifier outputs,CLP1 and CLP2, serve as the inverting input to thePWM comparator. CLP1 and CLP2 are externallyaccessible to provide frequency compensation for theinner current loops (Figures 3a and 3b). CompensateCEA_ such that the inductor current down slope, whichbecomes the up slope to the inverting input of the PWMcomparator, is less than the slope of the internally gen-erated voltage ramp (see the Compensation section).

PWM Comparator and R-S Flip-FlopThe PWM comparator (CPWM) sets the duty cycle foreach cycle by comparing the output of the current-erroramplifier to a 2VP-P ramp. At the start of each clockcycle, an R-S flip-flop resets and the high-side driver(DH_) turns on. The comparator sets the flip-flop assoon as the ramp voltage exceeds the CLP_ voltage,thus terminating the ON cycle (Figure 4).

2 x fs (V/s)

RAMP

CLK

CSP_

CSN_

GMIN

SHDN

CLP_

DRV_VCC

BST_

DH_

LX_

DL_

PGND

AV = 18

PWMCOMPARATOR

PEAK-CURRENTCOMPARATOR

112mV

S

R

Q

Q

Gm =500µS

Figure 4. Phase Circuit (Phase 1/Phase 2)

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16 ______________________________________________________________________________________

Differential AmplifierThe differential amplifier (DIFF AMP) facilitates outputvoltage remote sensing at the load (Figures 3a and 3b).It provides true differential output voltage sensing whilerejecting the common-mode voltage errors due to high-current ground paths. Sensing the output voltagedirectly at the load provides accurate load voltagesensing in high-current environments. The VEA pro-vides the difference between the differential amplifieroutput (DIFF) and the desired output voltage. The dif-ferential amplifier has a bandwidth of 3MHz. The differ-ence between SENSE+ and SENSE- regulates to thepreset output voltage for the MAX5038 and regulates to+1V for the MAX5041.

Voltage-Error Amplifier The VEA sets the gain of the voltage control loop anddetermines the error between the differential amplifieroutput and the internal reference voltage (VREF).

VREF equals VOUT(NOM) for the +1.8V or lower voltageversions of the MAX5038 and VREF equals VOUT(NOM)/2for the +2.5V and +3.3V versions. For MAX5041, VREFequals +1V.

An offset is added to the output voltage of theMAX5038/MAX5041 with a finite gain (RF/RIN) of theVEA such that the no-load output voltage is higher thanthe nominal value. Choose RF and RIN from theAdaptive Voltage Positioning section and use the follow-ing equations to calculate the no-load output voltage.

MAX5038:

MAX5041:

where RH and RL are the feedback resistor network(Figure 2).

Some applications require VOUT equal to VOUT(NOM) atno load. To ensure that the output voltage does notexceed the nominal output voltage (VOUT(NOM)), add aresistor RX from VCC to EAN.

Use the following equations to calculate the value of RX.

For MAX5038 versions of VOUT(NOM) ≤ +1.8V:

For MAX5038 versions of VOUT(NOM) > +1.8V:

For MAX5041:

The VEA output clamps to +0.9V (plus the common-mode voltage of +0.6V), thus limiting the average maxi-mum current from individual phases. The maximumaverage current-limit threshold for each phase is equalto the maximum clamp voltage of the VEA divided bythe gain (18) of the current-sense amplifier. This allowsfor accurate settings for the average maximum currentfor each phase. Set the VEA gain using RF and RIN forthe amount of output voltage positioning required asdiscussed in the Adaptive Voltage Positioning section(Figures 3a and 3b).

Adaptive Voltage PositioningPowering new-generation processors requires newtechniques to reduce cost, size, and power dissipation.Voltage positioning reduces the total number of outputcapacitors to meet a given transient response require-ment. Setting the no-load output voltage slightly higherthan the output voltage during nominally loaded condi-tions allows a larger downward voltage excursion whenthe output current suddenly increases. Regulating at alower output voltage under a heavy load allows a largerupward-voltage excursion when the output current sud-denly decreases. A larger allowed, voltage-step excur-sion reduces the required number of output capacitorsor allows for the use of higher ESR capacitors.

Voltage positioning and the ability to operate with multiplereference voltages may require the output to regulateaway from a center value. Define the center value as thevoltage where the output drops (∆VOUT/2) at one half themaximum output current (Figure 5).

R VR

VX CC

F

REF= − ×[ . ]1 6

R V VR

VX CC NOM

F

NOM= − + ×[ ( . )]2 1 2

R V VR

VX CC NOM

F

NOM= − + ×[ ( . )]0 6

VR

R

R R

RVOUT NL

IN

F

H L

LREF( ) = +

⎛

⎝⎜

⎞

⎠⎟ ×

+⎛

⎝⎜

⎞

⎠⎟ ×1

VR

RVOUT NL

IN

FOUT NOM( ) ( )= +

⎛

⎝⎜

⎞

⎠⎟ ×1 (3)

(4)

(5)

(6)

(7)

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Set the voltage-positioning window (∆VOUT) using theresistive feedback of the VEA. Use the following equa-tions to calculate the voltage-positioning window for theMAX5038:

Use the following equation to calculate the voltage-posi-tioning window for the MAX5041:

where RIN and RF are the input and feedback resistors ofthe VEA, GC is the current-loop gain and RS is the cur-rent-sense resistor or, if using lossless inductor currentsensing, the DC resistance of the inductor.

Phase-Locked Loop: Operation andCompensation

The PLL synchronizes the internal oscillator to theexternal frequency source when driving CLKIN.Connecting CLKIN to VCC or SGND forces the PWMfrequency to default to the internal oscillator frequencyof 500kHz or 250kHz, respectively. The PLL uses aconventional architecture consisting of a phase detec-tor and a charge pump capable of providing 20µA ofoutput current. Connect an external series combinationcapacitor (C25) and resistor (R4) and a parallel capaci-tor (C26) from PLLCMP to SGND to provide frequencycompensation for the PLL (Figure 1). The pole-zero paircompensation provides a zero at fZ = 1 / [R4 x (C25 +C26)] and a pole at fP = 1 / (R4 x C26). Use the follow-ing typical values for compensating the PLL: R4 = 7.5kΩ, C25 = 4.7nF, C26 = 470pF. If changing thePLL frequency, expect a finite locking time of approxi-mately 200µs.

The MAX5038/MAX5041 require compensation on PLLCMP even when operating from the internal oscilla-tor. The device requires an active PLL in order to gen-erate the proper internal PWM clocks.

MOSFET Gate Drivers (DH_, DL_)The high-side (DH_) and low-side (DL_) drivers drivethe gates of external N-channel MOSFETs (Figures 1and 2). The drivers’ high-peak sink and source currentcapability provides ample drive for the fast rise and falltimes of the switching MOSFETs. Faster rise and falltimes result in reduced cross-conduction losses. Formodern CPU voltage-regulating module applicationswhere the duty cycle is less than 50%, choose high-side MOSFETs (Q1 and Q3) with a moderate RDS(ON)and a very low gate charge. Choose low-sideMOSFETs (Q2 and Q4) with very low RDS(ON) andmoderate gate charge.

The driver block also includes a logic circuit that pro-vides an adaptive non-overlap time to prevent shoot-through currents during transit ion. The typicalnon-overlap time is 60ns between the high-side andlow-side MOSFETs.

BST_VDD powers the low- and high-side MOSFET drivers.Connect a 0.47µF low-ESR ceramic capacitor betweenBST_ and LX_. Bypass VCC to PGND with 4.7µF and0.1µF low-ESR ceramic capacitors. Reduce the PCboard area formed by these capacitors, the rectifierdiodes between VCC and the boost capacitor, theMAX5038/MAX5041, and the switching MOSFETs.

GR

CS

= 0 05.

∆VI R

G RR R

ROUTOUT IN

C F

H L

L

= ×× ×( ) × +

2

GR

CS

= 0 05.

∆VI R

G ROUT

OUT IN

C F=

×× ×2

LOAD (A)

VCNTR

NO LOAD 1/2 LOAD FULL LOAD

VOLTAGE-POSITIONING WINDOW

VCNTR + ∆VOUT/2

VCNTR - ∆VOUT/2

Figure 5. Defining the Voltage-Positioning Window

(8)

(9)

(10)

(11)

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Overload ConditionsAverage current-mode control has the ability to limit theaverage current sourced by the converter during a faultcondition. When a fault condition occurs, the VEA out-put clamps to +0.9V with respect to the common-modevoltage (VCM = +0.6V) and is compared with the outputof the current-sense amplifiers (CA1 and CA2) (seeFigures 3a and 3b). The current-sense amplifier’s gainof 18 limits the maximum current in the inductor orsense resistor to ILIMIT = 50mV/RS.

Parallel OperationFor applications requiring large output current, parallelup to three MAX5038/MAX5041s (six phases) to triplethe available output current. The paralleled convertersoperate at the same switching frequency but differentphases keep the capacitor ripple RMS currents to a mini-mum. Three parallel MAX5038/MAX5041 convertersdeliver up to 180A of output current. To set the phaseshift of the on-board PLL, leave PHASE unconnected for90° of phase shift (2 paralleled converters), or connectPHASE to SGND for 60° of phase shift (3 converters inparallel). Designate one converter as master and theremaining converters as slaves. Connect the master andslave controllers in a daisy-chain configuration as shownin Figure 6. Connect CLKOUT from the master controllerto CLKIN of the first slaved controller, and CLKOUT fromthe first slaved controller to CLKIN of the second slavedcontroller. Choose the appropriate phase shift for mini-mum ripple currents at the input and output capacitors.The master controller senses the output differential volt-age through SENSE+ and SENSE- and generates theDIFF voltage. Disable the voltage sensing of the slavedcontrollers by leaving DIFF unconnected (floating).Figure 7 shows a detailed typical parallel application cir-cuit using two MAX5038s. This circuit provides fourphases at an input voltage of +12V and an output volt-age range of +1V to +3.3V at 104A.

Applications InformationEach MAX5038/MAX5041 circuit drives two 180° out-of-phase channels. Parallel two or three MAX5038/MAX5041 circuits to achieve four- or six-phase opera-tion, respectively. Figure 1 shows the typical applicationcircuit for a two-phase operation. The design criteria fora two-phase converter includes frequency selection,inductor value, input/output capacitance, switchingMOSFETs, sense resistors, and the compensation net-work. Follow the same procedure for the four- and six-phase converter design, except for the input and outputcapacitance. The input and output capacitance require-ments vary depending on the operating duty cycle.

The examples discussed in this data sheet pertain to atypical application with the following specifications:

VIN = +12V

VOUT = +1.8V

IOUT(MAX) = 52A

fSW = 250kHz

Peak-to-Peak Inductor Current (∆IL) = 10A

Table 1 shows a list of recommended external compo-nents (Figure 1) and Table 2 provides component sup-plier information.

Number of PhasesSelecting the number of phases for a voltage regulatordepends mainly on the ratio of input-to-output voltage(operating duty cycle). Optimum output-ripple cancella-tion depends on the right combination of operating dutycycle and the number of phases. Use the followingequation as a starting point to choose the number ofphases:

NPH ≈ K/D

where K = 1, 2, or 3 and the duty cycle is D = VOUT/VIN.

Choose K to make NPH an integer number. For exam-ple, converting VIN = +12V to VOUT = +1.8V yields better ripple cancellation in the six-phase converterthan in the four-phase converter. Ensure that the outputload justifies the greater number of components formultiphase conversion. Generally limiting the maximumoutput current to 25A per phase yields the most cost-effective solution. The maximum ripple cancellationoccurs when NPH = K/D.

Single-phase conversion requires greater size and powerdissipation for external components such as the switch-ing MOSFETs and the inductor. Multiphase conversioneliminates the heatsink by distributing the power dissipa-tion in the external components. The multiple phasesoperating at given phase shifts effectively increase theswitching frequency seen by the input/output capacitors,thereby reducing the input/output capacitance require-ment for the same ripple performance. The lower induc-tance value improves the large-signal response of theconverter during a transient load at the output. Considerall these issues when determining the number of phasesnecessary for the voltage regulator application.

Inductor SelectionThe switching frequency per phase, peak-to-peak rip-ple current in each phase, and allowable ripple at theoutput determine the inductance value.

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CLKIN

CLKOUTSGNDPGND

IN

PHASE

DL2

LX2

DH2

DL1

LX1

DH1

VCC

VIN

EAOUT

EANDIFF

SENSE-

SENSE+

CSP2

CSN2

CSP1CSN1

VCC

VIN

VIN

CLKOUTSGNDPGND

IN

PHASE

DL2

LX2

DH2

DL1

LX1

DH1

EAOUT

EAN

CLKIN

CSP2

CSN2

CSP1CSN1

DIFF

VCC

VIN

VIN

CLKOUTSGNDPGND

IN

PHASE

DL2

LX2

DH2

DL1

LX1

DH1

EAOUT

EAN

CLKIN

CSP2

CSN2

CSP1CSN1

DIFF

VCC

VIN

VIN

TO OTHER MAX5038/MAX5041s

MAX5038/MAX5041

MAX5038/MAX5041

MAX5038/MAX5041

LOAD

*FOR MAX5041 ONLY.

*

*

Figure 6. Parallel Configuration of Multiple MAX5038/MAX5041s

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MAX5038(MASTER)

Q4

Q3

D2

Q2D1

VIN

4 x 22µFC8–C11

Q1

VIN

VIN = +12V

VCC

D4

D3

C400.1µF

C120.47µF

C384.7µF

C3–C75 x 22µF

L20.6µH

R21.35mΩ

L10.6µH

R11.35mΩ

DH1

LX1

DL1

BST1

VCC

DH2

LX2

DL2

BST2CSP2CSN2PGOODPHASECLKOUTSGNDPGNDCLP2CLP1

R9

PGOODVCC

R6

C35

C36

R5

C33

C34

C390.1µF

C1, C22 x 47µF

R32.2Ω

C31

C32

R4

CSP1CSN1SENSE+SENSE-INCLKINPLLCMP

R8

R7

EAOUT

EAN

DIFF

OVPIN

EN

C130.47µF

C14, C15,C41, C422 x 100µF

C16–C25,C43–C46

14 x 270µF

C26–C30,C37

6 x 10µF LOADVOUT = +1.1V TO+3.3V AT 104A

VCC

MAX5038(SLAVE)

Q8

Q7

D6

Q6D5

VINC52–C554 x 22µF

Q5

VIN

D8

D7

C640.1µF

C570.47µF

C654.7µF

C48–C515 x 22µF

L40.6µH

R111.35mΩ

L30.6µH

R101.35mΩ

DH1

LX1

DL1

BST1

VCC

DH2

LX2

DL2

BST2CSP2CSN2PHASESGNDPGNDCLP2CLP1

VCC

R15

C59

C58

R14

C60

C61

C470.1µF

R122.2Ω

C62

C63

R13

CSP1CSN1SENSE+SENSE-CLKININPLLCMPEN

R17

R16

EAOUT

EAN

DIFF

C560.47µF

R18

R19

VCC

RX

VCC

RX

Figure 7. Four-Phase Parallel Application Circuit (VIN = +12V, VOUT = +1.1V to +3.3V at 104A)

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Table 1. Component List

DESIGNATION QTY DESCRIPTION

C1, C2 2 47µF,16V X5R input-filter capacitors TDK C5750X5R1C476M

C3–C11 9 22µF, 16V input-filter capacitors TDK C4532X5R1C226M

C12, C13 2 0.47µF, 16V capacitors TDK C1608X5R1A474K

C14, C15 2 100µF, 6.3V, output-filter capacitors Murata GRM44-1X5R107K6.3

C16–C24, C33 10 270µF, 2V output-filter capacitors Panasonic EEFUE0D271R

C25 1 4700pF, 16V X7R capacitor Vishay-Siliconix VJ0603Y471JXJ

C26, C28, C30 3 470pF 16V capacitors Murata GRM1885C1H471JAB01

C27, C29 2 0.01µF 50V X7R capacitors Murata GRM188R71H103KA01

C31 1 4.7µF 16V X5R capacitor Murata GRM40-034X5R475k6.3

C32 3 0.1µF 16V X7R capacitors Murata GRM188R71C104KA01

D1, D2 2 Schottky diodes ON-Semiconductor MBRS340T3

D3, D4 2 Schottky diodes ON-Semiconductor MBR0520LT1

L1, L2 2 0.6µH, 27A inductors Panasonic ETQP1H0R6BFX

Q1, Q3 2 Upper-power MOSFETs Vishay-Siliconix Si7860DP

Q2, Q4 2 Lower-power MOSFETs Vishay-Siliconix Si7886DP

R1 1 2.2Ω ±1% resistor

R2, R3 4 Current-sense resistors, use two 2.7mΩ resistors in parallel, Panasonic ERJM1WSF2M7U

R4 1 7.5kΩ ±1% resistor

R5, R6 2 1kΩ ±1% resistors

R7 1 4.99kΩ ±1% resistor

R8, R9 2 37.4kΩ ±1% resistors

Table 2. Component SuppliersSUPPLIER PHONE FAX WEBSITE

Murata 770-436-1300 770-436-3030 www.murata.com

ON Semiconductor 602-244-6600 602-244-3345 www.on-semi.com

Panasonic 714-373-7939 714-373-7183 www.panasonic.com

TDK 847-803-6100 847-390-4405 www.tcs.tdk.com

Vishay-Siliconix 1-800-551-6933 619-474-8920 www.vishay.com

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22 ______________________________________________________________________________________

Selecting higher switching frequencies reduces theinductance requirement, but at the cost of lower efficien-cy. The charge/discharge cycle of the gate and draincapacitances in the switching MOSFETs create switchinglosses. The situation worsens at higher input voltages,since switching losses are proportional to the square ofinput voltage. Use 500kHz per phase for VIN = +5V and250kHz or less per phase for VIN > +12V.

Although lower switching frequencies per phase increasethe peak-to-peak inductor ripple current (∆IL), the ripplecancellation in the multiphase topology reduces the inputand output capacitor RMS ripple current.

Use the following equation to determine the minimuminductance value:

Choose ∆IL equal to about 40% of the output currentper phase. Since ∆IL affects the output-ripple voltage,the inductance value may need minor adjustment afterchoosing the output capacitors for full-rated efficiency.

Choose inductors from the standard high-current, surface-mount inductor series available from variousmanufacturers. Particular applications may require cus-tom-made inductors. Use high-frequency core materialfor custom inductors. High ∆IL causes large peak-to-peakflux excursion increasing the core losses at higher fre-quencies. The high-frequency operation coupled withhigh ∆IL, reduces the required minimum inductanceand even makes the use of planar inductors possible.The advantages of using planar magnetics include low-profile design, excellent current-sharing between phas-es due to the tight control of parasitics, and low cost.

For example, calculate the minimum inductance atVIN(MAX) = +13.2V, VOUT = +1.8V, ∆IL = 10A, and fSW =250kHz:

The average current-mode control feature of theMAX5038/MAX5041 limits the maximum peak inductorcurrent which prevents the inductor from saturating.Choose an inductor with a saturating current greaterthan the worst-case peak inductor current.

Use the following equation to determine the worst-caseinductor current for each phase:

where RSENSE is the sense resistor in each phase.

Switching MOSFETswhen choosing a MOSFET for voltage regulators,consider the total gate charge, RDS(ON), power dissipa-tion, and package thermal impedance. The product ofthe MOSFET gate charge and on-resistance is a figure ofmerit, with a lower number signifying better performance.Choose MOSFETs optimized for high-frequency switch-ing applications.

The average gate-drive current from the MAX5038/MAX5041 output is proportional to the total capacitanceit drives from DH1, DH2, DL1, and DL2. The power dis-sipated in the MAX5038/MAX5041 is proportional to theinput voltage and the average drive current. See the VINand VCC section to determine the maximum total gatecharge allowed from all the driver outputs together.

The gate charge and drain capacitance (CV2) loss, thecross-conduction loss in the upper MOSFET due to finiterise/fall time, and the I2R loss due to RMS current in theMOSFET RDS(ON) account for the total losses in the MOS-FET. Estimate the power loss (PDMOS_) in the high-sideand low-side MOSFETs using following equations:

where QG, RDS(ON), tR, and tF are the upper-switchingMOSFET’s total gate charge, on-resistance at +25°C,rise time, and fall time, respectively.

where D = VOUT/VIN, IDC = (IOUT - ∆IL)/2 and IPK = (IOUT + ∆IL)/2

I I I I ID

RMS HI DC PK DC PK− = + + ×( ) ×2 23

PD Q V f

V I t t fR I

MOS HI G DD SW

IN OUT R F SWDS ON RMS HI

−

−

= × ×( ) +

× × +( ) ×⎛

⎝⎜

⎞

⎠⎟ + ×

41 4 2. ( )

IR

IL PEAK

SENSE

L_

.= +0 0512

∆

Lk

HMIN =−( ) ×

× ×= µ

13 2 1 8 1 8

13 2 250 100 6

. . .

..

LV V V

V f IMININMAX OUT OUT

IN SW L=

−( ) ×× × ∆ (13)

(14)

(15)

(16)

(17)

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______________________________________________________________________________________ 23

For example, from the typical specifications in theApplications Information section with VOUT = +1.8V, thehigh-side and low-side MOSFET RMS currents are 9.9Aand 24.1A, respectively. Ensure that the thermal imped-ance of the MOSFET package keeps the junction tem-perature at least 25°C below the absolute maximumrating. Use the following equation to calculate maxi-mum junction temperature:

TJ = PDMOS x θJ-A + TA

Input CapacitorsThe discontinuous input-current waveform of the buckconverter causes large ripple currents in the inputcapacitor. The switching frequency, peak inductor cur-rent, and the allowable peak-to-peak voltage ripplereflected back to the source dictate the capacitancerequirement. Increasing the number of phases increas-es the effective switching frequency and lowers thepeak-to-average current ratio, yielding lower inputcapacitance requirement.

The input ripple is comprised of ∆VQ (caused by thecapacitor discharge) and ∆VESR (caused by the ESR ofthe capacitor). Use low-ESR ceramic capacitors withhigh ripple-current capability at the input. Assume thecontributions from the ESR and capacitor discharge areequal to 30% and 70%, respectively. Calculate theinput capacitance and ESR required for a specified rip-ple using the following equation:

where IOUT is the total output current of the multiphaseconverter and N is the number of phases.

For example, at VOUT = +1.8V, the ESR and inputcapacitance are calculated for the input peak-to-peakripple of 100mV or less yielding an ESR and capaci-tance value of 1mΩ and 200µF.

Output CapacitorsThe worst-case peak-to-peak and capacitor RMS ripplecurrent, the allowable peak-to-peak output ripple volt-age, and the maximum deviation of the output voltageduring step loads determine the capacitance and theESR requirements for the output capacitors.

In multiphase converter design, the ripple currents fromthe individual phases cancel each other and lower theripple current. The degree of ripple cancellationdepends on the operating duty cycle and the number ofphases. Choose the right equation from Table 3 to calcu-late the peak-to-peak output ripple for a given dutycycle of two-, four-, and six-phase converters. The max-imum ripple cancellation occurs when NPH = K / D.

C

IN

D D

V fIN

OUT

Q SW=

× −( )×

1

∆

ESRV

IN

IINESR

OUT L=

( )+⎛

⎝⎜⎞⎠⎟

∆∆2

I I I I ID

RMS LO DC PK DC PK− = + + ×( ) ×−( )2 2 1

3

PD Q V f

C V fR I

MOS LO G DD SW

OSS IN SWDS ON RMS LO

−

−

= × ×( ) +

× × ×⎛

⎝⎜⎜

⎞

⎠⎟⎟ + ×

23

1 42

2. ( )

Table 3. Peak-to-Peak Output RippleCurrent Calculations

NUMBER OFPHASES (N)

DUTYCYCLE (D)

EQUATION FOR ∆IP-P

2 < 50%

2 > 50%

4 0 to 25%

4 25% to 50%

4 > 50%

6 < 17%

∆IV D

L fO

SW= −

×( )1 2

∆IV V D

L fIN O

SW=

−( ) −( )×

2 1

∆IV D

L fO

SW= −

×( )1 4

∆IV D D

D L fO

SW= − −

× × ×( )( )1 2 4 1

2

∆IV D D

D L fO

SW= − −

× ×( )( )2 1 3 4

∆IV D

L fO

SW= −

×( )1 6

(18)

(19)

(20)

(21)

(22)

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24 ______________________________________________________________________________________

The allowable deviation of the output voltage during thefast transient load dictates the output capacitance andESR. The output capacitors supply the load step untilthe controller responds with a greater duty cycle. Theresponse time (tRESPONSE) depends on the closed-loopbandwidth of the converter. The resistive drop acrossthe capacitor ESR and capacitor discharge causes avoltage drop during a step load. Use a combination ofSP polymer and ceramic capacitors for better transientload and ripple/noise performance.

Keep the maximum output voltage deviation less thanor equal to the adaptive voltage-positioning window(∆VOUT). Assume 50% contribution each from the out-put capacitance discharge and the ESR drop. Use thefollowing equations to calculate the required ESR andcapacitance value:

where ISTEP is the load step and tRESPONSE is theresponse time of the controller. Controller responsetime depends on the control-loop bandwidth.

Current LimitThe average current-mode control technique of theMAX5038/MAX5041 accurately limits the maximum out-put current per phase. The MAX5038/MAX5041 sensethe voltage across the sense resistor and limit the peakinductor current (IL-PK) accordingly. The ON cycle ter-minates when the current-sense voltage reaches 45mV(min). Use the following equation to calculate maximumcurrent-sense resistor value:

where PDR is the power dissipation in sense resistors.Select 5% lower value of RSENSE to compensate for anyparasitics associated with the PC board. Also, select anon-inductive resistor with the appropriate wattage rating.

CompensationThe main control loop consists of an inner current loopand an outer voltage loop. The MAX5038/MAX5041 usean average current-mode control scheme to regulatethe output voltage (Figures 3a and 3b). IPHASE1 andIPHASE2 are the inner average current loops. The VEAoutput provides the controlling voltage for these currentsources. The inner current loop absorbs the inductorpole reducing the order of the outer voltage loop to thatof a single-pole system.

A resistive feedback around the VEA provides the bestpossible response, since there are no capacitors tocharge and discharge during large-signal excursions, RFand RIN determine the VEA gain. Use the following equa-tion to calculate the value for RF:

where GC is the current-loop gain and N is number ofphases.

When designing the current-control loop ensure that theinductor downslope (when it becomes an upslope at theCEA output) does not exceed the ramp slope. This is anecessary condition to avoid sub-harmonic oscillationssimilar to those in peak current-mode control with insuffi-cient slope compensation. Use the following equation tocalculate the resistor RCF:

For example, the maximum RCF is 12kΩ for RSENSE =1.35mΩ.

CCF provides a low-frequency pole while RCF provides amidband zero. Place a zero at fZ to obtain a phase bumpat the crossover frequency. Place a high-frequency pole(fP) at least a decade away from the crossover frequencyto achieve maximum phase margin.

Rf L

V RCFSW

OUT SENSE≤ × × ×

×2 102

GRC

S= 0 05.

RI R

N G VFOUT IN

C OUT=

×× × ∆

PDRR

SENSE= × −2 5 10 3.

RI

N

SENSEOUT

= 0 045.

CI t

VOUT

STEP RESPONSE

Q=

×∆

ESRV

IOUTESR

STEP= ∆ (23)

(24)

(25)

(26)

(27)

(28)

(29)

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______________________________________________________________________________________ 25

Use the following equations to calculate CCF and CCFF:

PC Board LayoutUse the following guidelines to layout the switchingvoltage regulator.

1) Place the VIN and VCC bypass capacitors close tothe MAX5038/MAX5041.

2) Minimize the high-current loops from the inputcapacitor, upper switching MOSFET, inductor, andoutput capacitor back to the input capacitor nega-tive terminal.

3) Keep short the current loop from the lower switch-ing MOSFET, inductor, and output capacitor andreturn to the source of the lower MOSFET.

4) Place the Schottky diodes close to the lowerMOSFETs and on the same side of the PC board.

5) Keep the SGND and PGND isolated and connectthem at one single point close to the negative termi-nal of the input filter capacitor.

6) Run the current-sense lines CS+ and CS- veryclose to each other to minimize the loop area.Similarly, run the remote voltage sense linesSENSE+ and SENSE- close to each other. Do notcross these critical signal lines through power cir-cuitry. Sense the current right at the pads of cur-rent-sense resistors.

7) Avoid long traces between the VCC bypass capaci-tors, driver output of the MAX5038/MAX5041, MOSFET gates and PGND pin. Minimize the loopformed by the VCC bypass capacitors, bootstrapdiode, bootstrap capacitor, MAX5038/MAX5041,and upper MOSFET gate.

8) Place the bank of output capacitors close to the load.

9) Distribute the power components evenly across theboard for proper heat dissipation.

10) Provide enough copper area at and around theswitching MOSFETs, inductor, and sense resistorsto aid in thermal dissipation.

11) Use at least 4oz copper to keep the trace induc-tance and resistance to a minimum. Thin copper PCboards can compromise efficiency since high cur-rents are involved in the application. Also, thickercopper conducts heat more effectively, therebyreducing thermal impedance.

Chip InformationTRANSISTOR COUNT: 5431

PROCESS: BiCMOS

Cf RCFFP CF

=× × ×

12 π

Cf RCFZ CF

=× × ×

12 π

28

27

26

25

24

23

22

21

20

19

18

17

16

15

1

2

3

4

5

6

7

8

9

10

11

12

13

14

CLKIN

CLKOUT

BST2

DH2

LX2

DL2

EN

PGND

IN

VCC

DL1

LX1

DH1

BST1

CSN1

CSP1

EAOUT

EAN

DIFF

SENSE-

SENSE+

CLP1

SGND

CLP2

PLLCMP

PHASE

CSN2

CSP2

SSOP

TOP VIEW

MAX5038AMAX5041A

Pin Configuration

(30)

(31)

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Maxim cannot assume responsibility for use of any circuitry other than circuitry entirely embodied in a Maxim product. No circuit patent licenses areimplied. Maxim reserves the right to change the circuitry and specifications without notice at any time.

26 ____________________Maxim Integrated Products, 120 San Gabriel Drive, Sunnyvale, CA 94086 408-737-7600

© 2004 Maxim Integrated Products Printed USA is a registered trademark of Maxim Integrated Products.

SS

OP

.EP

S

PACKAGE OUTLINE, SSOP, 5.3 MM

11

21-0056 CREV.DOCUMENT CONTROL NO.APPROVAL

PROPRIETARY INFORMATION

TITLE:

NOTES:1. D&E DO NOT INCLUDE MOLD FLASH.2. MOLD FLASH OR PROTRUSIONS NOT TO EXCEED .15 MM (.006").3. CONTROLLING DIMENSION: MILLIMETERS.4. MEETS JEDEC MO150.5. LEADS TO BE COPLANAR WITHIN 0.10 MM.

7.90H

L

0∞

0.301

0.025

8∞

0.311

0.037

0∞

7.65

0.63

8∞

0.95

MAX

5.38

MILLIMETERS

B

C

D

E

e

A1

DIM

A

SEE VARIATIONS

0.0256 BSC

0.010

0.004

0.205

0.002

0.015

0.008

0.212

0.008

INCHES

MIN MAX

0.078

0.65 BSC

0.25

0.09

5.20

0.05

0.38

0.20

0.21

MIN

1.73 1.99

MILLIMETERS

6.07

6.07

10.07

8.07

7.07

INCHES

D

D

D

D

D

0.239

0.239

0.397

0.317

0.278

MIN

0.249

0.249

0.407

0.328

0.289

MAX MIN

6.33

6.33

10.33

8.33

7.33

14L

16L

28L

24L

20L

MAX N

A

D

e A1 L

C

HE

N

12

B

0.068

Package Information(The package drawing(s) in this data sheet may not reflect the most current specifications. For the latest package outline information,go to www.maxim-ic.com/packages.

Documents

Dual-Phase, Parallelable, Average Current-Mode Controllers · PHASE = VCC 115 120 125 PHASE = unconnected 85 90 95 CLKOUT Phase Shift (at fSW = 125kHz) φCLKOUT PHASE = SGND 55 60