Directional Couplers from 30 to 140GHz in Silicon

Benjamin Laemmle 1, Klaus Schmalz 2, Christoph Scheytt 2, Alexander Koelpin 1, and Robert Weigel 1

1 Institute for Electronics Engineering,

University of Erlangen-Nuremberg, Cauerstr. 9, 91058 Erlangen, Germany2 IHP Microelectronics GmbH,

Im Technologiepark 25, 15236 Frankfurt (Oder), Germany

AbstractIn this paper directional couplers with reduced size,by lumped elements, inverted microstrip, and broadside coupledlines at 61, 110, and 122 GHz center frequency and up to 156-GHzbandwidth have been designed. The couplers show an isolation upto 40 dB. Different SiGe BiCMOS technologies with 250-nm and130-nm feature width and 5 to 7 metal layers have been used. Themeasurement results have been compared to simulation resultsand good agreement has been observed.

Index Termsmillimeter wave directional couplers, passivecircuits

I. INTRODUCTION

The increase of operating frequencies in silicon integrated

circuits enables the integration of passive microwave elements

with reasonable size on chip [1]. The wavelength of millimeter-

wave signals is smaller than the die size enabling the use of dis-

tributed elements in complex integrated circuits and systems.

Transmission lines, Wilkinson power dividers, or matching

networks are already standard elements in millimeter-wave sil-

icon designs. Even complex structures like bandpass filters [2],

ratrace couplers, or Lange Couplers have been integrated and

presented. While for III-V based millimeter-wave integrated

circuits (MMIC) an isolating microwave substrate is available,

this is not the case for silicon. Also silicon is often labeled as

substrate, although this refers to a semiconductor substrate for

fabrication of transistors. However, several metal layers exist

where a ground-plane and conductors for microwave structures

can be formed. With conductors on different metal layers even

3-D structures can be fabricated unlike to III-V based MMICs.

Integrated directional couplers have several applications

for integrated receivers, transmitters, or other circuits. The

couplers can be designed as quadrature hybrid couplers with

equal signal split and 90 phase difference. The application is

quadrature signal generation in the LO [3] or signal path [4]

of quadrature receivers, active IQ modulators [5], or reflection

type modulators [6]. The coupler has to feature low and equal

insertion loss and stable phase difference for the coupled and

direct port.

However, the use of the directive behavior of such couplers

in silicon technology has not been in focus for a while. The

design of reflectometers for different applications requires cou-

plers with high isolation and therefore directivity whereas the

properties listed above are of minor concern. Such directional

couplers are used to divide the incident and reflected wave and

are part of integrated vector network analyzers for readout of

integrated mm-wave sensors [7] or built-in test circuits [8].

In this paper different directional couplers have been de-

signed and presented in two different process technologies.

Several branchline couplers have been designed and measured

and a broadside coupled-line coupler with more than 150GHz

bandwidth is presented. The couplers are compared in area,

insertion loss, isolation, and susceptibility to process variation.

The passive structures have been simulated with SONNET

and very good agreement between simulation and measure-

ment is observed.

II. DIRECTIONAL COUPLERS

130fF130fF

130fF 130fF

135pH 135pH

92pH

92pH

85 85

85

85

0.275

0.275

0.4 0.4

110fF 110fF

110fF110fF

P1P1 P2P2

P3P3 P4P4

Fig. 1. Schematic of a lumped elements (left) and reduced size (right)branchline couplers at 61 GHz. The inductors and capacitors reactance (left)is equal to the characteristic impedance of the respective lines (35 or 50).

A. Reduced Size Branchline Couplers

Two reduced size branchline couplers at 61GHz and at

122GHz have been designed in different technologies. The

standard branchline coupler includes four quarterwave lines

(or branches) and therefore consumes a large chiparea. A

special technique enables the use of branches with shorter

physical length by the use of lines with a higher characteristic

impedance and capacitors at the junctions. According to [9]

the electrical length of a quarterwave line can be calculated

with sin = Z0/Z1,2 and the capacitors as C = cos /(Z0),where Z0 is the actual impedance (86) and Z1,2 is thedesired impedance (35 or 50) of the line. The schematic ofthe coupler is shown on the right side of Fig. 1, whereas the

layout of the coupler with the metal-insulator-metal (MIM)

capacitors is depicted in Fig. 2. The process variations of

these capacitors mainly determines the isolation of the cou-

pler. The effect on center frequency and phase difference

between coupled and direct port is low. The size of the

coupler depends on the physical length of the lines, which is

Proceedings of Asia-Pacific Microwave Conference 2010

Copyright 2010 IEICE

TH2F-5

806

inversely proportional to the characteristic impedance of the

lines. In order to increase the impedance of the line, the unit

capacitance has to be decreased by increasing the height h

(on the uppermost metal layer) and decreasing the conductor

width. However, most designers and simulators assume the

full embedding of the upper metal in the dielectric substrate.

In this process however, the lack of planarization after the

last deposition of SiO2 and the subsequent deposition of

the passivation layer results in a layer stack as shown in

Fig. 2. With smaller conductor width w the capacitance of

the sidewalls to the ground plane is increasing, but with this

topology the electric field is penetrating the passivation layer

and the air. A proper simulation setup is therefore required,

which has been setup by dielectric bricks in SONNET . The

characteristic impedance is decreasing from 86 to 82 withflat SiO2 layer in comparison to the actual layer stack.

MIM Capacitor

h

w

TopMetal2

Metal1

Passivation r=5

SiO2 r=4.1

Si epi r=11.9 =5 S/m

Si bulk r=11.9 =2 S/m

Fig. 2. Layer stack (left) and layout (right) of a reduced size branchlinecoupler with MIM-capacitors.

55 57 59 61 63 65 6740

30

20

10

0

Frequency (GHz)

Magnitude (

dB

)

S11

S21

S31

S41

Fig. 3. Deembedded S-Parameter measurements of the 61-GHz reducedsize branchline coupler. The measurement results of the coupler with thehighest isolation measured is shown, The variation of the capacitors result ina degradation of the isolation to 30 dB at the center frequency, but has nearlyno influence on insertion loss and the phase difference.

B. Lumped Elements Branchline Coupler

Another option to improve the area requirements of branch-

line couplers is the use of lumped elements. A quarterwave

line with characteristic impedance Z0 can be synthesizedby a Pi-network consisting of an inductor with L = Z0

and two capacitors with 1/C = Z0. The circuit and theirparameters are shown in Fig. 1. The inductors are designed

with SONNET and show a good agreement with simulation.

The use of a SiGe BiCMOS technology results in a layer

stack where am epitaxial-grown silicon layer (required for

bipolar transistor formation) with higher substrate resistivity

is introduced between he silicon and the isolating SiO2 layers.

The deembedded measurement results of the lumped ele-

ments coupler are depicted in Fig. 4. This coupler shows higher

losses compared to the reduced size coupler. A chipmicrograph

is shown in Fig. 5.

55 57 59 61 63 65 6725

20

15

10

5

0

Frequency (GHz)

Magnitude (

dB

)

S11

S21

S31

S41

Fig. 4. S-Parameter measurements of lumped elements branchline coupler.The isolation is high although all inductors couple into the same substrate.

C. Inverted Microstrip Branchline Coupler

Inverted microstrip lines have been presented in this tech-

nology for a low noise amplifier at 122GHz [10]. The ground

plane is formed on top of the silicon wafer and the conductor

placed directly below as shown in Fig. 6. This results in

lower ground losses due to the thicker ground metal, and

lower sensitivity to electromagnetic radiation fields due to the

Fig. 5. Chipmicrograph of 61-GHz reduced size (left) and lumped elements(right) branchline couplers. The size is nearly identical and both devicespossess the same pad-frame. Ground pads are shared between neighboringcouplers to reduce the footprint for a characterization. The branches of thereduced size coupler (left) are bent to save chip area. It has to be ensured,that the coupling between the bends and different branches is negligible. Theinductors and the MIM capacitors of the lumped elements couplers can beeasily identified. In the center of the coupler a dedicated ground return pathfor each inductor is drawn. The couplers are fabricated in a 250-nm SiGeBiCMOS technology with a 5-metal layer front-end from IHP.

807

shielding groundplane. However, the field of the line couples

to the conducting silicon substrate resulting in lower isolation.

An inverted microstrip coupler has been designed with

122GHz center frequency in a 7-metal 130-nm BiCMOS

process from IHP. The required area is four times the one

of a reduced size coupler at the same frequency.

h

w

TopMetal2

TopMetal1

SiO2 r=4.1

Si epi r=11.9 =5 S/m

Si bulk r=11.9 =2 S/m

Fig. 6. Layer stack of inverted microstrip branchline coupler showing thelarge ground plane on TopMetal2 and the conductor on TopMetal1. Theground plane is connected to a metal grid throughout the chip to ensureshielding of all structures. The distance h between conductor and groundplane is nearly four times smaller than the distance to the silicon substrate,therefore the unit capacitance to ground is much larger than to the substrate.

Deembedded measurement results from 90 to 140GHz are

shown in Fig. 8. The coupler shows better than 20 dB isolation

from 115 to 128GHz and a return loss below -40 dB at

131GHz. The coupler has better performance in measurement

than in simulation.

D. Broadside Coupled-Line Directional Coupler

Broadband directional couplers can be designed as coupled-

lines, in most cases drawn by parallel lines with a small

gap. However, higher coupling can be achieved by broadside

coupling in silicon technology. The vertical distance between

two conductors in the upper metal layers is in most cases

smaller than the permitted horizontal distance. Moreover, the

size w of both coupled-lines can be arbitrarily chosen. The

width of the lower conductor should be made smaller than

the upper one, as the distance to the substrate is lower. The

coupler can be further optimized for special applications by

introducing more asymmetries, e.g. a displacement of the

lower conductor. The structure has been placed three times

to enable 4-Port S-Parameters with a 2-Port measurement

setup. On each structure different ports are connected to

the pads and the remaining ports are terminated on-chip.

The micrograph of the three connected structures is shown

Fig. 7. Chipmicrograph of the inverted microstrip (left) and reduced size(right) branchline coupler at 122 GHz center frequency. The conductor of theinverted microstrip coupler can not be seen as the ground plane is formed ontop of the metal stack. The structures have been fabricated in a 130-nm SiGeBiCMOS process with 7-metal aluminum front-end from IHP.

90 100 110 120 130 14050

40

30

20

10

0

Frequency (GHz)

Magnitude (

dB

)

S21

meas

S31

meas

S41

meas

S11

meas

Fig. 8. Deembedded S-Parameter measurements of the inverted microstripbranchline coupler from 90 to 140GHz showing an isolation better than 20 dBand equal signal split from 115 to 128GHz. The optimum return loss is betterthan -40 dB at 131GHz.

Metal5

TopMetal1SiO2 r=4.1

Si epi r=11.9 =5 S/m

Si bulk r=11.9 =2 S/m

s

Fig. 9. Layer stack (left) and layout (right) of broadside coupled-linedirectional couplers. The lower metal layer is directly below the upperconductor and is not seen in the layout. The slit in the ground plane increasesthe coupling of the structure.

in Fig. 10. All S-Parameters can be measured in this way

except the transmission coefficient of the lower line. A THRU

deembedding structure has also been placed on the chip.

The connections on all three structures should be identical

(but attached to different ports). This is unfortunately not

possible, as two ports lie on lower metal layers. This requires

a via and a connection on the lower layer (with nonidentical

characteristic impedance). The magnitude of the transmission

of this connection is nearly identical but the phase is slightly

different. Simulations however show a difference in magnitude

and phase of only 0.028dB and up to 0.35 from 20 to

200GHz. The measurements, which are shown in Fig. 11, have

Fig. 10. Chipmicrograph of a broadside coupled directional coupler. Thecoupler has been placed three times to measure 4-Port S-Parameters with a2-Port VNA. The connections should be identical to simplify deembedding,but attached to different ports. The coupler has been fabricated in a 130-nm7-metal layer SiGe BiCMOS process from IHP.

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been performed from 20 to 115GHz with an Agilent 8510XF

VNA and from 90 to 140GHz with a Rohde & Schwarz

ZVB. The transmission coefficients at 110GHz are 4.15 dB.

The coupler has a 3-dB bandwidth of 156GHz with corner

frequencies at 32GHz (measured) and 188GHz (simulated).

The isolation is above 12 dB over the measured frequency

range even without optimizing the structure. The measured

phase difference between direct and coupled port is between

81 and 90 from 20GHz to 140GHz. Simulation results

of the S-parameters are also shown in Fig. 11. A very good

agreement can be observed in the total frequency range with

only a small step in the change of the measurement setup.

The return loss, however, shows a major step at 90GHz,

presumably due to calibration. The broadside coupled-line di-

20 60 100 140 18025

20

15

10

5

0

Frequency (GHz)

Magnitude (

dB

)

S21

meas

S31

meas

S41

meas

S11

meas

S21

sim

S31

sim

S41

sim

S11

sim

Fig. 11. S-Parameter measurements and simulated values of transmission pa-rameters for the broadside coupled-line directional coupler. The S-Parametershave been measured from 20 to 115GHz and from 90 to 140GHz withtwo different frequency extenders and setups. The measurements are in goodagreement with the simulation results.

rectional coupler can be further optimized for either broadband

quadrature generation or for improved isolation. The coupling

ratio of the broadside coupled-line can be arbitrarily chosen.

III. COUPLER COMPARISON

A comparison of the designed couplers can be found in

Table I with the center frequency, the size, the transmission

parameters of the direct and coupled port and the isolation at

the center frequency.

The reduced size coupler possesses the highest isolation,

but only in a small frequency range. It has good insertion

loss, low area and is therefore the best choice for narrowband

applications. The broadside coupled-line topology shows lower

insertion loss, lower sensitivity to process parameters, higher

bandwidth, and lower area requirements compared to the other

presented couplers. The isolation and therefore the directivity

is lower than the reduced size coupler. The remaining couplers

show lower performance in terms of insertion loss and area

requirement.

TABLE ICOMPARISON OF DIFFERENT COUPLERS

Coupler Type f0 size S21 S31 Iso

(GHz) (m2) (dB) (dB) (dB)

Reduced Size 62 180 x 205 -4.39 -4.52 39

Reduced Size 122 125 x 115 -4.8 -4.8 21

Lumped Elements 61 180 x 205 -5.23 -5.38 23

Inverted Microstrip 123 225 x 250 -4.8 -4.8 27

Broadside 110 95 x 160 -3.8 -3.8 13

IV. CONCLUSION

In this paper five different couplers have been presented and

compared with different center frequencies and bandwidths

in different technologies. A reduced size branchline coupler

at 61 and 122GHz, a lumped elements branchline coupler

at 61GHz, a normal size branchline coupler realized with

inverted microstrip lines at 122GHz, and a broadside coupled-

lines directional coupler have been designed and measured.

Further studies will concentrate on the optimization and the

behavior of broadside coupled directional couplers up to

325GHz with improved directivity.

ACKNOWLEDGMENT

The authors would like to thank Falk Korndorfer, Johannes

Borngraber, and Christian Wipf for measurement and chipmi-

crographs of the structures and IHP Microelectronics GmbH

for fabricating the chips.

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