Design optimisation of ring elements for broadbandreflectarray antennas

N. Misran, R. Cahill and V.F. Fusco

Abstract: Plane wave scattering from a flat surface consisting of two periodic arrays of ringelements printed on a grounded dielectric sheet is investigated. It is shown that the reflection phasevariation as a function of ring diameter is controlled by the difference in the centre resonantfrequency of the two arrays. Simulated and measured results at X-band demonstrate that thisparameter can be used to reduce the gradient and improve the linearity of the reflection phaseversus ring size slope. These are necessary conditions for the re-radiating elements to maximise thebandwidth of a microstrip reflectarray antenna. The scattering properties of a conventional dualresonant multilayer structure and an array of concentric rings printed on a metal backed dielectricsubstrate are compared and the trade-off in performance is discussed.

1 Introduction

A microwave reflectarray is a compact low profile antenna,consisting of a grounded flat array of resonant conductingelements and a primary source. A plane wavefront iscreated by controlling the scattering properties of theindividual microstrip patches, which are designed to give aprogressive reflection phase shift across the aperture [1].This type of antenna combines many of the best features ofarrays and conventional reflectors. For example, spatialfeeding eliminates the design complexity associated withconventional microstrip fed arrays and can reduce thedielectric absorption and ohmic losses responsible forreducing the gain of these antennas particularly at highfrequencies. The main disadvantage of the reflectarray is thelimitation on the available bandwidth, which is imposed byvarying the dimensions of each re-radiating element aboutits nominal resonant size. This is the simplest methodavailable to control the reflection phase; however, the phaseversus microstrip patch size slope is typically very large andnon-linear at the extremities [2]. The former implies thattight fabrication tolerances are necessary to achieve thedesired phase value, and the latter property restrictssatisfactory operation to a narrow band of frequencies. Athick substrate can be employed in the design to reduce thephase slope; however, as the Q factor decreases so does thetotal phase range. Ideally, values from 01 to 3601 should beavailable from the elements in a reflectarray, so a reductionin the phase range is detrimental to the pattern performanceof this class of antenna.

Encinar [3] has exploited the dual resonant response of atwo-layer grounded array of dissimilar size patch elementsto overcome the limitations associated with the use of thicksubstrates. In that work, a linear response was achievedover an 8% bandwidth from a five-layer sandwich structure

consisting of a ground plane, two spacers, and two periodicarrays with a constant stacked patch size ratio. This designexcludes the possibility of obtaining dual frequencyoperation since independent control of the dimensions ofthe two patches is normally required in each unit cell [4].

This paper considers a building block for a simpler andmore versatile architecture where the reflection phase iscontrolled by varying the diameter of resonant rings. Suchelements are widely employed as frequency selective surfaces(FSS) [5]. First, the electromagnetic behaviour of a two-layer grounded ring array (Fig. 1a) is studied, since theelements in the two screens are weakly coupled andessentially yield independent reflection coefficients [6]. It isshown that the rings can be nested (Fig. 1b) and printed onto a single planar surface to give the required dual resonantresponse. This structure is a superior alternative to otherbroadband element designs that have been reported becausethe periodic array can be printed on a single surface,eliminating the need for double side mask alignment. Thismakes construction of the antenna very much simpler andcheaper, and makes broadband dual frequency operationfrom a multilayer structure is possible. However, the phaserange obtainable from a grounded concentric ring screen isshown to be limited by the minimum required physicalseparation of the elements. The transmission line matrix(TLM) [7] method has been used to study the scatteringbehaviour of the periodic ring arrays, and the results at Xband are shown to be in close agreement with waveguidesimulator measurements [8].

2 Phase response of ring elements

Design methodologyPeriodic arrays of conductive ring elements on close packedsquare or triangular lattices have frequency selectiveproperties that have been exploited for a wide range ofterrestrial antenna and space instrument applications [9].The filter response of these frequency selective surfaces(FSS) is largely determined by the element diameter andperiodicity, and the electrical properties and thickness of thebacking substrate. The transmission and reflection bands ofsimple ring arrays are generally widely spaced with an edgeof band frequency ratio (where the loss is less than 0.5dB)

The authors are with High Frequency Electronics Laboratories, School ofElectrical and Electronic Engineering, Queen’s University Belfast, AshbyBuilding, Stranmillis Road, Belfast BT9 5AH, N. Ireland, UK

r IEE, 2003

IEE Proceedings online no. 20030857

doi:10.1049/ip-map:20030857

Paper first received 23rd January and in revised form 16th June 2003. Onlinepublishing date: 22 October 2003

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of 2.5:1 or more [5]. Planar arrays of printed concentricrings [10] have a usable passband between the resonances,which are generated by the individual rings, and for theseelements the band spacing can be as small as 1.2:1. HigherQ filtering where the passband and rejection band spacing isless than 1.07:1 has been achieved by making using of theinterference effects between two or more screens of simplering elements [11]. Generally a trade-off must be establishedbetween the polarisation purity, Q factor, bandwidth,sensitivity to angle of incidence, and the onset of gratinglobes. However, for a given application, well establisheddesign rules can be invoked to optimise the physical layoutof the periodic array.

A reflectarray is simply a printed FSS with a groundplane on the opposite surface of the substrate; therefore, thiscan be designed to give broadband performance bycontrolling the frequency response of the array in the sameway as a FSS. Broadbanding of the reflectarray elementshas been investigated by analysing the complex scatteringparameters of a unit cell consisting of one or more co-located resonant rings separated from the ground plane bya dielectric wafer. In this way, the infinite array model [1]can be used to establish the relationship between thediameter of the rings and the phase of the reflected energy.For this study, the transmission line matrix (TLM) packageMICROSTRIPES V5.5 [7] (MICROSTRIPES is a trademark software product by FLOMERICS) was used tomodel the response of the structures at X band. The three-

dimensional mesh was graded manually with a minimumsize of 0.005l in the region close to the array where the fieldsare highest. To reduce the run-time, symmetry wasexploited and only one quarter of the unit cell was meshed,and an infinite array created using suitable boundaryconditions. The excitation port was located l/4 above theelements and the detection port was located on the surfaceof the ground plane, which served as the phase reference forthe S11 predictions. Frequencies swept simulations of theinfinite array were performed by setting a zero cut-offfrequency in the waveguide structure, and thereforemodelling was performed at normal incidence where thelargest unattainable phase range has been shown to occur[12].

Double layer ring reflectarray elementsFigure 1a shows the configuration of the double layergrounded ring sandwich structure, which consists of twoperiodic arrays, two dielectric spacers and a ground plane.The 0.4mm wide conducting loops are centred on a 10mmsquare lattice, and the screens are separated from each otherand the ground plane by 1.524mm thick dielectric substratelayers of permittivity er ¼ 3:54 and tand¼ 0.0018. TACO-NIC [13] material with these values was used to provideexperimental verification of the computed results. The ratioof the upper and lower ring diameters (U/L), d2 and d1 inFig. 1a, was selected to be 0 (upper ring removed), 0.8, 0.93and 1 (upper and lower ring sizes the same). Computedplane wave reflection phase coefficients for TE normalincidence are shown in Fig. 2, where the lower ring diameteris varied between 4.0mm and 6.6mm. To provide a usefulcomparison, the curves in Fig. 2 have been plotted at thefrequency of resonance (fo) for a lower ring diameter of5.4mm, and relative to the 1801 reflection phase obtainedfrom a perfect conductor. The results suggest that theinterlayer coupling factor is small since scattering from theupper ring changes the resonant frequency by less than

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Figure 2 Computed and measured reflected phase of groundedtwo-layer stacked ring array structureUpper ring diameter/lower ring diameter (U/L)Resonance frequency (fo)

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100MHz about the nominal value of 8.55GHz for thesingle embedded ring (U/L¼ 0). For the upper ring, theeffective permittivity of the substrate is lower, and thereforeat this frequency a second resonance occurs when the loopdiameter is 6.4mm. Figure 2 shows that, in the range whereboth rings are capacitive (3.9–5.4mm), the size of the upperring has little effect and the reflection response is non-linearwith a phase range less than 1501. When the diameter of thelower ring is larger than 5.4mm the element is inductive, butthe upper ring is capacitive below its resonant diameter of6.4mm. Therefore, between these values the equivalentcircuit of the two-layer periodic screen is simply a shunt LCcircuit. At 8.5GHz, a second resonance (3601 in Fig. 2) isgenerated when the size of the lower ring is 8mm and6.4mm for U/L values of 0.8 and 1 respectively. Thesignificance of this is clear from Fig. 2, which shows that thelinear phase range of the slope between the two resonancesis critically dependent on the difference in the physical sizeof the lower ring at these two positions. For example, Fig. 2shows that when U/L¼ 0.8 the phase range is less than2001, and the non-linear response is similar to a singleembedded element (U/L¼ 0). However, when the diameterof the two rings are identical (U/L¼ 1), the curve is linearover a phase range which exceeds the 3601 needed toprovide a practical reflectarray design [1]. When the U/Lparameter is reduced slightly to 0.93, the phase slope is lesssteep, and therefore this array design is less sensitive tomanufacturing tolerances.

The scattering behaviour of the stacked rings can beexplained by observing the predicted swept frequencyreflection phase coefficients of the three arrays. This isshown in Fig. 3 relative to a perfect conductor where, foreach U/L parameter value, the lower ring diameter is5.4mm. These results can be mapped to those plotted inFig. 2 if it is assumed that the permittivity of the dielectric isindependent of frequency and also within the computedband the variation in electrical thickness does not modifythe reflection phase values. In Fig. 3 the phase variation issimilar in the range up to 9GHz and 10GHz for U/L valuesof 1 and 0.8 respectively, since, as shown in the TLMmodel, the surface current is confined mostly to the lowerring. However, at higher frequencies the phase variationresults mainly from current excitation in the upper ring andas a consequence can be tailored by choosing the relativesize of the upper element. For a U/L value of 0.93, thegradient is less steep than a stacked ring arrangement wherethe diameters are identical (U/L¼ 1). A further decrease inthe phase gradient can be achieved when the separation of

the two resonances increases further, as shown in Fig. 3 forthe element configuration U/L¼ 0.8. However, betweenthe resonant points (8.6GHz and 12.7GHz) in the range9.5–10GHz, the phase slope reduces since the currentamplitude in both rings is very low. This is shown in Fig. 4,where the periodic structure is around 80% transmissive (at11GHz) between the resonant frequencies, which areshifted upwards when the ground plane is removed. Thisexplanation is consistent with the formation of a plateau inthe phase plot in Fig. 2 for diameters 6mm to 6.6mm;however, beyond this range, the reflection phase convergesrapidly to 3601 when the lower ring diameter is 8mm.Clearly the ratio of the stacked ring diameters is animportant design parameter which can be used to minimisethe phase gradient without compromising the linearity ofthe response curve.

Concentric ring planar structureA double resonant response can also be obtained from aconcentric ring FSS where the conductive loops are nestedand printed on to a single surface [10]. The advantage ofthis approach is that the reflectarray structure can beconstructed simply by photo-etching the array pattern onone side of a conductivity coated dielectric substrate.Moreover, broadband dual frequency operation may beachieved by replacing the single ring elements in the five-layer sandwich structure described in the previous sectionwith concentric rings.

The critical design driver for multilayer reflectarrays is therelative size of the stacked elements, since this determinesthe separation of the two resonant frequencies. A U/L ringdiameter ratio of 0.93 was shown to provide a suitableresponse for broadband applications by resonating atfrequencies around 8.5GHz and 11.0GHz. The groundedconcentric ring array design shown in Fig. 1b behaves like aperfectly conducting screen at the same two frequencieswhen the inner to outer ring diameter ratio is I/O¼ 0.8,therefore this reflectarray element might be expected to givea linear phase variation similar to the one shown forU/L¼ 0.93 in Fig. 2. However, the computed results inFig. 5 show that the reflection phase change is small whenthe element size is increased from 7.0mm to 8.0mm. Twoother geometries were analysed with inner to outer ringdiameters I/O¼ 0.71 and 0.86, which are excited at similarresonant frequencies to the double layer structure withparameter values U/L¼ 0.8 and 1 respectively.

A comparison between the results in Figs. 2 and 5 showsthat it is not possible to achieve a linear response over aphase range 43301 from a concentric ring array by simplymapping the resonant frequencies of the two reflectarray

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structures. Although the swept frequency plots in Figs. 3and 6 show that the phase variation is similar around thefirst resonance (8.5GHz), the response of the inner and toprings is very different. For example, the computed results inFig. 6 show that the Q factor of the concentric rings is largerbecause of the formation of a wide region between the tworesonant frequencies where the phase slope is significantlyreduced (�1201 to�1801). Removing the ground plane andobserving the transmission coefficients (Fig. 7) of the FSSconfirms that each array is almost completely transmissiveover a range of frequencies which increases when theparameter value I/O decreases. Again, this behaviour isconsistent with the results plotted in Fig. 5, where the

plateau is formed in the region where the periodic array isweakly excited. Improving the linearity of the phase versuselement size plots requires modification of the classicaltransmission coefficients of the concentric ring array toremove or narrow the passband peak between the tworejection bands plotted in Fig. 7. Clearly this can beachieved by reducing the separation between these twobands; however, in a practical design this is difficult toimplement because of the need to avoid contact between theconductive tracks of the nested rings. For example, in ourcomputer model, the gap is only 0.34mm for the I/O¼ 0.86design, where the ring conductor width is reduced from0.4mm to 0.2mm. A limitation on the broadbandperformance obtainable from elements printed on thicksubstrate material is therefore imposed by the physicalinability to accommodate the nested rings.

3 Measured results

A series of waveguide simulator measurements was used tovalidate the simulated scattering parameters of the reflec-tarray elements. By inserting just a few unit cells across therectangular aperture of a waveguide it is possible to measurethe reflection phase of an infinite grounded rectangulararray of rings of a given size [8]. The resonant frequency ofthe testpieces, and phase variation with element size, weremeasured using a VNA, which provided the reflection phaserelative to a metal sheet which served as a reference plane inthe S11 test set-up. The electromagnetic behaviour of thereflectarray element in the TE plane is identical to thescattering that occurs in the waveguide when a TE10 waveimpinges on the testpiece; however, in the waveguide theangle of incidence is dependent on frequency. A standardWR-90 waveguide adapter was connected to a 200mm longtapered waveguide with a rectangular output section ofdimensions 100mm� 30mm� 10mm. For this waveguidesize the angle of incidence varies [8] between 351 and 271over the measurement frequency range 8.5GHz to11.1GHz. The ring patterns of the double layer array wereprinted on opposite sides of a 1.524mm thick er ¼ 3:54,tand¼ 0.0018 TACONIC RF-35-600 substrate, and asecond laminate with copper deposited on one side onlyprovided the ground plane and spacer as shown in Fig. 1a.The two substrates were clamped together around the edgesby nylon screws to remove air gaps and mounted in the30mm� 10mm rectangular waveguide aperture, whichcould accommodate three closely packed rings, thusensuring insensitivity to angle of incidence effects [5] whichwas confirmed using a Floquet modal technique [9]. The

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Figure 5 Computed and measured reflected phase of groundedconcentric ring array structureInner ring diameter/Outer ring diameter (I/O)Resonance frequency (fo)

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Figure 7 Computed frequency dependent transmission response ofconcentric ring array without ground plane at normal incidence(d1¼ 6.5 mm)

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three unit cells containing the concentric rings were simplyprinted on one side of the 1.524mm thick copper-coatedsubstrate as shown in Fig. 1b.

In Fig. 2, experimental and simulated results at twofrequencies are plotted for a double layer ring array with aU/L value of 0.93. At 8.65GHz, the input transition to thewaveguide simulator is close to its cut-off frequency,nevertheless good correlation is demonstrated in the linearregion of the plot at and beyond the 5.4mm resonantdiameter of the embedded rings where the reflectarrayantenna design data is obtained. This reflectarray elementconfiguration is predicted to give a similar response at11.1GHz, since at this frequency (fo) the resonant diameterof the upper ring is 5mm. In Fig. 2 measurements that wereperformed at this frequency, which is well above thewaveguide cut-off, are in closer agreement with thecomputed results in both the linear region of the plotsand the non-linear range, where both rings are inductive.Note that in Figs. 2 and 5 the experimental phase datapoints have been shifted by 1801 to enable the shape of theplots to be more easily compared. The equivalent differencein the resonant frequency (or conversely, the resonant ringsize at the measurement frequency) can be attributed to theprediction accuracy of the TLM computer model. Mea-sured and numerical data for a concentric ring array withan inner to outer diameter ratio of I/O¼ 0.8 are comparedin Fig. 5. The results confirm the existence of a smallplateau region centred at 7.5mm where there is incompletereflection from the periodic array.

4 Conclusions

It has been shown that the bandwidth of a reflectarrayantenna can be increased by exploiting the scatteringproperties of a dual resonant periodic array of printedrings above a ground plane. The ability to control thelinearity and Q factor of the reflection phase response hasbeen demonstrated by varying the diameter ratio of astacked double ring resonant element. To avoid non-linearity in the phase plot, it is necessary for the periodicarray to be strongly excited over the full range of ringdiameters used in the reflectarray design, and for this tooccur the individual resonances must be closely spaced. Fora given dielectric thickness, this imposes a limit on thegradient reduction obtainable. The optimum performancefor the stacked arrangement described in this paper isobtained when the resonant frequency of the upper ring isapproximately 1.28 times higher than the lower ring. For agiven periodicity, smaller elements generate a narrowerreflection band and the implication of this is an increase inthe resonant Q factor of the array and hence the phaseslope. However, for a stacked arrangement this is

compensated for by the larger separation between theground plane and the upper ring, and therefore independentcontrol of the Q factor is obtainable for the two rings. Adual resonant periodic array of nested rings has also beenshown to provide a broadband response with a linear phaserange in excess of 3301 at normal incidence. This is anattractive option because of the simplicity of printing thearray on a single surface, and furthermore by stacking twoscreens it should be possible to provide an independent andcontrollable broadband phase response at two frequencies.However, for optimum performance, the resonant fre-quency of the concentric rings must be closer than the tworings in the stacked arrangement, since as shown in Figs. 4and 7 the frequency selective properties of the periodicelements are quite different.

5 Acknowledgments

This work was funded by the UK EPSRC under grant no.GR/R15139/01 and GR/R54996/01. N. Misran is sup-ported by a studentship from the Universiti KebangsaanMalaysia. The authors acknowledge the support providedby Dr Rachid Aitmehdi of Flomerics.

6 References

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5 Cahill, R., and Parker, E.A.: ‘Crosspolar levels of rings in reflection at451 incidence: influence of lattice spacing’, Electron. Lett., 1982, 18,(24), pp. 1060–1061

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10 Cahill, R., and Parker, E.A.: ‘Concentric rings and Jerusalem crossarrays as frequency selective surfaces for a 45 degree incidencediplexer’, Electron Lett., 1982, 18, (8), pp. 313–314

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