IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 8, AUGUST 2014 4039

Design and Test of a 35-kJ/s High-Voltage CapacitorCharger Based on a Delta-Connected Three-Phase

Resonant ConverterSuk-Ho Ahn, Member, IEEE, Hong-Je Ryoo, Ji-Woong Gong, and Sung-Roc Jang

Abstract—This paper describes the design, implementation, andanalysis of a 35 kJ/s high-voltage capacitor charger based on adelta-connected three-phase series resonant converter that pro-vides a constant charging current with high efficiency and high-power density. In order to obtain the maximum output power forvarious charging voltages, each high-voltage transformer suppliedby a delta-connected resonant inverter is designed with two sec-ondary windings and voltage-doubled rectifiers. This configurationallows not only a flexible output current and voltage with fixed out-put power but also a high power factor on the input side. On thebasis of the analysis of the series-loaded resonant converter oper-ating at a discontinuous conduction mode, the details of the designprocedure for the resonant inverter are provided. Furthermore, theimplementation of the high-voltage transformers and rectifiers isalso explained while considering insulation and compactness. Ex-periments were carried out on the developed charger with differenttypes of capacitors, depending on their applications, and the resultsare discussed. In addition, malfunctioning tests were conducted forthe open, short, and misfiring during charging conditions. Finally,the developed high-voltage capacitor charger was shown to be veryreliable, even under faulty operating conditions in the system.

Index Terms—High-voltage capacitor charger, pulsed-powerapplications, series-loaded resonant converter.

I. INTRODUCTION

R ECENTLY, a considerable number of studies have beenconducted on high-voltage capacitor chargers for many

types of capacitive energy-storage-based pulsed-power applica-tions including laser accelerators, electrothermal chemical gun,rail gun, and plasma source ion implantation [1]–[14]. More-over, the concern with the importance of fast charging timeand efficiency as well as the compactness and reliability hasbeen growing over the proliferation of the application area.The advantages of the constant-current charging method overthe constant-voltage charging through a current-limiting resis-tor method are the charging rate and efficiency [1], [5], [11].

Manuscript received June 30, 2013; revised September 15, 2013; acceptedOctober 20, 2013. Date of current version March 26, 2014. Recommended forpublication by Associate Editor P. Tenca.

S.-H. Ahn and J.-W. Gong are with the Department of Energy Conver-sion, University of Science and Technology, Changwon 641120, Korea (e-mail:whiteyan@keri.re.kr; smarty@keri.re.kr).

H.-J. Ryoo and S.-R. Jang are with the Electric Propulsion Research Center,Korea Electro-technology Research Institute, Changwon 641120, Korea (e-mail:hjryoo@keri.re.kr; scion10@keri.re.kr).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TPEL.2013.2287709

Furthermore, many types of topologies have been proposed forthe resonant converter to generate a constant output current.In addition, the resonant converter allows high-frequency op-eration because of soft switching such as zero-voltage (ZV)or zero-current (ZC) switching of inverter switches. Hence, ahigh-frequency inverter with reduced switching loss helps toincrease the efficiency of the capacitor charger as well as thepower density with the compact design of magnetic and filtercomponents.

The resonant converter topology can be classified into series,parallel, and series–parallel resonant converters on the basisof a resonant-tank structure. Even though the parallel resonantconverter topology is suitable for a voltage step-up application,because a capacitor is connected parallel to the load [13], itprimarily acts as the voltage source. The output current of theseries–parallel resonant converter topology, which has the com-bined advantages of the other topologies, also depends on theoutput voltage [14], [15]. On the other hand, the series reso-nant converter in the discontinuous conduction mode (DCM)operation provides the pure current source characteristic thatrepresents a constant charging current independent of the ca-pacitor voltage. It facilitates the control of the charging currentby applying a fixed-switching frequency. Moreover, soft switch-ing at on and off transitions can be achieved for the full loadrange. Both the conduction loss and parasitic capacitance of thesecondary winding of the transformer, however, are pointed outas the main disadvantages of the series resonant converter in theDCM operation [2]. Besides efficiency, the input power factoris also an important performance factor of the capacitor charger.In order to achieve a high power factor, an optimized design ofLC filter components and a power-factor correction circuit hasbeen introduced [1], [7], [16]–[20].

In this paper, the enhanced scheme of a three-phase delta-connected series resonant inverter with a voltage-doubled recti-fier is proposed to reduce the conduction loss and parasitic ca-pacitance. Conventionally, 6 switches are required to configurea three-phase half-bridge series resonant inverter, as reportedin [21] and [22], and 12 switches are required to configure athree-phase full-bridge series resonant inverter. In this study,the proposed scheme uses only six switches to configure thethree-phase full-bridge series resonant converter by applyingthe delta connection. It also has an advantage in terms of costand structure. Furthermore, the controllable maximum output-voltage function by modifying the configuration of the six rec-tifiers is suggested to increase the charging current for rela-tively low-voltage applications. Furthermore, the flexible output

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4040 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 8, AUGUST 2014

current and voltage with fixed output power provides a highpower factor for many applications.

The analysis, design, and implementation of the series res-onant converter based on the proposed high-voltage capacitorcharger are explained in detail. Furthermore, experimental re-sults that verify the reliability of the charger under short andopen conditions and unforeseen severe fault conditions whilecharging such as short, opening, and unintended short are dis-cussed in this paper.

II. ANALYSIS AND DESIGN OF THE PROPOSED HIGH-VOLTAGE

CAPACITOR CHARGER

For the proposed high-voltage capacitor charger, a series res-onant converter in the DCM operation topology was adoptedbecause of its many advantages such as high-power density,current source characteristics, and high efficiency. For a high-voltage power supply, a substantial leakage inductance of thehigh-voltage transformer, which can adversely affect the op-eration of the converter, is inevitable because of the isolationdistance between the primary and the secondary windings. Nev-ertheless, this parasitic element can be used as a resonant induc-tance in the series resonant converter. Additionally, the seriesresonant converter topology with the DCM operation is suit-able for capacitor charging applications because of its currentsource characteristic, i.e., it provides a nearly constant currentregardless of the load voltage but depending on the frequency ra-tio between the switching and resonant frequencies. Therefore,when both frequencies are fixed, they give a constant currenteven when operating under shorting conditions.

In order to understand the design of the developed capacitorcharger, it is necessary to focus on the overall scheme of thecharger that is depicted in Fig. 1. It can be classified into fourmain parts: an input rectifier and filter for ac to dc conversionof the input, a three-phase resonant inverter for dc to ac con-version, a high-voltage transformer and rectifier portion, and anRC divider for sensing high voltages. It should be noted that thethree-phase output is connected as a delta scheme through theresonant tank and high-voltage transformer, which is advanta-geous for the operation.

In the delta-connected primary winding configuration, the in-put voltage of each transformer is the same as that of the inverter.In order words, the delta connection creates the three-phasefull-bridge scheme with just six insulated gate bipolar transis-tors (IGBTs) in comparison with the star connection, whichneeds 12 IGBTs for the three-phase full-bridge scheme. Thehigh-voltage transformer with the rectifier portion consists ofsix voltage-doubled rectifier circuits that are individually con-nected to each secondary winding.

Depending on the connection of these output rectifiers, themaximum values of the output voltage and current can be deter-mined with a fixed maximum output power to generate a higheroutput current in the case of a lower charging voltage for a widerange of applications. That is, when the output rectifiers fromsix secondary windings are connected in series, it can generatesix times the output voltage of one rectifier with the output cur-rent of one rectifier. On the other hand, if all the rectifiers are

Fig. 1. Scheme of the proposed delta connected three-phase capacitor charger.

TABLE ISPECIFICATIONS OF THE PROPOSED HIGH-VOLTAGE CAPACITOR CHARGER

connected in parallel, the value of the output current will be sixtimes that of one rectifier with the voltage of one rectifier. Fur-thermore, the R and C voltage divider gives a noiseless sensingsignal to the controller for the voltage control.

The specifications of the designed charger are summarized inTable I. The maximum charging power is designed to be 35 kJ/sfor a 440-V ac input condition with 10% line regulation. More-over, when all the high-voltage rectifier circuits are connectedin series, the maximum charging voltage is 25 kV. This can bechanged depending on the application. When a lower chargingvoltage with a high repetition rate is desired, the output currentcan be increased by simple reconnection of the high-voltagerectifier circuits. In this study, two types of rectifier connectionsare tested. The first is the original design, which is a series con-nection with 25-kV maximum voltage, and the second is theconnection where three rectifiers are connected in series andtwo in parallel with a maximum voltage of 12.5 kV, as shown

AHN et al.: DESIGN AND TEST OF A 35-kJ/s HIGH-VOLTAGE CAPACITOR CHARGER 4041

Fig. 2. Modified rectifier for low-voltage applications.

in Fig. 2. In addition, four kinds of protection are enforced forthe safe operation of the proposed capacitor charger. The first isover-voltage protection against excess voltage at the output sideto protect the high-voltage rectifier diodes from malfunctionsuch as open-circuit operation. Because the proposed schemeoperates as a current source, an open circuit may generate volt-ages above the level of the rectifier diode ratings, even if thereis a voltage controller. Then, over-current protection is requiredto protect the IGBTs from leg shorts. The detected input currentfrom the dc link is compared with the peak value of the resonantcurrent for fault conditions. The third protection is against an ex-cess rise in temperature due to switching and conduction lossesor circuit failures. The final protection is the charging-time pro-tection. As mentioned previously, the proposed topology oper-ates under short-circuit conditions without any problems. Thus,it is necessary to set the maximum charging time because thecharger will keep working even under fault conditions such aswhen a rectifier diode is broken or with output connection prob-lems. The power density of the proposed charger is measured tobe 335 W/l.

A. Operation Principle of the Proposed Topology

As mentioned previously, the proposed topology is a delta-connected three-phase resonant converter. The proposed con-verter, which is configured with only six switches, operates asa three-phase full-bridge converter. Each switch is turned ONunder the zero-current switching (ZCS) condition and turnedOFF under the zero-current zero-voltage switching (ZCZVS)condition. The operation principle of the proposed high-voltagecapacitor charger can be divided into six modes within eachoperating cycle. The operation-mode diagram and the opera-tion waveforms including the line current, which is the same asthe resonant current, phase current, and switching signal, whichis appropriate to the proposed topology upon each mode, areshown in Figs. 3 and 4, respectively. The principle of operationis discussed next.

Mode 1: Mode 1 begins when switch S1 is turned ON. Theinput voltage (Vdc) is applied to the resonant tank (Lr3, Cr3) ofTX3, and the input current flows through switches S1, S6, and(Lr3, Cr3). This forms a powering path. Further, the resonant-tank current of TX1 is freewheeling through D2 connected toswitch S2 in parallel because of the energy accumulated earlier.Then, switch S2 is turned OFF under the ZCZVS condition.

Mode 2: In mode 1, mode 2 begins when S5 is turned ONunder the ZCS condition. Vdc is applied to the resonant tank(Lr1, Cr3) of TX1, and the input current flows through switchS1, which is turned ON earlier, S5, and (Lr1, Cr3). Further,

Fig. 3. Operation waveforms of the proposed capacitor charger.

the resonant-tank current of TX2 is freewheeling through D6connected to switch S6. Switch S6 is turned OFF under theZCZVS condition.

Mode 3: When switch S3 is turned ON, mode 3 begins. The in-put current flows through switch S3, S5, and resonant tank (Lr2,Cr2) of TX2 forming the powering path. Further, a freewheelingpath is formed through switch S3, resonant tank (Lr3, Cr3) ofTX3, and diode D. Because of the current flowing through D1,switch S1 is turned OFF under the ZCZVS condition.

Modes 4–6 can be analyzed using a similar procedure as men-tioned previously. The resonant inductor current and capacitorvoltage have opposite polarity as those of modes 1–3.

B. Design of the Gate Drive Circuit

For the proposed converter, a gate drive circuit having a simplestructure is introduced. The proposed gate drive circuit and itsoperation principle are shown in Fig. 5. The proposed gate drivecircuit has a specific function of protection from leg short causedby unexpected noise, as explained next.

Mode 1: When a turn-on control signal is applied at the inputof the gate drive circuit, the current flows through R1, D1, andC1, which is charging. Then, the current flows through M1 thatis turned ON by the voltage across R1.

Mode 2: After M1 is turned ON, the current flows throughM1, R3 (charging gate of main switch) and R1, D1, and C1,which is also charging. As C1 is charging, the voltage acrossR1, which is the gate voltage of M1, decreases but not enoughto turn OFF the voltage of M1.

4042 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 8, AUGUST 2014

Fig. 4. Operation mode diagrams of the proposed capacitor charger.

Mode 3: In this mode, the main switch is turned ON and thecurrent flows through R1, D1, R2, D2, and the main switch. Thecharging voltage of C1 is clamped to the voltage across the mainswitch made by the turn-on resistance of the actual switch. Thisspecific state of the gate driver circuit is exited to protect againstleg short. If leg short occurs at this state, leg short current flowsthrough the main switch. Because of the turn-on resistance ofthe actual switch, the voltage across the main switch increases.Therefore, diode D2 is reverse biased and current flows onlythrough C1. Then, the gate voltage of M1 that is the voltageacross R1 is decreased to turn OFF the voltage of M1 becauseof the charging voltage of C1. As a result, the main switch ispromptly turned OFF and protected from the leg short current.

Mode 4: When the turn-off control signal is applied at theinput of the gate drive circuit, the current flows through R4, R3,and D3, thereby, discharging the gate of the main switch.

Fig. 5. Proposed gate drive circuit. (a) scheme of the proposed gate drivecircuit. (b) Operation principal of the proposed gate drive circuit.

C. Design of the Proposed Capacitor Charger

The first step of the capacitor charger design is the calcula-tion of the resonant tank parameters (resonant capacitance: Crand resonant inductance: Lr) based on the maximum chargingpower and input voltage. In order to determine the resonant tankparameters, it is necessary to identify the required peak value ofthe resonant current at the primary winding of the transformerto meet the output specifications (35 kJ/s, 25 kV). The valueof the output current (Iout) is determined from the followingequation:

Iout =2 × Pch,max

Vout,max= 2.8[A] (1)

wherePch,max maximum charging power;Wprimary maximum output voltage.

The initial design is based on the maximum output voltageconnection where all the high-voltage rectifiers are connectedin series, as shown in Fig. 1. Because of the characteristicsof the capacitor charging power supply and the voltage chargefrom zero to the maximum value, the output current was cal-culated to be 2.8 A, which is two times larger than the usual

AHN et al.: DESIGN AND TEST OF A 35-kJ/s HIGH-VOLTAGE CAPACITOR CHARGER 4043

continuous dc power supply. On the basis of the calculated dcoutput current, the peak value of the resonant current at the trans-former’s secondary side (Isec,peak) can be found to calculate thecharacteristic impedance (Zo) of the resonant tank. This can becalculated from the relationship between Iout and Isec,peak asexpressed in the following equation:

Iout =2π· Isec,peak · fs

fo+

2π· Isec,peak · fs

fo· Vdc,min − Vpri

Vdc,min + Vpri(2)

whereIsec,peak the peak value of the resonant current through trans-

former secondary winding;fs switching frequency;fo resonant frequency (fo = 1/2π

√Lr · Cr );

Vdc,min minimum dc-link voltage;Vpri transformer primary winding voltage.

The first term of (2) implies the average value of the positivehalf cycle resonant current and the second is that of the nega-tive half average value. Because the resonant current waveformdepends on many factors, including the switching frequency,resonant frequency, and input and transformer primary voltages,the step-by-step calculation of the aforementioned parametersis required to design the resonant tank components. The switch-ing frequency should be determined by considering the IGBTswitch characteristics such as conduction loss, switching loss,turning on and off delay time, and antiparallel diode reverserecovery time. Depending on the aforementioned factors, themaximum switching frequency has limitations even if it oper-ates with the zero current turn on and zero current and zerovoltage turn off. Finally, the switching frequency is selected as25 kHz due to the use of high ratings of IGBT (1200V, 200A)which has 400-A peak collector current (ICM) in maximumdevice junction temperature (Tj(max) , 150 ◦C). Moreover, thevalue of the resonant frequency should be greater than twice themaximum switching frequency from the definition of the DCMoperation. In this paper, the resonant frequency is determined tobe 3.5 times the switching frequency by considering the deadtime of each leg with a fixed pulse width. The minimum dc inputvoltage (Vdc,min) and the transformer primary voltage (Vpri) areclosely related to the voltage gain which represents the ratio ofboth voltages. Its value cannot exceed the unity owing to thebasic concept of series resonant converter topology which in-volves a divider circuit between the resonant tank impedanceand load. The voltage across the transformer’s primary windingis calculated as 0.95 times that of the minimum dc input voltagebecause the higher value of voltage gain provides less stresson the switch and resonant tank components but with a smallmargin to compensate for the voltage drop, such as in the high-voltage rectifier diode [5]. Finally, the peak value of the resonantcurrent at the transformer’s secondary winding (Isec,peak) canbe found based on the above parameters. In addition, the peakvalue of the resonant current at the primary winding (Ipri,peak)can be calculated by simple multiplication of the transformer’sturn ratio (n) and the identified peak current, as in the followingequation:

Ipri,peak = Isec,peak · Wsecondary

Wprimary= Isec,peak · n (3)

whereIpri,peak peak value of the resonant current through primary

winding of transformer;Wprimary turns of transformer primary winding;Wsecondary turns of transformer secondary winding;n transformer turn ratio.

In order to find the accurate value of the turn ratio with respectto the input and output voltage specifications, (4) can be used

n =Vsec

Vpri=

13· 12· Vout,max

Vpri. (4)

The one-third and one-half terms in (3) represent the threeseries connected transformers and voltage-doubled rectifier,shown in Fig. 1. From the calculated value of the turn ratioof the transformer, the Ipri,peak value was determined to be330 A. Therefore, the peak current rating of the selected 1200 V/200 A IGBT satisfies the calculated Ipri,peak . The character-istic impedance of the resonant tank is determined from therelationship between the peak value of the resonant currentand the characteristic impedance, as expressed in the followingequation:

Zo =Vdc,min + Vpri

Ipri,peak. (5)

Finally, on the basis of the two calculated parameters (charac-teristic impedance and resonant frequency) associated with theresonant tank components, the values of the resonant capacitorand inductor are determined as follows:

Lr =Zo

2π · fo

∼= 2.9 μH (6)

Cr =Lr

Z2o

∼= 1.2 μF. (7)

The second step of the proposed capacitor charger design pro-cedure is the design of a high-voltage transformer with a specificleakage inductance. For a compact design and low componentcount, the designed high-voltage transformer should have thesame leakage inductance as the designed resonant inductance.In addition, many factors such as the core saturation, insulationbetween the primary and secondary windings, and the coolingof the transformer and its dependence on the losses should beconsidered.

By considering the ease of design of the transformer bob-bins and the relatively high-power ratings of each transformer, aUU-shaped core was selected, and on the basis of the operatingswitching frequency, a PC40 material core was chosen. Further-more, from the calculation of the RMS current value throughthe primary winding and the required number of turns to preventsaturation, the size of the core was determined to be 120× 235×20 mm with 14 turns in the primary winding. The basic designof the transformer mentioned previously can be obtained withsimple calculations, but the specific leakage inductance shouldbe obtained by maintaining the insulation distance between theprimary and secondary windings.

For the determined core model, the value of leakage in-ductance (Lleakage) is proportional to the square of primary

4044 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 8, AUGUST 2014

Fig. 6. Structure of the designed transformer.

winding turns (Wprimary ), permeability of air (μ0), length ofeach primary winding turn (P ), and distance between eachwinding (Δ1−2) and is inversely proportional to the total wind-ing length of the primary winding (H) as expressed in thefollowing equation:

Lleakage = W 2primary · μ0 ·

P

H· Δ1−2 (8)

whereLleakage value of leakage inductance;μ0 permeability of air;P length per each turn of primary winding;H total winding length of primary winding;Δ1−2 distance between primary and secondary windings.

For the leakage inductance to reflect the designed resonantinductance value, only the value of Δ1−2 can be adjusted be-cause P,H , and Wprimary are fixed and depend on the core andpower ratings.

Therefore, a specially designed bobbin that provides a properdistance for leakage inductance and isolation between primaryand secondary windings is required. Fig. 6. shows an outline ofthe designed transformer with the bobbin.

The implemented parameters are summarized in Table II. Be-cause of a small difference between the design and measuredvalues of the leakage inductance, the resonant capacitance valuewas increased from 1.2 to 1.5 μF to provide the same resonantfrequency because the same ratio of each frequency generatesthe same output current. In this paper, the RMS current of theresonant capacitor was calculated to be about 80 A. To achievethe RMS current rating, ten 0.15 μF capacitors with 11-A max-imum RMS current are used in parallel considering the RMScurrent margin.

TABLE IISUMMARY OF DESIGNED PARAMETERS

Fig. 7. Picture of the developed capacitor charger.

Fig. 8. Switching signal waveforms with 25-kHz switching frequency of theproposed capacitor charger (20 V/div).

III. EXPERIMENTAL RESULTS

Based on the design parameters and by considering the insu-lation and cooling of the high-voltage components, the proposedhigh-voltage capacitor charger was developed by classifying theinput rectifier and three-phase resonant inverter as the first blockand the high-voltage transformer and rectifier parts as the sec-ond block. An outline of each block is shown in Fig. 7 and 8shows the proper switching signal waveforms of the proposedcapacitor charger.

A. Test Results for Different Load Conditions

The developed charger was tested with four types of capac-itors to ensure reliable operation regardless of the load capaci-tance and charging voltage values. Fig. 9 shows the waveform of

AHN et al.: DESIGN AND TEST OF A 35-kJ/s HIGH-VOLTAGE CAPACITOR CHARGER 4045

Fig. 9. 1.250-mF capacitor charging waveform (2 kV/div.).

Fig. 10. 0.47-μF capacitor charging waveform (5 kV/div., 100 A/div.).

Fig. 11. 1.236-mF capacitor charging waveform (1 kV/div).

the 1.25-mF capacitor with the charging voltage ranging from 0to 11.5 kV. The charging time was measured to be 4.7 s, and thisresult approximately matches the calculated result of 5 s. Themaximum charging voltage of this test is 25 kV because of theseries connection of all the high-voltage rectifiers. For the sameoutput voltage and current conditions, the 23-kV charging testwith a 0.47-μF capacitor was performed. The result is shownin the waveforms in Fig. 10, including the resonant current ofone inverter and the charging voltage. The measured chargingtime is 3.5 ms, and this almost matches the calculated value of3.8 ms. Fig. 11 shows the charging test result with a 12.5-kVmaximum charging voltage condition that was generated by us-ing three series and two parallel connections of the high-voltagerectifier. As depicted in Fig. 11, the 1.236-mF capacitor charged

Fig. 12. Resonant waveforms of 11-μF capacitor charging test according tothe charging voltage change (200 V/div., 100 A/div.).

up to 7 kV within 1.3 s. When this is compared with the twoaforementioned results, it is clear that the value of the charg-ing current is two times greater than that of the series-connectedoutput rectifier circuit. Finally, a 11-μF capacitor charging test isconducted to check the resonant current waveforms correspond-ing to the charging voltage, and the results are shown in Fig. 12.These results verify that the proposed configuration of the high-voltage rectifier circuit allows for a flexible output voltage andcurrent with a fixed maximum charging power. Hence, it canbe used for various kinds of low-to-high voltage pulsed-powerapplications.

The developed capacitor charger was also tested under theshorting and open conditions of the charger output terminal toassess its reliability. Fig. 13 shows the waveforms of the outputcurrent under a short-circuit condition. Because of the currentsource characteristics of the applied topology, the charger outputcurrent cannot increase to the specific level; this may break therectifier diode, even if it provides a relatively high-peak currentcompared with a normal operation. The charger was stoppedby the over-charging time protection. The open-circuit test wasperformed with just the high-voltage probe connected to thecharger output terminal, and the measured voltage waveform isdepicted in Fig. 14.

The output voltage increased instantaneously because of thehigh value of load resistance. Then, the constant current flowsthrough the high value of the sensing resistor and output filtercapacitor. However, the operation of the voltage control clampedthe output voltage to the reference voltage within few tens of

4046 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 29, NO. 8, AUGUST 2014

Fig. 13. Short-circuit operation waveform (10 A/div.).

Fig. 14. Open-circuit operation waveform (2 kV/div.).

microseconds. This test was performed many times and thecharger worked properly without any problems.

B. Test Results for the Practical Pulsed-Power Application

In order to guarantee the reliable operation of the developedcharger in practical applications, an experimental model wascreated to simulate malfunctions such as shorts, opening, andmisfiring during charging. The experimental results are givencase by case as follows.

1) Short-Circuit Test During Charging Operation: The short-circuit test was performed by closing switch 2 in Fig. 15(a)during the charging procedure to ensure that the charger is notdamaged in case of a short circuit. Fig. 15(b) shows the testresults. Switch 2 closed at a charging voltage of 6.2 kV, and aconstant supply of current was maintained after the short withoutany damage to the charger.

2) Open-Circuit Test During the Charging Operation: Theopen-circuit test during charging represented the most diffi-cult operating condition in the case of a constant-current typecapacitor charger. It involved opening switch 1, as shown inFig. 16(a), during the charging procedure. However, because of

Fig. 15. Short-circuit test during the charging procedure (500 ms/div). (a) Testprocedure. (b) Test waveform.

Fig. 16. Open-circuit test during the charging procedure. (a) Test procedure.(b) Test waveform.

higher charging currents, even when mechanical switching wasrevealed, the charging current kept flowing through the coronaarc path, and there was no interruption in the charging sequence.After the charging procedure was completed, the charging cur-rent stopped through the opening of switch 1, and there wasstill no damage to the capacitor charger. From this test result,our developed capacitor charger is shown to be very reliable inits operation, even though it was designed based on a currentsource converter.

3) Misfiring During the Charging Operation: The final testwas performed by triggering switch 4 during the charging pro-cedure. If the main triggering switch is fired during the chargingprocedure, the capacitor can be charged with a negative voltageof up to a maximum of 30% of the positive charging voltage.Because the output side of the capacitor charger is connectedto the rectifying diodes, this negative voltage can damage theinternal diodes of the charger through excessive freewheeling

AHN et al.: DESIGN AND TEST OF A 35-kJ/s HIGH-VOLTAGE CAPACITOR CHARGER 4047

Fig. 17. Misfiring test during the charging procedure. (a) Test procedure.(b) Test waveform. (c) Magnified current waveform (5 A/div).

currents. For a fault-free operation, a ballast resistor R1 of 40 Ωis connected at the output side of the capacitor charger.

After setting a charging voltage of 8 kV as the reference, themain SCR was triggered at a charging voltage of 7 kV duringcapacitor charging. All the capacitor energy was dumped intothe load and a negative voltage was shown in the capacitor.

As a result the maximum output current reached up to 27 A,but no damage was found at the output diodes because of theballast resistor, as shown in Fig. 17. The experimental resultsclearly show that the proposed capacitor charger can be usedfor wide-ranging pulsed-power applications that require reli-able operation even when the load operates under undesirableconditions.

IV. CONCLUSION

In this study, a high-voltage capacitor charger for pulsed-power applications was designed and tested. On the basis ofthe characteristics of the series resonant converter in the DCMoperation, a three-phase delta-connected inverter scheme wasproposed with six voltage-doubled rectifiers; this allowed fora flexible output voltage and current with fixed output powerat maximum ratings. The detailed design procedure of thethree-phase resonant converter, including the determination ofresonant tank parameters and the design of the high-voltagetransformer, was provided with respect to the input and out-put specifications. On the basis of the design parameters, the

proposed capacitor charger was developed to have a compactdesign. Moreover, the charger was tested with different capaci-tors for normal charging operation, and the measured chargingtime proved that the developed charger attained the given outputspecifications with high efficiency. Furthermore, two types ofoutput rectifier connections were tested to verify the flexibil-ity of the output current and voltage of the proposed capaci-tor charger. In addition, short-circuit and open-circuit operationtests were performed, and the developed capacitor charger wasshown to be undamaged. Finally, to ensure the reliable operationof the capacitor charger for practical pulsed-power systems, amalfunction test simulation model was generated. Three kindsof tests were performed for the short circuit, open circuit, andmisfiring during charging cases.

From the experimental results, the developed capacitorcharger shows a very stable and reliable operation, so it is con-firmed to be a good candidate for different kinds of pulsed-powerapplications.

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Suk-Ho Ahn (M’11) received the B.S. degree in elec-trical engineering from Incheon National University,Incheon, Korea, in 2009, and is currently workingtoward the M.S. and Ph.D. degrees at the Univer-sity of Science and Technology in Daejeon, Daejeon,Korea.

His research interests include the soft switchedresonant converter applications and battery chargersystems.

Hong-Je Ryoo received the B.S., M.S., and Ph.D. de-grees in electrical engineering from SungKyunkwanUniversity, Seoul, Korea, in 1991, 1995, and 2001,respectively.

From 2004 to 2005, he was with WEMPEC atthe University of Wisconsin-Madison, as a Visit-ing Scholar for his postdoctoral study. Since 1996,he has been with the Korea Electrotechnology Re-search Institute in Changwon, Korea. He is currentlya Principal Research Engineer in the Electric Propul-sion Research Division and a Leader of the Pulsed

Power World Class Laboratory at Korea Electro-tTechnology Research In-stitute, Changwon, Korea. Also, he has been a Professor in the Departmentof Energy Conversion Technology, University of Science and Technology,Daejeon, Korea, since 2005. His current research interests include pulsed-power systems and their applications, as well as high-power and high-voltageconversions.

Dr. Ryoo is a Member of the Korean Institute of Power Electronics, and theKorean Institute of Electrical Engineers.

Ji-woong Gong received the B.S. degree in electri-cal engineering from Chonnam National University,Gwangju, Korea, in 2012, and is currently workingtoward the M.S. degree at the University of Scienceand Technology in Daejeon, Daejeon, Korea.

His research interests include the soft switched res-onant converter applications and high-voltage pulsed-power supply systems.

Sung-Roc Jang was born in Daegu, Korea, in 1983.He received the B.S. degree from Kyungpook Na-tional University, Daegu, Korea, in 2008, and the M.S.and Ph.D. degrees in electronic engineering fromthe University of Science and Technology, Daejeon,Korea, in 2011.

Since 2011, he has been a Senior Researcherof the Electric Propulsion Research Center, Ko-rea Electrotechnology Research Institute, Changwon,Korea. His current research interests include high-voltage resonant converters and solid-state pulsed-

power modulators and their industrial applications.Dr. Jang received the Young Scientist Award at third Euro-Asian Pulsed-

Power Conference in 2010, and the IEEE Nuclear Plasma Science Society BestStudent Paper Award at IEEE International Pulsed-Power Conference in 2011.