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D5.1 – Report on preliminary THz RF-Frontend and Antenna, Phased array beamforming, baseband algorithms and optical RF-frontend ready for implementation in off-line tests
TERRANOVA Project Page 1 of 92
This project has received funding from Horizon 2020, European Union’s
Framework Programme for Research and Innovation, under grant
agreement No. 761794
Deliverable D5.1 Report on preliminary THz RF-Frontend and
Antenna, Phased array beamforming,
baseband algorithms and optical RF-frontend
ready for implementation in off-line tests Work Package 5 – THz System Technology
TERRANOVA Project
Grant Agreement No. 761794
Call: H2020-ICT-2016-2
Topic: ICT-09-2017 - Networking research beyond 5G
Start date of the project: 1 July 2017
Duration of the project: 30 months
D5.1 – Report on preliminary THz RF-Frontend and Antenna, Phased array beamforming, baseband algorithms and optical RF-frontend ready for implementation in off-line tests
TERRANOVA Project Page 2 of 92
Disclaimer This document contains material, which is the copyright of certain TERRANOVA contractors,
and may not be reproduced or copied without permission. All TERRANOVA consortium
partners have agreed to the full publication of this document. The commercial use of any
information contained in this document may require a license from the proprietor of that
information. The reproduction of this document or of parts of it requires an agreement with
the proprietor of that information. The document must be referenced if used in a
publication.
The TERRANOVA consortium consists of the following partners.
No. Name Short Name Country 1
(Coordinator)
University of Piraeus Research Center UPRC Greece
2 Fraunhofer Gesellschaft (FhG-HHI & FhG-IAF) FhG Germany 3 Intracom Telecom ICOM Greece 4 University of Oulu UOULU Finland 5 JCP-Connect JCP-C France 6 Altice Labs ALB Portugal 7 PICAdvanced PIC Portugal
D5.1 – Report on preliminary THz RF-Frontend and Antenna, Phased array beamforming, baseband algorithms and optical RF-frontend ready for implementation in off-line tests
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Document Information
Project short name and number TERRANOVA (761794) Work package WP5 Number D5.1 Title Report on preliminary THz RF-Frontend and
Antenna, Phased array beamforming,
baseband algorithms and optical RF-frontend
ready for implementation in off-line tests
Version V1.0 Responsible unit FhG Involved units FhG, PIC, ICOM, UOULU, UPRC, ALB Type1 R Dissemination level2 PU Contractual date of delivery 30.06.2018 Last update 30.06.2018
1 Types. R: Document, report (excluding the periodic and final reports); DEM: Demonstrator, pilot,
prototype, plan designs; DEC: Websites, patents filing, press & media actions, videos, etc.; OTHER: Software, technical diagram, etc. 2 Dissemination levels. PU: Public, fully open, e.g. web; CO: Confidential, restricted under conditions
set out in Model Grant Agreement; CI: Classified, information as referred to in Commission Decision 2001/844/EC.
D5.1 – Report on preliminary THz RF-Frontend and Antenna, Phased array beamforming, baseband algorithms and optical RF-frontend ready for implementation in off-line tests
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Document History
Version Date Status Authors, Reviewers Description
v0.1 20.03.2018 Draft Thomas Merkle (FhG-IAF) Initial definition of a
document structure
v0.2 27.03.2018 Draft Thomas Merkle (FhG-IAF) First version of Section 1
and Executive Summary
v0.3 04.03.2018 Draft Thomas Merkle (FhG-IAF) Revision of structure
based on input of all
partners
v0.4 13.06.2018 Draft Robert Elschner (FhG-HHI)
Georgia Ntouni (ICOM)
Francisco Rodrigues (PIC)
Draft of Section 3 (PIC),
Section 5 (HHI), and
Section 6 (ICOM)
v0.5 19.06.2018 Draft Dimitrios Kritharidis (ICOM)
Alexandros Katsiotis (ICOM)
Francisco Rodrigues (PIC)
A.-A. A. Boulogeorgos (UPRC)
Janne Lehtomäki (UOULU
Input to Section 2
(ICOM), Section 3 (PIC),
Section 4 (UPRC) and
Section 6 (UOULU)
v0.6 22.06.2018 Draft Georgia Ntouni (ICOM) Input to Section 6
v0.7 26.06.2018 Draft Thomas Merkle (FhG-IAF)
Robert Elschner (FhG-HHI)
Input to Section 2, 3, 4
and 5
v0.8 28.06.2018 Draft A.-A. A. Boulogeorgos (UPRC)
Georgia Ntouni (ICOM)
Janne Lehtomäki (UOULU)
Review / proofreading of
Section 1, 6
v0.9 29.06.2018 Draft Dimitrios Kritharidis (ICOM)
Georgia Ntouni (ICOM)
Robert Elschner (FhG-HHI)
Thomas Merkle (FhG-IAF)
Review / proofreading of
all sections, final editing
v1.0 30.06.2018 Final Thomas Merkle (FhG-IAF)
Angeliki Alexiou (UPRC)
Revision of all sections
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Acronyms and Abbreviations
Acronym/Abbreviation Description 5G Fifth Generation
A ACO Analog Coherent Optics ADC Analog-to-Digital Converter AFC Automatic Frequency Correction AFE Analogue FrontEnd AGC Automatic Gain Control AiP Antenna-in-Package AM Amplitude Modulation AMC Adaptive Modulation and Coding AP Access Point ASIC Application-Specific Integrated Circuit ATDE Adaptive Time Domain Equalizer AWG Arbitrary Waveform Generator AWGN Additive White Gaussian Noise AWV Antenna Weight Vector
B BB BaseBand BC Beam Combining BEOL Back End of Line BER Bit Error Rate BF BeamForming BPSK Binary Phase Shift Keying BS Base Station
C CAUI 100 gigabit Attachment Unit Interface CDR Clock and Data Recovery CFP C-Form Factor Pluggable CMOS Complementary Metal–Oxide–Semiconductor CoMP Coordination Multi-Point COTS Commercial Off-The-Shelf / Components Off-The-Shelf CPR Carrier Phase Recovery CRC Cyclic Redundancy Code CSMA/CA Carrier Sense Multiple Access with Collision Avoidance CW Continuous Wave DAC Digital to Analog Converter
D DC Direct Current DCH Data CHannel DDC Digital Down Conversion DEMUX DE-MUltipleXer DL DownLink DMT Discrete Multi-Tone DoA Direction of Arrival DoF Degree of Freedom DP-IQ Dual Polarization In-phase and Quadrature
D5.1 – Report on preliminary THz RF-Frontend and Antenna, Phased array beamforming, baseband algorithms and optical RF-frontend ready for implementation in off-line tests
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DPD Digital PreDistortion DSB Double Sideband DSP Digital Signal Processing DUC Digital Up Conversion DWDM Dense Wavelength Division Multiplexing
E E2E End-to-End EC European Commission E/O Electrical-Optical ETSI European Telecommunications Standards Institute eWLB embedded Wafer Level Ball grid array
F FEC Forward Error Correction FD Full Duplex FDD Frequency Division Duplexing FDMA Frequency Division Multiple Access FIFO First In First Out FM Frequency Modulation FPGA Field-Programmable Gate Array FEOL Front End of Line FSO Free-Space Optics FSPL Free Space Path Loss FTTH Fiber To The Home FWA Fixed Wireless Access
G GaAs Gallium Arsenide
H HEMT High Electron Mobility Transistor
I I/Q In-phase and Quadrature I2C Inter-Integrated Circuit IEEE Institute of Electrical and Electronics Engineers IF Intermediate Frequency IM/DD Intensity Modulation/Direct Detection ISI InterSymbol Interference ISM Industrial Scientific and Medical band ITU International Telecommunication Union ITU-R Radiocommunication sector of the International
Telecommunication Union IQD Indoor Quasi Directional
K KPI Key Performance Indicator
L LO Local Oscillator LoS Line of Sight LNA Low Noise Amplifier
M MAC Medium Access Control mHEMT Metamorphic High Electron Mobility Transistor MIMO Multiple Input Multiple Output MMIC Monolithic Microwave Integrated Circuit mmWave Millimeter Wave MUE Mobile User Equipment
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MUX MUltipleXer MZI Mach-Zehnder Interferometer
N NGPON2 Next-Generation Passive Optical Network 2 nLoS Non-Line Of Sight NR New Radio NRZ Non-Return to Zero
O OFDM Orthogonal Frequency Division Modulation OIF Optical Internetworking Forum OLT Optical Line Terminal ONUs Optical Network Units OOK On-Off Keying
P P2MP Point-to-Multi-Point P2P Point-to-Point PA Power Amplifier PAM Pulse Amplitude Modulation PCB Printed Circuit Board PDM Polarization-Division Multiplexing PDM-QAM Polarization Multiplexed Quadrature Amplitude
Modulation PER Packet Error Rate PHY PHYsical PIC Photonic Integrated Circuit PLL Phased Locked Loop PONs Passive Optical Networks PSP Pulse Shaping Filter PtMP Point-to-Multi-Point
Q QAM Quadrature Amplitude Modulation QoE Quality of Experience QoS Quality-of-Service QSFP Quad Small Form-Factor Pluggable
R RA Random Access RAT Radio Access Technology RAU Remote Antenna Unit RF Radio Frequency RoF Radio over Fiber RRM Radio Resource Management RSRP Reference Signal Received Power RSSI Received Signal Strength Indicator Rx Receiver
S SD-FEC Soft-Decision Forward-Error Correction SDM Space Division Multiplexing SDMA Space Division Multiple Access SDN Software Define Network SFF Small Form Factor SFP Small Form-Factor Pluggable SiGe Silicon-Germanium SISO Single Input Single Output
D5.1 – Report on preliminary THz RF-Frontend and Antenna, Phased array beamforming, baseband algorithms and optical RF-frontend ready for implementation in off-line tests
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SLS Sector Level Sweep SM Spatial Multiplexing SMF Single Mode Fiber SNR Signal to Noise Ratio SOTA State Of The Art SPI Serial Parallel Interface SRC Sample Rate Conversion SSB Single-SideBand SSW Sector SWeep SSW-FBCK Sector SWeep FeedBaCK STM-1 Synchronous Transport Module, level 1 STS Symbol Timing Synchronization
T TDD Time Division Duplexing TDM Time Division Multiplexing TDMA Time Division Multiple Access TERRANOVA Terabit/s Wireless Connectivity by Terahertz innovative
technologies to deliver Optical Network Quality of
Experience in Systems beyond 5G
THz Terahertz TIA TransImpedance Amplifier TWDM Time and Wavelength Division Multiplexed Tx Transmitter
U UL Uplink UE User Equipment
V VCO Voltage Controlled Oscillator VGA Variable Gain Amplifier VLC Visible Light Communication
W WLAN Wireless Local Area Network WDM Wavelength Division Multiplexing WiFi Wireless Fidelity WLBGA Wafer Level Ball Grid Array
X XG-PON 10 Gbit/s Passive Optical Network XPIC Cross Polarization Interference Cancellation
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Contents
1. Introduction ........................................................................................................................... 16
1.1 Scope ....................................................................................................................... 17
1.2 Structure .................................................................................................................. 18
1.3 References ............................................................................................................... 18
2. System Block Diagram REVIEW ............................................................................................. 19
2.1 Review of TERRANOVA Architecture Candidates .................................................... 19
2.2 Generic Architecture and Hardware Overview ....................................................... 19
2.3 Beamforming Demonstrator Block Diagram ........................................................... 20
2.4 References ............................................................................................................... 21
3. Optical Link and TERRANOVA Media Converter Design ........................................................ 22
3.1 Standardized Optical Transceivers .......................................................................... 22
3.1.1 IM/DD Transceivers ......................................................................................... 22
3.1.2 Coherent Transceivers ..................................................................................... 23
3.2 TERRANOVA Media Converter Design ..................................................................... 24
3.2.1 Proposal 1 – IM/DD using PICadvanced NG-PON2 technology ....................... 24
3.2.2 Proposal 2 – IM/DD using COTS 100G ............................................................. 26
3.2.3 Proposal 3 - IM/DD transceivers based on amplitude modulation ................. 26
3.2.4 Proposal 4 – Optical Coherent transmission ................................................... 28
3.2.5 Conclusion ....................................................................................................... 30
3.3 Concepts for Future Media Converter Integration ................................................. 30
3.4 Conclusions and Outlook ......................................................................................... 35
3.5 References ............................................................................................................... 35
4. RF Frontend and Antenna Prototypes ................................................................................... 36
4.1 THz Frontends for Point-to-Point Applications ....................................................... 36
4.1.1 Duplexing Techniques...................................................................................... 37
4.1.2 Transceiver Correction Schemes and Synchronization ................................... 38
4.1.3 Transceiver Architectures and Project Development Plan .............................. 40
4.2 New Process Technologies for III-V based MMICs .................................................. 41
4.2.1 Technology Overview ...................................................................................... 41
4.2.2 Motivation and Status of BEOL Development ................................................. 43
4.3 New MMIC Frontend Building Blocks ...................................................................... 46
4.3.1 Component Candidates for the TERRANOVA Media Converter...................... 47
4.3.2 Circuit Components for 220-320 GHz Transceivers ......................................... 49
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4.4 Conclusions and Outlook ......................................................................................... 52
4.5 References ............................................................................................................... 53
5. Baseband Digital Signal Processing for THz Systems ............................................................. 58
5.1 THz P2P transmission experiments ......................................................................... 58
5.1.1 Experimental setup .......................................................................................... 59
5.1.2 Used Digital Signal Processing ......................................................................... 60
5.2 First THz System Measurement Results .................................................................. 61
5.3 Conclusions and Outlook ......................................................................................... 67
5.4 References ............................................................................................................... 67
6. Phased Array Beamforming ................................................................................................... 68
6.1 State-of-the-Art in Phased Array Beamforming techniques ................................... 68
6.2 Mathematical Model of Phased Array Architectures .............................................. 69
6.3 Comparison of Beamforming Techniques ............................................................... 70
6.3.1 Conventional Beamforming ............................................................................. 70
6.3.2 Tapered Beamforming ..................................................................................... 71
6.3.3 Null-Steering Beamforming ............................................................................. 74
6.3.4 Adaptive beamforming .................................................................................... 74
6.3.4.1 Minimum Variance Distortionless Response (MVDR) Beamformer ............ 76
6.3.4.2 Linearly Constrained Minimum Bariance (LCMV) Beamformer .................. 76
6.4 Beamforming Implementation Issues of Demonstrators ........................................ 80
6.4.1 Available Phase Shifter Resolution for Analogue Beamforming ..................... 80
6.4.1 Differential Phase Noise in THz Phased Array Systems ................................... 82
6.4.2 Four Element Horn Antenna Array .................................................................. 85
6.4.3 Beam Search and Alignment ........................................................................... 87
6.5 Calibration Techniques for Phased Array Antennas ................................................ 88
6.6 Conclusions and Outlook ......................................................................................... 89
6.7 References ............................................................................................................... 90
7. Conclusions ............................................................................................................................ 92
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List of Figures
Figure 1: WP5 project plan and connection with WP2 and WP6. ........................................... 17
Figure 2: General overview of the components under investigation within WP5. ................. 19
Figure 3: Generic view on the TERRANOVA media converter. ................................................ 20
Figure 4: LO generation option for the digital beamformer. .................................................. 21
Figure 5: Typical IM/DD optical transceiver pluggable in XFP form factor. ............................ 22
Figure 6: Basic block diagram of an XFP IM/DD transceiver module. ..................................... 23
Figure 7: Coherent receiver architecture. ............................................................................... 24
Figure 8: Possible application of the IM/DD transceivers using PICadvanced
NG-PON2 technology. ............................................................................................................. 24
Figure 9: IM/DD solution based on PICadvanced NG-PON2 transceivers. .............................. 25
Figure 10: IM/DD solution based on CFP transceivers. ........................................................... 26
Figure 11: IM/DD solution based on amplitude linear transceivers. ...................................... 27
Figure 12: Possible application of the coherent solution. ....................................................... 28
Figure 13: Coherent solution based on single polarization THz radio interface. .................... 29
Figure 14: Coherent solution based on the dual-polarization THz radio interface with
optional DSP at the radio front end. ....................................................................................... 30
Figure 15: Functional block diagram of the media converter for coherent optical
transmission. ........................................................................................................................... 31
Figure 16: Illustration of a typical ACO module and the cage system on the host board [3-9] .
................................................................................................................................................. 31
Figure 17: Insertion loss of the board-to-board connector on the host board, taken from [3-
8]. ............................................................................................................................................. 32
Figure 18: Manufactured baseband amplifier test module for risk and problem
identification. .......................................................................................................................... 32
Figure 19: Measured transmission line loss using different printed circuit board materials. 33
Figure 20: Calculated frequency limitations of the centred striplines as a function of the
waveguide thickness. .............................................................................................................. 34
Figure 21: State-of-the-art of hybrid integrated optical transponder modules. .................... 34
Figure 22: Channel allocation plan of IEEE 802.15.3d-2017 standard. ................................... 37
Figure 23: Overview of design activities of M1-M12. ............................................................. 41
Figure 24: Comparison of different transistor technology options for THz frontend design. 42
Figure 25: Comparison of various BEOL processes, for different transistor technologies ...... 43
Figure 26: BEOL development and tests in TERRANOVA, (1) base process, (2) development
run 1, (3), final solution for final transceiver integration. ....................................................... 44
Figure 27: Left: SEM picture of manufactured mixer IP core component for 220 to 300 GHz
with the 3LPP process (before TFMS end passivation), right: corresponding optical
photograph. ............................................................................................................................. 45
Figure 28: Left: Layout and photograph of manufactured transmission lines for process
testing and monitoring, right: measured attenuation of a microstrip line of 10 µm width. .. 45
Figure 29: Integration density of different BEOL for broadband RF applications (to scale), see
also [4-65] for the 65nm CMOS amplifier. .............................................................................. 46
Figure 30: Chip photograph of the fabricated Kukielka amplifier, schematic representation,
and summary of the measured key performance parameters. .............................................. 47
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Figure 31: On-wafer measured performance of the manufactured Kukielka amplifier, (a) S-
parameters and noise figure, (b) output power compression characteristics at 10 GHz. ...... 48
Figure 32: Designed test package and photograph of the manufactured 4-channel Kukielka
amplifier chip. .......................................................................................................................... 49
Figure 33: Measured frequency response, left: magnitude in dB, right phase in deg for the 4-
channel Kukielka amplifier. ..................................................................................................... 49
Figure 34: First IP core library for the DL, MMIC for generating an LO signal that can be tuned
varied from 200 to 260 GHz. ................................................................................................... 50
Figure 35: On-wafer measured output power at 240 GHz for the LO chip ............................. 50
Figure 36: Chip photograph of manufacture 120 GHz driver amplifier and measured RF
performance (S-parameters and output power characteristics). ........................................... 51
Figure 37: Chip photograph of the fabricated broadband LO multiplier for selecting different
Rx/Tx channels from 220 to 290 GHz and measured output power characteristic at a fixed
input power level. .................................................................................................................... 51
Figure 38: Chip photograph of the fabricated 220 -260 GHz power amplifier for the DL
frequency band, left 3LPP design, right 4L design with MET4 layers (before MET4 processing)
. ................................................................................................................................................ 52
Figure 39: On-wafer measured RF performance of the 220 -260 GHz power amplifier, left S-
parameters, right output power at 240 GHz. .......................................................................... 52
Figure 40: Experimental setup for tests of THz P2P link ......................................................... 58
Figure 41: Photograph of first lab setup at Fraunhofer HHI.................................................... 58
Figure 42: Block diagram of the digital signal processing ....................................................... 61
Figure 43: BER vs. Digital DAC Amplitude (I and Q component) for 16-GBd using 16QAM
(without pre-emphasis, at maximum Tx output power, optimal Ry I/Q orientation, at a
carrier frequency of 306.36 GHz) ............................................................................................ 62
Figure 44: BER vs. Digital DAC Amplitude (I and Q component) for 16-GBd 16QAM (with pre-
emphasis, maximum Tx output power, optimal Rx I/Q orientation, 306.36 GHz carrier
frequency) ............................................................................................................................... 62
Figure 45: Received constellations under best case conditions. (a) 16 GBd 16QAM without
pre-emphasis @ BER = 1·10-3. (b) 16 GBd 16QAM with pre-emphasis @ BER = 6.5·10-4. ...... 63
Figure 46: BER vs. Carrier Frequency for 16 GBd 16QAM (no pre-emphasis, maximum Tx
output power, optimal Rx I/Q orientation, optimal digital DAC amplitude) ........................... 63
Figure 47: BER vs. Rx I/Q orientation for 16-GBd 16QAM (different carrier frequencies, no
pre-emphasis, maximum Tx output power, optimal digital DAC amplitudes) ........................ 64
Figure 48: Discussion of Rx I/Q orientation. (a) Schematic of Rx aligned QAM constellation
(orientation angle = 0°). (b) Schematic of Rx misaligned QAM constellation (orientation angle
= 45°). (c) Measured Rx constellation (only I component at Tx) at an orientation angle giving
the best case BER. (d) Measured Rx constellation (only I component at Tx) at an orientation
angle giving the worst case BER. ............................................................................................. 64
Figure 49: BER vs. THz attenuation for 16-GBd 16QAM (306.36 GHz carrier frequency, with
pre-emphasis, optimal digital DAC amplitudes, optimal Rx I/Q orientation) ......................... 65
Figure 50: BER vs. RX frequency offset for 16-GBd 16QAM (306.36 GHz carrier frequency,
with pre-emphasis, maximum TX output power, optimal digital DAC amplitudes) ............... 66
Figure 51: BER vs. Digital DAC Amplitude (I and Q component) for 32-GBd 16QAM (with pre-
emphasis, maximum Tx output power, optimal Rx I/Q orientation, 306.36 GHz carrier
frequency) ............................................................................................................................... 66
Figure 52: Received constellation under best case conditions for 32-GBd 16QAM with pre-
emphasis @ BER = 1.1·10-2. ..................................................................................................... 67
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Figure 53: Plane wave impinging on a ULA ............................................................................. 70
Figure 54: Beampatterns for conventional beamforming ....................................................... 71
Figure 55: Beampatterns for tapered beamforming with binomial distribution and 4 AEs ... 72
Figure 56: Beampatterns for tapered beamforming with various windows. The ULA consists
of 16 elements and its steering direction is at in (d) is the number of nearly
constant-level sidelobes adjacent to the mainlobe ................................................................ 73
Figure 57: Beampatterns for Null-steering beamforming with 16 AEs ................................... 75
Figure 58: Beampatterns for Null-steering beamforming with 4 AEs and ............ 76
Figure 59: Magnitude of the beamformers’ output signals for a ULA with 4 AEs................... 78
Figure 60: Power pattern of beamformers for a ULA with 4 AEs ............................................ 78
Figure 61: Power pattern (rectangular) of beamformers for a ULA with 4 AEs ...................... 79
Figure 62: Magnitude of the beamformers’ output signals for a ULA with 16 AEs ................ 79
Figure 63: Power pattern of beamformers for a ULA with 16 AEs .......................................... 80
Figure 64: Beampatterns for conventional beamforming with and without quantization
assuming 4 AEs ........................................................................................................................ 81
Figure 65: Beampatterns for conventional beamforming with and without quantization
assuming 16 AEs ...................................................................................................................... 82
Figure 66: Functional block diagram of the 4 channel LO beamformer with DDS based phase
shifting. The synchronous DAC output is achieved by a master trigger (omitted in this
drawing for clarity), [6-14]. ..................................................................................................... 83
Figure 67: Phase noise of the individual channels of the LO beamformer, measured at the
output carrier frequency of 8.333 GHz. In comparison, the phase noise of a commercial
frequency synthesizer is shown. Carrier power Pc = -2 dBm in all cases, [6-14]. ................... 83
Figure 68: Measured cumulative constellation diagrams and EVM of a single receive channel
at 4 Gbaud for increasing QAM modulation depths. 100 constellation diagrams of 4096
symbols were accumulated in each plot, [6-14]. .................................................................... 84
Figure 69: Measured cumulative constellation diagrams and EVM of a single receive channel
at 4 Gbaud for a 16-QAM modulation format, after initial calibration by null-steering. 100
constellation diagrams of 4096 symbols were accumulated in each plot, [6-14]. .................. 84
Figure 70: Horn antenna element ........................................................................................... 85
Figure 71: Directivity pattern of the horn antenna element ................................................... 85
Figure 72: Directivity pattern of the four elements of the ULA .............................................. 86
Figure 73: Directivity radiation pattern of a ULA with 4 AEs pointing at ................. 86
Figure 74: Directivity radiation patterns of a ULA with 4 AEs pointing at ............. 87
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List of Tables
Table 1: Main experimental parameters ................................................................................. 59
Table 2: Relation of clock and carrier frequencies used in the experiment ............................ 60
Table 3: Parameters for digital signal processing blocks ......................................................... 61
Table 4: SINR for various beamformers .................................................................................. 77
Table 5: Output noise power for various beamformers.......................................................... 77
Table 6: Interference power for various beamformers ........................................................... 77
Table 7: Horn antenna element dimensions ........................................................................... 85
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Executive Summary
The deliverable “D5.1 Report on preliminary THz RF-Frontend and Antenna, Phased array
beamforming, baseband algorithms and optical RF-frontend ready for implementation in off-
line tests” is the first of a series of three deliverables that accompany the research and
development work on the “THz System Technologies” of TERRANOVA. There are two overall
objectives in work package WP5. Firstly, research towards new software and hardware
components that address the peculiarities and challenges of hybrid optical-THz
communication systems. Secondly, practically implementing and providing components for
the integration in the two system demonstrators of WP6 (“THz Demonstrator
Implementation and Validation”). With the first demonstrator, beamforming algorithms and
concepts will be tested and validated. With the second demonstrator, the capacity bounds
of a hybrid optical-THz communication link will be experimentally explored. Different use-
case scenarios and suitable system architecture candidates for each scenario were identified
as part of deliverable D2.1 (“TERRANOVA System Requirements” and D2.2 (“TERRANOVA
System Architecture”). The system component design and implementation may reveal
challenges that were not anticipated during the early system concept phase of the project.
There are also open questions on what performance the THz system components can
achieve today and in future.
The design of preliminary THz system components means in the context of D5.1 that designs
“ready for implementation in off-line tests” are presented or in other words the technology
concept of the components is formulated. At this stage, the major hardware components of
interest are the hybrid optical-electrical interface, the THz Rx/Tx frontend at chip level and
the antenna. They define all together the optical-THz frontend for which also an integration
platform and future packaging strategies will be outlined in the course of the project. The
major software components are the baseband modem and beamforming algorithms. The
formulation of the hardware concepts is documented by a first set of prototype designs
“ready for implementation” which means ready for fabrication. Offline tests mean that the
algorithms can be tested in a simulation environment with artificial data or experimentally
recorded datasets. The successive deliverables D5.2 and D5.3 will provide proof of the
component concepts in a lab environment. This means that the functioning of the hardware
components will be experimentally tested and software components will be ready for the
implementation in a real-time environment. In WP6, the THz system components of WP5
will be integrated for demonstration of the TERRANOVA system concepts and the validation
of the technologies.
The main objectives of the deliverable are to progress in:
Development of a high level hybrid optical-THz wireless modem architecture, and
identification of the required key algorithms for point-to-point transmission;
Identification of concepts for Rx/Tx frontends and the design of integrated circuit
prototypes of the top candidates;
Methods for beamforming and calibration of THz Rx/Tx antenna arrays for future
use in line-of-sight beamforming schemes with pre-defined codebooks, and
Top-down design of the TERRANOVA media converter for the hybrid optical-THz link,
and identification of electro-optical THz frontend integration and packaging
solutions.
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1. INTRODUCTION
One of the key enabling technologies of TERRANOVA is embedding terahertz (THz) wireless
links into fibre optic communication networks. Several use case scenarios and system
architectures were previously identified in the deliverables of D2.1 and D2.2 of this project
[1-1], [1-2]. They all promise to extend the quality of service and experience (QoS and QoE)
from optical systems to the THz wireless domain, which also implies date rates in the Tbit/s
regime. The currently available wireless baseband modem technologies do not, however,
meet the requirements of such systems. Instead of wireless baseband modems, the
TERRNAOVA solutions use COTS (commercial-off-the-shelf) optical transceivers. Provided
that the THz wireless system components can be designed to comply with the optical
transceivers and optical links, the enormous technical progress of the fibre optical industry
can be exploited, e.g. powerful ASICs (application-specific integrated circuits) for DSP (digital
signal processing) and specific ADC/DACs having large analogue bandwidths, which
nowadays are able to support data rates of 100/200 and 400 Gbit/s. Work package (WP) 5 of
TERRANOVA investigates and develops the concepts of THz modems, frontends and
antennas, and the interface to embed them into different fibre optic links.
The TERRANOVA use case scenarios consider P2P (point-to-point) and P2MP (point-to-multi-
point) link architectures, which can be part of a static, reconfigurable or mobile network.
This leads to quite a number of design variations and alternatives for the required THz
system components. In order to handle the complexity within this project, the
implementation of two different system demonstrators was planned, which are a high
capacity P2P hybrid fibre optic - THz wireless link and a beamforming THz wireless link. Each
demonstrator focuses on specific challenges and innovations only. The first demonstrator
explores the capacity bounds of wideband channels at frequencies from 220-325 GHz. The
second demonstrator investigates phased array frontend architectures and their calibration
and control for beamforming.
WP5 develops component prototypes for implementing the two different system
demonstrators. This WP also addresses research challenges and innovations within the two
TERRANOVA pillars, i.e., “Tbit/s Wireless Connectivity” and “Co-Design of Optical and THZ
Wireless Links”, where some of the important interdisciplinary topics are:
Spectral efficient THz transceivers above 200 GHz: There is a significant functional
gap between the available THz wireless frontends of today and what is already
considered state-of-the-art (SOTA) at millimetre-wave (mmWave) frequencies.
TERRANOVA focuses on the integration of broadband analogue baseband functions,
the revision of transceiver architectures and analogue/digital correction schemes to
operate with higher order modulated signals at THz frequencies.
End-to-end optimized hybrid optical – THz wireless links: The hybrid fibre optic –
THz wireless link raises the question how the fibre optic baseband DSP (optical DSP)
has to be modified to cope with the THz wireless link. TERRANOVA investigates de-
emphasis, frontend correction and impairment mitigation for such hybrid links,
targeting Tbit/s connectivity.
Signal conversion between optical and electrical carriers: There exists quite a
number of research works into the implementation of RoF (radio-over-fibre)
solutions at THz frequencies. In contrast, developing an electrical baseband interface
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for mapping signals between the optical domain and the THz wireless domain, which
is consistent with standard COTS optical transceivers, has been little explored.
Coherency and synchronization: There is a need to accurately estimate the carrier
phase to guarantee coherency to the local oscillator. Digital schemes are quite
common for optical systems but there is little experience how those algorithms
perform for optical – THz wireless links.
THz antenna arrays and beamforming: There are no experimental studies available
beyond 200 GHz that explore the calibration and stability of phased array
communication systems and its applicability to different mobile use case scenarios.
TERRANOVA develops calibration schemes for the extraction of pre-defined
codebooks and models the application to different beamforming architectures.
All of the listed research and innovation areas are interdependent. The co-design of the THz
system components, hardware and digital signal processing, is inherently necessary to solve
the individual challenges. An overview of the high-level project plan of WP5 is provided in
Figure 1. The embedding of WP5 in between WP2 and WP6, and the interaction with WP3
and WP4 is vital for reaching the objectives.
Figure 1: WP5 project plan and connection with WP2 and WP6.
1.1 Scope
The objective of this deliverable is to formulate the concepts of the key THz system
components in detail, which means “ready for implementation”. In a first step, the system
block diagrams are broken down to the component level and suitable components and
concepts are identified. The main part of this deliverable identifies and develops first
prototype hardware components, suitable for fabrication, and algorithms suitable for off-
line testing. This requires setting up the design environment and the design flow towards
the implementation. The co-design of software and hardware components is considered and
highlighted where possible. Finally, based on simulation models, the relevant key
performance indicators are assessed for the investigated THz system components. Work
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packages WP2 and WP3 will use those simulation results to refine the overall system models
and system simulations.
1.2 Structure
The structure of this document is as follows:
Section 2 (System Block Diagram) reviews the current system block diagrams to
break down the TERRANOVA candidate architectures into units for physical
implementation.
Section 3 (Optical Link and TERRANVOA Media Converter Design) identifies
transponder solutions for the main TERRANOVA architectures and down-selects
candidates for the demonstrators in WP6, defines the interfaces and baseband
development, and identifies future integration and packaging solutions.
Section 4 (RF Frontend and Antenna Prototypes) investigates analogue frontend
chipsets using a new back-end-of-line process for the identified THz frontend
architectures, the interfaces to the digital baseband, all accompanied by prototype
designs relevant for the demonstrators in WP6.
Section 5 (Baseband Digital Signal Processing for THz Systems) derives from the
optical PHY (Physical Layer) a modified DSP architecture for the TERRANOVA optical-
THz modem, sets up the development and test environment, and investigates the
capacity limits of the existing solution experimentally
Section 6 (Phased Array Beamforming) identifies and develops digital beamforming
algorithms for THz phased array antennas, sets up a simulation environment for
proof of concept and models the currently existing solution as a first step.
Section 7 (Conclusions) summarizes the main achievements in D5.1, and outlines the
next steps towards the component implementation and testing.
Sections 3-6 reflect the four tasks of WP5, but in different order. Since the THz system
components need to comply with the optical link and its interface, this aspect is presented in
Section 3 first, which covers work in WP5.4 mostly. The architectures of the THz frontend
(WP5.1) and modem (WP5.3) have to be co-designed and their presentation order is
somewhat arbitrary for that reason (Sections 4 and 5). For example, there are aspects of
frontend impairment calibration and carrier phase synchronization, where the capabilities of
the PHY DSP influence the THz frontend architecture. Vice-versa, the frontend performance,
functionality and implementation possibilities have influence on the PHY DSP. The
presentation of the technical work ends in Section 6 with the calibration and control of
phased array frontends, investigated in WP5.2. The components for the beamforming
demonstrator depend also on the phased array frontend architectures and implementation
limitations, which are part of work in WP5.1.
1.3 References
[1-1] “D2.1: TERRANOVA System Requirements”, EU Project TERRANOVA (grant
agreement 761794), public deliverable, v1.0, December 2017, online: https://ict-
terranova.eu/
[1-2] “D2.2: TERRANOVA System Architecture”, EU Project TERRANOVA (grant agreement
761794), public deliverable, v1.0, March 2018, online: https://ict-terranova.eu/
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2. SYSTEM BLOCK DIAGRAM REVIEW
The TERRANVOA system architecture candidates for the identified P2P and P2MP use-case
scenarios, which were proposed in the deliverables D2.1 and D2.2 of this project, [2-1], [2-2],
lead to quite a number of different hardware and algorithm components to be investigated.
This section briefly refines the system block diagrams further and breaks them down into
sub-assemblies, which are composed of the physical hardware components to be
investigated.
2.1 Review of TERRANOVA Architecture Candidates
A conceptual block diagram of the TERRANOVA hybrid fibre-optical – THz wireless link was
presented in D2.1, which depicts a unidirectional link for simplicity. Figure 2 depicts the
required software and hardware components and their assignment to the different tasks of
WP5. Deliverable D2.2 distinguished between incoherent optical link architectures compliant
to Ethernet and/or PON standards, and coherent optical link architectures compliant to
CFP2-ACO standards, depending on the location of the link in the optical network. This lead
to different THz frontend architectures and baseband specifications. A bidirectional solution
increases the implementation complexity of the THz frontend and its interface to the optical
link further. Finally, the P2MP use cases lead to the need of THz frontend array
architectures. In order to handle the complexity within this project, the implementation of
two different system demonstrators was planned, which are (1) a high capacity P2P hybrid
fibre-optic - THz wireless link and (2) a beamforming THz wireless link.
Figure 2: General overview of the components under investigation within WP5.
2.2 Generic Architecture and Hardware Overview
There are several functions that have to be provided for both of the demonstrators and
which are common to the different architectures considered so far. The IQ direct conversion
frontend (see [2-2], page 76) is the most natural match to the coherent optical link but
requires the support of two polarizations, thus two IQ channels. The incoherent optical link,
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which was proposed in [2-2] lacks spectral efficiency. Incoherent wireless PAM4 modulators
may be a way out, consistent also with optical link COTS, but the improvement of the net
data rate is only moderate. In addition, this solution requires gearbox functions at the
electro-optical interface. Instead, an alternative incoherent optical link solution is discussed
in Section 3.2, which is compatible to the coherent IQ direct conversion THz wireless
frontend architecture. This scheme uses for each of the wireless I and Q components a
separate optical link, which is a highly scalable solution, for example when using optical
wavelength-division multiplexing over a single fibre. Thus, it can also support the 2-channel
IQ direct conversion wireless architecture by using 4 optical links for the two IQ pairs. In
conclusion, the 2-channel transceiver frontend is a very versatile component, which allows
the investigation of different optical link solutions within TERRANOVA. The corresponding
electro-optical interface must support four analog baseband signals for the 2-channel IQ
direct conversion transceiver. The overall generic block diagram is depicted in Figure 3. This
representation is a unidirectional representation. The bi-directional counterpart includes the
same scheme for the wireless receive path. Both, the receive and transmit path need to be
adjusted to the optical interface by the analog baseband converter, which requires in its
most simplistic form only signal conditioning functions but may also include analog equalizer
functions for example. The preferred interface solution is a DSP-less solution. Within WP5,
the components for the TERRANOVA media converter and the THz frontend will be
investigated. For signal conditioning, broadband baseband amplifiers are key components to
focus on at first. In addition, a low-loss packaging platform needs to be established, possibly
also consistent with MSA pluggable standards.
Figure 3: Generic view on the TERRANOVA media converter.
2.3 Beamforming Demonstrator Block Diagram
The beamforming demonstrator focuses at first on digital beamforming. This requires also a
solution supporting multiple IQ-channels, which can be composed of the discussed 2-
channel transceiver units. For the digital beamformer, the LO generation must maintain a
good phase synchronization between the individual channels which is heavily impaired by
frequency multipliers that amplify the relative phase fluctuations. For this reason, the LO
generation cannot be carried out individually for each channel. A multichannel LO signal
source is a flexible solution as depicted in Figure 4. The LO interface is selected at an
intermediate frequency that allows board level routing between the LO signal source and
the IQ direct conversion transceivers. The fourth sub-harmonic seems to be a good
compromise for this frequency, though a detailed analysis by sample designs is planned in
the next phase of the project.
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The beamforming demonstrator requires an analog baseband interface to a customized E-
band modem. Since all beamforming functions are implemented in the modem in this
solution, there are no control signals required between the modem and the frontend, except
for the correction of frontend imperfections.
Figure 4: LO generation option for the digital beamformer.
2.4 References
[2-1] “D2.1: TERRANOVA System Requirements”, EU Project TERRANOVA (grant
agreement 761794), public deliverable, v1.0, December 2017, online: https://ict-
terranova.eu/
[2-2] “D2.2: TERRANOVA System Architecture”, EU Project TERRANOVA (grant agreement
761794), public deliverable, v1.0, March 2018, online: https://ict-terranova.eu/
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3. OPTICAL LINK AND TERRANOVA MEDIA CONVERTER
DESIGN
This section covers the progress on planning and designing the components for the optical-
THz wireless link. Different concepts were considered in the first project phase and four
different proposals will be presented in the first part of this section as a result. The second
part reports on first steps towards the actual physical implementation on component level.
3.1 Standardized Optical Transceivers
3.1.1 IM/DD Transceivers
Generally, two main detection schemes can be used to convert an optical signal to the
electrical domain: direct detection and coherent detection. The first one is simply based on
intensity modulation and direct detection (IM/DD), i.e. only the total power of the optical
field is converted into an electrical signal by a simple photodiode at the receiver side. In the
second approach, the received signal is mixed and boosted with an optical source signal
provided by a local oscillator (LO), and the full optical field information of the signal (i.e.
amplitude, phase and polarization) can be converted into the electrical domain [3-1].
Most applications in data-centres, metro and access networks employ IM/DD pluggable
transceivers as they are more cost-effective as compared to coherent solutions. They
perform sufficiently well for the relative short distances that needs to be bridged.
Most common form-factors of the IM/DD transceivers are the well-known SFP+, QSFP28,
XFP and CFP2, with bit rates varying from 1 Gbps to 100 Gbps, and distances from hundreds
of meters to 80 km and more.
XFP transceivers
XFP transceivers began in point-to-point applications, but soon spread to Passive Optical
Networks (PON) applications. Since they are a cost-effective solution, they can support the
recent optical network technologies such as XG-PON, XGS-PON and NG-PON2.
Figure 5: Typical IM/DD optical transceiver pluggable in XFP form factor.
PICadvanced is specialized in XFP form-factor development for the new access network
technology NG-PON2, which supports 10G/10G for up-/downstream and up to 64 clients for
each of the 4 wavelengths enabling 40 Gbps aggregated traffic in downstream.
XFP transceivers (10 Gigabit Small Form Factor Pluggable) follow the Multi-Source
Agreement (MSA). Despite of the different technologies such as 10G Ethernet, SONET/SDH,
ITU-T 10G PON, XFP transceivers all have the same basic recommendations such as electrical
interface, management interface and power dissipation limits.
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Interface
As it is common to many other transceiver form factors, XFP transceivers use an I2C
management interface, in order to control and read internal diagnostics.
XFP transceivers use two lanes of 10 Gbps, i.e. transmitter (Tx) and receiver (Rx) data paths –
CML or LVPECL -, to provide 10 Gbps symmetrical service. When possible, they employ the
use of CDR and CTLE to improve even further the link budget and reach longer distances. A
basic block diagram of an XFP can be found in Figure 6.
Figure 6: Basic block diagram of an XFP IM/DD transceiver module.
3.1.2 Coherent Transceivers
Backbone applications need higher capacity, higher spectral efficiency and longer fibre reach
which the IM/DD technology cannot deliver. Coherent transceivers have superior optical
performance and can provide electronic equalization of fibre impairments, such as
chromatic dispersion or polarization-mode dispersion. Moreover, as the phase from the
signal is recovered, higher order QAM modulation schemes are allowed leading to the
support of higher bitrates with high spectral efficiency. However, coherent solutions are
more expensive when compared to common IM/DD transceivers as they comprise more
complex optical components and require DSP ASICs with high-speed DACs and ADCs. Current
transceivers in the market can already achieve net bitrates of up to 400 Gbps.
Pluggable transceivers for coherent applications can be divided in analog coherent optics
(ACOs) and digital coherent optics (DCOs) subcategories. Both contain all necessary optical
components for transmit and receive in a single package. ACO offer lower cost, and typically
smaller form factor, however DCOs already comprises the DSP ASIC with DAC and ADC inside
the package, while ACO needs DSP capabilities on a host board.
In the current/next generation, pluggable coherent optical transceivers use the CFP/CFP2
form factor and follow the Multi-source Agreement / OIF agreement for coherent optics and
all the inherent guidelines. CFP/CFP2 uses the MDIO management interface and can have
different types of data path interfaces, i.e. digital (DCO) or analog baseband (ACO). A typical
block diagram for a coherent receiver can be found in Figure 7.
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Figure 7: Coherent receiver architecture.
This detector employs polarization-diversity, where two phase-diversity configurations by
using two 90° optical hybrid components are combined to detect the in-phase and
quadrature signals of each polarization of the light. The polarization beam splitter (PBS) is
used to split both signal polarizations, and after four balanced detectors, the in-phase and
quadrature signals from each polarization can be sampled using ADCs. Therefore, a great
advantage of these receivers is the ability to detect the full information of the optical field,
enabling the use of advanced modulation formats with all four dimensions in the optical
transmission. Finally, advanced DSP can be employed to manipulate the digitized
information towards the compensation of the optical link distortions.
3.2 TERRANOVA Media Converter Design
In this section, we present several possible solutions for the TERRANOVA Media Converter
design. The TERRANOVA Media Converter enables the possibility to transmit from 100 Gbps
up to 800 Gbps from TERRANOVA radio front-end through the fiber. In order to transmit
such high data rates, we have to consider state of the art optical IM/DD and optical coherent
transmission technologies.
3.2.1 Proposal 1 – IM/DD using PICadvanced NG-PON2 technology
The first solution presented comprises the use of a time-wavelength division multiplexing
(TWDM) transceiver (included in PICadvanced portfolio) that allows the transmission of
multiple 10 Gbps links with different wavelengths in a single fiber. It can be viewed as a
lower cost solution when compared to commercial IM/DD 100 Gbps solutions. A block
diagram of a possible application of this proposal is presented in the Figure 8.
Figure 8: Possible application of the IM/DD transceivers using PICadvanced NG-PON2 technology.
For this type of application, the Figure 9 depicts the proposed optical architecture with the
TERRANOVA Media Converter. Therefore, the idea behind the system is to multiplex a
defined number of 10 Gbps modulated optical carriers in order to achieve the desired
aggregated data rate over the optical link. For the XFP solution, the non-return-to-zero (NRZ)
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modulation format must be used due to the integrated subcomponents, such as limiting
amplifiers and transimpedance amplifiers (TIAs), or the clock and data recovery (CDR)
subsystem.
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Figure 9: IM/DD solution based on PICadvanced NG-PON2 transceivers.
For the “central office-to-radio” direction, N parallel bit streams reach a Serializer /
Deserializer (SerDes) device, which converts the N parallel 10 Gbps / 25 Gbps data signals
into 10 Gbps signals. The de-serialized 10 Gbps NRZ signal lanes will then feed the different
XFP modules which will modulate different optical carriers.
Each XFP transmits on a different optical wavelength, resulting in multiple wavelengths at
10 Gbps each (WDM network). The wavelength multiplexer (WM) is used to combine all
wavelengths, and at the receiver side each one is then filtered in order to be directly
detected at the optical transceiver level. After photo-detection, 10 Gbps electrical NRZ
signals can then be connected to a DSP. Another option is to first convert to a different line
rate using a SerDes device in order to match the connections. At the DSP, no digital
equalization is required for the NRZ signals, therefore they can be directly linked to the DSP.
The DSP is only used to generate the in-phase and quadrature electrical signals to be
transmitted over the THz radio system.
For the “radio-to-central office” direction, the radio signal is affected with dispersion or
phase noise, and therefore, before being transmitted over the optical link, it must be
compensated in the DSP. The ADCs are used to digitize the in-phase and quadrature
electrical signals. After the DSP, the signal can then be sent to the optical link using multiple
10 Gbps electrical NRZ signals. The optical transceiver solution is also based on multiple
XFPs, each one transmitting on a separate optical wavelength, and the WM is used to
combine all wavelengths.
At the fibre, the “radio-to-central office” direction signal is composed by multiple 10 Gbps
NRZ wavelengths. Note that the “central office-to-radio” direction wavelengths must be
different from the “radio-to-central office” direction wavelengths in order to avoid non-
linear crosstalk. At the receiver side, each wavelength is firstly filtered using an optical WM
and then detected by the XFPs.
The advantages of this technology are the following:
Low cost solution. Since the idea is to parallelize a high electrical bandwidth signal
(100G/800G) into multiple 10 Gbps signals, the transceiver solution is cost-effective.
In addition, the transceiver is fully based on IM/DD components;
High flexible optical solution. The aggregated data rate can be easily extended by
increase the number of parallel XFPs.
And the disadvantages are:
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Low spectral efficiency. Since the modulation format is NRZ, and a wavelength guard
band must be used to separate the different wavelengths (e.g. 100 GHz as the used
in the NG-PON2 technology), the optical spectral efficiency is low;
Fibre length links up to 20/40 km. In case the NG-PON2 standard is used as baseline,
maximum reach would be 40 km as the band of transmission exhibits chromatic
dispersion.
3.2.2 Proposal 2 – IM/DD using COTS 100G
Commercial off the shelf transceivers for 100 Gbps applications with IM/DD already exist in
the market - such as CFP to CFP4 form factors - and present a concurrent solution for
Proposal 1. Although this solution is available, it can be more expensive than solution for
Proposal 1, when applications for 10 km+ or 100 G+ are needed. A possible architecture of
this Proposal is depicted in Figure 10.
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N x 10/25G N RZ100G N RZ
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Figure 10: IM/DD solution based on CFP transceivers.
Although quite simple, the solution does not offer high flexibility. A commercial off the shelf
transceiver is employed, and thus the solution is fixed at 100 Gbps, not allowing
parallelization since the wavelength is fixed. The purpose of the DSP is to generate and
demodulate the in-phase and quadrature signals from the radio system, since the optical
interface is based on NRZ signals. This solution can be also limited at <20 km due to the
chromatic dispersion.
3.2.3 Proposal 3 - IM/DD transceivers based on amplitude modulation
Proposal 3 comprises a similar scenario to Proposal 1, however in this case direct amplitude
modulation is applied from the IQ radio signals to the optical transceiver. The main
advantage for this application is that the IQ signals from the THz antenna can be directly
mapped into the optical transceivers, but this comes at the expense of fiber distance due to
dispersive distortions. The optical link would be reduced in comparison to the 20/40 km
presented for Proposal 1.
Figure 11 shows the proposed optical architecture. The idea is to use two optical
wavelengths per direction to transmit both in-phase and quadrature signals. Highly linear
IM/DD transceivers must be used to achieve full amplitude modulation.
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Radio System
SSMF
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Figure 11: IM/DD solution based on amplitude linear transceivers.
For the “central office-to-radio” direction, the DSP is used to generate the in-phase and
quadrature electrical signals, and each signal is modulated in a different wavelength by using
for example a Mach-Zehnder Modulator (MZM). The WM is used to combine both
wavelengths, and at the receiver side both are direct detected. After optical detection, the
in-phase and quadrature signals are available to be transmitted to the radio system.
For the “radio-to-central office” direction, the concept is similar, and therefore another two
wavelengths are used to transmit the in-phase and quadrature radio signals. Since both
radio signals reach the optical system affected with dispersion or phase noise, at the
receiver side both signals must be sampled and compensated in DSP. The DSP could be
located directly after the radio system or after the optical system, in both cases it is required
to demodulate the in-phase and quadrature signals.
This technology is cost-effective since only four IM/DD optical transceivers are required and
has a great advantage, which is its easy scalability for higher symbol rates. Modulation
schemes of higher order such as 256QAM are in principle transparent for this technology, if
the linearity of the IM/DD transmission is sufficient. Furthermore, since both in-phase and
quadrature radio components are high bandwidth signals (>16 GBd), they are affected by
chromatic dispersion in the optical link thus the typical optical reach for such scenario would
be less than 10 km.
For the TERRANOVA project, and to the best of our knowledge, there is no commercial off
the shelf solution available for this approach. Thus, this proposal requires further
investigation, since most of the similar configurations in the literature are based on the radio
over fibre (RoF) applications using for instance orthogonal frequency-division multiplexing
(OFDM). In [3-2], experimental results of transmitting several digital 16QAM RoF signals with
high spectral efficiency were discussed, based on Sub-Carrier Multiplexing (SCM) techniques.
At the transmitter, a simple MZM is employed, and at the receiver, the optical signal is
converted to the electrical domain by a photo receiver. Due to the SCM, the IM/DD system
corresponds actually to a heterodyne detection, and the transmitted signal is available in
both amplitude and phase in the electrical domain at the receiver. Therefore, no DSP
subsystem is required for optical equalization and only filtering subsystems to improve the
signal-to-noise ratio (SNR) are used.
Since the idea of our proposed solution is based on transmitting two electrical baseband
signals with amplitude modulation, i.e. both I/Q radio signals carrying multi-level modulation
(which were previously slight distorted over the radio channel), an approach widely
investigated that can be compared to the proposed is based on transmitting pulse amplitude
modulation (PAM) signals using IM/DD transceivers. For instance, in [3-3] a single optical
carrier with a single polarization 56 GBd PAM8 signal combined with raised cosine shaping
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and pre-emphasis is propagated over 80 km fibre using simple IM/DD transceivers. However,
to mitigate the chromatic dispersion induced over the 80 km link as well as bandwidth
limitations, a 3-tap feed-forward equalizer (FFE) is applied at the receiver. If the symbol rate
is decreased and lower distances were used, the FFE may be avoided. In addition, in [3-4]
and [3-5], Nyquist-shaped PAM signals are also employed using IM/DD transceivers to
achieve single data rate lines of >100 Gbps for shorter-reach applications. Also, in [3-6] 56
Gbps PAM signals are demonstrated for inter-data centre connection optical networks over
a 100 km optical link. All these configurations are using amplitude equalizers (e.g. FFE) at the
receiver side to improve the eye diagram performance. This could also be applied to this
proposal since the radio signal should be restored (and equalized) in the DSP located at the
central office.
3.2.4 Proposal 4 – Optical Coherent transmission
In applications where fibre cannot be employed and an “over the air” fibre extension is
needed (TERRANOVA “fibre extender” application), analog coherent transmission becomes
an interesting solution. This solution potentially reaches up to hundreds of kilometres. In a
best case scenario, it only requires the DSP+ADC/DACs to be located in the central office to
jointly recover the signals and mitigate wireless and optical impairments. While these kind of
transceivers and DSP for coherent optical links are usually more expensive than the
incoherent counterpart e.g. in Proposal 3, this disadvantage might be compensated by the
superior performance allowing to bridge higher distances and use higher bit rates and
spectral efficiencies. Furthermore, the relative DSP complexity required for the pure THz link
in comparison to the hybrid optical – THz link is still to be explored.
The Figure 12 shows a possible application of the coherent optical solution.
Figure 12: Possible application of the coherent solution.
In order to evaluate the optical system configuration, Figure 13 depicts details of the
technology for a single-polarization (single IQ) THz frontend. At the optical transmitter side,
the in-phase and quadrature signals are directly connected to one of the two IQ modulators
of the coherent ADO transceivers pluggable.
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SSMF
Optical System
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Figure 13: Coherent solution based on single polarization THz radio interface.
During the propagation over the optical link, the signal is affected by polarization rotations.
Therefore, from the “central office-to-radio” direction, the signal generally reaches the
optical coherent receiver as a dual-polarization signal, i.e. distributed over both
polarizations. Before being transmitted to the radio channel, the single-polarization signal
must be reconstructed in a coherent DSP. Traditionally, the coherent DSP includes the
following subsystems [3-7]:
Pre-processing subsystems with a matched or a low-pass filter in order to mitigate
noise interferences;
Amplitude normalization subsystem to improve the dynamic range of the DSP;
Clock recovery algorithm towards to compensate the ideal sampling instant of the
ADCs;
Static equalization to compensate the chromatic dispersion;
Adaptive equalization to compensate the rotations of the state-of-polarization (SOP)
of the optical signal;
Carrier frequency and phase recovery to compensate the frequency and phase noise
between the transmitted and the received local oscillators lasers, respectively.
For the “radio-to-central office” direction, the DSP located on the central office side is
expected to compensate both radio and optical interferences.
A second approach of the coherent system relying on a dual-polarization (double IQ) THz
frontend is shown in Figure 14. From the “central office-to-radio” direction, instead of
transmitting a single polarization signal, a dual-polarization signal from the DSP directly
feeds the two optical IQ modulators of the ACO pluggable, and both signals are propagated
over the optical link in separated polarizations. At the optical receive side, the two received
optical polarizations can then be in principle directly connected to the dual-polarization THz
front-end without a coherent DSP, while the joint impairments of the optical-THz link are
compensated at the central office DSP, including the polarization rotations in the optical link.
Optionally, a coherent DSP can be still used at the THz frontend.
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SSMF
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Figure 14: Coherent solution based on the dual-polarization THz radio interface with optional DSP at the radio front end.
3.2.5 Conclusion
Each of these proposals have different application scenarios, so the choice of the application
will directly choose the media converter type for TERRANOVA. Proposal 3 is the simplest
solution, but it may be limited by the fibre reach and symbol rate operation, and due to the
lack of commercial off the shelf transceivers requires further investigation. Proposal 1 may
be the cheaper solution by using multiple XFP transceivers, but a DSP between the radio and
the optical system is required to enable an on-off keying (OOK) optical system. A similar
issue is observed with the Proposal 2, but it is further limited in flexibility providing a fixed
100 Gbps lane.
For high data rate throughput, the coherent solution may be the most attractive solution,
providing high spectral efficiency by using advanced modulation formats, highly extended
reach by using advanced DSP, and high sensitivity and wavelength selectivity provided by the
coherent detection, while this might come at the cost of a higher implementation
complexity, in particular with regard to the DSP with the associated ADC and DAC devices.
3.3 Concepts for Future Media Converter Integration
One of the objectives of WP5 is to investigate how to co-integrate a state-of-the-art high
bitrate optical transponder and a THz wireless frontend in a compact and cost-efficient unit.
In that context, it is also of interest to explore advanced electro-optical packaging solutions,
considering also the progress of PICs (Photonic Integrated Circuits). Integration aspects of
the coherent optical transmission (Proposal 4) are discussed in this section since it targets to
use optical components of the shelf with little modifications. The coherent hybrid wireless-
optical link promises the highest spectral efficiency by using two polarizations in the THz
wireless domain. However, the incoherent optical transmission of the IQ baseband signals
can be also expanded to four optical channels (corresponding to four wavelengths). In this
case, the coherent and incoherent optical transmission lead to the same analog THz
frontend architecture, though the specifications on the electrical analog interface deviate in
detail. The incoherent optical link and its interface is scientifically more interesting since it
allows to develop customized solutions which exploit the progress of PICs and new
packaging technologies, and may be in that sense more disruptive.
The block diagram in Figure 15 shows the assignment of the functions of the TERRANOVA
media converter to the host board and transponder module. The THz wireless frontend sits
on the host board and emulates some of the functions that a bridge would have. The host
board is not part of a switch but is actually part of an antenna feed. Physical requirements
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on the size of this “THz frontend host board” will be investigated in the next phase of the
project, as part of the antenna design. The illustration of the CFP2-ACO module is taken from
[3-8], where more details on electrical and mechanical specifications can be found.
Figure 15: Functional block diagram of the media converter for coherent optical transmission.
Figure 16 shows the cage system of the host board as proposed in [3-9]. The connectors on
the host board are specified as part of the OIF Implementation Agreement for CFP2-ACO
modules. The pluggable is guided by the cage and guarantees repeatable and reliable
contact with the board-to-board connector.
Figure 16: Illustration of a typical ACO module and the cage system on the host board [3-9] .
The typical accepted performance of the connector is also specified in [3-9], derived from
measurements on test board. The reported small signal transmission (S21) of the de-
embedded connector is depicted Figure 17. From those specifications, the connector can be
used up to about 20 GHz, which suits the FDD frequency plan of TERRANOVA (see also
Section 4) with 40 GHz of RF bandwidth per channel. When targeting baseband bandwidths
>20G, plugs and host connectors need to be used for the 56 GBd standard with no cross
mating. Further experimental development work of this electro-optical integration approach
has started as part of work package WP6.
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Figure 17: Insertion loss of the board-to-board connector on the host board, taken from [3-8].
Suitable board materials and transmission line technologies for the electro-optical
integration platform were considered. As soon as the interconnections extend over longer
distances of more than a few centimetres, the transmission line losses become important
and severely limit the system bandwidth. Figure 18 shows a test module, which was
designed and manufactured for characterizing the developed 4-channel baseband amplifier
chips of Section 4 using digital test signals. The module was used to get experimental
experience of the detailed problems. An insertion loss of 10 dB was measured at 30 GHz.
However, the 1-dB bandwidth should reach at least 20 GHz in TERRANOVA, the 3-dB
bandwidth should reach 40 GHz to be compliant with the CFP2 insertion loss requirements
and its future evolution. This sets the specifications for the new integration platform,
considering also the need to comply with the CFP2 board-to-board connectors (in terms of
thickness and trace metallization).
Figure 18: Manufactured baseband amplifier test module for risk and problem identification.
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The module in Figure 18 was assembled using standard Ultralam material from Rogers
Corporation of 100 µm thickness (further referred to as “LCP100”). For the selection of
future board materials and the development of the integration platform, the transmission
line losses of different board materials are compared with on-chip transmission line losses in
Figure 19. Focusing on LCP50 and LCP100 material (microstrip lines on 50 and 100 µm thick
Ultralam single sided boards), which was previously used as antenna material up to 100 GHz,
it becomes obvious that the board thickness (or the ground-to-signal spacing) is the most
dominating parameter that determines the losses of a given transmission line technology.
This is also noticeable for on-chip transmission lines, e.g. coplanar transmission lines on
GaAs substrates, using a ground-to-ground spacing of 20 µm and 50 µm (GaAs20 and
GaAs50 in Figure 19).
Figure 19: Measured transmission line loss using different printed circuit board materials.
From this experimental study for risk mitigation, there are two routes that will be further
investigated in the second phase of WP5 for the integration platform of the TERRANOVA
media converter.
The first approach is using thick board materials compliant with the CFP2 standard. This
requires the use of different transmission line technologies, instead of microstrip lines,
dielectrically filled centred striplines will be investigated. For further reducing the losses of
filter sections, suspended striplines will be also considered. Both transmission line types
have frequency limitations, mainly set by the width (a) of the rectangular waveguide and the
associated cut-off frequency, which is a function of the height (b) for a>b. The height
determines the transmission line losses to first order, and a typical width-to-height ratio a/b
> 1.5 is targeted. Figure 20 compares the cut-off frequency of the centred striplines and
suspended striplines, when using LTCC (Low Temperature Co-Fired Ceramics) and Rogers
PCB materials (RO4350, 5880). All the materials have different dielectric constants. LTCC is
an attractive option due to its possibilities to realize 2.5D packages and the thermal and
mechanical stability. For future extension of the baseband signal bandwidth, the cut-off
frequency should be at least 60 GHz, limiting the minimum achievable losses.
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Figure 20: Calculated frequency limitations of the centred striplines as a function of the waveguide thickness.
The second approach further investigated is to miniaturize and co-integrate all functions as
close as possible. Hybridly integrated transponder modules are typically surface mountable
on PCB boards, integrating PICs for wavelength multiplexing and laser modulator integration
with high speed driver electronics chips and passive filter components. [3-10] gives an
overview of the history and state-of-the art in packaging photonic components up to 2015.
The photograph of Figure 21 from [3-11] shows a typical image of a TOSA / ROSA module
(transmitter/receiver optical sub assembly) as a reference design, which is still widespread in
use. This approach is good start for prototyping and validating first system concepts. In a
next step, the test modules can be re-designed using more advanced integration concepts.
The next steps in WP5 will be to develop hybrid modules for testing in back-to-back
configuration the TERRANOVA media converter with the two outlined packaging
approaches.
Figure 21: State-of-the-art of hybrid integrated optical transponder modules, from [3-11].
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3.4 Conclusions and Outlook
This section has addressed the WP5 objective of the proposal to explore different concepts
for the optical link between a baseband unit and a remote THz antenna. Four proposals
were considered in more detail. The coherent optical link and the transmission of the IQ
signals of the THz wireless link by an IM/DD analog optical link were identified as the most
promising candidates for the implementation in the next phase of the project. Another
related objective of the work package is the evaluation and testing of approaches for the co-
integration of optical transponders and THz wireless frontends. One approach is to work
with COTS transponder modules for which implementation details and specifications were
developed. The second approach is to use or develop customized optical-electrical
components. This approach suits the exploitation plan of the consortium partners very well.
In the next phase of the work package, proof-of-concept test modules for both ideas will be
designed.
3.5 References
[3-1] I. P. Kaminow, T. Li, and A. E. Willner, Optical Fiber Telecommunications. B: Systems
and Networks, 2008, vol. V.
[3-2] P. P. Monteiro et al., "Mobile fronthaul RoF transceivers for C-RAN applications,"
2015 17th International Conference on Transparent Optical Networks (ICTON),
Budapest, 2015, pp. 1-4.
[3-3] M. Chagnon et al., "Single Carrier 168 Gb/s PAM8 over 80 km Below HD-FEC Using
Simple Receiver Equalization for Data Centre Interconnects," 2017 European
Conference on Optical Communication (ECOC), Gothenburg, 2017, pp. 1-3.
[3-4] N. Kikuchi et al., "Intensity-modulated / direct-detection (IM/DD) Nyquist pulse-
amplitude modulation (PAM) signaling for 100-Gbit/s/λ optical short-reach
transmission," 2014 The European Conference on Optical Communication (ECOC),
Cannes, 2014, pp. 1-3.
[3-5] M. Xiang et al., "Single-Lane 145 Gbit/s IM/DD Transmission With Faster-Than-
Nyquist PAM4 Signaling," in IEEE Photonics Technology Letters, vol. 30, no. 13, pp.
1238-1241, July1, 1 2018.
[3-6] S. Yin et al., "100-km DWDM Transmission of 56-Gb/s PAM4 per λ via Tunable Laser
and 10-Gb/s InP MZM," in IEEE Photonics Technology Letters, vol. 27, no. 24, pp.
2531-2534, Dec.15, 15 2015.
[3-7] S. J. Savory, "Digital filters for coherent optical receivers," Optics Express, vol. 16, no.
2, pp. 804-817, January 2008.
[3-8] OIF Optical Internetworking Forum, “Implementation Agreement for CFP2-Analogue
Coherent Optics Module”, IA # OIF-CFP2-ACO-01.0, January 2016.
[3-9] “CFP2 Hardware Specification,” CFP Multi-Source Agreement (MSA), Rev. 1.0, July
2013.
[3-10] Z. Zhang, et. al.,“Packaging investigation of optoelectronic devices,“ Journal of
Semiconductors, vol. 36, no. 10, 2015.
[3-11] Kish F A, Nagarajan R, Joyner C H, et al. „100 Gb/s (10 _ l0 Gb/s) DWDM photonic
integrated circuit transmitters and receivers,“ Conference on Lasers & Electro-Optics
(CLEO), CMGG3, 2005.
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4. RF FRONTEND AND ANTENNA PROTOTYPES
TERRANOVA’s research and development on RF frontend components is driven by the
objective to increase the spectral efficiency and the integration density of THz wireless
transceivers. Both aspects are crucial for their commercial success, considering the progress
of wireless technologies at lower frequencies, for example at the E-band (71-76, 81-86 GHz),
which promise longer distances and the use of higher order modulation schemes. For this
reason, operating wireless systems at THz frequencies aims to the exploitation of enormous
spectral resources available for LMS/FS (land-mobile / fixed services) at higher integration
densities as of today’s available frontend solutions.
A new generation of THz frontend MMICs (Monolithic Microwave Integrated Circuits) will be
developed in the course of the project for that reason. The circuit designs use the
InAlAs/InGaAs metamorphic HEMT (high electron mobility transistor) process of the
Fraunhofer IAF. The transistor gate length is 35 nm. Currently, this node is the most
advanced transistor node at the Fraunhofer IAF though its BEOL (Back End of Line)
interconnection possibilities are rather limited. In the first phase of WP5, from M1-M12, the
objective of increasing the integration density was addressed by the development of a new
4L BEOL process that replaces the traditional 2-layer/ airbridge technology found in standard
commercial III-V technologies at lower frequencies. Two wafer fabrication runs were
dedicated to that process development. For that purpose a wafer split was introduced,
meaning that wafers of the batch were taken out of the standard fabrication flow for
process development. Motivation and aspects of this work as well as first circuit results will
be presented.
The initial frequency plan of TERRANOVA considers FDD with DL and UL frequencies at 220 –
260 GHz and 260 – 300 GHz, respectively. The design of guard bands is subject to further
investigation. The spectrum allocation as part of the new IEEE 802.15.3d standard, plans for
multiple frequencies from 252 – 321 GHz [4-1], [4-2]. This standard was also put into ITU
WRC-2019 (World Radiocommunication Conference) for consideration. The system
architectures of D2.2 will be further investigated and detailed in WP2, while WP5 will
develop a flexible analogue IP core library based on the new BEOL process. The second
phase of this work package (WP5), M13-M24, will look into the application of this library to
specific transceiver designs based. The simulation and measurement results will also feed
into system simulations in WP2 and WP3. Another major objective of WP5 in that context is
the research on the electro-optical interface for the hybrid optical-wireless architecture. A
first study has focused on the design of analogue baseband amplifiers from DC to 50 GHz
and variable gain functions.
4.1 THz Frontends for Point-to-Point Applications
The first frequency plan of TERRANOVA, reported in deliverable D2.2, considers the
frequency bands 220 – 260 GHz for the DL and 260 – 300 GHz for the UL, while part of them
will be used for guard bands. More details on the frequency plan will be derived within WP2.
In comparison, the spectrum allocation as part of the new IEEE 802.15.3d-2017 is depicted in
Figure 22, allocating 69 channels between 252.72 and 321.84 GHz that may be used. There
are ongoing discussions, as part of the ITU WRC-2019 preparation, on how this frequency
plan can be introduced into the planned regulation of the frequency spectrum 275 –
450 GHz. Due to a conflict with RAS (radio astronomy services) and EESS (earth exploration-
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satellite services), it is possible that only a sub-spectrum in 252.72 – 292 GHz will be
available, instead of the full spectrum up to 321 GHz.
Figure 22: Channel allocation plan of IEEE 802.15.3d-2017 standard.
The specific design options presented in Section 3 for the implementation of the hybrid
optical-wireless link lead to different THz frontend architectures and baseband interfaces.
The incoherent optical link with analogue IM/DD transceivers for transmitting the I/Q signals
of the THz link facilitates the exploitation of polarization for realizing duplex operation of the
wireless link. However, the minimization of transmitter (Tx) to receiver (Rx) interference by
means of antenna design and signal processing, is complex. Improved system models are
required before this is addressed in more detail in WP2. The optical link with coherent
transmission (Proposal 4) exploits the polarization of the wireless link by multiple-input
multiple-output (MIMO) signal processing to increase the capacity. This is a more natural
extension of the transmission scheme of the optical link to the THz wireless link. The
corresponding frontend architecture needs to support two I/Q channels while only one I/Q
channel is required for the incoherent case.
In summary, the THz frontend design needs to address different frequency band options and
different frontend architectures. The idea of this work is to develop a rather standardized
analogue IP core library of functional blocks suitable for the implementation of the various
transceiver options of TERRANOVA. In parallel, supporting the idea of co-design of the
wireless and optical link as part of WP2, the establishment of behavioural simulation models
of all functional blocks is also targeted. The following sub-sections review and analyse
transceiver correction schemes, to be considered for the final transceiver solutions, and
derives the block diagrams of the different transceiver architectures considered in the next
phase of WP5.
4.1.1 Duplexing Techniques
The separation between DL (Down Link) and UL (Uplink) is based on a duplexing scheme:
either FDD (Frequency Division Duplexing) or TDD (Time Division Duplexing) may be
employed. In FDD, a different frequency is used for each direction, whereas in TDD, both
directions use the same frequency and DL/UL separation is achieved in time, with the frame
being divided in DL and UL sections respectively. It is apparent that FDD offers lower latency,
albeit through higher cost due to the double transceivers used, compared to TDD.
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TDD basically emulates a full duplex communication over a half-duplex communication link.
TDD also offers advantages in asymmetric communications, i.e. where
the uplink and downlink data rates differ. In such cases, when the uplink/downlink traffic
increases, capacity can be dynamically allocated, and as the traffic load decreases, capacity
can be taken away. Another important advantage of TDD is the fact that since both
directions share the same frequency, due to the principle of channel reciprocity, they face
similar channel characteristics. This is especially useful in techniques that require channel
state information at the transmitter.
On the other hand, FDD is more efficient in the case of symmetric traffic, as it avoids the
overhead of switching over from Tx to Rx that TDD would require. Another advantage of FDD
is that it simplifies radio planning. By transmitting and receiving in the same frequency, the
probability of interference is minimised. Respectively, a TDD system would have to maintain
guard times between neighbouring systems, unavoidably reducing spectral efficiency.
Synchronisation must also be imposed by guaranteeing common transmit/receive reference
times, something that increases complexity and cost.
Another duplexing technique that is recently gaining attention, is the FD (Full-duplex)
communication, with simultaneous transmission and reception at the same carrier
frequency. FD communication has for long time been seen as a very promising way to boost
spectrum efficiency, but its application was not practical due to the severity of self-
interference effects. A powerful transmitter is pushing its receiver to saturation, since they
both operate at the same frequency. Recent breakthroughs in advanced self-interference
cancellation (SIC) techniques, enable the implementation of FD communications,
demonstrating an almost doubled spectral efficiency.
4.1.2 Transceiver Correction Schemes and Synchronization
The fast evolution of wireless communication systems is driving the design and
implementation of modern flexible and software-configurable radio transceivers [4-1]-[4-5].
By definition, flexible radios are characterized by the ability to operate over multiple-
frequency bands, the support of different types of waveforms, and the compatibility with
current and future air interface technologies. The terms multi-mode and multi-band are
commonly used in this context. Furthermore, TERRANOVA is expected to support high data-
rate applications and services that require efficient and low-cost wideband radio designs for
the mobile terminal [4-6], [4-7].
In this context, the well-known direct-conversion architecture (DCA) has become
instrumental for realizing compact, low-power, and low-cost transceiver designs for
wideband radio [4-8]. In direct-conversion transceivers, quadrature mixing is used, which
theoretically provides infinite attenuation of the image band and removes the need for
analogue image-rejection filtering. However, in practice, the DCA is sensitive to
imperfections of the analogue radio frequency (RF) frontend sections of the transceiver, due
to fundamental physical limitations. Such imperfections are in-phase and quadrature
imbalance (IQI), phase noise (PHN) and amplifier non-linearities [4-5], [4-9]-[4-17]. Next, we
analyse these imperfections and we provide a state-of-the-art review of the compensation
techniques that can be employed.
IQI: stems from the unavoidable amplitude and phase differences between the physical
analogue in-phase (I) and quadrature (Q) signal paths. This problem arises mainly because of
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the finite tolerances of the capacitors and resistors used in the implementation of the
analogue RF frontend components. Although a perfectly balanced quadrature down-
conversion corresponds to a pure frequency translation, IQI introduces a frequency
translation that results in a mixture of image and desired signals. In more detail, I/Q
mismatches decrease the theoretically infinite image rejection ratio (IRR) of the receiver
down to 20 − 40 dB, resulting in crosstalk or interference between mirror frequencies [4-6],
[4-18]. Consequently, IQI degrades the effective signal-to-interference-plus-noise ratio
(SINR) and causes performance degradation. The impact of IQI is more severe in systems
employing high-order modulations and high coding rates, such as Wireless Local Area
Networks (WLANs), Worldwide Interoperability for Microwave Access (WiMAX), Long-Term
Evolution (LTE), and Digital Video Broadcasting (DVB), among other standards [4-19]. Hence,
effective IQI compensation is essential for the design of high data-rate communication
systems employing the DCA.
Various approaches have been proposed so far to eliminate, compensate, and mitigate the
effects of IQI using baseband signal processing techniques, see [4-4], [4-6], [4-20]-[4-32], and
references therein. For example, in [4-28], the authors proposed a number of pilot designs
for channel estimation in orthogonal frequency domain modulation (OFDM) systems in the
presence of I/Q mismatches at both the transmitter and the receiver. Moreover, estimation-
based system-level algorithms, including least square equalization, adaptive equalization,
and post-fast Fourier transform least squares, were proposed in [4-32] to compensate the
distortions caused by IQI. Furthermore, blind (non-data-aided) digital signal processing-
based compensation of IQI for wideband multi-carrier systems was studied in [4-4], [4-6], [4-
20], [4-33], [4-25]. Specifically, in [4-20], a digital compensation method was proposed for
MIMO systems employing space-time block coding, which is based on the algebraic
properties of the received signal combined with a suitable pilot structure, while interference
cancellation-based and blind source separation-based compensation methods were
presented in [4-25].
All previously mentioned works deal with IQI as a source of impairment that should be
compensated. In contrast to this approach, IQI at the Tx may also be seen as a source of
diversity, due to the Tx-induced mirror-frequency interference. This diversity can be fully
exploited via joint maximum likelihood (ML) detection of the signals received in the mirror
subcarriers, or partially exploited by other sub-optimal nonlinear detection techniques such
as successive interference cancellation (SIC), as was demonstrated experimentally for OFDM
in [4-34], and later confirmed in [4-35]. Still, when weighed against the implementation
complexity of nonlinear Rxs, the small achievable signal-to-noise ratio (SNR) improvement
may prove to be too expensive [4-36]. Moreover, as pointed out in several prior works
including [4-34], Rx IQI is detrimental for the outage and error performance of wireless
communication systems, regardless of the utilized detector. The reason for this is that Rx IQI
affects both the received signal and the noise; hence, it is commonly believed that Rx IQI
should be compensated [4-4], [4-6], [4-20], [4-21], [4-11], [4-25], [4-31]. Finally, [4-37]
provides a low-complexity technique that achieves a diversity gain in the presence of Rx IQI.
PHN: Noise is of major concern in local oscillators (LOs), because introducing even small
noise into an LO leads to dramatic changes in its frequency spectrum and timing properties.
This phenomenon, peculiar to LOs, is known as phase noise or timing jitter, and was
identified as one of the major performance limiting factors of communication systems in
several studies, see for example [4-12], [4-38]-[4-53]. Generally, the disturbance of the
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oscillator’s output amplitude is marginal. As a result, most influence of the oscillator
imperfection is noticeable in random deviation of the oscillator’s output frequency. These
frequency deviations are often modelled as a random excess phase, and therefore referred
to as phase noise. Phase noise will increasingly appear to be a performance limiting factor
especially in the case of multi-carrier communications, when DCA implementations or
systems with high carrier frequencies are considered, since, in those cases, it is harder to
produce an oscillator with sufficient stability.
Amplifier non-linearities: Feedback, feedforward, and predistortion are the most common
techniques for the compensation of the nonlinear transmitter [4-54], [4-55], [4-56]. The
principle of the feedback technique is to force the power amplifier (PA) output signal to
follow the input signal by feeding the output signal back to the input [4-57]. The
disadvantage of the feedback technique is its sensitivity to the delays introduced by the
amplifier and associated signal processing components [4-54]. For this reason, it is not
suitable for the compensation of modern wide bandwidth transmitters. However, wider
bandwidths can be handled using, for example, a Cartesian feedback technique [4-55].
Seidel has presented the first modern feedforward compensation scheme, which is based on
the early concept of Black from 1928. The principle behind this is to have an extra error
amplifier, whose output is subtracted from the main output signal. The error amplifier is
driven by the distortions produced in the main amplifier. The feedforward scheme is
applicable to wider bandwidth signals than the feedback technique. The power efficiency of
the entire Tx may be reduced, due to the use of the extra amplifier [4-58]. Likewise, time
mismatches between the parallel branches may reduce the performance. However, research
on feedforward compensation is still ongoing.
Macdonald has demonstrated the predistortion concept in audio communication as an
alternative to the feedback technique. The feedback technique was inconvenient for the
compensation of the nonlinear distortions of the loudspeaker. In the predistortion concept,
a nonlinear functional block, i.e., a predistorter, is inserted prior to the amplifier so that the
combined transfer function of these two components is almost linear. In other words, the
predistorter behaves as an inverse of the amplifier. In the same paper, Macdonald also
discusses postdistortion, i.e., the compensator is located after the amplifier. The
postdistortion approach was not suitable for the compensation of the loudspeaker either [4-
59].
Predistortion can be accomplished in the analogue domain at radio, intermediate, or
baseband frequencies, or in the digital domain at baseband [4-60], [4-56]. The analogue
predistortion has some advantages such as low cost and less complex DSP. For very high
bandwidth radio over fibre (RoF) applications such as fibre cable television systems,
analogue predistorters are also applicable, see e.g., [4-61] and references therein. However,
more flexibility is obtained using a digital predistorter handled by a digital signal processor
(DSP) and, in addition, an adaptive predistorter may be difficult to implement in the
analogue domain. For that reason, the digital predistortion approach has gained much
attention in the recent literature [4-62], [4-63].
4.1.3 Transceiver Architectures and Project Development Plan
Different basic transceiver architectures and reference designs were reviewed in deliverable
D2.2 ([4-64], Section 4). This leads to the development of common key building blocks for
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the implementation of integrated frontend MMICs. The development of a standardized well-
understood analogue IP core library facilitates the implementation of the different
transceiver schemes, reduces risks and provides system designers with behavioural models
for their research. Those behavioural models are developed and used within WP2, WP3 and
WP4. The basic development strategy of the project is illustrated in Figure 23. The design
and investigation of new frontend components is based on the development of a new 4-
layer BEOL interconnection process. Apart from the process development, new transmission
line and interconnection test structures were investigated, characterized and validated up to
320 GHz. Test transceiver components were developed in the first period of this work
package for the DL frequency band from 220 – 260 GHz: The focus was on the DL for risk
mitigation reasons and easier experimental characterization.
Figure 23: Overview of design activities of M1-M12.
The 240 GHz frontend test core is comprised of mixer, amplifier and multiplier components,
which are typically most critical in the design process. The components were tested with
different BEOL variations. In the next step higher integrated transceiver MMICs will be
developed and tested for the DL. In the same iteration, the building blocks for the UL will be
developed.
4.2 New Process Technologies for III-V based MMICs
This section reports on the progress of the BEOL process development. The first part gives
an overview of available FEOL (Front End of Line) transistor technologies and typically
available BEOL options, as a reference of the state-of-the-art. The second part provides first
results.
4.2.1 Technology Overview
The technical motivation for using a III-V based HEMT technology for prototyping the system
components in TERRANOVA is summarized with the help of Figure 24. Current RF transistor
technologies for mmWave and THz applications (status September 2017), which are either
operated in a foundry mode or in a qualified process mode with a fixed FEOL process (both
are referred to as “quasi foundry mode”), were considered in this survey. In addition, some
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emerging technology candidates in research are shown. For the devices in “foundry mode”,
measured data in their BEOL environment were considered, since relevant for the final RF
performance. The device size was chosen to approximate typical device sizes used in
mmWave amplifier designs operating above100 GHz.
Figure 24: Comparison of different transistor technology options for THz frontend design.
In the representation of Figure 24, the MSG (maximum stable gain) region and the transition
frequency (“k=1” stability point) to the MAG (maximum available gain) region, plotted on a
logarithmic frequency scale are used as a key performance indicator. The inset shows the
theoretical slope in the two regions, 10 dB/dec in the MSG region and 20 dB/dec in the MAG
region. This MSG/MAG curve is a practical upper boundary for the design of broadband
systems, if they are required to be immune to process variations. Please note, no measured
data of the “k=1” point were available for the silicon technologies, and for that reason they
represent merely the highest measured points (65 GHz or 110 GHz related to the
measurement system limits) but not the actual performance limit. However, the actual
“k=1” point must be located along the slope of 10 dB/dec containing the highest measured
MSG. In this transistor technology comparison, the III-V based transistor technologies stand
out (both HEMT and HBT). The HBTs have a higher MSG but a lower “k=1” stability point,
which can be improved by a common base configuration instead of a common emitter
configuration. The 35nm HEMT process can operate between 220 – 320 GHz in its MSG
region for typical devices sizes. This leads not only to superior noise figures but also to wider
system bandwidths for a given gain. In TERRANOVA, the focus is on wideband systems,
exploiting the frequency spectrum from 220 to 300 GHz with high spectral efficiency.
Broadband noise figure and output power are key performance metrics, and for this reason,
the 35 nm mHEMT transistor technology is a good candidate for the implementation of
those systems with superior performance.
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On the downside, the standard BEOL of current III-V HEMT processes limits the integration
density for systems operating above 100 GHz. Figure 25 presents a survey of typical BEOL
processes developed for RF applications on silicon (SiGe BiCMOS, SOI CMOS), and III-Vs.
Figure 25: Comparison of various BEOL processes, for different transistor technologies
While the III-V HBT process has a rather flexible RF BEOL, it lacks the complexity that silicon
technologies offer for the implementation of additional digital functions. The standard BEOL
of III-V HEMTs uses traditionally the semi-insulating GaAs (or InP) substrate for the
implementation of transmission lines. For more complex routing, and the implementation of
coplanar transmission lines, an airbridge technology is available. All other BEOL processes
use thin film transmission lines on the front side. This leads to more routing flexibility but
also to higher losses in tendency. For isolated circuit functions, e.g. a stand-alone amplifier
or mixer MMICs, the III-V HEMT BEOL is inexpensive and offers very low losses, which
benefits power and low-noise applications. Silicon BEOLs use SiO dielectrics, while the III-V
processes shown in Figure 25 use BCB (benzocyclobutene), a polymer based dielectric that
can be structured either by dry chemical etching or photolithographic patterning. The
standard Fraunhofer IAF BEOL for the 35 nm mHEMT process employs two thin BCB layers
but still uses the airbridge technology. This complicates the design of higher integrated
MMICs, and has also compatibility issues with standard RF surface mount packaging
technologies used by silicon foundries nowadays. In contrast, the standard rectangular
waveguide packaging solution often used with the III-V HEMT BEOL is well established up to
more than 300 GHz for prototyping and offers excellent and reliable performance suiting lab,
outdoor and space applications.
4.2.2 Motivation and Status of BEOL Development
TERRANOVA considers new advanced electro-optical packaging approaches for the
integration of the media converter between the optical and wireless link. The development
of a new BEOL process is one important practical step towards increasing the integration
density of today’s existing THz solutions based on the Fraunhofer IAF 35 nm mHEMT
process. The requirements on packaging demand also the replacement of the airbridge
technology for mechanical reasons and an end passivation of the top metal interconnection
lines.
Within the reporting period, different BEOL options were investigated, which are
summarized in the schematic illustrations of Figure 26. The 3L mHEMT BEOL with the
traditional airbridge technology was replaced by the 4L mHEMT BEOL. In this first step, the
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airbridge technology was replaced by a layer of BCB, filling and supporting the former
airbridge with dielectric material. This allows us to use the former airbridge metal layer
(TFMS) as a full additional routing layer for RF signals. In the next step, the deposition of a
thick BCB layer on top was investigated, a via process and the deposition of a thin top metal
(ANT) to be used as a layer for antenna integration were developed for that purpose. This 4L
mHEMT BEOL was compared to another variation, the 3LPP mHEMT BEOL. The 3LPP uses a
planarizing thin BCB layer on top of the TFMS metal layer instead of the thick BCB layer.
Figure 26: BEOL development and tests in TERRANOVA, (1) base process, (2) development run 1, (3), final solution for final transceiver integration.
First investigations of antenna prototypes showed that an even thicker dielectric layer than
in the 4L process would be necessary for broadband low-loss operation. This is addressed by
the idea of transferring a separate thick antenna substrate on top. The thickness of the
antenna substrate can be adjusted according to the optimal thickness required for optimal
antenna performance. In addition, the antenna substrate can be metallized on both sides
before the substrate transfer. In the end, it was found that the 3LPP mHEMT BEOL offers
more flexibility and reduces the complexity of the BEOL process (before antenna substrate
transfer). In addition, the 3LPP process is also compatible with the standard GaAs wafer
backside process, which comprises wafer thinning to 50 µm and backside via processing
through the GaAs wafer. The backside process keeps compatibility with standard waveguide
packaging technologies for prototyping and allows to address different markets.
As part of the process development, different test circuits were designed and manufactured,
though not fully optimized, since measurement-based passive component models were not
available at this stage. For that reason, it is expected that the presented circuit level results
of Section 0 will be further improved in the next three design and manufacturing cycles
scheduled as part of TERRANOVA. A SEM image of a manufactured 3LPP test circuit is shown
in Figure 27. Since BCB is nearly transparent optically, the corresponding chip photograph
does not allow us to distinguish the different metal layers well. As the BCB is non-conductive
and charges accumulate, some of the features appear distorted in the SEM picture.
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Figure 27: Left: SEM picture of manufactured mixer IP core component for 220 to 300 GHz with the 3LPP process (before TFMS end passivation), right: corresponding optical
photograph.
Based on transmission line test structures, in parallel to the process development, passive component models were extracted. At first, those models were used for monitoring the repeatability of the process and for benchmarking. At a later stage, these test structures were used to develop models for circuit designs. As an example, the measured attenuation of some of the transmission line types are compared in Figure 28, normalized to the length in millimetres. As a reference for the state-of-the-art performance of the 3L process, coplanar transmission lines on GaAs having a ground-to-ground spacing of 20 µm and 50 µm (CPW20 and CPW50 GaAs) are also added to the graph. The attenuation of microstrip lines in the 3LPP BEOL using the TFMS signal layer are compared before (3LPP) and after final passivation (3LPP-Pass). The measured performance of the 3LPP microstrip lines is comparable to coplanar transmission line references of the 3L process, and can be ranked between a CPW50 and CPW20 transmission line. Although the final passivation slightly increases the losses, the effective dielectric constant is also increased, which reduces the effective electrical length. The thin film microstrip line of the 3L process is also presented, which has the highest attenuation of all reference lines in the graph.
Figure 28: Left: Layout and photograph of manufactured transmission lines for process testing and monitoring, right: measured attenuation of a microstrip line of 10 µm width.
Summarizing the status of the BEOL development, while the initial target was the 4L BEOL,
the 3LPP BEOL turned out to be more flexible. The first results indicated that more
experience should be collected to optimize all process design rules, which has to be acquired
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by more manufacturing cycles and circuit designs with increasing complexity. The chip size
can be reduced by about a factor of 4, as demonstrated on the example of three different
driver amplifiers, which operate at similar frequency. The layouts are depicted to scale in
Figure 29 with the dimensions referring to the photograph outline (including pads).
Figure 29: Integration density of different BEOL for broadband RF applications (to scale), see also [4-65] for the 65nm CMOS amplifier.
The first example at 90-110 GHz uses a 100 nm mHEMT with the 3L BEOL. The second
example at 120-150 GHz uses a 65 nm CMOS with its typical RF BEOL. These are compared
to the result of this work, at 100-145 GHz, which uses the 35 nm HEMT technology with the
developed 3LPP BEOL. The results highlight that the integration density with the 3LPP
process reaches the one of silicon CMOS designs. The 35 nm mHEMT leads to superior
linearity and higher gain per stage, which also translates into the superior noise
performance of this technology. Looking at the output power, however, due to similar
breakdown voltage Vbreak and maximum drain currents Id,max, the 65nm CMOS design
achieves an even slightly higher saturated output power than the 35nm mHEMT design. The
gain per stage of the 65 nm CMOS design corresponds reasonably well to the technology
comparison in Figure 26. For this reason, with the 35nm mHEMT, more gain can be traded
for bandwidth while maintaining the two key performance parameters, linearity and noise
figure.
4.3 New MMIC Frontend Building Blocks
The following measurement results comprise the first circuit designs for the DL, from 200-
260 GHz using the 3LPP BEOL. The building blocks tested were used to assist the
development of the 3LPP BEOL process and for setting up a design library. This design library
of functional building blocks will be the basis for the integration of the different transceivers
for the high capacity demonstrator and the beamforming demonstrator. In the first
subsection, initial test designs of baseband amplifiers for the TERRANOVA media converter
are presented. A test package was also designed for the fabricated circuits, which is reported
in more detail in Section 3.3 in the context of the co-integration of the THz wireless frontend
with the optical transponder solutions. The second subsection presents some of the
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measurement results of the first designs that highlight the progress towards the WP5
objectives.
4.3.1 Component Candidates for the TERRANOVA Media Converter
For the implementation of the TERRANOVA media converter, some of the critical analogue
interface circuits were investigated, along with the possibility to integrate them using the 35
nm mHEMT process. In the current 35 nm HEMT process version, only normally-on
transistors are available. As a result of the negative threshold voltage, the gate-source
voltage required for class-A operation of a common-source amplifier is only about 100 mV.
The optimum operating drain-source voltage of a single stage is at least 800 mV. Hence, the
resulting DC potential difference between two cascaded common-source amplifier stages, is
greater than 500 mV (Fig. 1). This voltage has to be either compensated by a level-shifter
circuit or reduced by choosing specific types of circuit topologies. As a result of the low gate-
source voltage of only 100 mV, an efficient level-shifting using a multistage source follower
is not feasible. Therefore, resistively coupled amplifiers, a diode-level-shifter and a Kukielka
amplifier were investigated. In the course of TERRANOVA, the Kukielka solution was finally
selected as the solution to be further investigated and integrated. The manufactured
Kukielka test amplifier and its schematic circuit representation are shown in Figure 30.
Figure 30: Chip photograph of the fabricated Kukielka amplifier, schematic representation, and summary of the measured key performance parameters.
The Kukielka concept is a two-stage broadband Cherry-Hooper amplifier with multiple
feedback loops. The design equations based on the Kukielka configuration have been
studied in [4-65]. In contrast to the standard topology, the local series feedback loop at the
common-source input stage was omitted. This simplifies the input matching of the used
mHEMT and reduces the DC potential at the drain of transistor M1. Hence, the direct
coupling of the input stage and the output stage in the Darlington configuration is achieved
without shifting the DC level. Instead, the DC potential at the Darlington stage is increased
by the series feedback resistor Rs2.
The on-wafer measured S-parameters and noise figure of the fabricated test circuits are
presented in Figure 31. An important advantage of the Kukielka amplifier is the low noise
figure of less than 4 dB between 1 and 50 GHz, the achieved gain of 21 dB and the P-1dB
compression point of 2.5 dBm, which requires a very small chip area of 185 x 270 µm (0.05
mm²), in comparison to the other investigated concepts. The 3-dB bandwidth is 50 GHz.
State-of-the art amplifiers realized in CMOS, SiGe and InP HBT technologies achieve gain
levels above 30 dB since higher numbers of gain stages are used [4-67], [4-68]. The required
chip area of [4-68], [4-69] is larger than 0.2 mm2, since inductive peaking techniques were
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used, which in contrast were omitted in the Kukielka solution. One of the critical aspects is
the DC power consumption of 200 mW per amplifier, which becomes a severe problem in
arrays using digital phase shifting, since it scales with the number of channels. For this
reason, the overall power consumption of one channel of the THz transceiver is significantly
increased by the baseband amplifiers.
(a)
(b)
Figure 31: On-wafer measured performance of the manufactured Kukielka amplifier, (a) S-parameters and noise figure, (b) output power compression characteristics at 10 GHz.
In a further step, the possibility to use the Kukielka concept in the 2-channel receiver of the
coherent optical link (with two IQ receive signals), was investigated. As part of a risk
mitigation approach in the project, a separate test chip was designed and fabricated for that
purpose. The test chip targets to get information at an early phase of the project on the
homogeneity and unexpected parasitic effects that may occur in this multi-channel
configuration. In addition, test packages were developed (see Section 3.3 for test results). A
photograph of the fabricated test chip and a CAD drawing of the test package are shown in
Figure 32. The pitch of the 4-channel amplifier was chosen to be consistent with the pitch of
the planned 2-channel receiver. The DC bias supply in the current test chip emulates the DC
supply situation in the receiver chip, where multiple-channels will share one common bias
supply. The measured small-signal S-parameters in Figure 33 demonstrate that the phase of
the frequency response of all four channels was nearly identical (less than 2 deg variation),
and the corresponding magnitude showed variations of less than 1 dB, using a single bias
supply for all 4 amplifiers.
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Figure 32: Designed test package and photograph of the manufactured 4-channel Kukielka amplifier chip.
Figure 33: Measured frequency response, left: magnitude in dB, right phase in deg for the 4-channel Kukielka amplifier.
4.3.2 Circuit Components for 220-320 GHz Transceivers
The following subsection presents some of the first components that were designed,
manufactured and tested in the context of the 3LPP BEOL development. The components
comprise different frontend functions that need to be accurately implemented and fine-
tuned for a successful transceiver design in the next step. All functions were measured on-
wafer, and can be used to develop behavioural models for system simulations. In addition,
the measured results can be used to improve the passive and active device models for the
next circuit design phase.
The motivation for the design of a library containing all major frontend functional blocks in a
standardized way, consistent with the objective to develop phased array transceivers for
220-320 GHz, is depicted in Figure 34. The floorplan of the multi-channel transceiver suits a
minimum antenna array pitch of 500 µm, which is half a free-space wavelength at 300 GHz.
For this reason, all functional blocks operating at 220-320 GHz and the first 1st subharmonic
of 110-160 GHz were designed to have a height of 225 µm. A corridor of 25 µm is reserved
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for the bias supply of the functional blocks. Functions operating at the 2nd subharmonic (55-
80 GHz) are designed to fit into a height of 450 µm respectively 900 µm since they will drive
multiple channels in array architectures and can consume more space for that reason. This
leads to a grid plan that simplifies the implementation of different transceiver architectures.
Figure 34: First IP core library for the DL, MMIC for generating an LO signal that can be tuned varied from 200 to 260 GHz.
The on-wafer measured performance of the MMIC for generating a local oscillator signal
between 200 and 260 GHz is depicted in Figure 35. The maximum output power at 240 GHz
was about 8 dBm at a DC drain bias voltage of 1.2 V.
Figure 35: On-wafer measured output power at 240 GHz for the LO chip
The measured performance of the 120 GHz driver amplifier and the frequency doubler from
120 to 240 GHz are shown in Figure 36 and Figure 37. In a similar way, all other functional
blocks were characterized as additional breakout circuits.
The measured S-parameter and output power characteristics of the 3-stage 120 GHz driver
amplifier are shown in Figure 36. The amplifier provides 28 dB of small signal (SS) gain from
105 - 140 GHz. The input / output matching was better than 10 dB over the defined
frequency band. The gain flatness deviates slightly between scalar measurements and vector
network analyser measurements due to different source power levels and the on-wafer
calibration limitations by parasitic mode excitation and lack of probe isolation. Transistors in
common source configuration were employed. The output stage had a total gate width of
200 µm for the targeted operation resulting in a saturated output power of 12 dBm and a P-
1dB output compression power of >10 dBm, at a DC drain bias of 1.2 V.
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The on-wafer measured performance of the driver amplifier, followed by a standardized
frequency doubler function, is shown in Figure 38 and Figure 39. The combination of both
functions was verified to operate in 220-290 GHz, allowing to select arbitrary channels in the
IEEE 802.15.3 channel plan, as well as the TERRANOVA FDD frequency plan.
Figure 36: Chip photograph of manufacture 120 GHz driver amplifier and measured RF performance (S-parameters and output power characteristics).
Figure 37: Chip photograph of the fabricated broadband LO multiplier for selecting different Rx/Tx channels from 220 to 290 GHz and measured output power characteristic at a fixed
input power level.
The 3-stage topology of the 120 GHz driver amplifier was also selected for the
implementation of a 240 GHz output amplifier for a Tx chip. The amplifier consumes an area
of 500 x 225 µm. A photograph of the dedicated on-wafer test structure is shown in Figure
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38. The calibration reference plane is at the probe tips, both for the S parameter
measurements and the scalar power measurements with an Erickson PM4 calorimeter. The
total gate width of the output stage was reduced to 120 µm. The SS performance of the 240
GHz power amplifier exhibited a gain of 20 dB with a flatness of better than 1 dB from 210 -
250 GHz (Figure 39). A 3 dB bandwidth of nearly 60 GHz with an input/ output matching of
better than 10 dB was achieved. The compression characteristic of the output power versus
the input power is provided also in Figure 39 for a frequency of 240 GHz. The impact of
different DC drain bias conditions on the output power and P-1dB output referred
compression point, are compared in this graph. The saturated output power at the probe
tip, can be increased by 1 dB to 10 dBm when increasing the DC drain voltage from 1.0 V to
1.2 V. The loss due to the access lines from the probe tip to the amplifier cell was estimated
to be 1.0 dB. However, this access line for testing is replaced by a similar transmission line in
future integrated transceiver MMICs to connect to antennas or waveguide transitions. For
this reason, the measured output power equals the power that is also expected in a
transceiver circuit. The P-1dB output referred compression point was better than 4 dBm.
Figure 38: Chip photograph of the fabricated 220 -260 GHz power amplifier for the DL frequency band, left 3LPP design, right 4L design with MET4 layers (before MET4 processing)
.
Figure 39: On-wafer measured RF performance of the 220 -260 GHz power amplifier, left S-parameters, right output power at 240 GHz.
4.4 Conclusions and Outlook
The presented work in progress has covered three objectives of WP5 towards new THz
frontend technologies. The first objective is the development a new BEOL process that
addresses the requirement to increase the functional integration density of current THz
frontends. A first version of this BEOL process was established but needs further testing at
circuit design level to eliminate all potential bugs. Instead of the initially proposed 4L BEOL,
the 3LPP BEOL was established, after first experiments with the 4L BEOL. The 3LPP BEOL has
actually several advantages in comparison to the initial proposal. The main advantage is its
compatibility to various packaging processes including split-block waveguide packages. The
second objective was the development of first building blocks for the implementation of
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new transceiver architectures. Here, accompanying the BOEL process development,
functions for the DL were developed in a first step. In a next step, their modelling and the
design of a first library of RF building blocks for the UL will be designed and tested. The final
step will be the implementation of the transceiver solutions for the different proposed
demonstrators of WP6, including the beamforming demonstrator. From a timeline, this
seems to fit quite well, since the system proposals are gaining more detailed specifications,
while at the same time the THz IP core library is growing by all components required. The
third objective is the investigation of the analogue baseband interface for the TERRANOVA
media convert. First circuit candidates were designed, manufactured and compared. The
most promising candidate was the Kukielka amplifier concept, which was also tested as a 4-
channel version.
4.5 References
[4-1] IEEE Std 802.15.3d-2017, “Amendment 2: 100 Gb/s wireless switched point-to-point
physical layer,” IEEE Standards Association, 2017.
[4-2] T. Kuerner, and S. Rey, "IEEE 802.15.3d and other activities related to THz
Communications. Where to go next?," Towards Terahertz Communications
Workshop, European Commission, 7 March 2018.
[4-3] X. Li and M. Ismail., "Multi-Standard CMOS Wireless Receivers: Analysis and Design,"
The Springer International Series in Engineering and Computer Science., no. 675,
2002.
[4-4] Y. Tsai, C.-P. Yen and X. Wang, "Blind frequency-dependent I/Q imbalance
compensation for direct-conversion receivers," IEEE Trans. Wireless Commun., vol.
9, no. 6, pp. 1976-1986, 2010.
[4-5] A.-A. A. Boulogeorgos and G. K. Karagiannidis, "Low-cost Cognitive Radios against
Spectrum Scarcity," IEEE Technical Committee on Cognitive Networks Newsletter,
vol. 3, no. 2, pp. 30-34, 2017.
[4-6] L. Antilla, M. Valkama and R. M., "Circularity-based I/Q im- balance compensation in
wideband direct-conversion receivers," IEEE Trans. Veh. Technol., vol. 57, no. 4, pp.
2099-2113, 2008.
[4-7] A.-A. A. Boulogeorgos, A. Alexiou, T. Merkle, S. C., R. Elschner, A. Katsiotis, P.
Stavrianos, D. Kritharidis, P.-K. Charsias, J. Kokkoniemi, M. Juntii, J. Lehtomaki, A.
Teixeira and F. Rodreigues, "Terahertz Technologies to Deliver Optical Network
Quality of Expereience in Wireless System Beyond 5G," IEEE Communications
Magazine, 2018.
[4-8] H. Zamat and C. R. Nassar, "Introducing software defined radio to 4G wireless:
Necessity, advantage, and impediment," J. Commun. Netw., vol. 4, no. 4, pp. 1-7,
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[4-9] A.-A. A. Boulogeorgos, P. C. Sofotasios, S. Muhaidat, M. Valkama and G. K.
Karagiannidis, "The effects of RF impairments in Vehicle- to-Vehicle
communications," in IEEE 25th International Symposium on Personal, Indoor and
Mobile Radio Communications - (PIMRC): Fundamentals and PHY (IEEE PIMRC 2015 -
Fundamentals and PHY), Hong Kong, P.R. China, 2015.
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5. BASEBAND DIGITAL SIGNAL PROCESSING FOR THZ
SYSTEMS
In this Section 5, we describe the work in progress and the first experimental results for Task
5.3 entitled “Baseband signal and code design for THz systems”. In particular, the main goals
of the first investigations were: (1) the basic characterization of the THz frontend from a
system point of view, (2) the validation of the prototype single-carrier digital signal
processing algorithms for the THz link, and (3) the assessment of the maximum achievable
bitrate over the THz P2P link with the current module generation.
5.1 THz P2P transmission experiments
As a first step, experiments at Fraunhofer HHI were performed for the simplest case, i.e. for
the pure THz P2P link without additional optical links. In this investigation, the currently
available THz module generation from Fraunhofer IAF was used. In the following, the
experimental setup, the DSP used for evaluating the system, and the experimental results
will be presented.
Figure 40: Experimental setup for tests of THz P2P link
Figure 41: Photograph of first lab setup at Fraunhofer HHI
DAC Att.
Att.
Tx module x12 x3
f = 8.11 – 8.71 GHz
f = 2.625 GHz
φ d link
Rx module
Scope
x12 x3
Offline DSP
10 MHz Reference
Att
.
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5.1.1 Experimental setup
The construction of the first THz setup is illustrated in the block diagram in Figure 40 and the
photograph in Figure 41. The main experimental parameters are summarized in Table 1.
Table 1: Main experimental parameters
Symbol rate 16 GBd 32 GBd
Modulation format 16QAM
Overhead 28% (FEC + Framing)
Forward Error Correction SD-FEC with threshold BER of 2·10-2
Pulse shape RRC 0.35
Spectral width 21.6 GHz 43.2 GHz
Gross bit rate 64 Gb/s 128 Gb/s
Net bit rate 50 Gb/s 100 Gb/s
The baseband transmitter comprised a digital-to-analog converter (DAC) operating at 84
GS/s (8 bit, 3-dB bandwidth of about 25 GHz) which generated two electrical signals
representing the in-phase (I) and quadrature components (Q) of a single-polarization
quadrature amplitude modulation (QAM) signal. Throughout the investigations, 16QAM
modulation was used due to its higher spectral efficiency as compared to 4QAM (4 bits per
symbol for 16QAM vs. 2 bits per symbol for 4QAM) and the higher vulnerability to
component nonlinearities. The symbol rates under test were chosen to be 16 and 32 GBd,
which correspond to 64 Gb/s and 128 Gb/s raw bit rate, respectively. Assuming the use of a
soft-decision forward-error correction (SD-FEC) scheme with a threshold BER of 2·10-2, the
net bit rates correspond to 50 and 100 Gb/s, respectively. To limit the spectral extent of the
modulated signal without imposing a large peak-to-average power ratio (PAPR), a root-
raised cosine (RRC) pulse shape was used with a moderate roll-off value of 0.35. Digital pre-
emphasis could be applied to compensate for transmitter bandwidth limitations.
The DAC output signals were then applied to the I and Q inputs of the TX module comprising
a direct-conversion I/Q mixer. The output swing was in the order of 500mVpp. In order to
operate the mixer in the linear regime, fixed 3 dB attenuators were placed at the output of
the DAC. For characterization purposes, the drive signal swing could be further attenuated
by digitally reducing the DAC output amplitude at the cost of DAC resolution.
For the generation of the local oscillators, a tunable clock signal was upconverted to
approximately 300 GHz using a frequency multiplier-by-12 followed by frequency multiplier-
by-3. Initially, the clock signal was split and fed both into the Tx and Rx path to have a stable
clock relation. This allowed, by means of a manually-set phase delay φ, to control the
mapping of the transmitter I/Q components to the receiver I/Q components in most
experiments. In later experiments, the setup was changed to use two independent clock
generators in order to also test the performance with free-running carrier frequencies (not
shown in Figure 40).
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Table 2: Relation of clock and carrier frequencies used in the experiment
Clock signal Generated carrier frequency
8.11 GHz 291.96 GHz
8.15 GHz 293.4 GHz
8.21 GHz 295.56 GHz
8.31 GHz 299.16 GHz
8.41 GHz 302.76 GHz
8.51 GHz 306.36 GHz
8.61 GHz 309.96 GHz
8.71 GHz 313.56 GHz
The THz P2P transmission link comprised two horn antennas with 23 dBi gain each and a 58
cm-long free-space link. In order to emulate longer link distances, a manually tunable
attenuator could be used. In the Rx module, the received signal was converted back to
baseband using a direct-conversion I/Q mixer. The I and Q outputs were captured by a digital
storage oscilloscope with 80 GS/s (8 bit, 32-GHz bandwidth), which was synchronized with
the DAC using a 10-MHz reference clock.
Finally, digital signal processing was performed offline as described in the next section.
5.1.2 Used Digital Signal Processing
The block diagram of the utilized digital signal processing is shown in Figure 42. The
transmitter DSP comprised blocks for the generation of a random bit sequence, the mapping
to the 16QAM constellation, the insertion of training sequences [5-1] at the beginning of
each frame, pulse shaping (i.e. RRC with 0.35 roll-off) and pre-emphasis. The receiver DSP
comprises blocks for receiver-side front-end I/Q corrections [5-2], a training-aided block [5-
1] including: frame synchronization, carrier-frequency offset compensation and T/2-spaced
equalization (i.e. feed-forward, 51 taps), carrier-phase estimation (i.e. blind-phase search [5-
3] with 32 angles and 128 symbols per block), T-spaced equalization of Tx I/Q impairments
[5-4] (i.e. LMS adaption, 201 taps), demapping, decision and, finally, BER counting. The main
parameters for the DSP are also summarized in Table 3.
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Figure 42: Block diagram of the digital signal processing
Table 3: Parameters for digital signal processing blocks
Parameter Value
Sequence length per frame 215 symbols
Length of training sequence 448 symbols
Tap number of T/2 spaced equalizer 51 taps
Block length for carrier-phase estimation 128 symbols
Number of test angles for carrier-phase estimation 32 angles
Tap number of T-spaced I/Q equalizer 201 taps
5.2 First THz System Measurement Results
The following subsection presents the first measurement results. Throughout the
experiments, the antennas were optimally aligned. In the first step, a symbol rate of 16 GBd
was chosen, corresponding to a net bit rate of 50 Gb/s. Figure 43 shows the measured BER
as a function of the digital DAC amplitudes of I and Q (127 was the maximum value for a
resolution of 8 bit). The curves were extracted without utilizing pre-emphasis and at a carrier
frequency of 306.36 GHz. The Tx output was maximized (no additional THz attenuation) and
the phase between the Tx and Rx local oscillator was optimized (optimal Rx I/Q orientation).
The curves indicate that the optimum DAC amplitudes are 75 and 70 for the I and Q
components, respectively, indicating a small Tx I/Q imbalance. In this case, a BER of 1·10-3
was achieved which is well below the threshold BER of the SD-FEC, i.e. error-free
transmission with 50 Gb/s net rate can be expected after FEC decoding. For higher DAC
amplitudes, the performance degrades, which is attributed to nonlinear distortion due to
saturation of the direct-conversion mixer. For lower DAC amplitudes, the degradation is
attributed to a lower SNR.
Map
pin
g
Trai
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ence
inse
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n
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Pre
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el
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DA
C
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orr
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on
s
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e S
ynch
ron
izat
ion
Car
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-off
set
com
pen
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on
Trai
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ided
T/2
-sp
aced
Eq
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izer
Car
rier
-Ph
ase
Esti
mat
ion
Car
rier
-Ph
ase
Esti
mat
ion
T-sp
aced
I/Q
Eq
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Figure 43: BER vs. Digital DAC Amplitude (I and Q component) for 16-GBd using 16QAM (without pre-emphasis, at maximum Tx output power, optimal Ry I/Q orientation, at a
carrier frequency of 306.36 GHz)
Figure 44: BER vs. Digital DAC Amplitude (I and Q component) for 16-GBd 16QAM (with pre-emphasis, maximum Tx output power, optimal Rx I/Q orientation, 306.36 GHz carrier
frequency)
In Figure 44, results from similar measurements are plotted, but now a digital pre-emphasis was applied in order to flatten the Tx transfer function. In this case, the optimal digital DAC amplitudes are shifted towards higher values, and the BER is improved to 6.5·10-4, i.e. by almost a factor of 2. The shift of the optimal DAC amplitudes is attributed to the smaller average power of the digital waveform, as the digital pre-emphasis is realized by attenuating the lower frequency components in order to raise the level of the higher frequency components.
1,00E-04
1,00E-03
1,00E-02
55 65 75 85 95 105
BER
Digital DAC Amplitude (Quadrature Component)
60
65
70
75
80
85
90
95
100
105
Digital DAC Amplitude(Inphase Component)
1,00E-04
1,00E-03
1,00E-02
55 65 75 85 95 105
BER
Digital DAC Amplitude (Quadrature Component)
60
65
70
75
80
85
90
95
100
105
Digital DAC Amplitude(Inphase Component)
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16 GBd 16 QAM without pre-emphasis 16 GBd 16 QAM with pre-emphasis
Figure 45: Received constellations under best case conditions. (a) 16 GBd 16QAM without pre-emphasis @ BER = 1·10-3. (b) 16 GBd 16QAM with pre-emphasis @ BER = 6.5·10-4.
Figure 46: BER vs. Carrier Frequency for 16 GBd 16QAM (no pre-emphasis, maximum Tx output power, optimal Rx I/Q orientation, optimal digital DAC amplitude)
Figure 46 shows the dependence of the optimal BER on the carrier frequency, without pre-
emphasis. There are two distinct minima at 295.56 GHz and 306.36 GHz, while the BER is
slightly increased within this frequency range. Outside of this range, the BER is increasing
monotonically. The operation window with a BER below 1·10-2 is about 20 GHz, i.e. from
293.4 GHz to 313.56 GHz.
1,00E-03
1,00E-02
1,00E-01
290,00 295,00 300,00 305,00 310,00 315,00
BER
Carrier frequency [GHz]
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Figure 47: BER vs. Rx I/Q orientation for 16-GBd 16QAM (different carrier frequencies, no pre-emphasis, maximum Tx output power, optimal digital DAC amplitudes)
Figure 47 shows the dependence of the BER on the Rx I/Q orientation angle, without pre-
emphasis, as controlled by the manual phase shifter in the Tx local oscillator path. It can be
observed that the BER is strongly depends on the orientation angle while the angles with
best and worst performance are independent of the carrier frequency. The reason for this is
the Rx mixer saturation as the peak amplitudes of the I and Q components at the Rx depend
on the I/Q orientation. This is schematically depicted in subfigures (a) and (b) of Figure 48,
where it is shown how the received I and Q peak amplitudes are increased by a factor of
in case of a Tx-Rx misalignment with an orientation angle of 45°, i.e. the peak power
increases by a factor of 2. In Figure 48 (c) and (d), measured RX constellations are shown at
an orientation angle giving the best case BER (c) and the worst case BER in (d). To illustrate
the effect, only the I component was transmitted in both cases. As expected from the
discussion above, the measured orientation angle corresponds to 0° when a best case BER is
achieved, while for the worst case BER, the orientation angle equals 45°.
Figure 48: Discussion of Rx I/Q orientation. (a) Schematic of Rx aligned QAM constellation (orientation angle = 0°). (b) Schematic of Rx misaligned QAM constellation (orientation angle = 45°). (c) Measured Rx constellation (only I component at Tx) at an orientation angle giving the best case BER. (d) Measured Rx constellation (only I component at Tx) at an orientation
angle giving the worst case BER.
1,00E-03
1,00E-02
1,00E-01
0 0,2 0,4 0,6 0,8 1 1,2 1,4 1,6 1,8 2
BER
RX I/Q Orientation Angle [rad/(pi/4)]
291,96
293,4
295,56
299,16
302,76
306,36
309,96
313,56
Carrier Frequency[GHz]
A I
Q
A A
I
Q
A
(a) (b) (c) (d)
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Figure 49: BER vs. THz attenuation for 16-GBd 16QAM (306.36 GHz carrier frequency, with pre-emphasis, optimal digital DAC amplitudes, optimal Rx I/Q orientation)
Figure 49 shows the dependence of the BER on the additional attenuation in the THz free-
space link for different digital DAC amplitudes (each optimized in terms of Tx I/Q imbalance)
and with digital pre-emphasis. It can be observed that, due to the high margin towards the
threshold BER of the SD-FEC, significant additional attenuation can be tolerated, indicating
the potential for an increased free-space distance.
To further investigate the performance of the THz system, the setup was reconfigured in
such a way that two independent synthesizers were connected to the transmitter and
receiver modules, respectively. Since the frequency of the synthesizers’ clock signals drifts
slightly over time, random phase noise is introduced on the received constellation resulting
in a time-dependent RX I/Q orientation angle. Its compensation requires a carrier phase
estimation algorithm. Furthermore, considering that both devices are not synchronized,
there usually exists a small difference in the frequency both transmitter and receiver
modules operate on, which is denoted as frequency offset. It translates into a fast rotation
of the received constellation over time. In order to be able to receive the transmitted signal
correctly, this offset has to be compensated by means of a frequency-offset compensation
algorithm. The performance the applied DSP compensation methods for both phase noise
and carrier frequency offset is depicted in Figure 50. It can be observed that the DSP is able
to correctly compensate up to a discrepancy of almost 4 GHz between the operating
frequencies of the transmitter and receiver modules, resulting in a slight BER degradation
when compared to the previous experiments. All DSP parameters remained the same except
for two: the block length for carrier phase estimation was set to 2048 symbols and the
number of test angles for carrier-phase estimation was changed to 64. This was done to
better compensate the increased phase noise experienced by the data samples in this case
and improve the overall performance of the system.
1,00E-04
1,00E-03
1,00E-02
1,00E-01
27,532,537,542,547,5
BER
Attenuator setting
70/60
75/65
85/75
85/75
85/80
90/85
95/90
100/95
105/100
105/105
Digital DAC
Amplitudes (I/Q)
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Figure 50: BER vs. RX frequency offset for 16-GBd 16QAM (306.36 GHz carrier frequency, with pre-emphasis, maximum TX output power, optimal digital DAC amplitudes)
Finally, Figure 51 shows measurement results for the transmission of a 32-GBd 16QAM
signal with a net data rate of 100 Gb/s. For this experiment, however, the setup was rolled
back to the one-synthesizer configuration, where a single device generates the clock signal,
which is then fed both to the transmitter and receiver modules. Pre-emphasis was applied in
order to optimize the transmit spectrum. Compared to the results for 16 GBd shown in
Figure 44, the optimal digital DAC amplitude is shifted towards higher values with optimal
performance at the maximum available amplitude. Thus, the performance might be further
optimized by removing the fixed 3-dB attenuators at the DAC output. This shift in required
DAC amplitude is attributed to the enhanced pre-emphasis as compared to the case for 16-
GBd. At the optimal DAC amplitudes of 125/125, a BER of 1.1·10-2 was measured. The
corresponding constellation is shown in Figure 52. This means that an error-free
transmission with a net data rate of 100 Gb/s was shown in this experiment.
Figure 51: BER vs. Digital DAC Amplitude (I and Q component) for 32-GBd 16QAM (with pre-emphasis, maximum Tx output power, optimal Rx I/Q orientation, 306.36 GHz carrier
frequency)
1,00E-02
1,50E-02
2,00E-02
2,50E-02
3,00E-02
75 80 85 90 95 100 105 110 115 120 125 130
BER
Digital DAC Amplitude (Quadrature Component)
80
85
90
95
100
105
110
115
120
125
Digital DAC Amplitude(Inphase Component)
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Figure 52: Received constellation under best case conditions for 32-GBd 16QAM with pre-emphasis @ BER = 1.1·10-2.
5.3 Conclusions and Outlook
In this section, we documented the first TERRANOVA experiments on single-carrier high
symbol rate transmission with high spectral efficiency over a THz free-space link. As a main
result, the applicability of the prototype single-carrier DSP to the THz wireless transmission
was demonstrated under realistic experimental conditions. In particular, the DSP algorithms
were shown to efficiently combat bandwidth limitations, I/Q impairments, frequency offset
and phase noise. Along with the basic characterization results, several issues with current
THz module generation were found, e.g. the dependence of the BER on the carrier frequency
as well as on the Rx I/Q orientation angle, which will be addressed in the next generation
chips as well as through further calibration measurements and procedures. Nevertheless, an
error-free 100 Gb/s THz transmission using a 32-GBd 16QAM signal on a 300 GHz carrier was
demonstrated with the current module generation, showing the high potential of this
technology to reach the envisioned project goals.
5.4 References
[5-1] R. Elschner, F. Frey, C. Meuer, J. K. Fischer, S. Alreesh, C. Schmidt-langhorst, L. Molle,
T. Tanimura, and C. Schubert, “Experimental demonstration of a format-flexible
single-carrier coherent receiver using data-aided digital signal processing,” Opt. Exp.,
vol. 20, no. 27, pp. 28786–28791, Dec. 2012.
[5-2] I. Fatadin, S. Savory, and D. Ives, “Compensation of quadrature imbalance in an
optical QPSK coherent receiver,” IEEE Photon. Technol. Lett. 20(20), 1733–1735
(2008).
[5-3] T. Pfau, S. Hoffmann, and R. Noé, “Hardware-efficient coherent digital receiver
concept with feedforward carrier recovery for m-QAM constellations,” J. Lightwave
Technol. 27(8), 989–999 (2009).
[5-4] C. R. S. Fludger and T. Kupfer, “Transmitter Impairment Mitigation and Monitoring
for High Baud-Rate, High Order Modulation Systems”, Proc. ECOC 2016, p.Tu.2.A.2.
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6. PHASED ARRAY BEAMFORMING
6.1 State-of-the-Art in Phased Array Beamforming techniques
Beamforming emphasizes signals in the desired direction while suppresses signals in other
directions. An array of antenna elements (AEs) is required in order to implement
beamforming and concentrate the transmitted/received power towards a particular
direction and, thus, to increase the antenna gain. The ideal baseband digital beamforming
requires one distinct radio frequency (RF) chain per antenna element. However, this
requirement is prohibitive in terms of complexity, because mmWave as well as THz
communications potentially utilize large antenna arrays to compensate the severe path
losses with highly directional transmissions [6-1]. To this end, hybrid beamforming has
gained the interest of the scientific community with the ideal requirement of using as many
RF chains as the desired data streams. Hybrid beamforming combines digital with analogue
beamforming and can achieve a performance close to the ideal fully digital beamforming
with the benefit of reduced hardware complexity and power consumption [6-2], [6-3], [6-4].
In the analogue domain, the phase shifters more commonly found in the literature are
digital, due to their immunity to noise present in voltage control lines. Interestingly, Chen et
al. in [6-5] stated that graphene transmission lines have low insertion losses in the THz
frequencies. There is ongoing research on the phase shifters to be utilized at high carrier
frequencies [6-6], while the key challenges include the insertion losses, the cost and the
complexity of their design [6-7]. Analogue beamforming is cheaper, simpler and less power
consuming than digital beamforming, but cannot be applied at wideband systems due to the
frequency-dependent nature of the phase shifters. Digital beamforming can be performed
over wide bandwidths, both in the time domain with tapped delay line filters and in the
frequency domain by using a fast Fourier transform. The research interest in digital
beamforming concentrates on algorithms to improve the computation time and the signal
tracking.
In [6-8], it has been proven that the optimal architecture for hybrid beamforming should
meet a trade-off between complexity and performance, while it highly depends on the
application and the channel conditions. Interestingly, determining the optimal number of RF
chains is a complex, but very challenging optimization problem [6-3]. In the literature, there
exist two main hybrid beamforming architectures: a) the full-connected, where each RF
chain is connected to all the AEs and b) the sub-array, where each RF chain is connected to a
group of AEs [6-3], [6-1]. Another possible hybrid beamforming architecture referred to as
virtual sectorization structure can be found in [6-8]. This structure creates multiple “virtual
sectors” in the digital domain, so that each set of RF chains is processed separately in
baseband in order to reduce overhead and complexity. Interestingly, 3GPP Release 13 for
LTE-Advanced Pro [6-9] is related to the sub-array hybrid beamforming architecture, while
Release 14 [6-10] includes developments that are closer to the 3D hybrid beamforming, i.e.
higher resolution.
The THz beamforming demonstrator will most likely consist of four horn antenna elements.
Since hybrid beamforming is a meaningful compromise in terms of complexity when the
number of antenna elements is large, digital or analogue beamforming are more suitable in
our case. Besides, motivated by the fact that using four RF chains is not prohibitive, digital
beamforming in the baseband seems to be a promising technique for the demonstrator. As
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already mentioned, digital beamforming results to better performance, offers more degrees
of freedom and is superior in terms of reliability compared to analogue beamforming.
6.2 Mathematical Model of Phased Array Architectures
The THz beamforming demonstrator consists of an RF-frontend that operates in the
frequency band above 275 GHz. The relationship of the signal bandwidth with the carrier
frequency determines whether a system is considered as wideband or narrowband. When
the bandwidth of the signal is adequately small compared to the carrier frequency, the
system is considered narrowband. This assumption is not well defined and many variations
exist in the literature. In general, it holds true when the signal bandwidth is less than one
percent of the carrier frequency. In our case, the signal bandwidth, B, of the mmWave
modem is inherently small compared to the carrier frequency, fc (which is around 300 GHz),
and thus the system is considered narrowband. Interestingly, this assumption allows us to
approximate the propagation delays of the impinging signal between the antenna elements
with phase shifts.
For the beamforming demonstrator, the study is concentrated on uniform linear arrays
(ULAs), where the AEs are located along a line and placed in equal distance one from the
other. This inter-element spacing, d, is considered the spatial equivalent of the time interval
and, similar to the Nyquist–Shannon sampling theorem, it should hold that d ≤ λ/2, in order
to avoid the formulation of grating lobes [6-11], [6-12]. Grating lobes are lobes that have the
same radiated power as the main lobe but in a different direction. However, lowering the
element spacing conflicts with the desire to have as large aperture as possible for a fixed
number of elements. To this end, we generally set d = λ/2, as in the case of a ULA the
aperture equals the distance between the first and the last element of the array.
The angle depicted in Figure 53, is the direction from which the signal is received, while
the propagation speed equals the light speed c. Since the antenna elements are equally
spaced, the propagation paths between two adjacent elements differ by d sin that results
in a time delay equal to . For a narrowband system, this delay corresponds to a
phase shift equal to 2 π d sin / λ.
In order to ease the analysis, the centre point of the phased antenna array has been utilized
as the reference point. Therefore, for a ULA with M AEs, the array response vector or
steering vector [6-12], if M is even, equals
where T stands for the transpose.
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Figure 53: Plane wave impinging on a ULA
6.3 Comparison of Beamforming Techniques
The following beamforming techniques equally concern transmit (Tx) and receive (Rx)
beamforming. It should be mentioned that adaptive beamforming is more commonly
applied at the receiver, while it can also be applied at the transmitter in the presence of
feedback. For uniformity reasons, we selected to focus on Rx beamforming in the presented
simulations. The desired signal corresponding to a specific direction, i.e. the angle , is
extracted by implementing a certain weighting on the received signals before adding them.
In the following, beamforming techniques candidates for implementation in the
demonstrator are presented and compared.
6.3.1 Conventional Beamforming
Conventional beamforming utilizes fixed and predefined phase shifts that are chosen
independent of the signal received by the antenna array [6-11]. The beamforming weight
vector is the conjugate transpose of the steering vector at divided by , i.e.,
where H stands for the conjugate or Hermitian transpose. The division with is introduced
to satisfy the following equation . It should be noted that in the case of a ULA,
the beamformer is a spatial filter having a direct analogy to an FIR frequency-selective filter
at the time domain. In order to evaluate the performance of a beamformer, we utilize the
beampattern which is equal to , where . In Figure 54, the
beampatterns for different and values are depicted. The larger lobe of the
beampattern is steered at and we refer to it as mainlobe, while the lower lobes are
known as sidelobes. Due to selection of antenna spacing, grating lobes are not present.
Conventional beamforming would be perfectly sufficient, if the only signal present at the
receiver aside from the additive thermal noise was the signal of interest from . However,
in many cases, there exist signals propagating in the same carrier frequency and impinging
on the array from different angles. We refer to these signals as interference. In many cases,
we utilize adaptive methods in order to identify and overcome these interferers. However,
conventional beamforming can be further optimized in order to mitigate the effect of
interference to some extent.
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Figure 54: Beampatterns for conventional beamforming
6.3.2 Tapered Beamforming
The phases of the weight vector steer the mainlobe to the desired direction and, thus,
emphasize impinging waves at this angle. It has been proven that the sidelobe levels of the
beampattern can be further reduced by tapering the weights of the conventional
beamformer vector. More specifically, a vector t with non-negative real values is utilized and
the tapered beamformer vector equals
,
where represents the Hadamard product, i.e., the element-by-element multiplication of
the two vectors. The vector t is determined by the selection of a proper window from the
existing ones in the literature, where the window length is set equal to the number of
antenna elements M. Besides, the taper vector values may be distributed according to the
binomial distribution. Utilizing the binomial distribution has the advantage of completely
eliminating the sidelobes with the drawback of significantly increasing the mainlobe width
(see Figure 55).
Fixed windows, such as the Hanning and the Hamming windows, result in a fixed main
sidelobe level, which does not depend on the number of elements. However, there also exist
windows, such as the Taylor, the Kaiser and the Dolph-Chebyshev windows, where the main
sidelobe level is a design parameter and can be selected. In Figure 56, various windows
are implemented and the corresponding beampatterns are extracted assuming a ULA with
16 AEs and a desired beamsteering angle . Interestingly, a tradeoff exists between
the main sidelobe level and the width of the mainlobe. More specifically, as the main
sidelobe level decreases, the mainlobe width increases. This interplay is one of the
important issues during the selection of the appropriate window, which is mainly driven by
the actual application.
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Figure 55: Beampatterns for tapered beamforming with binomial distribution and 4 AEs
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Figure 56: Beampatterns for tapered beamforming with various windows. The ULA consists of 16 elements and its steering direction is at in (d) is the number of nearly
constant-level sidelobes adjacent to the mainlobe
Below, there are some characteristics of the most useful windows.
The Hanning window corresponds to a constant -32 dB main sidelobe level, while it
reduces the number of the sidelobes.
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The Hamming window corresponds to a -43 dB main sidelobe level, but has more
sidelobes than the Hanning window.
The Blackman-Harris window almost eliminates the sidelobes, but results in a wide
mainlobe.
The Taylor window offers the ability to select the number of nearly constant-level
sidelobes adjacent to the mainlobe expressed by the symbol in Figure 56.
However, the adjacent sidelobes have lower level than the more distant ones.
The Dolph-Chebychev window has an “equiripple” behavior, which means that it
results in sidelobes that all have the same desired level.
The Kaiser window decreases the level of all the sidelobes without changing their
number. Moreover, it results in a lower mainlobe width increase when compared to
the Taylor and the Dolph-Chebyshev windows using the same desired main sidelobe
level.
Consequently, applying a window, results in a wider mainlobe with respect to the
conventional beamforming. Hence, the sidelobe level reduction can be achieved only at the
expense of the resolution. Furthermore, utilizing windows seems to be more efficient in the
case where many sidelobes exist and the mainlobe width is narrow. This holds true
particularly when an antenna array with an adequate number of antenna elements is
utilized.
6.3.3 Null-Steering Beamforming
When the direction of interference is known, should be selected so that nulls are
introduced in the direction of the interfering signals. If is the desired signal direction and
are the directions of the L interfering signals, should be selected to satisfy
These constraints can be rewritten
as , where and are defined as follows
By using the method of Lagrange multipliers, can be obtained from the following equation
In Figure 57, the derived beampattern of the null-steering beamforming with 16 AEs is
depicted. It can be easily observed that a null has been introduced at , both in the
case where the beamsteering direction is and . Interestingly, null-steering
beamforming may not always be a good design. More specifically, if the null direction is
close to the desired mainlobe direction and/or the number of antenna elements is small, the
steering capability of the ULA is tremendously reduced. This can be easily verified from
Figure 58, where introducing perfect nulls at the direction of interference has a detrimental
effect on the mainlobe, which does not point exactly at the desired direction.
6.3.4 Adaptive beamforming
In adaptive beamforming, the weight vector is derived by maximizing the theoretical signal-
to- interference-plus-noise ratio (SINR) and is implemented in the digital domain (i.e, digital
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beamforming). This design is considered more efficient in the sense that it balances between
the minimization of noise and interference from signals impinging on the array from
directions different than the desired one [6-12]. Maximizing the SINR is identical to
minimizing the interference-plus-noise power, which assuming a weight vector , equals
, where stands for the interference-plus-noise correlation matrix. In
practice, a priori knowledge of the second order statistics of the array data is not feasible. To
this end, techniques to estimate the unknown statistics from the received array signals are
required. An estimate of the interference-plus-noise correlation matrix requires that
no desired signal is present. To this end, the desired signal transmission is muted and
samples of the array received vector are collected. The estimate of the correlation matrix is
given by the equation below
,
where selecting a higher value for results in a better estimate of the correlation matrix.
Figure 57: Beampatterns for Null-steering beamforming with 16 AEs
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Figure 58: Beampatterns for Null-steering beamforming with 4 AEs and
6.3.4.1 Minimum Variance Distortionless Response (MVDR) Beamformer
The maximization of the SINR is achieved by solving the following optimization problem
where the constraint preserves the desired signal. The solution of this constrained
optimization problem is found by using Lagrange multipliers and is given by
6.3.4.2 Linearly Constrained Minimum Bariance (LCMV) Beamformer
Some applications may require additional conditions on the beamformer, which in most
cases concern the rejection of interference signals received from specific angles. If is the
desired signal direction and are the directions of the L interfering signals,
should satisfy the following equations
Thus, the LCMV beamforming minimizes the undesired signal output power of the
antenna array, while preserving or nulling the power in selected directions. The set of
constraints can be formulated in a proper equation as where is the constraint
matrix and the constraint response vector, which in our case are defined as follows
Hence, the constrained optimization problem can be expressed as
while the LCMV beamforming vector is given by
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In the following, the conventional beamformer is compared to the adaptive MVDR and
LCMV beamformers, when ULAs with 4 and 16 AEs are utilized, respectively. We assume that
the desired direction is at , while there exist two interference signals at
and with mean power 0.1 and 0.2, respectively. Finally, it has been assumed that
the existence of interference at is known, while the one at is not. To
this end, for the LCMV beamformer, it holds that
From Figure 59−Figure 63, it becomes evident that the adaptive beamformers outperform
the conventional one in the presence of interference. The SINRs of the two configurations
for the three different beamformers are listed in Table 4. It should be noted that the LCMV
beamformer completely eliminates the incoming interference at . Besides, when
compared to the MVDR beamformer, LCMV beamformer completely eliminates the
interference at but does not combat the interference at , as efficiently
as the MVDR beamformer (see Figure 61).
Table 4: SINR for various beamformers
SINR [dB] Conventional MVDR LCMV
4 AEs 4.58 4.99 5.16
16 AEs 12.05 12.35 12.35
Table 5: Output noise power for various beamformers
Output noise power Conventional MVDR LCMV
4 AEs 0.1149 0.1476 0.1314
16 AEs 0.0207 0.0221 0.0221
Table 6: Interference power for various beamformers
Interference power Conventional MVDR LCMV
4 AEs 0.1571 0.0267 0.0492
16 AEs 0.0058 0.0001 0.00002
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Figure 59: Magnitude of the beamformers’ output signals for a ULA with 4 AEs
Comparing Figure 59 and Figure 60 with Figure 62 and Figure 63, it is obvious that the
efficiency of the LCMV beamformer is higher compared to that of the MVDR beamformer,
when the number of utilized antenna elements is low. More specifically, it has been
observed that when 16 AEs are utilized, the MVDR and LCMV beamformers perform almost
identically. It should be emphasized that with adaptive beamforming, a balance between the
rejection of interference and the output thermal noise is achieved. Thus, the resulting
output noise does not cause a reduction in the SINR (see Table 4, Table 5 and Table 6).
Figure 60: Power pattern of beamformers for a ULA with 4 AEs
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Figure 61: Power pattern (rectangular) of beamformers for a ULA with 4 AEs
Figure 62: Magnitude of the beamformers’ output signals for a ULA with 16 AEs
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Figure 63: Power pattern of beamformers for a ULA with 16 AEs
6.4 Beamforming Implementation Issues of Demonstrators
6.4.1 Available Phase Shifter Resolution for Analogue Beamforming
Assuming that a discrete codebook has been designed to cover a large range for the desired
angle direction, each codeword of the codebook corresponds to a set of beamforming
phases, which should be applied by phase shifters [6-13]. When analogue beamforming is
implemented, the utilized phase shifters are of finite limited resolution. This indicates that
the applied phase shifts are not always equal to the ideal ones; instead the closer possible
value that is supported by the phase shifter is applied. As expected, this causes a variation in
the performance of the beamformer often referred to as quantization error, whose
importance is related to the resolution of the phase shifters. Hereby, it should be
emphasized than an -bit phase shifter has a resolution or step equal to .
From the beampatterns of Figure 64 and Figure 65, it is obvious that the phase shifter
resolution does not affect all the desired directions to the same extent, i.e., for the
beampattern is not affected at all because the optimal phase shifts are supported by the
phase shifters, i.e., they are integer values of . Furthermore, when the mainlobe width is
large, as in the case of 4 AEs, the difference in the performance can be considered negligible.
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Figure 64: Beampatterns for conventional beamforming with and without quantization assuming 4 AEs
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Figure 65: Beampatterns for conventional beamforming with and without quantization assuming 16 AEs
6.4.1 Differential Phase Noise in THz Phased Array Systems
The role of the local oscillator phase noise in phased array systems was experimentally
studied using a 4-channel testbed for the frequency range from 275 GHz to 325 GHz at the
initialization of the project [6-14]. The testbed itself was developed in the national funded
project TERAPAN. In this work, local oscillator phase shifting was initially identified as an
interesting solution for broadband phase shifting at THz frequencies. Although this leads to a
similar frontend architecture as required for digital phase shifting of baseband signals, the
main difference is that it would require only a single data stream and a single ADC/DAC. This
becomes attractive for THz high-data rate applications.
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Since low-phase noise fundamental oscillators at THz frequencies are difficult to realize, the
more promising approach is the frequency multiplication of a low frequency LO reference
signal. The generation of a local oscillator signal of set phase relation to a stable reference
crystal oscillator can be realized with the help of direct digital synthesis (DDS) circuits. By
synchronizing DDS circuits, a multi-channel local oscillator signal can be generated at very
low frequencies with set phase difference. Those LO signals can be multiplied in frequency
with integrated frequency multipliers to 300 GHz. The block diagram of this scheme is shown
in Figure 63 for the used 4-channel phased array testbed. The output frequency of the DDS
circuits was 2.083 GHz.
Figure 66: Functional block diagram of the 4 channel LO beamformer with DDS based phase shifting. The synchronous DAC output is achieved by a master trigger (omitted in this
drawing for clarity).
The measured phase noise of the individual channels at 8.333 GHz after frequency
multiplication-by-4 is shown in Figure 67.
Figure 67: Phase noise of the individual channels of the LO beamformer, measured at the output carrier frequency of 8.333 GHz. In comparison, the phase noise of a commercial
frequency synthesizer is shown. Carrier power Pc = -2 dBm in all cases.
The phase noise of the individual channels was nearly identical. While the single channel
phase noise was comparable to the phase noise of a commercial frequency synthesizer, the
differential phase noise between different channels, which becomes amplified by the
frequency multiplication, is very critical for the stability of phased array systems. Although
splitting the LO at very low frequencies is a rather worst case scenario, interconnect losses at
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THz frequencies do not allow the splitting and distribution of LO signals over longer
distances. The question on what differential phase noise can be tolerated, and what the
lowest frequency is to best split the LO signal to the individual channels is currently
unresolved and subject to the next step of this work. This question is for LO phase shifting
and digital phase shifting of the baseband signals the same.
The differential phase noise degrades the beamforming efficiency by modifying the shape of
the mainlobe. Eventually, the effect of phase noise is obvious in the reduction of the
achievable SINR and consequently, in the decrease of the achievable data rate. For the
current four channel receiver at 300 GHz, data transmission experiments with 1, 2, 3, and 4
receive channels were conducted at a symbol rate of 4 Gbaud. The performance of a single
channel is shown in Figure 68. For the transmission experiments with multiple channels, the
transmit power was reduced by a factor of 4. Figure 69 shows that the beamforming
efficiency is reduced and phase noise effects start to evolve with increasing the number of
receive channels though the EVM is numerically the same as in the single channel case
(about 13%rms).
Figure 68: Measured cumulative constellation diagrams and EVM of a single receive channel at 4 Gbaud for increasing QAM modulation depths. 100 constellation diagrams of 4096
symbols were accumulated in each plot.
Figure 69: Measured cumulative constellation diagrams and EVM of a single receive channel at 4 Gbaud for a 16-QAM modulation format, after initial calibration by null-steering. 100
constellation diagrams of 4096 symbols were accumulated in each plot.
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6.4.2 Four Element Horn Antenna Array
In the following, the horn antenna array utilized by the TERRANOVA demonstrator was
simulated in MATLAB. It should be emphasized that the carrier frequency is set equal to 200
GHz, which is the maximum permitted in MATLAB, as the actual value (i.e., 300 GHz) is not
supported. Therefore, these results are only indicative of the way the demonstrator antenna
array is expected to work. The dimensions of the antenna elements are summarized in Table
7, while the inter-element distance is set equal to C+0.25 mm.
Table 7: Horn antenna element dimensions
Figure variables Dimension Value
B Horn aperture width 3 mm
C Horn aperture height 1 mm
F Horn flare length 3.577 mm
D WR3 wave guide width 0.8640 mm
E WR3 wave guide height 0.4320 mm
d Inter-element spacing C+ 0.25 mm = 1.25 mm
Figure 70: Horn antenna element
Figure 71: Directivity pattern of the horn antenna element
In Figure 71, the directivity pattern of a horn antenna element is depicted, when its horn
aperture width is placed along the z axis. Placing the horn antennas as depicted in the left
bottom part of Figure 73 and steering the array at 0⁰, the directivity pattern of every single
element is depicted in Figure 72, while that of the array is depicted in Figure 73. Obviously,
the directivity patterns of the middle and the edge elements are symmetric to each other,
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respectively. Moreover, beamsteering is implemented in the elevation plane as depicted in
Figure 74. However, given that λ=1mm at 300GHz, the fact that the inter-element distance is
higher than λ/2 degrades the efficiency of the beamforming as grating lobes are created.
Through these indicative simulations, it became evident that the steering range of the
demonstrator’s array is not expected to be higher than [-45⁰, +45⁰].
Figure 72: Directivity pattern of the four elements of the ULA
Figure 73: Directivity radiation pattern of a ULA with 4 AEs pointing at
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(a) Coordinate system: polar
(b) Coordinate system: rectangular
Figure 74: Directivity radiation patterns of a ULA with 4 AEs pointing at
6.4.3 Beam Search and Alignment
In order to implement beamforming, it is required that the transmitter establishes a link
with the receiver, i.e. their beams are steered pointing at each other. Communication links
established at THz frequencies are directional, which results in the need for a different
procedure that the ones commonly used in the microwave systems where synchronization
signals are broadcasted omnidirectionally. As the demonstrator consists of 4 AEs in a ULA,
the beamsteering will take place either in the azimuth or the elevation direction. Moreover,
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the scanning range of the array to significantly less than 180⁰ due to the inter-element
spacing and the utilization of directional horn antennas as AEs.
Designing a codebook, which contains multiple beams in order to cover the feasible angular
space (i.e., the scanning range of the array), makes a directional search possible. To this end,
an exhaustive search will be implemented by the demonstrator [6-15]. More precisely,
utilizing all the codewords one-by-one, the demonstrator will scan the whole space and
determine the codeword, and thus the direction, that results in the highest received SINR
value. Although this method is regarded as time consuming, it is efficient to be utilized by
the demonstrator mainly due to the small codebook size. Interestingly, as the half power
beamwidth of the four element horn array is expected to be close to 15⁰, the number of
codewords will probably be less than 15.
Hence, the beam search procedure significantly depends on the beamforming algorithm that
will be implemented at the demonstrator.
6.5 Calibration Techniques for Phased Array Antennas
In this section, a survey on calibration techniques for phased antenna arrays operating at
mmWave and lower frequencies is presented. It should be noted that techniques for the
calibration of systems operating at THz frequencies have not been presented in the
literature yet. During the calibration, the amplitude and phase degradations of each array
element are measured and aligned in order for the array to match the requirements in
accuracy. Thus, the general goal is to ensure that the amplitude and phases applied to each
element are corrected in order to derive the desired radiation pattern.
The calibration technique described in [6-16] is called mutual coupling method (MCM) and
can be implemented in arrays with uniformly spaced elements that have symmetric
radiation patterns. MCM measures the mutual coupling between all adjacent elements and
aligns the insertion attenuation and phase of the array. The rotating element electric field
vector (REV) method presented in [6-17] is a practical self-calibration method for a phased
array, based on measurements of the array’s power variation while each element phase is
shifted. A faster, more steady and accurate method has been proposed in [6-18], where
calibration is achieved by sequentially measuring the near-field mutual coupling signal
between each active antenna element and the passive element used for calibration. The
measured data, i.e. the ones calculated in orbit, is compared with the stored data, i.e. the
reference values supplied at the manufacturing, to make the required compensations.
Two novel on-site calibration algorithms that compensate the mutual coupling effect as well
as gain, phase and location errors of active antenna arrays in satellite communications have
been presented in [6-19]. The experimental demonstration has been conducted at the L-
band (around 1.6 GHz). Besides, the calibration technique in [6-20] is based on digital signal
processing of the received signal and is intended for a satellite acquisition system. The array
composed of eight radiating elements operating in the S-band (i.e., 2.2–2.3 GHz). The
realization of a specific radiation pattern is achieved by properly adjusting the elements’
phase shifters, attenuators, and the division ratios of the power dividers to provide the
required excitations in phase and amplitude. A genetic algorithm (GA) with an antenna
measurement facility has been implemented in [6-21] for the optimization of the radiation
patterns by using the actual phase and amplitude states of the RF components.
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A cost-efficient calibration concept based on semi-passive backscatter transponders and an
internal power detection circuitry has been proposed in [6-22]. Using external feedback
through BTs, a fully flexible array design is achieved that scales well for an increasing
number of antenna elements. Thus, this approach is suited for online calibration of massive
MIMO systems. In [6-23], an amplitude-only measurement method for phased array
calibration has been proposed, measuring the signals (i.e., complex electric field) of
individual antenna elements. This method has the advantage of minimum measuring time,
while the demonstrated system is operating at 2 GHz. In [6-24], all the antenna element
signals are simultaneously measured by an auxiliary antenna as the antenna element phase
settings change. The calibration method is based on calculating the antenna elements’
excitations by solving linear equations.
Low-cost manufacturing tests and calibration procedures of phased arrays are critical to
enable applications at mmWave and higher frequency bands. A built-in self-test technique
for phased arrays has been proposed in [6-25]. Orthogonal code modulation is applied to
each antenna element and parallel measurements are conducted. The test signals are then
down-converted to a baseband interference signal consisting of code-modulated complex
cross correlations between all element signals. Using orthogonal code products, each cross
correlation is extracted from the interference signal and used to obtain amplitude and phase
data of each element. The demonstrated system of [6-25] is operating at 57 to 67 GHz. A 60
GHz phased array with four elements employing LO phase shifting has been studied in [6-
26]. Beamforming calibration has been achieved by performing gain equalization, I/Q
calibration and successive –approximation phase tuning.
Digital beamforming provides enhanced calibration capabilities, resulting in ultralow
sidelobes and wideband equalization [6-27]. A wideband closed-loop adaptive calibration
method for digital beamformers has been presented in [6-28] for operation from L- to X-
band. The channels are equalized relative to a selected channel before the implementation
of beamforming, using an internally distributed and injected additive white Gaussian noise
source. In [6-29], hybrid beamforming is implemented at the receiver side with a calibration
technique referred to as image injection (IMI). In IMI, the nonlinear behavior of the mixer is
utilized and the inherent LO distribution network is multiplexed as the injection path of the
pilot signal. The IMI calibration is implemented in real time at the IF stage and targets the
electrical performance variation of the RF circuit, thus, no dedicated transmitter is required.
6.6 Conclusions and Outlook
Digital and analogue phase shifting techniques have been discussed and evaluated through
simulations. Although the most prominent technique in the recent literature is the hybrid
beamforming, this technique is not appropriate the THz beamforming demonstrator based
on the existing antenna array. Besides, analogue phase shifters for the frequency band
between 275 and 300 GHz are not fully studied in the literature and are prone to severe
phase noise. This is expected to degrade the system performance in terms of both the
beamforming efficiency and the achievable data rate. To this end, the effect of phase noise
and other impairments introduced by the analogue components will be further examined in
the study to be conducted in the forthcoming months. Moreover, the existing system setup
and the conducted simulations reveal that digital beamforming is expected to perform
better for the THz beamforming demonstrator. This approach proposes utilizing element-
level processing with a dedicated RF chain at each antenna element. Based on the
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conducted evaluation and the derived results, digital beamforming algorithms will be further
investigated. It should be emphasized that the restrictions introduced by the digital and
analogue components will be considered in order to implement the most efficient
beamforming technique for the demonstrator.
6.7 References
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analog and digital beamforming for millimeter wave 5G,” IEEE Commun. Mag., vol.
53, no. 1, pp. 186–194, Jan. 2015.
[6-2] J. G. Andrews, T. Bai, M. Kulkarni, A. Alkhateeb, A. Gupta, and R. W. Heath,
“Modeling and Analyzing Millimeter Wave Cellular Systems,” IEEE Trans. Commun.,
vol. 65, no. 1, pp. 403–430, Jan. 2017.
[6-3] S. Kutty, S. Member, and D. Sen, “Beamforming for millimeter wave
communications: An inclusive survey,” IEEE Commun. Surv. Tutorials, vol. 18, no. 2,
pp. 949–973, Secondquarter 2016.
[6-4] T. E. Bogale and L. B. Le, “Beamforming for Multiuser Massive MIMO Systems: Digital
versus Hybrid Analog-Digital,” in Proc. IEEE Glob. Commun. Conf., Austin, TX, 2014,
pp. 4066–4071.
[6-5] P. Y. Chen, C. Argyropoulos, and A. Alu, “Terahertz antenna phase shifters using
integrally gated graphene transmission-lines,” IEEE Trans. Antennas Propag., Vol. 61,
No. 4, 1528–1537, Apr. 2013.
[6-6] F. Sohrabi and W. Yu, “Hybrid digital and analog beamforming design for large-scale
antenna arrays,” IEEE J. Sel. Topics Signal Process., vol. 10, no. 3, pp. 501–513, Apr.
2016.
[6-7] S. Payami, M. Ghoraishi, and M. Dianati, “Hybrid Beamforming for Large Antenna
Arrays With Phase Shifter Selection,” IEEE Trans. Wirel. Commun., vol. 15, no. 11, pp.
7258–7271, Nov. 2016.
[6-8] A. F. Molisch, V. V. Ratnam, S. Han, Z. Li, S. L. H. Nguyen, L. Li, and K. Haneda, "Hybrid
Beamforming for Massive MIMO: A Survey," IEEE Communications Magazine, vol.
55, no. 9, pp. 134-141, Sept. 2017.
[6-9] H. Ji, Y. Kim, J. Lee, E. Onggosanusi, Y. Nam, J. Zhang, B. Lee, and B. Shim, “Overview
of Full-Dimension MIMO in LTE-Advanced Pro,” IEEE Commun. Mag., vol. 55, no. 2,
pp. 176–184, Feb. 2017.
[6-10] 3GPP, “Release 14,” 2017. [Online]. Available: http://www.3gpp.org/release-14
[6-11] H. L. Van Trees, “Part IV of estimation and modulation theory: Optimum array
processing,” John Wiley and Sons, 2004.
[6-12] L. C. Godara, “Smart antennas,” CRC Press, 2004.
[6-13] J. Singh and S. Ramakrishna, “On the Feasibility of Codebook-Based Beamforming in
Millimeter Wave Systems With Multiple Antenna Arrays,” IEEE Trans. Wirel.
Commun.,
[6-14] T. Merkle, et. al., “Testbed for phased array communications from 275 to 325 GHz,”
in IEEE Compound Semiconductor IC Symposium (CSICS), USA, Oct. 2017.vol. 14, no.
5, pp. 2670–2683, May 2015.
[6-15] A. Alkhateeb, Y. H. Nam, M. S. Rahman, J. Zhang and R. W. Heath “Initial Beam
Association in Millimeter Wave Cellular Systems: Analysis and Design Insights”, IEEE
Trans. Wirel. Commun., vol. 16, no. 5, pp. 2807–2821, May 2017.
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[6-16] H. M. Aumann, A. J. Fenn, F. G. Willwerth, "Phased array antenna calibration and
pattern predication using mutual coupling measurements", IEEE Trans. Antennas
Propag., vol. AP-37, no. 7, pp. 844-850, Jul. 1989.
[6-17] T. Takahashi, H. Miyashita, Y. Konishi, S. Makino, "Theoretical study on
measurement accuracy of rotating element electric field vector (REV) method",
Electron. Commun. Jpn., vol. 90, no. 1, pp. 22-33, 2006.
[6-18] R. Li, B. Tian, C. Li, Y. Li and B. Li, "A novel on-orbit calibration technique for large
phased array antenna," in Proc. International Applied Computational
Electromagnetics Society Symposium (ACES), Suzhou, 2017, pp. 1-2.
[6-19] M. A. Salas-Natera, R. M. Rodríguez-Osorio, L. de Haro Ariet and M. Sierra-Pérez,
"Novel Reception and Transmission Calibration Technique for Active Antenna Array
Based on Phase Center Estimation," IEEE Trans. Antennas Propag., vol. 65, no. 10,
pp. 5511-5522, Oct. 2017.
[6-20] A. Antón, I. García-Rojo, A. Girón, E. Morales and R. Martínez, "Phase Response
Calibration of a Distributed Antenna Array for Satellite Acquisition," IEEE Antennas
Wireless Propag. Lett., vol. 15, pp. 1731-1734, Feb. 2016.
[6-21] H. T. Chou and D. Y. Cheng, "Beam-Pattern Calibration in a Realistic System of
Phased-Array Antennas via the Implementation of a Genetic Algorithm With a
Measurement System," IEEE Trans. Antennas Propag., vol. 65, no. 2, pp. 593-601,
Feb. 2017.
[6-22] P. Gröschel et al., "A System Concept for Online Calibration of Massive MIMO
Transceiver Arrays for Communication and Localization," IEEE Trans. Microw. Theory
Techn., vol. 65, no. 5, pp. 1735-1750, May 2017.
[6-23] R. Long, J. Ouyang, F. Yang, W. Han and L. Zhou, "Fast Amplitude-Only Measurement
Method for Phased Array Calibration," IEEE Trans. Antennas Propag., vol. 65, no. 4,
pp. 1815-1822, Apr. 2017.
[6-24] R. Long, J. Ouyang, F. Yang, W. Han and L. Zhou, "Multi-Element Phased Array
Calibration Method by Solving Linear Equations," IEEE Trans. Antennas Propag., vol.
65, no. 6, pp. 2931-2939, Jun. 2017.
[6-25] K. Greene, V. Chauhan and B. Floyd, "Built-In Test of Phased Arrays Using Code-
Modulated Interferometry," IEEE Trans. Microw. Theory Techn., vol. 66, no. 5, pp.
2463-2479, May 2018.
[6-26] L. Wu, H. F. Leung, A. Li and H. C. Luong, "A 4-Element 60-GHz CMOS Phased-Array
Receiver with Beamforming Calibration," IEEE Trans. Circuits Syst. I: Regular Papers,
vol. 64, no. 3, pp. 642-652, Mar.2017.
[6-27] C. Fulton, M. Yeary, D. Thompson, J. Lake and A. Mitchell, "Digital Phased Arrays:
Challenges and Opportunities," Proc. IEEE, vol. 104, no. 3, pp. 487-503, March 2016.
[6-28] J. P. Bruckmeyer, A. Hinshaw, C. E. Otero and I. Kostanic, "Adaptive Digital
Beamformer Autocalibration Using an Injected Noise Source," IEEE Trans. Instrum.
Meas., vol. 65, no. 8, pp. 1796-1803, Aug. 2016.
[6-29] L. Xie, X. Yin, C. Lu, L. Yang, H. Zhao and S. Li, "Hybrid Analog–Digital Antenna Array
With Built-in Image Injection Calibration," IEEE Trans. Antennas Propag., vol. 62, no.
11, pp. 5513-5523, Nov. 2014.
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7. CONCLUSIONS
The first deliverable of work package WP5 has reported on the advances in the design of
preliminary system components to be handed over to WP6 for implementation. The main
Sections 3-6 address the major objectives and individual tasks, which were proposed initially
at the beginning of the project.
In Section 3, the initial ideas for the hybrid optical THz wireless link, as developed in D2.2
were refined and further detailed. Part of the challenge was to identify among many
different options the most promising candidates and start investigating and implementing
first functions of the TERRANOVA media converter. In that context, most promising
integration approaches for the co-integration of state-of-the-art optical transponder
modules or functions with the THz wireless frontend were also identified. This work was
supported by some experimental implementation of critical components for risk mitigation.
The work in progress towards new THz frontend integrated circuit technologies was
presented in Section 4. Different options for a BEOL addressing the requirements of THz
applications were experimentally explored. Limitations of the 4L BEOL process were
identified and a compromise, the 3LPP BEOL process, was proposed and developed.
Although the 3LPP BEOL needs further experimental experience at circuit design level for
establishing the final design rules, first promising test circuits for the TERRANOVA downlink
(220-260 GHz) were already manufactured and tested using this BEOL.
Section 5 has experimentally investigated an existing THz link demonstrator of the first
generation. The detailed characterization of this link allowed the validation of existing signal
processing algorithms and the identification of the required improvements. The maximum
data rate for the current transceiver modules was determined, and for the first time an error
free 100 Gbit/s data transmission could be demonstrated using 16-QAM signals at a symbol
rate of 32 Gbd. The next steps will include modelling and design of an improved generation
of algorithms and developing new frontend hardware specifications.
Digital and analog phase shifting were evaluated in Section 6 by simulations. Since complex
hybrid analog-digital beamforming architectures cannot be implemented currently, the
focus is on small linear arrays with 4-8 elements. This work helped to narrow-down the
demonstrator hardware options and to identify the algorithms that will be explored in the
next phase of the project in more detail.
In the first phase of the project, the ongoing work covered all major objectives of WP5 as
planned. Among many options the most promising components and architectures were
identified for further implementation in the next project phase, which was actually the main
goal of this first deliverable.