Data Stream

On-Chip Low Profile Metamaterial Antennas

for Wireless Millimetre-wave Communications

A thesis submitted to The University of Manchester for the degree of

Doctor of Philosophy

in the Faculty of Engineering and Physical Sciences

2012

Ying Peng

School of Electrical and Electronic Engineering

On-Chip Low Profile Metamaterial Antennas for Wireless Millimetre-wave Communication

2

List of Contents

List of Contents .................................................................................................................................................. 2

List of Figures ..................................................................................................................................................... 6

List of Tables .................................................................................................................................................... 11

Abstract .............................................................................................................................................................. 12

Declaration ....................................................................................................................................................... 13

Copyright Statement..................................................................................................................................... 14

Acknowledgment ........................................................................................................................................... 15

List of Abbreviations .................................................................................................................................... 16

Chapter 1 ........................................................................................................................................................... 18

Introduction ..................................................................................................................................................... 18

1.1 Background .......................................................................................................................................... 18

1.2 HD Standard ........................................................................................................................................ 20

1.3 Wireless HD Communication System Design Considerations ........................................ 26

1.3.1 Power Margin ............................................................................................................................. 27

1.3.2 Antenna Technology ................................................................................................................ 29

1.3.3 Integrated Circuit Technology ............................................................................................. 32

1.4 Project Objectivities and Specification ..................................................................................... 34

1.5 Thesis Overview ................................................................................................................................. 37

Chapter 2 ........................................................................................................................................................... 40

Literature Review .......................................................................................................................................... 40

2.1 Millimetre-wave Communications ............................................................................................. 40

2.1.1 Bandwidth and Capacity ........................................................................................................ 41

2.1.2 Narrow Directional Beam ...................................................................................................... 42

2.1.3 Low cost Licensing and Matured Development ........................................................... 43

2.2 Millimetre-wave Antenna .............................................................................................................. 44


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2.3 Metamaterial Antenna ..................................................................................................................... 46

2.4 On-chip Antenna Investigation .................................................................................................... 52

Chapter 3 ........................................................................................................................................................... 58

Antenna Design Analysis ............................................................................................................................ 58

3.1 Antenna Background ....................................................................................................................... 58

3.2 Antenna Parameters ........................................................................................................................ 60

3.2.1 Radiation Power Density ....................................................................................................... 60

3.2.2 Power Gain and Directivity ................................................................................................... 61

3.2.3 Radiation Impedance and Efficiency ................................................................................. 63

3.2.4 Antenna Field Zones ................................................................................................................ 64

3.2.5 Radiation Pattern ...................................................................................................................... 65

3.2.6 VSWR Parameter ....................................................................................................................... 69

3.2.7 Polarization ................................................................................................................................. 70

3.2.8 Antenna Bandwidth ................................................................................................................. 71

3.3 Antenna Types .................................................................................................................................... 72

3.3.1 Microstrip Patch Antenna ...................................................................................................... 74

3.3.2 Slot Antenna ................................................................................................................................ 77

3.3.3 Dipole Antenna .......................................................................................................................... 78

3.4 Conclusion ............................................................................................................................................ 81

Chapter 4 ........................................................................................................................................................... 82

Wideband Planar Antenna Investigation ............................................................................................. 82

4.1 Background .......................................................................................................................................... 82

4.2 U-shaped Slot Antenna Design and Optimisation ................................................................ 83

4.2.1 Design of a Wide Bandwidth U-shaped Slot Patch Antenna .................................... 84

4.2.2 Wave Port Determination in HFSS ..................................................................................... 85

4.2.3 High Gain and Directivity Structure .................................................................................. 89

4.2.4 Ground Effect .............................................................................................................................. 95


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4.2.5 Feed Position Effect.................................................................................................................. 98

4.2.6 Length of Rectangular Metal ................................................................................................ 99

4.2.7 Rectangular Metal Size............................................................................................................ 99

4.2.8 U-shaped Slot Gap Width ..................................................................................................... 102

4.2.9 Thickness of SiO2 Layer Effect ............................................................................................ 106

4.2.10 Final U-shaped Slot Design Structure .......................................................................... 108

4.3 Folded Dipole Antenna .................................................................................................................. 111

4.3.1 Simple Folded Dipole Antenna Simulation ................................................................... 113

4.4 Conclusion .......................................................................................................................................... 115

Chapter 5 ......................................................................................................................................................... 116

AMCs for Millimetre-wave Antenna Application ............................................................................ 116

5.1 Introduction ....................................................................................................................................... 116

5.2 Antenna with Backed Metal Cavity .......................................................................................... 118

5.3 Antenna in AMC Cavity .................................................................................................................. 122

5.3.1 HIS Mechanism ........................................................................................................................ 124

5.3.2 HIS design................................................................................................................................... 128

5.3.3 HIS Fabrication by PCB Technology ................................................................................ 134

5.3.4 Folded Dipole Antenna with HIS Cavity Backed Simulation in HFSS ................ 136

5.4 Low Profile Patch Antenna with Micro-patterned Artificial Lattice Plane .............. 144

5.4.1 Non-conducting Via AMC Structure Design ................................................................. 145

5.4.2 On-chip AMC Structure Antenna Design and Fabrication ...................................... 149

5.5 Low Profile On-chip Antenna with Dog-bone and UC-PBG Structure Plane ........... 156

5.5.1 Dog-bone shaped AMC structure applied to on-chip antenna ............................. 156

5.5.2 UC-PBG Structure Applying to On-chip Antenna ....................................................... 159

5.5.3 AMC and UC-PBG On-chip Antenna Design and Fabrication ................................ 162

5.6 Conclusion .......................................................................................................................................... 167

Chapter 6 ......................................................................................................................................................... 170


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On-chip Antenna Fabrication and Measurement ........................................................................... 170

6.1 Measurement Setup ........................................................................................................................ 170

6.2 Reflection Coefficient Changes with Temperature ............................................................ 171

6.2.1 Temperature Measurement of One Layer AMC Structured Antenna ................ 172

6.2.2 Temperature Measurement of Dog-bone AMC and UC-PBG Structured Antenna

................................................................................................................................................................... 176

6.3 Gain Measurement of On-chip Antenna ................................................................................. 178

6.4 Conclusion .......................................................................................................................................... 186

Chapter 7 ......................................................................................................................................................... 187

Conclusion and Future Work .................................................................................................................. 187

7.1 Summary of the Work .................................................................................................................... 187

7.2 Suggestions for Future Research .............................................................................................. 191

References ................................................................................................................................................. 195

Appendix ......................................................................................................................................................... 204

List of Publications ................................................................................................................................. 204


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List of Figures

Fig.1-1: International frequency spectrum arrangement [11] .............................. 20

Fig.1-2: Cellular network range distribution [12] ................................................ 20

Fig.1-3: Typical WVAN system structure [15] ................................................... 24

Fig.1-4: Tx-Rx antenna gain relates to a target capacity [16].............................. 29

Fig.1-5: Basic wireless system structure [18] ...................................................... 30

Fig.1-6: Advances of terrestrial wireless communication systems and frequency

against data rate [18] .................................................................................... 31

Fig.2-1: Beam pattern of millimetre-wave and microwave [30]. ........................ 42

Fig.2-2: Aperture-coupled single element of microstrip antenna [32] ................ 45

Fig.2-3: (a) Radiation array placing between patches (b) Feeding structure on the

back of substrate [32] ................................................................................... 45

Fig.2-4: 60 GHz CPW-fed patch antenna [33] .................................................... 46

Fig.2-5: (a) PBG structure with lattice of holes in dielectric layer (b) layout of

the first PBG structure proposed in1991 (c) a 2-D micro cavity laser made

by Oscar Painter [38] ................................................................................... 47

Fig.2-6: Different types of metamaterial simulating in waveguide [50] ............. 49

Fig.2-7: Specifications of the two CLL element deep unit layer and the overall

configuration [54]......................................................................................... 50

Fig.2-8: Configuration of the metamaterial antenna [56]. ................................... 51

Fig.2-9: Top and side view of the proposed patch antenna [58] .......................... 51

Fig.2-10: Silicon wafer from IBM in different sizes [62]. ................................... 54

Fig.2-11: S-parameter plot of transmission line with standard silicon substrate

[63] ............................................................................................................... 54

Fig.2-12: S-parameter plot of transmission line with 1000 Ω·cm resistivity

substrate [63] ................................................................................................ 55

Fig.2-13: (a) top view layout (b) cross-sectional view illustration [26] .............. 55

Fig.2-14: Photograph of antenna (left) and cavity (right) [21]. ........................... 57

Fig.2-15: (a) Designed 60 GHz dipole antenna (b) Substrate layers of 130 nm Si

CMOS technology [68] ................................................................................ 57

Fig.3-1: Typical components of wireless communication system in transmitter,

Tx (top) and receiver, Rx (bottom). ............................................................. 58

Fig.3-2: Antenna region, near field and far field. ................................................ 65

Fig.3-3: (a) elevation pattern (b) azimuth pattern (c) combined radiation

pattern [69] ................................................................................................... 66


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Fig.3-4: Radiation pattern in E-H plane ............................................................... 67

Fig.3-5: (a) isotropic and omni-directional antenna; (b) directional antenna ...... 67

Fig.3-6: Radiation pattern with main beam and side lobes in normalized relative

signal strength [70] ...................................................................................... 69

Fig.3-7: Linear polarization of antenna [71] ........................................................ 70

Fig.3-8: Right hand circular polarization of antenna [71] ................................... 71

Fig.3-9: Structure of patch antenna with microstrip feed .................................... 74

Fig.3-10: Top view and cross-section of patch antenna with electric field around

it. .................................................................................................................. 76

Fig.3-11: waveguide slot antenna working from 2 to 24 GHz [78] ..................... 77

Fig.3-12: Slot antenna in infinite ground plane [78] ............................................ 78

Fig.3-13: A simple dipole antenna with feeder .................................................... 79

Fig.3-14: (a) Sketch of electric field around the dipole ....................................... 80

Fig.3-15: Factor A against the wavelength to thickness ratio [79] ...................... 80

Fig.4-1: Simple structure of a U-shaped slot patch antenna element .................. 83

Fig.4-2: (a) Side view of antenna layout with parameters indicated (b) Top view

structure of the antenna ................................................................................ 84

Fig.4-3: Building of radiation air-box .................................................................. 86

Fig.4-4: Electric field distribution on wave port .................................................. 87

Fig.4-5: Comparison between different wave port sizes (a) plot of S11 (b) plot of

VSWR Plot sizes of 10×fw, 8×fw and 12×fw are shown in red, blue and green

traces respectively. ....................................................................................... 88

Fig.4-6: (a) Variables (L1, g1, g2, g3) of U-shaped slot antenna; .......................... 89

Fig.4-7: (a) Radiation pattern of gain at E-plane and (b) Reflection coefficient

S11 when L1 = 0.1 mm, 0.14 mm, 0.2 mm .................................................... 91

Fig.4-8: (a) Gain and (b) S11 plots when g1 = g2 = g3 =0.03 mm, 0.04 mm, 0.07

mm ............................................................................................................... 92

Fig.4-9: Radiation pattern of gain when L1 = 0.14 mm and g1 = g2 = g3 = 0.04

mm ............................................................................................................... 93

Fig.4-10: Radiation pattern of directivity when L1 = 0.14 mm and g1 = g2 = g3 =

0.04 mm ....................................................................................................... 94

Fig.4-11: Reflection coefficient plot when L1 = 0.14 mm and g1 = g2 = g3 = 0.04

mm ............................................................................................................... 94

Fig.4-12: Ground planar location (a) ground plane on top of silicon layer (ground

location in this design) (b) ground plane on the bottom of silicon layer ..... 96

Fig.4-13: Ground varying in HFSS structure design ........................................... 96

Fig.4-14: m1, the highest gain at E-plane when gw = 0.25 mm and gl = 0.5 mm . 97


8

Fig.4-15: Blue trace, VSWR plot when gw = 0.25 mm and gl = 0.5 mm ............. 97

Fig.4-16: Reflection coefficient and VSWR varying at three feeding position of -

0.25 mm, 0 mm and 0.25 mm ...................................................................... 98

Fig.4-17: (a) Reflection coefficient and (b) VSWR when w1 = 0.58 mm, w2 = 0.4

mm, w3 = 0.58 mm ..................................................................................... 101

Fig.4-18: U-shaped slot gaps with width g1, g2 and g3 ...................................... 102

Fig.4-19: (a)(b)(c)(d) Reflection coefficient S11 varying with different slots g1, g2

and g3. ........................................................................................................ 104

Fig.4-20: VSWR when g1 = g3 = 0.05 mm, g2 = 0.005 mm .............................. 105

Fig.4-21: Red trace, g2 = 0.001 mm; Purple trace, g2 = 0.01 mm; Green trace, g2 =

0.05 mm ..................................................................................................... 105

Fig.4-22: (a) Radiation pattern of gain when H = 100 μm ................................ 107

Fig.4-23: VSWR of U-shaped slot antenna when H = 100 μm ......................... 107

Fig.4-24: Radiation pattern of antenna power gain in E, H plane with a 40 μm

substrate. .................................................................................................... 109

Fig.4-25: Radiation pattern of antenna directivity in E, H plane with a 40 μm

substrate. .................................................................................................... 109

Fig.4-26: Reflection coefficient and VSWR of the final structure U-shaped slot

antenna ....................................................................................................... 110

Fig.4-27: Folded dipole 2 m band antenna [81] ................................................. 111

Fig.4-28: Parameters of a folded dipole antenna ............................................... 112

Fig.4-29: HFSS simulation layout of a simple folded dipole antenna ............... 113

Fig.4-30: (a) Directivity at 60 GHz; (b) Gain at 60 GHz................................... 114

Fig.5-1: Metal cavity structure in HFSS ............................................................ 119

Fig.5-2: Cross section of the metal cavity ......................................................... 119

Fig.5-3: Return loss of the folded dipole antenna backed by a metal caivty ..... 120

Fig.5-4: 3D plot of the radiation E-field ............................................................ 120

Fig.5-5: (a) Gain, 5.58 dB of the antenna with metal-cavity at resonate frequency

.................................................................................................................... 121

Fig.5-6: Radiation pattern with the effect of a reflector .................................... 122

Fig.5-7: Antenna needs to be a quarter wavelength distance away from the

conductive reflector. .................................................................................. 123

Fig.5-8: Cross section of a high impedance electromagnetic surface [44] ........ 125

Fig.5-9: Top view of a hexagonal HIS structure [44] ........................................ 125

Fig.5-10: Single LC equivalent circuit ............................................................... 126

Fig.5-11: LC equivalent circuit for HIS structure .............................................. 126

Fig.5-12: E and H fields on magnetic conductor surface ................................... 127


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Fig.5-13: An antenna placed closely to a HIS sheet .......................................... 127

Fig.5-14: TE mode surface wave propagating on a HIS [44] ............................ 128

Fig.5-15: Cross section and top view of a hexagonal HIS structure [91] .......... 129

Fig.5-16: A section of a hexagonal HIS structure design .................................. 131

Fig.5-17: Process flow of making mushroom-shaped structures on PCB

technology .................................................................................................. 135

Fig.5-18: Folded dipole antenna with a HIS cavity backed in HFSS structure . 137

Fig.5-19: Corrugated metal slab structure [44] .................................................. 138

Fig.5-20: Hexagonal HIS structure acting as reflector of the antenna ............... 138

Fig.5-21: Top view of the HIS cavity ................................................................ 139

Fig.5-22: HFSS simulation result for the reflection coefficient, S11.................. 140

Fig.5-23: Radiation pattern of the antenna gain in dB ....................................... 141

Fig.5-24: Radiation pattern of the antenna directivity in dB ............................. 142

Fig.5-25: Far field 3D plot of the antenna directivity ........................................ 143

Fig.5-26: Improved patch antenna with two strips ............................................ 145

Fig.5-27: Eight metal layer Si CMOS process layout........................................ 146

Fig.5-28: Top view of square AMC unit structure ............................................ 146

Fig.5-29: Distinct microwave propagating path ................................................ 148

Fig.5-30: Simulated phase plot of S11 ................................................................ 149

Fig.5-31: Simulated phase sketch of wave propagating .................................... 149

Fig.5-32: (a) Simulate structure of patch antenna with AMC plane. (b) Side view

of structure layout ...................................................................................... 150

Fig.5-33: Return loss S11 of the patch antenna .................................................. 152

Fig.5-34: Simulated antenna gain at 85 GHz ..................................................... 152

Fig.5-35: Simulated antenna directivity at 85 GHz ........................................... 153

Fig.5-36: (a) On-chip antenna measured by probe station system (b) Fabricated

on-chip antenna with AMC structure by 8 layer 0.13 μm Si CMOS process

.................................................................................................................... 154

Fig.5-37: Measured reflection coefficient of on-chip antenna........................... 155

Fig.5-38: Unit element of dog-bone structure.................................................... 156

Fig.5-39: Equivalent circuit of dog-bone structure plane .................................. 156

Fig.5-40: 0.18 μm Si CMOS process from TSMC foundry model cross section

[110] ........................................................................................................... 157

Fig.5-41: Dog-bone structure layout on the Si CMOS technology process ...... 158

Fig.5-42: Dimensions of the dog-bone shape working at 65 GHz .................... 158

Fig.5-43: Simulated S11 phase plot of dog-bone unit using HFSS Ver.12 ........ 159

Fig.5-44: UC-PBG unit cell structure [112] ...................................................... 160

Fig.5-45: On-chip patch antenna with dog-bone AMC and UC-PBG model .... 162


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Fig.5-46: S11 simulated result ............................................................................ 163

Fig.5-47: E-field existing on patch antenna and UC-PBG patterns ................... 164

Fig.5-48: Radiation pattern of on-chip AMC and UC-PBG antenna gain at 65.5

GHz ............................................................................................................ 165

Fig.5-49: Radiation pattern of on-chip traditional patch antenna gain at 65.5 GHz

.................................................................................................................... 165

Fig.5-50: Fabricated on-chip AMC and UC-PBG structured antenna layout .... 166

Fig.5-51: Reflection coefficient results.............................................................. 167

Fig.6-1: Measurement apparatus of on-chip antenna ......................................... 170

Fig.6-2: Temperature controller ......................................................................... 171

Fig.6-3: Single layer AMC structure antenna measuring in probe station ........ 172

Fig.6-4: Reflection coefficient at room temperature (20 Celsius) ..................... 173

Fig.6-5: Reflection coefficient at 0 Celsius ....................................................... 173

Fig.6-6: Reflection coefficient result at -57 Celsius .......................................... 174

Fig.6-7: Reflection coefficient result at 75 Celsius ........................................... 174

Fig.6-8: Reflection coefficient result at 145 Celsius ......................................... 175

Fig.6-9: Dog-bone AMC and UC-PBG antenna in temperature measurement . 176

Fig.6-10: S11 plot at 15 Celsius temperature ...................................................... 176

Fig.6-11: Reflection coefficient against temperature plot ................................. 177

Fig.6-12: On-chip antenna and probe station ..................................................... 178

Fig.6-13: Far field determination ....................................................................... 179

Fig.6-14: Antenna measurement position plot ................................................... 180

Fig.6-15: S21 measurement result at different position ...................................... 182

Fig.6-16: S11 measurement result at different position ...................................... 182

Fig.6-17: Simulation model of gain observing in HFSS Ver.12........................ 183

Fig.6-18: S11 and S21 plot in simulation ............................................................. 184

Fig.6-19: Simulated radiation pattern at the plane with P18 of Ant 2 at 65 GHz 184

Fig.7-1: Jerusalem Crosses AMC structure under patch antenna ...................... 191

Fig.7-2: Snowflake shape AMC structure with folded dipole antenna .............. 191

Fig.7-3: Snowflake shape AMC structure with patch antenna .......................... 192


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List of Tables

Table1-1: Specifications of HD Wireless Communication Network Device ...... 24

Table1-2: Data Rate Requirement for different resolutions, frame rates and

numbers of bits per channel per pixel for HDTV standard [16]. ................. 25

Table1-3: Requirement for uncompressed HD video steaming application [16] 27

Table1-4: Comparison between SiGe and Si CMOS technologies ..................... 34

Table 3-1: Classification of antenna .................................................................... 73

Table 4-1: Performance changes with L1 and slot gap width ............................... 90

Table 4-2: L1, L2 varying effects .......................................................................... 99

Table 4-3: Final parameter values chosen with a SiO2 substrate height of 40 μm

.................................................................................................................... 108

Table 5-1: Sheet capacitance correction factor in various geometries .............. 133

Table 6-1: Average in-band S11 and bandwidth relationship with temperature . 175

Table 6-2: Average in-band reflection coefficient S11 changing with temperature

.................................................................................................................... 177

Table 6-3: Data collection from twenty different positions ............................... 181


12

Abstract

The aim of this work is to design and realise millimetre-wave low profile on-chip antennas for 60 GHz

short-range wireless communication systems. For this application, it is highly desirable that the antenna

can be compatible with standard silicon complementary metal oxide semiconductor (Si CMOS)

technology for high level integration and mass production a low cost. Firstly, millimetre-wave antennas

on normal dielectric substrates and cavities were studied in detail in order to better understand how the

antenna parameters could have effects on their performance at millimetre-wave spectrum. On-chip 60

GHz antennas based on Si CMOS technology were then proposed, designed, fabricated and characterised.

A millimetre-wave U-shaped slot antenna with wide bandwidth was first investigated, simulated and

designed. The simulation results reveal that this antenna can operate at millimetre-wave frequencies with

1 GHz bandwidth at 73.5 GHz and 76.5 GHz, respectively. A 60 GHz folded dipole antenna was also

studied and designed. A metal cavity was added on the back of a folded dipole antenna to act as reflector.

Simulated results show that a folded dipole antenna with a metal cavity can achieve a radiation efficiency

of 97.9% at its resonant frequency. Compared to the gain obtained for the folded dipole antenna without a

cavity, the antenna gain with metal cavity can be enhanced by 3.58 dB.

The main challenges of making high gain and high efficiency Si CMOS on-chip antennas at millimetre-

wave spectrum come from two sources; the thin silicon dioxide (SiO2) layer (maximum 10 m) and

silicon substrate loss (10 cm). The thin SiO2 layer prevents the use of an elevated ground plane, which

could significantly reduce the silicon substrate loss, due to the imaging current effect. Si CMOS

substrates normally have resistivity of 10 cm, which is very lossy at millimetre-wave spectrum. To

tackle these challenges, metamaterial structures, named artificial magnetic conductor (AMC) structures,

were studied and utilised for low profile Si CMOS on-chip antenna design and realisation.

AMC forms high impedance on its surface, reflecting the incident wave without phase reversal so as to

enhance the radiation efficiency. The AMC folded dipole antenna was designed with a mushroom-shaped

structured metamaterial cavity. Simulation results show that the gain increased 1.5 dB in the antenna with

AMC structure, while the distance to the metamaterial surface was reduced by 90% compared to that of

the pure metal cavity. Additionally, two low profile Si CMOS on-chip antennas with novel planar AMC

structures were designed, fabricated and characterised. They were manufactured by 0.13 μm Si CMOS

technology from Chartered foundry and 0.18 μm Si CMOS technology from TSMC, respectively. The

techniques proposed in these two antennas provide valuable alternatives to the existing approaches. The

measurement results show that bandwidth of the on-chip antenna with a micro-patterned artificial lattice

is approximately 10 GHz. The one with a dog-bone shape and uniplanar compact photonic band gap

(UC-PBG) structures managed a 1.6 dB gain and 1 GHz bandwidth enhancement compared to that

without AMC structures.


13

Declaration

No portion of the work referred to in the thesis has been submitted in support of an

application for another degree or qualification of this or any other university or other

institute of learning.


14

Copyright Statement

i. The author of this thesis (including any appendices and/or schedules to this

thesis) owns certain copyright or related rights in it (the “Copyright”) and s/he has

given The University of Manchester certain rights to use such Copyright, including

for administrative purposes.

ii. Copies of this thesis, either in full or in extracts and whether in hard or

electronic copy, may be made only in accordance with the Copyright, Designs and

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This page must form part of any such copies made.

iii. The ownership of certain Copyright, patents, designs, trade marks and other

intellectual property (the “Intellectual Property”) and any reproductions of copyright

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described in this thesis, may not be owned by the author and may be owned by third

parties. Such Intellectual Property and Reproductions cannot and must not be made

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Intellectual Property and/or Reproductions.

iv. Further information on the conditions under which disclosure, publication and

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Reproductions described in it may take place is available in the University IP Policy

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University Library, The University Library‟s regulations (see

http://www.manchester.ac.uk/library/aboutus/regulations) and in The University‟s

policy on presentation of Theses.


15

Acknowledgment

I appreciate the help of many people for supporting and assisting my PhD study.

Firstly I would like to thank my supervisor Dr. Zhirun Hu for his tireless support and

encouragement in both my study and my life.

I would also like to thank Dr. Rob Sloan for his help in my undergraduate final year

project, which determined my research direction in microwave communication filed. I

appreciate the help from Prof. Zhipeng Wu for his support and advices. I express my

gratitude to many staff members in our group, Mr. Keith Williams, Dr. Abdallah,

M.A, Dr. Saswata Bhaumik and Dr. Zhongwen Jin for their assistance, discussion and

advice over my PhD. Special thanks to Prof. Haiying Zhang, Prof. Haigang Yang, Dr.

Zhiqiang li and Dr. Tongqiang Gao for their help on fabrication.

I also appreciate Mark Bentley, Sherri McLain, Graham Kean, Swee Kim Ang, Peter

Tran, Lara Meredit, Mousumi Roy, Emerson Sinulingga, Warit P.M, and Xin Niu to

help me with this thesis proofreading.

Finally, I would also like to thank my parents from the bottom of my heart for all of

their support and concerns in everything over these years. I am appreciated my lovely

friends and dear Liang for being with me in U.K. and making my PhD strudy

unforgettable.


16

List of Abbreviations

A/V Audio to Video

ADC Analogy Digital Conversion

AMC Artificial Magnetic Conductor

BB Base Band

BER Bit Error Rate

BiCMOS Bipolar CMOS

CLL Capacitively Loaded Loop

CPS Coplanar Strip line

CPW Coplanar Waveguide

CSRR Complementary Split-ring Resonators

DRC Design Rule Check

EBG Electromagnetic Band Gap

FCC Federal Communications Commission

FNBW First Nulls Beam Width

FSS Frequency Selective Surface

GaAs Gallium Arsenide

HD High-definition

HDTV High-definition Television

HF High Frequency

HIS High Impedance Surface

HPBW Half Power Beam Width

IC Integrated Circuit

IF Intermediate Frequency

InP Indium Phosphide

LOS Line Of Sight

LTCC Low Temperature Co-fired Ceramic

MMIC Monolithic Microwave Integration Circuit

NF Noise Figure

NLOS Non Line Of Sight

P2P Point-To-Point

PBG Photonic Band Gap

PCB Printed Circuit Board

PHY Physical Layer

RF Radio Frequency

SD Standard Definition


17

Rx Receiver

Si CMOS Silicon Complementary Metal Oxide Semiconductor

SiGe Silicon Germanium

SiO2 Silicon Dioxide

SNR Signal to Noise Ratio

TE mode Transverse Electric mode

TM mode Transverse Magnetic mode

Tx Transceiver

UC-PBG Uniplanar Compact Photonic Band Gap

UHF Ultra-high Frequency

VSWR Voltage Standing Wave Ratio

WLAN Wireless Local Area Network

WPAN Wireless Personal Area Network

WVAN Wireless Video Area Network

Chapter 1 Introduction

18

Chapter 1

Introduction

1.1 Background

In 2001, the Federal Communications Commission (FCC) allocated 7 GHz within the

57 – 64 GHz frequency band for unlicensed use. During the past few years,

substantial knowledge about the 60 GHz millimetre-wave channel has been

accumulated and a great deal of work has been done on the development of

millimetre-wave communication systems for commercial use. In principle there is

nearly 7 GHz of bandwidth available for the use of wireless local communications. In

2007 an international frequency spectrum arrangement that shows in Fig.1-1 was

published, indicating clearly that most countries have different allocations for this

unlicensed frequency band [1]. Nevertheless the 60 GHz frequency band has attracted

a lot of interest for all kinds of short-range wireless communications, especially the

wireless high definition home entertainment market.

For the last decade, great progress has been made in the high-definition (HD) media

production market. Many products have been developed to meet the need of HD

system such as high-definition television (HDTV) sets. For example in 2002, a project

aiming to record and play HD content was officially announced as Blu-Ray [1], and

later on 18 July 2006, the first mass-market Blu-Ray rewritable disc drive was


19

released by Sony [2]. This kind of high-speed data communication demands a huge

data file transmission to support real-time HD video streaming. Therefore based on

the studies of many systems under the proposal of IEEE 802.15.3c Task Group, a

gigabit wireless link has been proposed in a 60 GHz millimetre-wave spectrum [3].

The opening of a big chunk of free spectrum formed a technology push and a market

pull for 60 GHz millimetre-wave communication. Besides the announcement of a 7

GHz unlicensed frequency band, 60 GHz millimetre-wave communication has other

advantages. The first one is that at 60 GHz, oxygen has a property of high

atmospheric absorption. Transmission power attenuates heavily along the propagating

path. Therefore, among the propagating range of different wireless systems, the 60

GHz wireless system is aimed for use in short-range indoor systems, as shown in

Fig.1-2. Moreover, the heavy propagating attenuation decreases the interruption

between neighbour systems and increases the privacy protection.

On the other hand, substantial knowledge of 60 GHz millimetre-wave channels has

been accumulated for commercial applications. Electronic devices for millimetre-

wave communication application are mainly made from group semiconductors such

as gallium arsenide (GaAs) and Indium phosphide (InP) because they have high-speed

operating properties [4-7], They are however expensive. In the past few years,

alternative semiconductor technologies have been explored and the IBM engineers

have demonstrated the first experimental 60 GHz transmitter and receiver chips using

a high-speed alloy of silicon germanium (SiGe) [8-10]. However the high cost of this

material makes it difficult to apply in areas of mass production such as transceivers

and RF Front ends. Therefore, in order to drag down the price, it‟s been proposed that


20

the mass production of transceiver components for use at 60 GHz utilises Si CMOS

technology.

Fig.1-1: International frequency spectrum arrangement [11]

Fig.1-2: Cellular network range distribution [12]

1.2 HD Standard

HD techniques are rapidly developing in wireless communication. In order to increase

the video quality, more data information needs to be stored. Most of the HD devices

are desiged to upgrade their data holding ability, which can be called capacity.

Compared with the standard definition (SD) technique that uses 480 lines of pixels

from top to bottom, HD technique uses 720 or 1080 lines of pixels, which is almost 6

times as many as SD. If HD is introduced into the TV broadcast application, a modern


21

digital system is required to carry enough data within a required bandwidth. Dating

back to 1969, the first consumer HDTV with a 5:3 aspect ratio, which was a slightly

wider screen format than the usual 4:3 standard, was developed by Japanese state

broadcaster NHK. However, introducing new technology to the public is always a big

challenge due the lack of compatible technology already in the system. Finally by the

early 2000s, the HDTV compression technology had progressed to deliver sufficient

data capacity and processing power. The technology now supports compression

algorithms powerful enough to make HDTV affordable for consumers.

According to the study carried out by IEEE 802.15.3c Task Group, a gigabit wireless

link can be realised in 60 GHz millimetre-wave communication. With the millimetre-

wave technology, a large amount of data can be held in the point-to-point (P2P)

transmission channel system. In a short-range wireless communication system, HD

devices can provide gigabit data transmission. Therefore real video streaming can be

formed among different HD sources and displayers. On the other hand, to make more

efficient use of the 7 GHz unlicensed frequency band, wireless communication have

also aimed to increase the digital compression [13].

In the mid1990s, the cellular communication industry went through a period of

explosive growth and wireless communication networks became much more

pervasive after the cellular concept was first developed. From the years 2001 to 2006,

worldwide cellular and personal communication subscribers have increased from 600

million to 2 billion. Why does the network grow that rapidly? The main reason is that

governments throughout the world provided an increased number of new radio

spectrum licenses for personal communication services with a frequency band from

1800 MHz to 2000 MHz [14]. Today, the new unlicensed frequency band around the


22

60 GHz range gives another chance for the wireless communication technology

revolution.

The development of the wireless local area network (WLAN) brought the chance to

replace cables in single rooms, buildings, and offices, using technology that could

transmit data back and forward between computers on the same network through

standardisation under the IEEE 802.11 series. In the WLAN system, all the devices in

the network can be connected together to exchange data. Similar to WLAN, wireless

personal area network (WPAN) has the same function of data exchange but within a

much smaller area, for example the Bluetooth technique and wireless PC headphones.

In our work, we are more concerned with combining the small range wireless

communication network technique with the HD displaying technique to provide a

high quality data transfer and display in a short-range area. This system can be

referred to as an HD wireless system and the main difference between this and others

systems is the data propagation speed and transmission bandwidth. In other words, the

data rate and capacity has to increase to support the HD streaming data transfer. To

implement this idea in the market and satisfy the criteria of HD quality, a

wirelessHDTM

standard has been published, defining the specification for the wireless

digital network interference [15].

WirelessHDTM

is built by several world leading technology companies. They

organised an industry-led standardisation effort to define a next-generation wireless

digital interface specification for consumer electronics and PC products. Specifically,

WirelessHDTM

emerges as a new, industry-led special interest group with the stated

goal of enabling wireless connectivity for streaming HD content between source

devices and HD displays. The latest specification was architected and optimized in


23

Jan 2008 for wireless display connectivity [15]. The specification defined the first

generation implementation of high-speed rates as capable of reaching up to 3 Gbps

within 10 meters for the use of consumer electronics, PC products, and portable

device segments.

Specifications announced by WirelessHDTM

are based on a new wireless

communication interface protocol. It is a new network, which combines video and

audio data transmission between devices and is called wireless video area network

(WVAN). Specifications of WVAN are shown as follows:

•Stream uncompressed audio and video at up to 1080p resolution, 24 bit colour

at 60 Hz refresh rates

• Deliver compressed Audio/Video (A/V) streams and data

• Advanced A/V and device control protocol

•Unlicensed operation at 60 GHz with a typical range of at least 10 m for highest

resolution HD A/V

• Smart antenna technology to enable non line of sight (NLOS) operation

• Data privacy for user generated content

A typical WVAN system structure model is shown in Fig.1-3, where the HDTV

display in the centre acts as the termination of this network and it is surrounded by

other source devices. In order to display HD media pictures fluently, the data rate in

the WVAN system is important. Devices in this network can be mainly classified into

source and sink. Table1-1 lists the data rate specifications of some representative

source and sink devices that could be used in the HD WVAN


24

Fig.1-3: Typical WVAN system structure [15]

Table1-1: Specifications of HD Wireless Communication Network Device

Data rates 3.0 – 1.5 Gbps 1.0 Gbps 40 Mbps

Source HD A/V source Data source Audio source

Set top box

HD-DVD player

HD-DVD recorder

Blu-Ray disc player

Blu-Ray disc player

Personal video recorder

Broadcast HD receiver

Personal media players

Digital video cameras

Digital still cameras

Digital audio players

HD A/V audio

source

Stereo tuner

Broadcast radio

receiver

Sink HD A/V sink Data sink Audio sink

Flat panel display

Blu-Ray disc recorder

HD-DVD recorder

Personal video recorder

Satellite receiver

Personal media players

Digital video cameras

Digital still cameras

Digital audio players

Speakers

Audio receiver

Audio Amp


25

In Table1-1 the device that requires the highest data rate is the HD A/V source and

sink. In the HD A/V device, data is uncompressed. To make the picture quality a

reality, the uncompressed source transmission requires a greater data rate than the

compressed one. Therefore to guarantee enough bandwidth and speed for HD data

transmission, 3.0 Gbps is the minimum data rate possible. However, some large

capacity technologies such as Blu-Ray discs and players need more bandwidth to

guarantee the transmission performance quality. Table1-2 shows the data rate

requirement for different HDTV resolutions [16].

Table1-2: Data Rate Requirement for different resolutions, frame rates and numbers of

bits per channel per pixel for HDTV standard [16].

Table1-1 and Table1-2 indubitably indicated that the HD devices require high data

rates. However, the data rate is not the only parameter that affects the data

transmission performance, and sufficient use of the transmission power and the


26

unlicensed frequency band are other factors to consider. In addition, the antenna plays

an important role in the system. With an efficient antenna in the system, power can be

fully utilised in RF transmission and the unlicensed frequency bands can also be used.

1.3 Wireless HD Communication System Design

Considerations

Millimetre-wave technology has been established for many decades now and has

mainly been deployed in military applications. Millimetre-wave is classified as an

electromagnetic spectrum that spans from 30 GHz to 300 GHz, which corresponds to

wavelengths from 10 mm to 1mm. In recent years this technology has started being

taken into the civil market due to the availability of the 7 GHz unlicensed frequency

band. Based on the millimetre-wave technology, the 60 GHz wireless communication

technique focused more on the market requirement. In March 2005, the IEEE

802.15.3c Task Group (TG3c) was formed to develop an millimetre-wave based

alternative physical layer for the existing IEEE 802.15.3 WPAN Standard 802.15.3-

2003 [3]. This is the first standard addresses multi-gigabit wireless system that forms

a solution to the multimedia distribution application. WirelessHDTM

requests a bit rate

of 3 Gbps to link HDTV sets to disc players, video cameras, game consoles, PCs and

other devices in the system as discussed in Section 1.1.

In achieving this data rate of 3 Gbps and data capacity, challenges occurs in three

sections: 1) power margin, 2) antenna technology, 3) circuit integrated technology.


27

1.3.1 Power Margin

Generally, there are two propagation methods: line-of-sight (LOS) with the evaluating

scheme, namely Ricean Distribution, and non-line-of-sight (NLOS) with the

evaluating scheme, namely Rayleigh distribution [17]. For the indoor environment,

signal propagation paths are easily blocked by furniture and humans. A power margin

depends on the indoor wireless channel properties (LOS or NLOS). Table1-3

recapitulates key requirements for uncompressed HD video streaming for short-range

WVAN [16].

Table1-3: Requirement for uncompressed HD video steaming application [16]

Applications Data

rate

BER Data type Environment K of

LOS/NLO

S

Uncompressed

HD video

streaming

0.05 –

5.5

Gbps

1.00E-12 Isochronous Home

5 – 10 m

1.55 / 2.44

Conference room

20 m

1.77 / 3.83

The type of propagation path defines different antenna and system requirements. As

Table1-3 shows, factors of LOS and NLOS cases change with the environment

operation range. In a conference room of 20 m width, LOS must have the very least a

Ricean factor K of 1.77, while NLOS must have a Rayleigh factor K of 3.83 in order

to guarantee the quality of HD video streaming [16].

Using Shannon‟s Channel Capacity Theorem to determine the upper data rate

limitation, C = B log2 (SNR +1), where SNR is signal to noise ratio, the maximum

achievable capacity can be computed. Channel capacity, C, can be increased via

corresponding increases in either the bandwidth, B, or the SNR. The main

responsibility of power is supplying enough energy to transmit through the


28

propagating path and down convert into the receiver. Compared with an increase in

the bandwidth of the channel, improving the SNR of the receiver is a more direct

method of increasing the energy efficiency. The SNR at the receiver can be calculated

as follows [16]:

𝑆𝑁𝑅 = 𝑃𝑇 + 𝐺𝑇 + 𝐺𝑅 − 𝑃𝐿0 − 𝑃𝐿(𝑑) − 𝐼(𝐿) − (𝐾𝑇 + 10 log10 𝐵 − 𝑁𝐹) (1)

where 𝐺𝑇 and 𝐺𝑅 refer to the transmitter and receiver antenna gain, respectively. 𝑃𝑇 is

the power transmitted out; 𝑃𝐿0 and 𝑃𝐿(𝑑) are propagating path loss; 𝐼(𝐿) is the

insertion loss; 𝐾𝑇 is the thermal loss, and 𝑁𝐹 is the noise figure. To improve SNR and

reduce interference, antennas with narrow and focused patterns are required. In fact,

transmission losses that include path loss and insertion loss are much higher at 60

GHz than at other low frequencies. Therefore in the WPAN area, transmission loss

limits the wireless system performance, and detrimental effects take place on the

system coverage and relays unless an extra repeater is connected to strengthen the

transmission power. Since using directional antennas can enhance the power transmit,

high gain antennas are highly desirable.

Influenced by the noise added in the propagation path, the signal that arriving at the

receiver device has a high transmission loss, and this phenomenon is worsened by an

increase in the transmission distance. Thus the antenna at the transmitter should be

efficient and with a directional radiation pattern to strengthen the transmission power

so as to overcome the fading margin. In a conference room environment with a

propagating distance of 20 m, a fixed 10 dBm transmit power, bandwidth 1.5 GHz

and an implementation loss of 6 dB, engineers measured the relationship between


29

transceiver antenna gains and capacity value using two path cases, as shown in Fig.1-

4 [16].

Therefore to meet the HD requirement of a 3 Gbps capacity, for LOS path the antenna

gain is required to be 25 dB, while for NLOS path, it is required reach at least 37 dB.

To sum up, challenges in the power margin mainly involve overcoming the path loss

and increasing efficiency to achieve the target channel capacity or bit error rate (BER)

within a required distance.

Fig.1-4: Tx-Rx antenna gain relates to a target capacity [16]

1.3.2 Antenna Technology

In the wireless communication system, antenna plays an important role. It acts as the

input and output interface between the RF transmission channel and the wireless

equipment as shown in Fig.1-5. Modern wireless system consists elements such as


30

base band (BB), radio frequency (RF), intermediate frequency (IF) and antenna, Ant.

[18].

Fig.1-5: Basic wireless system structure [18]

Antennas operating in millimetre-wave communication systems should be, firstly,

well matched to the transmitters/receivers. They also need to provide gain and

directivity that satisfy the requirements. Some additional functions can be added to

the antenna system such as applying the beam steering techniques to save the

transmission and receiving power and appropriately controlling radiation patterns to

increase system capacity.

Different wireless communication systems require different data transmission rates.

Fig.1-6 shows that a 60 GHz antenna requires high transmission speeds exceeding 1

Gbps [18]. To successfully deploy 60 GHz short-range wireless networks, it is highly

desirable that the systems are small size, of lightweight, highly efficient and of low

cost. Recently, there have been some proposed structures operating around 60 GHz

frequencies with small size and lightweight [19].


31

Fig.1-6: Advances of terrestrial wireless communication systems and frequency

against data rate [18]

Increasing the antenna gain can bring down the receiver‟s noise figure. However,

without using a smart antenna system in the transmission path, LOS is the most

suitable propagating mode to save power and increase SNR. It has been calculated that

between two 20 dB transmitters and receiver antennas with LOS operation, the system

can support 1.25 Gbps for path range within 50 meters. To meet the bandwidth

requirement at 60 GHz, antenna radiation efficiency needs to be improved. This can

be done by integrating antennas into an array with a beam steering function applied.


32

The size of an antenna at 60 GHz is so small that it can be embedded on a standard

chip package. With the implementation of array, the antenna gain, system directivity

and bandwidth will be improved [20]. Though increasing the antenna gain is a method

to cutting down the fading margins in the link budget, antennas with steerable beam

focus on the direction can choose a path to propagate. To find the destination, such

antennas run through a searching algorithm and then, either mechanically or using

several antenna elements, construct a phased array with optimised radiation patterns.

With this technique, the directivity of antenna and transmission systems becomes high

and meets the small form-factor requirement for WPAN or WLAN devices.

1.3.3 Integrated Circuit Technology

In wireless communication systems, the integrating of system components is

important. To integrate system elements all together and make them compatible with

each other, technology of different materials should be carefully chosen. Basically the

choice of integrated circuit (IC) technology depends on the requirement of the system

and the aspect implementations. That‟s to say, the choice of technologies should

consider issues such as power consumption, efficiency, dynamic range, linearity

requirements and integration level. This is because they are related to the transmission

rate, cost and size, modulation scheme, transmit power and bandwidth. At frequencies

above 60 GHz, there are three main technologies ready for use.

Group III and IV semiconductor technology such as GaAs and InP;

SiGe technology;

Silicon technology used in CMOS technology.


33

All these technologies can now be used around 60 GHz up to 100 GHz. None of them

can totally meet all the objectives required in HD wireless communication systems.

That‟s because the market requires an integrated technique with efficiency both from

technological and economical perspectives. Each of those three technologies has its

own drawbacks. For example, GaAs technology allows fast, high gain, and low noise

implementation but suffers from poor integration and expensive implementation.

On the other hand, SiGe technology is a cheaper alternative to the GaAs with

comparable performance but it is still expensive compared to Si CMOS technology.

Si CMOS technology provides a high integration solution, a low price and a small

size compared with other technologies, but the main challenge is the substrate loss

and power handling.

A major part of today‟s 60 GHz technology depends on relatively expensive material

such as GaAs semiconductor material. It was only when researchers from IBM, the

University of California, Los Angeles, and Berkeley proved that silicon chips could

transmit signals with a low power and cost, that the 60 GHz wireless communication

market become promising [21]. Table1-4 is a comparison table with Si CMOS

technique and SiGe technique properties.

According to Table1-4, Si CMOS substrate is active only during transition between

states. This characteristic reduces power consumption so as it is suitable for use in

large ICs.


34

Table1-4: Comparison between SiGe and Si CMOS technologies

SiGe Si CMOS

Switch Always on Current pass active

Transistor Speed High Low

Passive device performance High Poor

Match chip Right on Integrated

Design Simple Difficult

Integration Level Easy Easy

Application Provide 1 Gbps

within 8m

Replacing other WPAN sub

technique

Cost Very high Low

Furthermore, the negligible gate current in Si CMOS is advantageous in analogy

digital conversion (ADC) circuits. In comparison to other technologies, Si CMOS

technology can provide much cheaper ICs per wafer. Si CMOS is also a good thermal

conductor allowing efficient removal of power dissipated as heat. The main

technology challenge is integrating this Si CMOS into devices of the wireless system

together with the antenna system and the down stage IF part. Another factor that

restricts the Si CMOS technology structure design is the relatively low transmission

speed. In order to overcome these drawbacks, bandwidth should be increased to

provide electrons flow. Although the switching mode plays a positive effect on the

power consumption, Si CMOS substrate absorbs some of the transmit power as a loss

at the signal sending port. To decrease the substrate power absorbability to the lowest

level and radiate most power is the issue that needs to be resolved.

1.4 Project Objectivities and Specification

Scanning widely around the wireless communication fields, much progress has been

made towards the millimetre-wave antenna design for 60 GHz WPAN applications. A

millimetre-wave radio front-end implemented as an assembly of monolithic

microwave integrated circuits (MMICs) in GaAs technology has been carried out, but


35

has proven rather expensive [22]. A 60 GHz linearly and circularly polarised antenna

array on liquid crystal polymer substrate was developed and provides 10 dB return

loss and bandwidth greater than 2 GHz [23]. A cost-efficient 60 GHz planar patch

array with 8×8 elements with 128 μm RT Duroid substrate was proposed for P2P

connections [24]; however, the liquid crystal polymer and RT Duroid material are still

not compactable with any IC technologies. In order to reduce the cost, studies mainly

are taken in the fields of millimetre-wave antenna that are based on the low cost of Si

CMOS technology. Linear tapered slot antenna [25] and Quasi-Yagi antenna [26, 27]

were also developed and fabricated; however, these antennas were not fabricated with

standard Si CMOS technologies, they were processed on thicker SiO2 substrates in

order to reduce the substrate loss. In other words the techniques employed for these

antennas cannot be realised using standard Si CMOS technologies, resulting in less

impact on high level integration in order to drive down the costs. The efficiency of the

quasi-yagi antennas is as low as 5.6 %; therefore, reducing the influence from the

lossy silicon substrate and fabricating high quality antennas based on standard Si

CMOS technology have been and still are challenging for us.

The aim of this work is to propose a millimetre-wave antenna for HD wireless

communication system applications. Recent millimetre-wave antennas are based on

expensive but high resistivity materials such as GaAs, InP and SiGe [22]. To reduce

the cost, we decided to design this antenna based on standard Si CMOS technology.

Planar antennas were chosen due to their easy fabrication properties.

The lossy silicon substrate, however, causes a big challenge in improving the antenna

efficiency. There are two methods to layout the on-chip antenna: first, the ground

plane of the antenna is printed at the bottom of silicon bulk, which makes the lossy


36

silicon acts as the substrate of antenna. In this case, energy will be absorbed by the

lossy substrate instead of radiating out. The second location of the ground plane is on

the top of the silicon bulk, which isolates the lossy Si substrate. In this case the

dielectric layer with high resistivity becomes the substrate of antenna. However, the

Si CMOS technology limits the thickness of dielectric layers to a maximum value of

approximately 10 μm, which is too thin for practical applications due the image

current effect. Metamaterial AMC structures are proposed, design and realised to

tackle these two challenges.

An AMC structure forms a surface with high impedance because when RF waves

propagate on such surfaces, no phase reversal is generated. There are many AMC

structures applied on antenna designs to improve the antenna efficiency and

bandwidth. In our work, AMC structure design needs to satisfy the following two

demands:

AMC generates the forbidden frequency band at 60 GHz;

It is easy to fabricate on standard Si CMOS technology.

The AMC structure will be applied to the antenna reflector instead of the normal

ground plane. The HIS property stops the surface wave from propagating and

improves the radiation efficiency of the antenna. Hence the main objectives of this

works are:

(1) Study of millimetre-wave antenna fundamentals and special needs for 60 GHz

short-range wireless networks.

(2) Study and investigation of AMC structures for low profile millimetre-wave

antenna applications.


37

(3) Study and investigation of Si CMOS properties for low profile millimetre-

wave applications, especially for on-chip antennas.

(4) Design, optimisation and realization of 60 GHz low profile Si CMOS on-chip

AMC antennas.

(5) Characterisation of 60 GHz low profile Si CMOS on-chip AMC antennas.

1.5 Thesis Overview

This thesis starts with the introduction chapter with HD system millimetre-wave

antenna requirement and Si CMOS technology. A summary of the contents of the

individual chapters from Chapter 2 of this thesis is presented below:

Chapter 2: Literature Review

In this chapter, a review of recent researches related to this work is presented and

discussed. Discussion is mainly focused on four parts of the development: millimetre-

wave communication development, millimetre-wave antenna, metamaterial antenna,

and on chip antenna investigation.

Chapter 3: Antenna Design Analysis

The background and the relevant theories of the antenna design are firstly given in the

beginning of this chapter. Theories of related parameters of antenna such as radiation

power density, radiation gain and directivity, radiation impedance and efficiency,

radiation zones, radiation pattern, VSWR parameters, polarisation and antenna

bandwidth are presented. Finally, different antenna types are introduced. Since the


38

planar antenna is easy to design and fabricate on chip, this type of antenna is chosen

to our work. Mechanisms of microstrip patch, slot and dipole antennas will also be

presented in this chapter.

Chapter 4: Wideband Millimetre-wave Planar Antenna Investigation

This chapter mainly presents a planar U-shaped slot antenna applied to millimetre

wave applications based on a SiO2 substrate. Full wave simulation by using HFSS

Ver.12 is taken to optimise a U-shaped antenna structure. Analysis of how each part

of the structure affecting the antenna radiation performance is taken. The analysis

result shows that the thickness of substrate between the antenna and ground plane

limits the bandwidth performance. To avoid the thickness limitation, a folded dipole

antenna is introduced instead of a U-shaped slot.

Chapter 5: AMC for Millimetre-wave Antenna Applications

In the beginning of this chapter, two folded dipole antennas with different resonant

cavities are presented and simulated. One of the cavities is made of a metamaterial,

AMC. Basic theory of how AMC works and affects electromagnetic wave is

presented. In this chapter, five AMC structures are studied and designed. They are

mushroom structure, corrugated slab, micro-patterned artificial lattice, dog-bone

structure, and UC-PBG structure. Three different antennas with a combination of

these AMC structures are analysed. Simulation results show that the cavity consisting

of mushroom and corrugated slab structure can improve the antenna radiation gain by

1.5 dB. However, to implement the mushroom structure, the conducting via need to

be built inside the silicon bulk. As it is hard to drill a well in the silicon bulk for the

mushroom-shaped cavity, this structure cannot be fabricated by standard Si CMOS

technology. The other two antennas with micro-patterned artificial lattice, dog-bone


39

and UC-PBG structures are proposed, designed and fabricated. Measurement results

show that the ones with dog-bone and UC-PBG structures can effectively avoid the

silicon substrate loss.

Chapter 6: On-chip Antenna Fabrication and Measurements

The procedure of fabricated on-chip antennas measurement is introduced in the

beginning of this chapter. Two fabricated antennas in Chapter 5 are measured using a

Cascade on-wafer probe station. The effects of temperature on antenna performance

were also measured. Since the probe station is a hundred times larger than on-chip

antenna, it is difficult to measure the radiation pattern in a chamber. We detect the

power receiving performance by placing two antennas apart to act as transmitter and

receiver. The transmission coefficient S21 is detected in different positions around

transmitting antenna, enabling us to analyse the antenna transmission properties

experimentally.

Chapter 7: Conclusion and Future Work

The objective of this chapter is to draw conclusions from all the topics discussed in

the PhD study. The chapter ends by proposing future work and discussing

improvement to the aforementioned structures.

Chapter 2 Literature Review

40

Chapter 2

Literature Review

2.1 Millimetre-wave Communications

Millimetre-wave generally corresponds to the radio spectrum between 30 GHz to 300

GHz, with the wavelength between one and ten millimetres. However, in the context

of wireless communications, the term generally corresponds to a few bands of

spectrum near 38, 60 and 94 GHz, and more recently to a band between 70 GHz and

90 GHz (also referred to as E-Band) that have been allocated for the purpose of

wireless communications in the public domain.

Though relatively new in the world of wireless communications, the history of

millimetre-wave technology goes back to the 1890‟s when J.C. Bose was

experimenting with millimetre-wave signals at just about the time when his

contemporaries such as Marconi were inventing radio communications [28].

Millimetre-wave technology remained within the confines of university and

government laboratories for almost half a century after Bose‟s research. The

technology started so see its early applications in Radio Astronomy in the 1960‟s,

followed by applications in the military in the 70‟s. In the 80‟s, the development of


41

millimetre-wave ICs created opportunities for mass manufacturing of millimetre-

wave products for commercial applications. In the 1993, an automotive collision

avoidance radar operating at 77 GHz marked the first consumer-oriented use of

millimetre-wave frequencies above 40 GHz [29]. In 2002, the FCC opened the

spectrum between 57 and 64 GHz for unlicensed wireless communication, resulting in

the development of a plethora of broadband communication and radar equipment for

commercial application. In 2003, the FCC authorised the use of 71-76 GHz and 81-86

GHz for licensed point-to-point wireless communications, creating a fertile ground for

new industries developing products and services in this band.

2.1.1 Bandwidth and Capacity

One of the key advantages of millimetre-wave communications is the large amount of

spectral bandwidth. The bandwidth available in each unlicensed frequency bands

more than 5 GHz. With such wide bandwidth available, millimetre-wave wireless

links can achieve capacities to as high as 5 Gbps full duplex, which is unlikely to be

matched by any lower frequency RF wireless technologies. The availability of this

extraordinary amount of bandwidth also enables the capability to scale the capacity of

millimetre-wave wireless links as demanded by the market needs. Typical millimetre-

wave products commonly available today operate with spectral efficiency close to 0.5

bits/Hz. However, as the demand arises for higher capacity links, millimetre-wave

technology will be able to meet the higher demand by using more efficient

modulation schemes.


42

2.1.2 Narrow Directional Beam

Unlike the microwave links, which provide a wide radiation beam, millimetre-wave

links cast very narrow beams, as illustrated in Fig.2-1. The narrow beams of

millimetre-wave links increased the frequency band reuse in a specific geographic

area and allowed for development of multiple independent links in close proximity.

For example, by using an equivalent antenna, the beamwidth of a 70 GHz link is four

times as narrow as that of an 18 GHz link. A key benefit of the highly narrow beam

millimetre-wave links is the scalability of their deployments. For example,

millimetre-wave is well suited for network topologies such as point-to-point mesh, a

dense hub-and-spoke or even a ring. Other wireless technologies often reach their

scalability limit due to cross interference before the full potential of such network

topologies can be realised.

Fig.2-1: Beam pattern of millimetre-wave and microwave [30].


43

2.1.3 Low cost Licensing and Matured Development

One of the key benefits of the millimetre-wave frequency bands is that they are

licensed, giving both the users and the service a use protection. However, as opposed

to the microwave bands, in which licensing costs require significant investment, the

cost of licensing E-band links is exceptionally low, with less than $500 per link for 10

years of interference protected use. The traditional form of spectrum licensing has

been a challenge for those who own licenses as well as for those who do not. For the

owner of a license, it often represents a significant upfront investment combined with,

in certain cases, legal obligations to make some specific use of the spectrum. For

those who do not own this license, it represents a barrier to competitive entry into this

particular market.

Millimetre-wave technology has a strong history and technological evolution behind

it. The characteristics of millimetre-wave have already been well understood for many

decades. With many decades of military and government-funded research, millimetre-

wave technology has reached a level of maturity compared to microwave radio

technologies.

In 2002, when the unlicensed spectrum around 60 GHz came out, a discussion about

the challenge of building a new generation of short-range MMW wireless

communication system was published. In the paper [31], features of affordability,

scalability, modularity, extendibility and interoperability were presented and

discussed. In addition, user convenience and network deployment efficiencies are

important prerequisites for market success. Along with the RF communication

development in the past few years, substantial knowledge of about 60 GHz


44

millimetre-wave channels has developed. A great deal of work for the commercial use

of millimetre-wave has also been carried out.

2.2 Millimetre-wave Antenna

In the recent decade, with the announcement of unlicensed frequency band,

millimetre-wave antennas have rapidly developed.

As discussed in Chapter 1, one of the attractions of 60 GHz is the high attenuation

along the propagating path. The heavy absorption of atmosphere at 60 GHz spectrum

can reduce the wave propagating diffraction effect and increase the high free space

loss. Therefore, at 60 GHz, the wireless propagating path between the transmitter and

receiver can easily be disturbed. Early in 1996, C. Peixeiro and his colleagues

designed a planar microstrip patch antenna array for the mobile station operating from

62 GHz to 63 GHz [19]. The antenna array was built up on the substrate of RT/Duriod

5880 and had obtained a maximum gain of 10.8 dB. However, the feeding network

presented in this antenna array could radiate energy in the millimetre frequency band,

causing the cross polarisation and side lobe radiation. In order to reduce these effects,

researchers in [32] used the slot-coupled printed antenna array to separate the feeding

network from the ground. As shown in Fig.2-3, a ground plane with slot in the centre

separates the radiating element from feeding line.

The array is finally arranged as illustrated in Fig.2-13, which substantially increased

the antenna radiation performance. This millimetre-wave antenna has the advantage

of low cost, but it is not compatible with other millimetre-wave products.


45

Fig.2-2: Aperture-coupled single element of microstrip antenna [32]

Fig.2-3: (a) Radiation array placing between patches (b) Feeding structure on the back

of substrate [32]

Microstrip antennas are popular in the application of millimetre-wave communication

for their low profile, lightweight, conformability and ease of integration with

millimetre-wave devices. In paper [33], a 60 GHz coplanar waveguide (CPW) fed

patch is proposed. Fig.2-4 shows the antenna structure the implementation of a high

dielectric constant substrate with related permittivity of 9.9. The CPW stubs in the

centre of dipole antenna improved the radiation efficiency.


46

Fig.2-4: 60 GHz CPW-fed patch antenna [33]

2.3 Metamaterial Antenna

The requirements of design tasks, such as the efficiency, bandwidth, directivity,

weight, and cost effectiveness, have been challenging for antenna engineers with

traditional schemes. Antennas engineers have been persistently seeking for new

technologies that can meet these requirements. The recent development on

metamaterial inspired antennas seems very promising to address these issues.

Metamaterials are artificial materials engineered to have properties that may not be

found in nature. The structure of metamaterials determine their properties, which can

be stated as using small inhomogeneities to create effective macroscopic behaviour

[34]. The primary research in metamaterials investigates materials with negative

refractive index. Negative refractive index materials appear to permit the creation of


47

superlenses, which can have a spatial resolution lower than that of the wavelength. A

form of 'invisibility' has been demonstrated at least over a narrow wave band with

gradient-index materials. Potential applications of metamaterials are diverse, which

include remote aerospace applications, sensor detection and infrastructure monitoring,

smart solar power management, public safety, radomes, high frequency battlefield

communication and lenses for high gain antennas, ultrasonic sensors, and even

shielding structures from earthquakes [35, 36].

A type of metamaterial called photonic band gap (PBG) structure was firstly made by

Eli Yablonovitch in 1991 at Bell Communications in New Jersey [37]. His team

mechanically drilled a complex diamond-shaped 3D array of millimetre-sized air

holes into a transparent material, which shows in Fig.2-5(a) and Fig.2-5(b).

Fig.2-5: (a) PBG structure with lattice of holes in dielectric layer (b) layout of the first

PBG structure proposed in1991 (c) a 2-D micro cavity laser made by Oscar Painter [38]

This 3D structure could block frequencies in the microwave region. Since then,

researchers have been incredibly inventive in devising all kinds of techniques to


48

generate band-gaps at different frequencies in a wide range of materials from silicon

to plastics.

When the PBG structures were applied for electromagnetic wave application, they

were also known as electromagnetic band gap (EBG) structures. Recently, several

authors proposed the application of EBG structures in [39-42] as superstrates in order

to improve the antenna performances. Typically, an EBG array, which consists of

dielectric elements and characterised by stop/pass bands is employed as a cover for

antennas to enhance the gain of a single patch antenna. Frequency selective surface

(FSS) was also been proposed as an alternative to dielectric EBGs for gain

enhancement [43]. The FSS offers similar transmission and reflection characteristics,

but is thinner than the EBG configuration. However, the distance between the FSS

superstrate and the ground plane, which determines the resonant frequency, needs to

be carefully designed.

In recent years, AMCs have attracted much attention from the academic society and

industry. AMC structures are typically formed by periodic patterns based on dielectric

substrates. An AMC structure has a property of high impedance surface (HIS). It was

first reported in Sievenpiper‟s work, which has a forbidden frequency band over

which waves cannot been propagated [44]. Once a suitable AMC structure is applied

to antenna, unwanted ripples at AMC forbidden frequency band can be eliminated.

HIS works within the frequency range, where the tangential magnetic field is tiny,

leaving only large electric field [44]. Most recently, AMC structures have reported to

used for designing low profile high gain planar antennas [45, 46] In addition to AMC


49

structures, other Metamaterial structures have also demonstrated their benefits for

higher directivity, low profile and planar antennas [47-50].

Fig.2-6: Different types of metamaterial simulating in waveguide [50]

In [50], many types of metamaterail structures were studied. Structures as shown in

Fig.2-6 are several published metamaterial unit element structures that have been used

to enhance the antenna performance.

In [51], The authors show how a flat slab made of a metamaterial engineered to have

a small negative index of refraction can be used to reshape radiation emitted from an

isotropic source and produce a highly directional output beam. In [52], a multilayer

grid structure that can significantly increase the antenna directivity and gain was

extended to millimetre-wave frequencies.


50

In addition of increase to the directivity and gain, it has also been reported that low

profile antenna radiation characteristics can be improved by using AMC [53-55]. In

[54], a printed electric dipole antenna was intergraded into a volumetric metamaterial-

based AMC block, as shown in Fig.2-7. It was demonstrated numerically in HFSS

that resonant modes could be excited to produce either large front-to-back ratios or

large broadside directivities. The proposed 3D AMC block can be constructed with

only two unit layers of capacitively loaded loop (CLL) elements that are

symmetrically positioned along the x-axis. The printed dipole was designed to be

symmetric both in the x and z directions. The block layers are symmetric about the

printed dipole layer in the z direction. This structure works at 9.45 GHz and obviously,

the structure of AMC is not easy to fabricate.

Fig.2-7: Specifications of the two CLL element deep unit layer and the overall

configuration [54].

Besides CLL multilayer structure, more simple planar AMC structure can be used to

increase low profile antenna efficiency. Researchers in [56] presented characteristics

of microstrip patch antennas on metamaterial substrates loaded with complementary

split-ring resonators (CSRRs). The proposed CSRR structure is placed in the ground


51

plane of the substrate. Simulation results were verified by experimental results

confirming that the CSRR loaded patch antenna achieves size reduction as well as

bandwidth improvement. Fig.2-8 shows the antenna structure.

Fig.2-8: Configuration of the metamaterial antenna [56].

Metamaterial structure can even be applied to antenna itself, i.e, the antenna itself is

made of metamaterials [57, 58], where patch antenna worked at 7 GHz with

bandwidth enhancement from 200 MHz to 3.2 GHz. The structure is shown in Fig.2-9.

Fig.2-9: Top and side view of the proposed patch antenna [58]


52

2.4 On-chip Antenna Investigation

The IC is one of the most important developments within the electronics arena. The

history of the integrated circuit shows that the IC developed as a result of the need for

very small electronic assemblies. Engineers and scientists saw the possibilities of

much greater levels of miniaturisation. IC came out for the need of improving military

equipment. The Second World War had conclusively proved the value of electronics

beyond all doubt. In conventional wireless or radar systems, antenna and circuit are

separated to the subsystem. A circuit antenna module was used to be the interference.

MMIC is a type of IC device that operates at microwave frequencies. MMIC typically

perform functions such as microwave mixing, power amplification, low noise

amplification, and high frequency switching. As both inputs and outputs on MMIC

devices are matched to a characteristic impedance of 50 ohms, MMIC is easier to be

used without an external matching network interference. MMICs were originally

fabricated using compound semiconductor, such as GaAs, InP and SiGe. These

materials have properties of high propagating speed but high costly [4-7, 59].

The exploration of alternative semiconductor technologies demonstrated the first

experimental millimetre-wave transmitter and receiver chips using a high-speed alloy

of SiGe by IBM engineers [8-10].

The primary advantage of Si technology is its lower fabrication cost compared with

GaAs, InP and SiGe. Although silicon wafer diameters are typically 8 inch or 12 inch,

which is larger compared with 4 inch or 6 inch for GaAs, the total costs are lower,


53

contributing to a less expensive IC. With the efficient and identical fabrication

process as standard Si CMOS, devices such as RF Front-end circuit can be massly

produced. Since the IEEE 802.15.3C Task Group has been formed to standardise

millimetre-wave radios, consumer requirements push the millimetre-wave developing

towards low-cost and high-integrated property. In order to lower the price, the use of

silicon technology to build 60 GHz transceiver components is a good alternative.

From copper oxide to germanium and silicon, the materials were systematically

studied in the 1940s and 1950s. The mainstay in analog and digital mixed

implementation technology is CMOS. Nowadays CMOS technology is widely used

because it provides density and power saving on digital side and also provides a good

combination of components for analogy design. Today, silicon monocrystals are the

main substrate used for ICs to integrate the transceiver and antenna with the digital

signal process blocks using Si CMOS technology [27, 60, 61]. Here in Fig.2-10

illustrates the layout of different sizes of silicon wafers.

The main challenge, however, of building antenna on the Si CMOS substrate is the

substrate loss. A standard Si CMOS substrate has low resistivity (typically ～10

Ω·cm) and high permittivity, which dissipates electromagnetic power, hence

significantly decreases the antenna radiation efficiency.


54

Fig.2-10: Silicon wafer from IBM in different sizes [62].

The plot in Fig.2-11 shows two S-parameters, S11 and S21 of a 1 mm microstrip

transmission line based on silicon substrate [63]. As silicon has low resistivity, the

reflection coefficient from 0 to 100 GHz remains lower than -10 dB and the

transmission coefficient stays below -3 dB. Fig.2-12 shows the S-parameter result of a

substrate with 1000 Ω·cm resistivity substrate. It can be easily concluded that the

energy loss is due to the low resistivity property of the silicon.

Fig.2-11: S-parameter plot of transmission line with standard silicon substrate [63]


55

Fig.2-12: S-parameter plot of transmission line with 1000 Ω·cm resistivity substrate [63]

Several antennas built on different type of silicon substrates have been reported. For

instance, in 2007, a quasi-yagi antenna operating at 100 GHz with silicon integrating

technology was reported.

Fig.2-13: (a) top view layout (b) cross-sectional view illustration [26]

Fig.2-13 shows the structure of quasi-Yagi antenna from top and cross-sectional

views. This antenna consists of one driver, two directors and a truncated ground plane


56

which acts as a reflector. It is fed by a uniplanar broadband microstrip-to-coplanar

strip transition etching on the top aluminium layer and is shown in Fig.2-13(a). As the

cross-sectional layout shows Fig.2-13(b), main the antenna is formed by two 2 μm

aluminium layers separating by a 2 μm thick SiO2 layer. Besides the feeding structure,

directors and driver are etched on the top aluminium layer while the reflector is

etched on the bottom layer.

Yagi-uda antenna is famous for its high directivity so this quasi-yagi antenna was

fabricated on silicon substrates for low resistivity 10 Ω·cm using the post back end of

line (post-BEOL) process. Post-BEOL processing indicates Beck End of line. It is a

fabrication for transistors, capacitors and resistors. It‟s a package get interconnected

with wafer. As a result, this antenna obtained a bandwidth from 89 GHz to 104 GHz,

8.2 dB return loss and 5.7 dB antenna gain at 100 GHz. However, the dielectric layer

used in this design is 20 m, which is not compatible with any standard Si CMOS

technologies.

In order to reduce the silicon substrate loss, the micromachining technique and proton

implantation process are applied to Si CMOS process [64, 65]. For example in 2007,

D. Liu and his colleagues presented a Si-based packing technology to enlarge the

bandwidth of millimetre-wave system [21]. Fig.2-14 illustrates the layout of antenna

proposed in [21]. The silicon substrate of antenna was implanted to a high resistivity

of 1000 Ω·cm. The antenna is fabricated using a 1.2 μm Cu damascene process. The

cavity is 400 μm deep and metallized with an 8 μm thick Cu layer. However, the

technology is again not compatible with any standard Si CMOS technologies


57

Fig.2-14: Photograph of antenna (left) and cavity (right) [21].

To design the antenna with standard on-chip technology, the main challenge is that

they are not efficient due to the low resistivity silicon [27, 66, 67] substrate. Most

recently there are some progresses on Si CMOS on-chip antennas. For instance, a new

dipole antenna was presented and obtained a large bandwidth covering from 57 GHz

to 64GHz allocated free band [68], In this paper, a 60 GHz antenna was produced by

130 nm Si CMOS process. A -10 dB matching bandwidth of 11% was also revealed.

The total simulated efficiency was found to be 3%. Fig.2-15 shows the dipole antenna

and substrate layers of 130 nm Si CMOS technology [68].

Fig.2-15: (a) Designed 60 GHz dipole antenna (b) Substrate layers of 130 nm Si

CMOS technology [68]

Chapter 3 Antenna Design Analysis

58

Chapter 3

Antenna Design Analysis

3.1 Antenna Background

Fig.3-1: Typical components of wireless communication system in transmitter, Tx (top)

and receiver, Rx (bottom).


59

Antenna is an electrical device, which converts electrical currents into radio waves,

and vice versa. It is usually applied with a radio transmitter or receiver as shown in

Fig.3-1. In the wireless transmission system, analogy signal generated by carrier wave

source is used to carry digital information and it is converted into radio wave to

radiate out into free space through antenna. As Fig.3-1 shows, the input signal flows

into transmit device Tx and mixes with an analogy signal as a carrier, after running

through filter and a high pass amplifier, signal propagates into the antenna and

radiates out in to free space. On the receiver, Rx, RF signal is detected by the antenna

and converted into subsystem by decoding. This is the typical way how the digital

signal is transmitted in a wireless system. In this system, the components and devices

can be classified into active and passive parts. Devices such as amplifiers with power

supply required are defined as active components while others are passive

components. Insertion loss is generated when the power running through the passive

devices. Besides insertion loss, this system has other factors affecting the signal

transmission quality.

The skin effect caused by the changing of magnetic field on the metal surface;

The absorption of RF signal, which is produced by the air;

The reflection, which takes place along the transmission path. If the RF signal

is blocked by any solid objects, reflection appears, the signal propagation path

changes its direction;

Matching is important for a circuit in the system. It makes power radiation

efficiently. VSWR is a way to measure how system matches and give a value

of 1 at perfect match. Besides that, the reflection coefficient of S-parameter is

another way to observe matching situation.


60

3.2 Antenna Parameters

Antenna is the interface between the device and RF wave transmission environment.

The main function of antenna is either sending or receiving RF energy between the

circuit and free space. The antenna changes electrical current into airborne wave which

radiates out into space. Fundamentally, any metallic objects about the size of their

wavelength works as an antenna to radiate RF energy. Radiation is generated by the

varying of velocity of charges such as speed, direction and oscillation. There are

mainly four parameters describing the performance of an antenna. In this section,

parameters are studied and their relationships with radiation will also be presented.

3.2.1 Radiation Power Density

Radiation power density shows an average power radiating out from an antenna. It

can be estimated by integrating power surrounded the antenna. However the equation

is based on another concept, which is radiation intensity. Radiation intensity in a

given direction has specified the power radiated from an antenna per unit solid angle.

Solid angle is a sphere angle with its vertex at the centre of a sphere of radius r, which

refers to a steradian. Since it is a far-field parameter, the equation can be expressed as

follows:

𝑈 = 𝑟2𝜔𝑟𝑎𝑑 (2)

where 𝑟 is radius of sphere and 𝜔𝑟𝑎𝑑 is the radian of solid angle.

The radiation power density can be obtained as

(3) SS S

avradavrad dsHEdandsPP )Re(2

1ˆ *


61

The radiation power density takes the average power of the integration power within

an infinitesimal area of closed surface. E and H are the electric and magnetic field

generating in the radiation area, 𝜔𝑟𝑎𝑑 is the radian of solid angle and 𝜔𝑎𝑣 is the radian

of an average angle. Real part of /2 represents the average power density.

3.2.2 Power Gain and Directivity

The power gain and directivity are parameters that define the ability of antenna

concentrating power in particular direction. It may be regarded as the ability of the

antenna to direct radiated power in a given direction. This is an inherent property of

the antenna that only includes the ohmic or dissipative losses arising from

conductivity of metal and dielectric loss. As an antenna radiates power in sphere, 3D

radiation pattern can be divided by two perpendicular planes, which are elevation and

azimuth plane. Details of radiation pattern will be presented in Section 3.2.5.

However, the power gain can be measured in a specified direction (𝜃, 𝜙), where 𝜃 is

the scanning angle in elevation plane and 𝜙 is that angle in azimuth plane.

𝐺 𝜃, 𝜙 = 4𝜋× 𝑝𝑜𝑤𝑒𝑟 𝑟𝑎𝑑𝑖𝑎𝑡𝑒𝑑 𝑝𝑒𝑟 𝑢𝑛𝑖𝑡 𝑠𝑜𝑙𝑖𝑑 𝑎𝑛𝑔𝑙𝑒 𝑖𝑛 𝑑𝑖𝑟𝑒𝑐𝑡𝑖𝑜𝑛 (𝜃 ,𝜙)

𝑇𝑜𝑡𝑎𝑙 𝑝𝑜𝑤𝑒𝑟 𝑎𝑐𝑐𝑒𝑝𝑡𝑒𝑑 𝑓𝑟𝑜𝑚 𝑠𝑜𝑢𝑟𝑐𝑒 (4)

Similar to the power gain expression, directivity, D indicates the ability of the

directional antenna propagate power to a required direction. Nevertheless, different

from gain, directivity does not count in the dissipative losses. It is a ratio of radiation

intensity in a given direction to the average radiation intensity. Directivity, 𝐷 𝜃, 𝜙

can be calculated through an equation

)( *HE


62

𝐷 𝜃, 𝜙 = 4𝜋× 𝑝𝑜𝑤𝑒𝑟 𝑟𝑎𝑑𝑖𝑎𝑡𝑒𝑑 𝑝𝑒𝑟 𝑢𝑛𝑖𝑡 𝑠𝑜𝑙𝑖𝑑 𝑎𝑛𝑔𝑙𝑒 𝑖𝑛 𝑑𝑖𝑟𝑒𝑐𝑡𝑖𝑜𝑛 (𝜃 ,𝜙)

𝑇𝑜𝑡𝑎𝑙 𝑝𝑜𝑤𝑒𝑟 𝑟𝑎𝑑𝑖𝑎𝑡𝑒𝑑 𝑏𝑦 𝑎𝑛𝑡𝑒𝑛𝑛𝑎 (5)

The total power per solid angle in direction can be represented by the radiation

intensity 𝑈 𝜃, 𝜙 . Hence the gain 𝐺 𝜃, 𝜙 and directivity 𝐷 𝜃, 𝜙 equations can be

expressed as

𝐺 𝜃, 𝜙 = 4𝜋× 𝑈 𝜃 ,𝜙

𝑃𝑠𝑜𝑢𝑟𝑐𝑒 (6)

𝐷 𝜃, 𝜙 = 4𝜋× 𝑈 𝜃 ,𝜙

𝑈 𝜃 ,𝜙 𝑠𝑖𝑛𝜃𝑑𝜃𝑑𝜙𝜋

02𝜋

0

(7)

Dissipative losses arise when power flowing through the antenna structure such as

circuit mismatching, substrate loss and thermal loss. Since this is the only difference

between the antenna gain and directivity, they are closely related with each other. A

coefficient 𝜂 is used to simply show the relationship between them. 𝜂 indicates the

ratio of the total power radiated by antenna over the total power accepted from source,

which can be expressed as:

𝜂 = 𝐺 𝜃 ,𝜙

𝐷 𝜃 ,𝜙 (8)

If the efficiency 𝜂 reaches its maximum value of 1, which indicates all the power out

of source is radiated out. As a reference, there is a peak value of those quantities

coincides with the direction of the principal lobe radiated by the antenna. Peak values

are usually taken in decibel relative to a short current element, a thin lossless half

wavelength dipole, or an ideal isotropic radiator. Also the peak directivity of the

antenna can be expressed as

𝑃𝑒𝑎𝑘 𝐷𝑖𝑟𝑒𝑐𝑡𝑖𝑣𝑖𝑡𝑦 = 𝑝𝑒𝑎𝑘 𝑝𝑜𝑤𝑒𝑟 𝑟𝑒𝑐𝑖𝑒𝑣𝑒𝑑

𝑎𝑣𝑒𝑟𝑎𝑔𝑒 𝑝𝑜𝑤𝑒𝑟 𝑟𝑒𝑐𝑖𝑒𝑣𝑒𝑑 (9)


63

3.2.3 Radiation Impedance and Efficiency

Impedance is important for the antenna system design, especially for matching.

Antenna is the interface between electromagnetic device and air. In order to reduce

the return loss and improve the radiation efficiency, antenna should be perfectly

matched. Antenna input impedance can be considered to be the sum of self-

impedance and mutual impedance, which can be expressed as:

𝐼𝑛𝑝𝑢𝑡 𝑖𝑚𝑝𝑒𝑑𝑎𝑛𝑐𝑒 = 𝑠𝑒𝑙𝑓 𝑖𝑚𝑝𝑒𝑑𝑎𝑛𝑐𝑒 + 𝑚𝑢𝑡𝑢𝑎𝑙 𝑖𝑚𝑝𝑒𝑑𝑎𝑛𝑐𝑒 (10)

As equation 10 shows, self impedance is the impedance that can be measured at the

input terminals of the antenna and mutual impedance is reactance generate by

influence of coupling to the antenna from other sources. The self-impedance is

consisted of antenna resistance and self-reactance, arising from reactive energy in

near-field region as shown in the following equation:

𝑆𝑒𝑙𝑓 𝑖𝑚𝑝𝑒𝑑𝑎𝑛𝑐𝑒 = (𝑎𝑛𝑡𝑒𝑛𝑛𝑎 𝑟𝑒𝑠𝑖𝑠𝑡𝑎𝑛𝑐𝑒) + 𝑗(𝑠𝑒𝑙𝑓 𝑟𝑒𝑎𝑐𝑡𝑎𝑛𝑐𝑒) (11)

The antenna resistance indicates the sum of radiation resistance Rr and a loss

resistance RL. The sum of Rr and RL refers to the antenna equivalent resistance as

equation 12 shows. It indicates the dissipated power consist radiated power and ohmic

losses.

𝐴𝑛𝑡𝑒𝑛𝑛𝑎 𝑟𝑒𝑠𝑖𝑠𝑡𝑎𝑛𝑐𝑒 = 𝑅𝑟 + 𝑅𝐿 12


64

According to equation 12, radiation efficiency 𝜂 can be defined as

𝜂 = 𝑅𝑟/ (𝑅𝑟 + 𝑅𝐿) 13

Therefore, for an efficient antenna, it is importance to have radiation resistance which

is greater than loss resistance, and the higher the Rr compared with RL, the higher the

efficiency.

3.2.4 Antenna Field Zones

Since electromagnetic field exists surrounding the radiating source and varies with

distance from an antenna, two principle regions located in free-space are defined

according as distance away from the antenna. They are called near field or Fresnel

zone and far field or Fraunhofer zone. Near field is the area close to antenna itself

while far field is distance, R, apart from antenna. Distance, R formed a boundary

between two regions

𝑅 =2𝐿2

𝜆

(14)

where L is the maximum dimension of the antenna and 𝜆 is the wavelength at antenna

resonant frequency. In near field, the power varies with the distance to the antenna.

However, in the far field, all power flow is directed radially outward to infinity and

the shape of field pattern is independent of the distance. Normally far field is a

measurable field which is used to measure antenna parameters such as power gain,

directivity and radiation pattern. Fig.3-2 shows sketch of radiation regions around an

antenna.


65

Fig.3-2: Antenna region, near field and far field.

3.2.5 Radiation Pattern

The mathematical and graphical parameter to represent the magnetic or electric field

around the antenna is called radiation pattern. For an antenna, the term radiation

pattern can be referred to near-field pattern or far-field pattern. The near-field pattern

refers to the positional dependence of the electromagnetic field. The near-field pattern

is commonly obtained by measuring a plane placed in front of the source, or a

cylindrical, or a spherical surface enclosing it. The far-field pattern is measured over

the antenna propagating range where the electromagnetic field does not change with

different positions. Far-field pattern shows a plot of the antenna energy radiation

distribution. This is the pattern that graphically shows a number of important

variables regarding the antenna performance. For example, power flux density,

radiation intensity, directive gain and polarization.


66

A full Radiation pattern is a 3D plot which combines elevation pattern and azimuth

pattern [69]. Fig.3-3 illustrated elevation and azimuth pattern, respectively and a

standard radiation pattern. The elevation pattern is the graph as if measurements were

taken from the side of antenna radiation port scanning over an angle of θ. The

azimuth pattern is the graph as if measurements were taken directly above the antenna

scanning over an angle of ϕ. To design an antenna, we always observe the radiation

pattern in both simulation and measurement. Agilent HFSS is the software that has

been used to simulate antenna structures in our work. Radiation pattern is plotted

based on a circular chart. As shown in Fig.3-4, numbers around circular chart

represents different angles of θ. The chart shows a full 360° of elevation plane.

However, the angle ϕ in azimuth plane affects the shape of the radiation pattern. The

radiation pattern at E and H planes of a patch antenna is shown in Fig.3-4. The red

trace is the plotted when ϕ = 0° for E plane while the blue trace is plotted when ϕ =

90° and describes H plane. More radiation pattern in different angles can be obtained

in HFSS simulation in order to describe the radiation closer to the real 3D plot.

Fig.3-3: (a) elevation pattern (b) azimuth pattern (c) combined radiation pattern [69]


67

Fig.3-4: Radiation pattern in E-H plane

Antennas also can be classified according to their radiation patterns. Isotropic antenna,

which has an ideal point source, radiates the same amount of energy in all spherical

directions. The radiation pattern of isotropic antenna can be seen as a sphere with

antenna centred. Isotropic antenna is an ideal point antenna and this is only the

theoretical case. Normally isotropic antenna is used as a reference radiator. In fact, a

similar spherical radiation pattern can be generated from omni-directional antennas.

They radiate and receive energy equally in all directions around themselves. Since

radiation signal has the same strength within a sphere area around source, this type of

antenna can be used to broadcasting a signal to all points. Fig.3-5(a) shows the plot of

isotropic and omni-directional antenna pattern.

Fig.3-5: (a) isotropic and omni-directional antenna; (b) directional antenna

90 degree

0 degree

Angle θ in

elevation

plane from

-180° to 180°


68

Different radiation formation was applied in directional antenna. Directional antenna

radiates RF energy in a desirable direction, and it also has higher signal strength along

the desirable direction than omni-directional antenna when both of them radiate the

same amount of power. Ideally, directional antenna only has its energy field pattern at

the request sending or receiving directions as Fig.3-5(b) shows. In that plot, taken a

slice of a plane through the 3D pattern, antenna parameters can be measured. For

example, gain can be measured in decibel over either a dipole or a theoretical

isotropic radiator, which is normally called antenna power gain or directional gain.

In a directional antenna pattern, energy field presents as lobes around the source.

There is a main beam for directional antenna, which contains most power flowing out

from the antenna source. At the same time, not all the power can be leading to the

same required direction. A part of power transmits to other directions with an angle or

even backward. These existing lobes are called side lobe and back lobe. Fig.3-6

illustrates a sketch of directional radiation pattern detailed with some determined

variables and parameters as an example.

As shown in Fig.3-6, a main beam with maximum energy of 0 dB, two -20 dB side

lobes and some back lobes can be seen in this directional antenna radiation pattern.

Combining the structure with a normalized relative signal strength margin plane, half

power beam width (HPBW) and beam width between first nulls beam width (FNBW)

can be obtained from Fig.3-6. HPBW refers to the angel where the radiated power

strength reaches half of total radiation power. FNBW refers to the angle between two

nulls from each side of the main radiation beam.


69

Fig.3-6: Radiation pattern with main beam and side lobes in normalized relative signal

strength [70]

3.2.6 VSWR Parameter

Voltage standing wave ratio (VSWR) is the parameter that can estimate the

bandwidth and the quality of antenna impedance match. In the case of the microstrip

antenna, it is usually the impedance, rather than the pattern which affects the

bandwidth of the antenna. Range of frequencies within which the value of VSWR is

less than certain value, S is defined as the bandwidth of the antenna. The bandwidth of

the antenna is related to the total quality factor as:

(15)

where is the total quality factor and in some applications the value of S is taken as

2, which corresponds to a return loss of 9.5 dB or 11% reflected power.

100( 1)( ) %

T

SBandwidth BW

Q S

TQ


70

Most commercial antennas, however, are specified to be 1.5 : 1 or less over some

bandwidth. Based on a 100 watt radio, a 1.5 : 1 VSWR equates to a forward power of

96 watts and a reflected power of 4 watts, or the reflected power is 4.2% of the

forward power.

3.2.7 Polarization

Polarization is a parameter of an electromagnetic wave propagating direction at

resonant frequency. It indicates the shape and orientation of the locus of the field

extremities vectors as a function of time. With this parameter, transmitting antenna

and receiving antenna can match the mode of polarization to superposition and

enlarge the propagating power. Generally polarization can be divided into linear and

circular two types. Besides these two types, cross polarization is radiation orthogonal

to the desired polarization. For instance, the cross polarization of a vertically

polarized antenna is the horizontally polarized fields. It is unwanted polarization,

which is generated by feeding network.

Fig.3-7: Linear polarization of antenna [71]


71

Fig.3-8: Right hand circular polarization of antenna [71]

The linear polarizations that show in Fig.3-7 consist of vertical, horizontal and

oblique planes while circular types that show in Fig.3-8 include Circular Right Hand,

Circular Left Hand, Elliptical Right Hand and Elliptical Left Hand [71].

3.2.8 Antenna Bandwidth

Antenna Bandwidth indicates the range of frequencies within which the antenna

operates. There are two methods of expressing the relative bandwidth for narrowband

and wideband antenna, respectively.

For narrowband antennas, percentage bandwidth is used to define their narrow

bandwidth. It is expressed as,

𝐵𝑎𝑛𝑑𝑤𝑖𝑑𝑡𝑕 =𝑓𝐻−𝑓𝐿

𝑓𝑐= 2

𝑓𝐻−𝑓𝐿

𝑓𝐻+𝑓𝐿 (16)

where 𝑓𝐻 , 𝑓𝐿 are the higher and lower frequency where half power is radiated out,

respectively. 𝑓𝑐 is the centre frequency. Equation 16 shows that the bandwidth is the


72

percentage of the frequency difference over the centre frequency. It is theoretical limit

to 200%, which occurs when 𝑓𝐿 = 0

For wideband antennas, fractional bandwidth is used. The wideband antenna

bandwidth is expressed as the ratio of high frequency to low frequency,

𝐵𝑎𝑛𝑑𝑤𝑖𝑑𝑡𝑕𝑓𝑟𝑎 =𝑓𝐻

𝑓𝐿 . (17)

and it is usually presented in form of B : 1, where B is the value of 𝐵𝑎𝑛𝑑𝑤𝑖𝑑𝑡𝑕𝑓𝑟𝑎 .

There are many methods dealing with the antenna bandwidth enhancement. Take the

microstrip antenna as an example; the bandwidth can be increased by applying either

a very low dielectric constant substrate or a relatively thick dielectric layer on the

antenna [72]. A superstrate planar can be used for wider the bandwidth [73]. Resonant

cavity can be employed to improve antenna performance [74]. AMC has also been

used to widen microstrip antenna bandwidth [58, 75].

3.3 Antenna Types

The history of antenna design was dated back to the late 19th

century when the first

radio experiments were proposed based on classical electromagnetic field theory [76].

Since then, many different types of antennas have been studied and designed to meet

different needs for both military and commercial use. Nowadays, antennas can be

very broadly classified either by the frequency spectrum in which they are commonly

applied or the structure appearance or based on the mode of radiation. Here we

classified antennas based on their different applications therefore they can be divided

into four categories. Each antenna structures are grouped as listed in Table 3-1.


73

Table 3-1: Classification of antenna

Antenna Types Structures

Wire Dipole; monoploe Loop Yagi-Uda

Aperture Horn

Reflector

Lens

Open-ended waveguide

Planar Microstrip Dipole Slot

Array Aperture Yagi-Uda Patch

Wire antennas including dipoles, monopoles, loops, Yagi-Uda and other similar

structures, they are widely used at lower frequencies from high frequency (HF) up to

ultra-high frequency (UHF) in buildings, cars, aircrafts, ships and etc. Wire antennas

generally have advantage of lightweight, low cost and simple design, while the

radiation gains generated from them are low.

Aperture antennas such as horns, lens, reflectors and open-end waveguides are

normally applied in microwave and millimetre-wave frequencies. They have

moderate to high radiation gain and mainly used in aircraft or spaceship, since they

can be very conveniently flush-mounted in the skin of those aircrafts.

Planar antennas including microstrip patch antennas, planar dipoles and planar slots

are also used in microwave and millimetre-wave frequencies. Planar antennas can be

made with photolithographic methods, such as, printed circuit board (PCB) and Si

CMOS technologies. Planar antennas are also easily arrayed for high gain purpose.

Antenna array consists of a regular arrangement of antenna elements with a feed

network. Antenna elements need to be regular and easy to arrange, such as planar

antennas. The main function of antenna array is controlling the pattern characteristics

such as beam pointing angle and sidelobe levels by adjusting the amplitude and phase

distribution of the array elements.


74

For a 60 GHz wireless network, it is highly desirable to have the antenna embedded in

the chip. Therefore, in this research project, planar antenna will be the choice. In the

following sections, two kinds of planar antennas and a dipole antenna will be

discussed in details.

3.3.1 Microstrip Patch Antenna

Microstrip antennas are often chosen to be applied in millimetre-wave wireless

communications due to their properties of small size, lightweight, low cost and easy

to install. Microstrip antennas are typically formed by etching the metal antenna

element pattern to an insulating dielectric substrate with a metal plane attached to the

bottom of the substrate that functions as a ground plane.

The basic microstrip antenna is the patch antenna. Radiation part of a patch antenna

has many shapes such as square, rectangular, circular and elliptical. There are many

methods to feed patch antennas by using microstrip line, coaxial probe, aperture

coupling and proximity coupling.

Fig.3-9: Structure of patch antenna with microstrip feed


75

Take rectangular patch antenna that shows in Fig.3-9 as an example; it is fed by a

microstrip line. The antenna itself has the main structure of a half-wavelength shaped

metal patched on a dielectric substrate with a relative permittivity, εr. This rectangular

patch is usually designed to operate near the resonant frequency. At resonant

frequency, the imaginary part of the impedance is zero. When signal feeds into the

patch element, a fringing field was produced due to the edge effect of the metal patch.

However, the fringing field acts as an additional length to the patch. Therefore, when

the length of the patch is slightly less than a half wavelength of RF signal propagating

the dielectric substrate media, this device will become a radiative microstrip antenna.

Since the value of length depends on the substrate media as well as the height and

width of the patch then an approximation formula to get the resonant length and it is

shown as:

(18)

where is the free-space wavelength, d is the wavelength in the dielectric, and r is

the substrate dielectric constant.

Electric field distribution was plotted out to represent the fringing field around the

patch and it is shown in Fig.3-10. As shown in the top view of the patch, its radiation

can be represented by two slots, separated by a transmission line of length L and open

circuited at both ends. Along the length of the patch, there are maximum voltage and

minimum current because two ends of the patch can be considered as open circuits.

The fields at edges can be resolved into normal and tangential components with

r

dL

49.049.0


76

respect to the ground plane, which is shown in the patch antenna cross-section in

Fig.3-10.

Top View

Cross-section view

Fig.3-10: Top view and cross-section of patch antenna with electric field around it.

Seeing by the cross-section, the electric field along the edges of the patch has

different direction but with the same magnitude. Since two edges that are separated by

the length L is half-wavelength apart then this can be considered as 180 degree out of

phase. Therefore the total fringing field along length L is null. Seeing from top,

electric fields at two edges are in phase with the same magnitude values. Hence the

maximum radiation field pattern is normal to the surface of the patch antenna.


77

These antennas have the advantages of low profile with conformability for planar and

non-planar surfaces, low-cost fabrication, ease of integration using modern printed-

circuit technology and compatible with MMIC designs. Bandwidth of the patch

antenna itself can be expressed as 𝐵. 𝑊. ∝ 𝜀𝑟−1

𝜀𝑟2

𝑤

𝐿𝑕, where w and L are the width

and length of patch, h is the substrate height[77]. According to this equation, height of

the substrate controls the bandwidth value. Therefore bandwidth will decrease as the

substrate thickness decreases. As the consequences planar microstrip patch antenna

provides narrow bandwidth for the thin substrate at millimetre-wave frequency range.

When the patch antenna is fabricated on a relatively thin substrate, it typically has

only approximately 5% bandwidth with respect to the centre frequency. According to

the requirement of high data transmit rate, simple patch antenna cannot meet the

needs.

Many techniques have been used to increase the bandwidth of microstrip patch

antenna, such as using a thicker substrate with lower relative permittivity, or by using

a T-shape feed probe.

3.3.2 Slot Antenna

Slot antenna is widely used in microwave and millimetre-wave communication

systems for its feature of omni-direction. It radiates power around azimuth with

horizontal polarization. In Fig.3-11 a waveguide slot antennas operating from 2 GHz

to 24 GHz is shows with its structure of series slots aligned together.

Fig.3-11: waveguide slot antenna working from 2 to 24 GHz [78]


78

In fact, if a thin slot is located in the centre of an infinite ground plane, it can be seen

as the complement to a dipole antenna in free space, which is described by H.G.

Booker [78]. Fig.3-12 shows the equivalent structures for both slot and dipole

antennas. Both of them have same dimensions but only different in the E-field and H-

field directions which are swapped. Therefore, the slot can be called as a magnetic

dipole. When wave propagating through a slot antenna, it has a 90 degree polarization

due to its E-H field arrangement.

Fig.3-12: Slot antenna in infinite ground plane [78]

3.3.3 Dipole Antenna

An ideal dipole is built with two wires acting as arm of dipole and the total length of

this arm is half-wavelength, which is shown in Fig.3-13. If the RF wave propagating

into a dipole, a sine wave current will be formed along the dipole and it has a

maximum distribution in the centre while the minimum distributes at the two ends.

This is the reason why most of the feeders used for this type of antenna are placed in

the centre.


79

Fig.3-13: A simple dipole antenna with feeder

As the ideal feed impedance of an half-wavelength antenna is 75 Ω in the free space

[76], then feeder with an impedance of same value should be chosen to match the

antenna. The position of the feeder affects the total input resistance as well. The feed

point resistance will be higher if the dipole is not fed at the centre [79]. For example,

if there is a distance d from one end of the dipole to the feed point, then the resistance

can be expressed as:

Rr =75

sin2 2πd

λ

(19)

For a basic dipole antenna, power is radiated out along the plane, which is

perpendicular to the antenna surface. As it has two poles, an electric field is formed

around the dipole poles as shown in Fig.3-14 (a). Current flows through the arm of the

dipole in a same direction and radiates out. The electric field finally ends at both ends

of the dipole. Fig.3-14 (b) shows the radiation pattern of a simple dipole antenna

without grounded plane. The pattern radiates into two opposite directions. Ideally, it

has a maximum radiation along the axis of centre feed point.


80

Fig.3-14: (a) Sketch of electric field around the dipole

(b) Radiation pattern of the simple dipole

An antenna that is defined with dipole structure should have an electrical length of

half wavelength. However it is not exactly the value of half wavelength or multiple of

that since the exact length is affected by a factor A. The different factor A is given by

the ratio of the antenna length to the thickness of the metal. A plot of their

relationships is displayed in Fig.3-15.

Fig.3-15: Factor A against the wavelength to thickness ratio [79]


81

Since the resonant frequency depends on the physical length of dipole arm, it is easily

chosen for more accurate length based on the equation 20 and factor A as described in

Fig.3-15 [79].

𝑙𝑒𝑛𝑔𝑡𝑕 𝑜𝑓 𝑑𝑖𝑝𝑜𝑙𝑒 𝑚𝑒𝑡𝑒𝑟 =150 × 𝐴

𝑟𝑒𝑠𝑜𝑛𝑎𝑛𝑡 𝑓𝑟𝑒𝑞𝑢𝑒𝑛𝑐𝑦 𝑀𝐻𝑧 (20)

3.4 Conclusion

Background and relevant theories of the antenna design are firstly given in the

beginning of this chapter. In this work, planar antenna is the first choice in order to

designing an on-chip antenna that can operate at 60 GHz. As discussed previously in

the chapter, planar antenna can be fabricated using Si CMOS technology and can be

fully integrated with the RF front-end. Mechanism of how planar antenna works has

been presented in Section 3.3.1 to 3.3.3. Microstrip patch antenna is easy to fabricate

but has a narrow bandwidth while slot antenna is also easy to fabricate and it has

feature of wider bandwidth since the slot can be cut out in different shapes for the

aforementioned purpose.

In this work, planar slot antenna has been chosen as main design due to the bandwidth

requirement. More specifically, the U-shaped slot antenna can provide wider

bandwidth. The slot is cut out of a planar patch and it can generate more than one

resonant frequency by changing the dimension of “U”. By mixing multi resonant

frequencies together, bandwidth becomes wider than single slot or patch antenna. This

structure will be discussed in details in Chapter 4. Besides U-shaped slot antenna,

other planar antennas have also been considered and designed and these will be

discussed in Chapter 5 and Chapter 6.

Chapter 4 Wideband Planar Antenna Investigation

82

Chapter 4

Wideband Planar Antenna Investigation

4.1 Background

Planar antenna is widely used in millimetre-wave antenna design. It is easy to design

and fabricate. However, one single planar antenna such as microstrip patch provides a

narrow bandwidth. This antenna should have a bandwidth of at least 3 GHz to support

the transmission of HD data, with efficiency of greater than 70%. There are many

methods to increase the antenna radiation bandwidth, which can be achieved by

decreasing the dielectric constant of the substrate, increasing the thickness of

substrate, adding superstrate planar upon it, adding resonant cavity or changing the

antenna structure with tapped slots.

In this chapter, U-shaped slot antenna was designed, simulated and analysed. This

antenna is based on a microstrip patch with a U-shaped slot added, which enable it to

work on different frequencies. Different parameters of this U-shaped slot antenna

structure will be simulated and optimised. Since resonant frequency depends on the

antenna structure, different frequencies will be arranged together by varying the

antenna structure.


83

However, the U-shaped slot antenna structure performance can be degraded due to the

limitation of substrate height because it is easily affected by the ground distance.

Therefore, a folded dipole antenna was introduced. Folded dipole antenna has a wider

bandwidth than normal dipole and it has an arm length of half wavelength. A full

wave simulation of the folded dipole antenna will be presented here.

4.2 U-shaped Slot Antenna Design and Optimisation

A U-shaped slot is developed for its wide bandwidth by creating two or more closely

adjacent resonant frequencies. The U-shaped slot is cut approximately symmetrical in

the centre of the patch. With this structure, the antenna gives a bandwidth of

approximately 40% for VSWR < 2.

Fig.4-1: Simple structure of a U-shaped slot patch antenna element

A U-shaped slot patch antenna is simple, as shown in Fig.4-1. It consists of a U-

shaped slot and a microstrip feeding. Due to the coupling between the slot and

rectangular metal planar, more than one resonant frequencies are generated. It was

found that the U-shaped slot patch antenna can be designed to attain 50% impedance

bandwidth as well as 30-40% gain of bandwidth [80].


84

4.2.1 Design of a Wide Bandwidth U-shaped Slot Patch Antenna

(a)

(b)

Fig.4-2: (a) Side view of antenna layout with parameters indicated (b) Top view

structure of the antenna

Fig.4-2 shows the top and side view layout of antenna with parameters that affect its

radiation performance. The side view illustrated in Fig.4-2(a) shows that the U-shaped

antenna structure consists of four layers, namely the silicon layer, ground layer, SiO2

layer and U-shaped slot patch layer locating from bottom to top. Other than the U-


85

shaped slot patch layer, each layer has its own varied width, known as Wsub, Wgnd and

m, respectively. In Fig.4-2(b), the area of m×n refers to the silicon layer and

Wsub×Lsub refers to the SiO2 layer size, which are varied by changing Sy and Sx.

Between those two layers is the ground layer. It has a width of Wsub and a length of

Lsub. The gl and gw are the difference between the dimension of ground plane and SiO2

layer. Therefore the ground size can be changed by varying gl and gw. The metal area

formed by U-shaped slot affects the energy absorbsion and radiation as well. Varying

Sw and Sl can change the slot location. By focusing only on the U-shaped slot, three

gap sizes, rectangular metal width and length could all affect the antenna radiation

performance. The effects from these different parameters on energy radiation will be

discussed later in details. The thickness of SiO2 layer is of 40 μm with a relative

permittivity εr of 4 and the silicon layer is of 150 μm with a relative permittivity εr of

11.9. In this structure, a microstrip line was chosen as feeding. Based on the height of

SiO2 layer, the feed length fl was calculated by the microstrip line calculator defined

as 0.681 mm, which made a match of 50 Ω.

4.2.2 Wave Port Determination in HFSS

The structure of U-shaped slot antenna covered by an air-box with full wave radiation

boundary in HFSS Ver.12 is shown in Fig.4-3. Ansoft HFSS assumes that all

structures are completely encased in a conductive shield with no energy propagating

through it by default. Wave ports are defined in HFSS to indicate the area where

energy enters and exits the conductive shield. With those ports, HFSS assumes that

each of them is connected to a semi-infinite long waveguide that has the same cross

section and same electrical materials known as the port. When the wave port is


86

defined correctly, there is a perfectly matched condition at the port in simulation

mode and S-parameters are normalized to frequency only dependent impedance. This

is important for the structural design and simulation. Unsuitable wave port size could

cause inconsistent results that affect the accuraciest of the interaction between the

wave ports and characteristic impedance of the structure. Once the wave port is

correctly designed, a uniform cross section with a certain length is formed. Within the

uniform cross section model the signal of non-propagation modes dies off so that the

accuracies could be improved. The wave port for the microstrip line feed shown in

Fig.4-4 always has a shape of rectangular planar with the correct width and height.

Since an electric field is formed at the surface of the wave port as illustrated in Fig.4-

4, the surface size is taken into consideration. If the wave port size is made smaller

than the waveguide, some RF energy that is unable to propagate into the structure

through the waveguide will be reflected. Overly large wave port is undesirable

otherwise it would weaken the electric field on the feeding cross section surface.

However, there is a range of wave port width within which the resonant frequency

remains constant.

Fig.4-3: Building of radiation air-box


87

Fig.4-4: Electric field distribution on wave port

To find a suitable wave port size, the effects with various width are investigated. By

changing the relationship between a and fw, with a being the wave port width and fw

being the feed cross section width, simulation is carried out by observing the return

loss S11. With "Solve ports only" in HFSS solution setup setting, results can be

verified to ensure propagation of the right modes. Amongst many port sizes tested,

only three were chosen for full structure simulation.

Fig.4-5 shows the results tested in full wave mode with three different port sizes of

8×fw, 10×fw and 12×fw. Return loss S11 and VSWR plots against frequency were

shown separately in (a) and (b). It was obvious that the size of 10×fw as shown in red

trace gives a best result in terms of return loss or VSWR. Meanwhile, when the width

has a value greater than 15×fw, there was no power radiated. Notably, between the

port width size of 10×fw and 12×fw, resonate frequency resulted in a slight change,

indicating that it is the correct wave port size range.


88

(a)

(b)

Fig.4-5: Comparison between different wave port sizes (a) plot of S11 (b) plot of VSWR

Plot sizes of 10×fw, 8×fw and 12×fw are shown in red, blue and green traces respectively.


89

4.2.3 High Gain and Directivity Structure

Fig.4-6: (a) Variables (L1, g1, g2, g3) of U-shaped slot antenna;

(b) Dimensions of each variable for an antenna gain of 5.29 dB

The structure of a U-shaped slot patch antenna is expected to be closely associated

with the performance of antenna radiation. Among those variables shown in Fig.4-6,

length L1 and U-shaped slot width g1, g2 and g3 could directly determine the antenna

radiation gain performance.

The structure built on a 40 μm SiO2 substrate shown in in Fig.4-6(b) gives a gain of

5.29 dB, directivity of 7.29 dB and a reflection coefficient below -20 dB. Simulation

was carried out on variables of L1, g1, g2 and g3 with different values, some results

with comparable changes are shown in Table 4-1. It was observed that the length of

L1 can affect the antenna radiation gain. Once the L1 reached 0.2 mm, the gain

dropped to nearly half to that of directivity, meaning that the radiation efficiency is

below 50%. This was due to partial power reflection at input of antenna as indicated

by the reflection coefficient which stayed on -11 dB.


90

Table 4-1: Performance changes with L1 and slot gap width

L1(mm) g1 =g2=g3 (mm) Gain (dB) Directivity (dB) S11 (dB)

0.20 0.06 3.56 6.78 -11

0.14 0.06 5.29 7.29 -20

0.10 0.06 5.12 7.09 -19

0.14 0.03 5.28 7.18 -23

0.14 0.04 5.35 7.21 -28

0.14 0.07 5.23 7.23 -18

Fig.4-7 shows the radiation gain at E-plane and the reflection coefficient S11 plot

when L1 was designed with three different lengths. Fig.4-7(a) illustrates the radiation

pattern at E-plane of antenna gain. The values of the antenna parameter L1 are

represented in different coloured lines. Fig.4-7(a) shows that RF energy propagates

towards the range where θ varies from -90° to 90°. Maximum gain is achieved when θ

= 0°.

(a)


91

(b)

Fig.4-7: (a) Radiation pattern of gain at E-plane and (b) Reflection coefficient S11 when

L1 = 0.1 mm, 0.14 mm, 0.2 mm

In Fig.4-7(b), the reflection coefficient, S11, when L1 = 0.2 mm, 0.14 mm and 0.1 mm

is shown in blue, red and purple traces, respectively. The results showed that the

highest gain and lowest S11 could be achieved when L1 = 0.14 mm, indicating that the

antenna radiates maximum power out into the directional patch.

Meanwhile, a slight change of gaps in U-shaped slot also affected the radiation results,

as shown in Fig.4-8. Simulations were made when three slots, g1, g2 and g3 have the

width of 0.03 mm, 0.04 mm and 0.07 mm. The radiation pattern of antenna radiation

gain and the reflection coefficient are shown in Fig.4-8(a) and (b), respectively. With

g1 = g2 = g3 = 0.04 mm, the highest gain of 5.35 dB and low reflection coefficient at

two frequencies were achieved.


92

(a) Radiation pattern of gain at E-plane

Blue trace: g1 = g2 = g3 = 0.03 mm

Red trace: g1 = g2 = g3 = 0.04 mm

Purple trace: g1 = g2 = g3 = 0.07 mm

(b)

Blue trace: g1 = g2 = g3 = 0.03 mm

Red trace: g1 = g2 = g3 = 0.04 mm

Purple trace: g1 = g2 = g3 = 0.07 mm

Fig.4-8: (a) Gain and (b) S11 plots when g1 = g2 = g3 =0.03 mm, 0.04 mm, 0.07 mm


93

Therefore, when the length L1 is approximately 0.14 mm and the U-shaped slot has a

gap width of 0.04 mm, a maximum gain of 5.35 dB with -28 dB reflection coefficient

can be obtained, whereas the bandwidth shown in S11 plot is only 1 GHz with a

resonant frequency at 75.7 GHz.

From Fig.4-8 it was observed that the U-shaped slot antenna worked at three

frequencies at 68 GHz, 75.7 GHz and 94.75 GHz. Radiation patterns in Fig.4-9 and

Fig.4-10 illustrate radiation gain and directivity at the same resonant frequency at

75.7 GHz. Two radiation patterns also show that gain is 5.34 dB, directivity is 7.21

dB and the maximum power gain occurs at the peak of radiation pattern.

Fig.4-9: Radiation pattern of gain when L1 = 0.14 mm and g1 = g2 = g3 = 0.04 mm


94

Fig.4-10: Radiation pattern of directivity when L1 = 0.14 mm and g1 = g2 = g3 = 0.04

mm

Fig.4-11: Reflection coefficient plot when L1 = 0.14 mm and g1 = g2 = g3 = 0.04 mm


95

Fig.4-11 shows the reflection coefficient plot when the length L1 = 0.14 mm. The plot

indicates that the U-shaped patch antenna resonate at two different frequencies, where

the marker m1 and m3 are. At each resonant frequency, the 10 dB radiation bandwidth

shown in Fig.4-11 is around 1 GHz.

There are other parameters of the U-shaped slot antenna structure that affect radiation

gain and bandwidth, which will be discussed in the following sections.

4.2.4 Ground Effect

As shown in the cross-section layout of U-shaped slot antenna substrate, the ground is

located between two kinds of semiconductor material, Si and SiO2. The height, H,

from the antenna metal layer to the ground is the thickness of SiO2 layer. To work at

frequency as high as 60 GHz, this value H is so small that the two metal layers are too

close to each other. In this case, a waveguide is formed due to two closely placed

metal planes. This waveguide takes some of the RF energy away towards other sides

while not radiating out. Designers put the reflector plane at the bottom of the silicon

layer to prevent the formation of waveguide while increase the radiation bandwidth as

shown in Fig.4-12(b). However, silicon substrate in standard Si CMOS technology

has a low resistivity of 10 Ω·cm, so the silicon substrate could absorb energy and

consequently causing energy loss. To avoid RF energy from propagating into the

silicon material, one of the alternatives is to place the ground plane on the top of

silicon layer, as shown in Fig.4-12(a).


96

(a) (b)

Fig.4-12: Ground planar location (a) ground plane on top of silicon layer (ground

location in this design) (b) ground plane on the bottom of silicon layer

Another way to decrease the energy loss is by reducing the ground plane size. As

shown in Fig.4-13, when the ground plane gets smaller, RF energy could be stopped

at the edge of the substrate and consequently radiate out due to the edge effect.

Fig.4-13: Ground varying in HFSS structure design

Simulations were taken at two different ground sizes and the results are shown in

Fig.4-14 and Fig.4-15. When the ground size parameters gw = gl = 0 mm, radiation

gain of this antenna is 1.64 dB as shown in Fig.4-14, marker m2. However, a radiation

gain of 2.18 dB is achieved as marker m1 in Fig.4-14 shows, when gw = 0.25 mm and

gl = 0.5 mm.


97

Fig.4-14: m1, the highest gain at E-plane when gw = 0.25 mm and gl = 0.5 mm

m2, the highest gain at E-plane when gw = gl = 0 mm

Fig.4-15: Blue trace, VSWR plot when gw = 0.25 mm and gl = 0.5 mm

Red trace, VSWR plot when gw = gl = 0 mm


98

The ground with gw = 0.25 mm and gl = 0.5 mm gave the gain of 2.18 dB while the

other only has a gain of 1.6 dB. In VSWR plot, blue trace refers to the smaller ground

planar which gives a wider bandwidth than the red one.

4.2.5 Feed Position Effect

Matching is an important element of the design. Only when the circuit is well

matched, the energy can be radiated sufficiently. Impedance matching is a way that is

used in the circuit matching technique. In order to have a successful impedance

matching, the impedance of each part should be calculated or simulated out. To

measure different points along the edge of U-shaped slot, the impedance is different

unless the structure is exactly the same at both sides. However, the reflection

coefficient S11 enables the estimation of the position where the perfect circuit

matching occurs.

Fig.4-16: Reflection coefficient and VSWR varying at three feeding position of -0.25

mm, 0 mm and 0.25 mm


99

As shown in Fig.4-16, when the position was at the centre 0 mm (purple trace), it

showed only one resonant frequency at 77 GHz with a reflection coefficient of -6 dB.

At -0.25 mm (red trace), there was another resonant frequency at 90 GHz in addition

to the resonation at 77 GHz. The wide bandwidth at 90 GHz can be used for large

amount of data transfer in wireless communication. With the help of resonant

frequency at 77 GHz, this antenna became a dual-band.

4.2.6 Length of Rectangular Metal

The varying length of rectangular metals mainly change the impedance of the U-

shaped slot, thereby affecting the circuit matching. As shown in Table 4-2, the highest

gain and widest bandwidth is obtained when L1 = 0.05 mm and L2 = 0.09 mm.

Table 4-2: L1, L2 varying effects

L1 (mm) L2 (mm) Gain (dB) Bandwidth (GHz)

0.01 0.09 2.56 4.5

0.05 0.09 2.44 5.5

0.10 0.09 3.17 0.2

0.20 0.09 2.54 0.4

0.05 0.12 1.76 2.0

4.2.7 Rectangular Metal Size

Three blocks of rectangular metals are the main makeup of U-shaped slot structure.

The one in the middle has the main impact on the antenna performance as it couples

with those surrounding slots to radiate energy. Each of the rest two rectangular metals

only has one edge coupling with slots so they don't have much effect to antenna

performance as the middle metal patch.


100

According to Fig.4-2, w1, w2 and w3 are three parameters of metal patch width. If they

have the value of 0.58 mm, 0.4 mm and 0.58 mm, respectively, the reflection

coefficient and VSWR are plotted in Fig.4-17, showed three single ripples occuring at

three frequencies separately.

(a)

(b)


101

(c)

(d)

Fig.4-17: (a) Reflection coefficient and (b) VSWR when w1 = 0.58 mm, w2 = 0.4 mm,

w3 = 0.58 mm

(c) Reflection coefficient and (d) VSWR when w1 = 0.4 mm, w2 = 0.3 mm, w3 = 0.4 mm


102

The simulation results in Fig.4-17 also showed that by varying the width of metal

sizes, either a wide bandwidth antenna or a multiple band antenna could be designed.

Finally when w1 = 0.4 mm, w2 = 0.3 mm, w3 = 0.4 mm, a bandwidth of 5.5 GHz was

obtained.

4.2.8 U-shaped Slot Gap Width

Comparing with length, the width of the slot gap is quite small. As Fig.4-18 shows,

variables g1, g2 and g3 determine the dimension of the “U” structure. In this section,

the gap width g1, g2 and g3 of “U” slots and their impacts on the bandwidth and

reflection coefficient are discussed.

Fig.4-18: U-shaped slot gaps with width g1, g2 and g3

When g1 = g2 = g3, as shown in Fig.4-19(b), the bandwidth around two resonant

frequencies had the same width of 2 GHz. When g1 and g2 were different, a wider

bandwidth was formed at one resonant frequency. For example in Fig.4-19(c) where

g1 = g3 = 0.03 mm and g2 = 0.005 mm, the bandwidth around resonant frequency 77

GHz was much wider than the one at 90 GHz. Once the value of g1 was greater than

0.03 mm, reflection coefficient showed three resonant frequencies with different

bandwidth. VSWR plot in Fig.4-20 improved this phenomena when g1 = g3 = 0.05

mm and g2 = 0.005 mm.


103

(a) g1 = g3 = 0.005 mm, g2 = 0.03 mm

(b) g1 = g3 = 0.005 mm, g2 = 0.005 mm


104

(c) g1 = g3 = 0.03 mm, g2 = 0.005 mm

(d) g1 = g3 = 0.05 mm, g2 = 0.005 mm

Fig.4-19: (a)(b)(c)(d) Reflection coefficient S11 varying with different slots g1, g2 and g3.


105

Fig.4-20: VSWR when g1 = g3 = 0.05 mm, g2 = 0.005 mm

In addition, the gap g2 could change the bandwidth as well, as shown in the VSWR

plot in Fig.4-21 where g2 = 0.001 mm, g2 = 0.01 mm and g2 = 0.02 mm, respectively.

It was observed that at high frequency around 90 GHz, the bandwidth achieved 6.6

GHz when g2 = 0.01 mm.

Fig.4-21: Red trace, g2 = 0.001 mm; Purple trace, g2 = 0.01 mm; Green trace, g2 = 0.05

mm


106

4.2.9 Thickness of SiO2 Layer Effect

By increasing the thickness of SiO2 from 40 μm to 100 μm, the gain of the antenna

could have a better improvement. The gain at E and H plane is shown in Fig.4-22.

The highest gain marked as m1 achieved 5.15 dB while the highest directivity marked

as m2 reached 6.27 dB.

When the SiO2 layer thickness is 100 μm, less power was dissipated from the

substrate edge. Simultaneously, the signal returned back decreased to -20 dB and a

broadband was obtained as shown in Fig.4-23. More than 3 GHz frequency range

stayed below the standard, VSWR = 2, meaning that the antenna had a bandwidth of 3

GHz.

(a)


107

(b)

Fig.4-22: (a) Radiation pattern of gain when H = 100 μm

(b) Radiation pattern of directivity when H = 100 μm

Fig.4-23: VSWR of U-shaped slot antenna when H = 100 μm


108

4.2.10 Final U-shaped Slot Design Structure

The final design came out when the above parameters were considered into the

performance of the antenna. The parameters chosen in the final design are outlined in

Table 4-3

Table 4-3: Final parameter values chosen with a SiO2 substrate height of 40 μm

Parameters Value (mm)

Ground planar gw 0.25

gl 0.5

Feed a 10×fw

fx -0.25

“U” shape L1 0.14

L2 0.9

w1 0.4

w2 0.3

w3 0.4

Gaps g1 0.005

g2 0.001

g3 0.03

Fig.4-25 and Fig.4-25 show the maximum directivity and the gain of the antenna at

6.25 dB and 2.44 dB in both E and H plane, respectively. The antenna efficiency was

not high enough to radiate all the power out. The gain of this antenna was less than

half to that of the directivity. This was mainly caused by the narrow space, 40 μm

between the U-shaped slot patched antenna and the ground plane.


109

Fig.4-24: Radiation pattern of antenna power gain in E, H plane with a 40 μm substrate.

Fig.4-25: Radiation pattern of antenna directivity in E, H plane with a 40 μm substrate.


110

Fig.4-26 shows the reflection coefficient and bandwidth of the U-shaped slot antenna.

It was observed that the antenna had dual resonant frequencies very close to each

other.

Fig.4-26: Reflection coefficient and VSWR of the final structure U-shaped slot antenna

The simulation results shown in Fig.4-26 suggested that the U-shaped slot antenna

could operate at two resonant frequencies, 73.5 GHz and 76.5 GHz. However, the

bandwidths of these two frequencies were 1 GHz, which could not meet the

requirement of 3 Gbps HD data transmission. The main reason of narrow bandwidth


111

was due to the thickness of the substrate. The U-shaped slot antenna was based on the

microstrip patch antenna; hence the ground size and the substrate height significantly

affected the radiation efficiency. When the ground plane is too close to the patch, with

a distance much smaller than a quarter of the wavelength, the electricmagnetic field

distribution would be totally different than the usual patch antenna and causing power

loss by the image current, and ultimately decreasing the radiation efficiency.

However, the main dielectric structure of Si CMOS technology typically has a heavy

loss of silicon bulk with height of 250 μm and a SiO2 layer with maximum height of 8

μm. Therefore the challenge remains in designing high efficiency on-chip antennas

based on Si CMOS technology.

4.3 Folded Dipole Antenna

Folded dipole antenna is based on dipole antenna, which has been discussed in

Chapter 3. This antenna generates electromagnetic field along the half-wavelength

arm while not cooperating with ground plane. Besides, it provides wider bandwidth

than dipole.

Fig.4-27: Folded dipole 2 m band antenna [81]


112

Folded dipole antenna is based on dipole while both sides of the arm are ended with

elements, as shown in Fig.4-27. It has been reported that folded antennas can provide

a bandwidth enhancement of more than 50% [82, 83]. The basic structure of the

folded dipole antenna could be proposed on chip is shown in Fig.4-28. Some

investigation results proved that this antenna structure has the same wide bandwidth

as its two ends folded as a loop antenna [84].

The difference between d1 and d2 affects the input dipole impedance. Once the ratio d1

/ d2 = 1 / 3, a 100 Ω input impedance is obtained [84]. In order to match the

impedance at the feeding point, the feeder is chosen as a 100 Ω coplanar strip line

(CSP).

Fig.4-28: Parameters of a folded dipole antenna

To work at 60 GHz, the length of folded arm is required to be half wavelength, which

is 1.25 mm according to the SiO2 substrate. Strip width of the folded arm has a value

of d1 = 0.02 mm, d2 = 0.06 mm, respectively.


113

4.3.1 Simple Folded Dipole Antenna Simulation

In the beginning, a simple folded dipole antenna based on the SiO2 substrate with a

relatively dielectric constant of 11.9, was simulated in Ansoft HFSS Ver.12.

Dielectric material SiO2 has a non-crystalline structure. It has electrical properties

such as low dielectric constant and low dielectric loss [85]. Thus, the loss tangent at

60 GHz can be ignored. In order to meet the needs of commercial use, centre

frequency of the antenna was designed to be 60 GHz. HFSS gave an analysis of 20

passes with the numerical delta of 0.02. The numerical delta in HFSS means the final

numerical error tolerance, which can be set by the user. The smaller the numerical

delta, the accurate the results will be. However, smaller numerical delta could

significantly prolong the computation time. Numerical delta of 0.02 was suggested by

the user manual. Antenna model simulated in HFSS Ver.12 was shown in Fig.4-29. It

was fed by a 100 Ω CPS with a space between the strips of 0.03 mm and a substrate

thickness of 40 μm. However the wave port needs to be designed with suitable size,

which will be discussed in Section 4.2.2. In this folded dipole antenna, it has a width

that is 2.5069 times of the feeder width.

Fig.4-29: HFSS simulation layout of a simple folded dipole antenna


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Simulation was taken against the frequency swept from 50 GHz to 70 GHz and the

resulted bandwidth of 3 GHz from 60 GHz to 63 GHz was obtained. The far field

radiation pattern of antenna directivity and gain were displayed in Fig.4-30 (a) and (b),

respectively.

(a)

(b)

Fig.4-30: (a) Directivity at 60 GHz; (b) Gain at 60 GHz


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The radiation patterns showed a standard dipole pattern in E-field and a monopole

pattern in H-field because no ground plane was added on the bottom of the substrate.

In order to increase the antenna radiation gain and bandwidth, a cavity can be added

at the back of antenna, which will be discussed in Chapter 5.

4.4 Conclusion

In this chapter, a U-shaped slot antenna with 1 GHz bandwidth was investigated and

designed. Effects of U-shaped slot antenna dimension variables have been analysed

and discussed. According to the simulation results, the length of U-shaped slot patch,

L1 determined the radiation gain, the gap size, and the distance between two gaps

affect the bandwidth of antenna. By generating multiple resonant frequencies nearby,

the antenna bandwidth can be enhanced. However, distance between operating U-

shaped slot patch antenna and ground plane, i.e., the substrate height, also has effects

on the antenna bandwidth performance. Given that most Si CMOS technologies have

only a few micrometers thick SiO2 layer, this U-shaped slot antenna may have limited

applications for on-chip antenna realisation.

Folded dipole antenna was presented in the end of this chapter and proves to be able

to provide wider bandwidth. Further design of this folded dipole antenna with

metamaterials will be purposed in Chapter 5.

Chapter 5 AMCs for Millimetre-wave Antenna Application

116

Chapter 5

AMCs for Millimetre-wave Antenna

Application

5.1 Introduction

In Chapter 2, it has been indicated that recent millimetre-wave antenna design focuses

on the property of low cost and high performance. Si CMOS is the technology that is

used for mass production due to its low cost and high compatibility properties. It is

the foundation for a modern, digital world. Computer memory, CPUs, digital signal

processors and many other functional chips are made on Si CMOS. We wouldn‟t be

where we are now if there was no Si CMOS technology. Recently Si CMOS

technology has advanced from low frequency to microwave/millimetre-wave

applications. Si CMOS technology builds multiple metal layers on silicon bulk

substrate. Each metal layer is separated by a SiO2 layer. The multilayered structure of

this technology can be utilised to make a very compact circuit layout with the

integration of planar antennas, making it a very attractive alternative to conventional

GaAs MMIC technology for microwave/millimetre-wave applications. Si CMOS on-

chip antennas for millimetre-wave wireless communications, especially for 60 GHz


117

home entertainment networks are highly desirable as they can significantly reduce the

cost. However, Si CMOS on-chip antennas suffer from high substrate loss, resulting

in low efficiency. Innovative design approaches will be key to successfully utilising

the Si CMOS technology for on-chip antenna applications.

As we discussed in Chapter 4, the primary limitation of on-chip antennas is the

bandwidth, which depends on the thickness of the structure with respect to the centre

wavelength. A U-shaped slot antenna may not be suitable for on-chip antenna

applications due to its restricted ground distance requirement. To reduce the influence

from the ground plane, two approaches can be taken: to have an antenna which is not

influenced by ground or to replace the ground metal plane with another reflective

material. The folded dipole antenna, however, generates electric current without the

help of a ground plane. We will study this antenna in details here.

In this chapter, we also introduce the AMC structure to the millimetre-wave antenna

design. AMC structure is a new type of metallic electromagnetic structure that has

been developed to have a HIS. This structure was first presented in 1999 [44]. It is

made of continuous metal but does not conduct AC current within a forbidden

frequency band. The difference between AMC and normal conductors is that this new

surface does not support propagating surface waves and also reflects electromagnetic

waves without phase change. All these properties make AMC suitable for replace the

normal conductive ground for reflecting electromagnetic waves without cancellation

caused by phase reversal. This chapter will discuss how AMC structure can be

applied to millimetre-wave antennas.


118

In Sections 5.2 and 5.3, two cavities are added to the back of a half-wavelength folded

dipole antenna. They are a metal cavity that is made of a normal conductor and an

AMC cavity that consists of HIS structures. The theory as to how the AMC cavity

works and its applications will also be presented and discussed. Considering that an

antenna used at 60 GHz should have high efficiency, compatibility and low cost

properties, two novel antennas with newly reported AMC structures are also proposed,

designed and presented in Sections 5.4 and 5.5.

5.2 Antenna with Backed Metal Cavity

As discussed in Chapter 4, the U-shaped slot patch antenna results revealed that the

thickness of the dielectric substrate affects bandwidth performance. A better antenna

performance can be achieved by placing the antenna in a resonant cavity. In this

chapter, two types of cavities are designed to provide a resonant effect to a half-

wavelength folded dipole antenna that works at around 60 GHz. One of the reflector

cavities is formed by the AMC structure. The other is formed by a simple metal cavity.

AMC is built up with a well-arranged periodic metallic unit pattern and forms a pure

magnetic conducting plane with high surface impedance.

A resonant cavity is usually used as an electromagnetic resonator, in which the RF

signal propagating inside keeps vibrating. Since the cavity‟s surfaces are enclosed and

it only has one entrance, a specific cavity size causes a reflection to the wave at a

specific frequency. At that specific frequency, the RF signal incident is bounced

backwards and forwards within the cavity with low loss. Therefore resonant occurs.


119

The standing wave intensity is increased during the resonant phenomenon, so that

more power will be radiated out and the antenna gain enhanced.

Bandwidth can be improved if the radiation device is suspended a distance H above

the ground plane [86]. Fig.5-1 gives the structure of the antenna backed by a metal

cavity. The folded dipole antenna is attached to the bottom of a 250 μm thickness

SiO2 substrate and placed in the centre of a 2.9 mm × 4 mm cavity.

Fig.5-1: Metal cavity structure in HFSS

Fig.5-2: Cross section of the metal cavity

Silicon Dioxide


120

Fig.5-2 shows the cross section of the metal cavity structure. To improve the matching

state, a U-shaped slot is created in the cavity wall near the feeding position. In the cavity,

the RF signal radiates out from the antenna and is reflected back due to the metal baffle.

The cavity height H should be around one quarter of the wavelength so that the metal

ground at the cavity bottom acts as an in-phase reflector. As a result, bandwidth and

radiation efficiency are increased. A bandwidth enhancement of 4 dB was obtained in the

HFSS simulation as shown in Fig.5-3.

Fig.5-3: Return loss of the folded dipole antenna backed by a metal caivty

Fig.5-4: 3D plot of the radiation E-field


121

(a)

(b)

Fig.5-5: (a) Gain, 5.58 dB of the antenna with metal-cavity at resonate frequency

(b) Directivity, 5.77 dB of the antenna with metal-cavity at resonate frequency


122

Fig.5-5 showed a radiation pattern of the antenna backed by a metal cavity at 62 GHz.

The simulated radiation efficiency at resonant frequency is 97.9%. Compared to the

gain obtained in Chapter 4, Gain increased from 2.44 dB to 5.58 dB.

The radiation pattern in Fig.5-5 indicates a better directivity compared to that without

the metal cavity, increasing to 5.77 dB. 3D plot of the E-field shows in Fig.5-6 was

generated by ten divisions of angle ϕ in azimuth plane of Fig.3-3 in order to speed up

the full wave simulation. Therefore it is not smooth and plump enough to present full

360 degree of angle ϕ. The cavity ground acts as a reflector and it reflects the

radiation power back to the feeder. As shown in Fig.5-6, the radiation pattern changed

its shape and gain is enhanced.

Fig.5-6: Radiation pattern with the effect of a reflector

5.3 Antenna in AMC Cavity

As discussed in Section 5.2, conductive surfaces such as the metal cavity can be used

as a reflector in antenna design [70].


123

The reflector redirects nearly half the radiation wave in the opposite direction so that

the gain has a 3 dB (from 2.44 dB to 5.58 dB) improvement as Section 5.2 proved.

However, one must very careful when using a conductive surface to reflect the waves.

It reverses the phase of reflected waves by 1800 for a good conductor and forms a

zero electric field at the surface. Therefore when the electromagnetic wave travels

through the surface, the reflected wave changes its phase. This property results in the

image currents on the conductive sheet cancelling those currents in the antenna and

decreasing the radiation performance, if the distance between the antenna and the

reflector is very short. That said, the distance between the radiator (antenna) and the

reflector is restricted.

Fig.5-7: Antenna needs to be a quarter wavelength distance away from the conductive

reflector.

A method for solving the phase shift problem is to keep the antenna a quarter

wavelength from the reflector as indicated in Fig.5-7. A wave radiating out of the

antenna travels at a quarter wavelength and has a phase shift of π/2 when it reaches

the ground conductor. When the wave is reflected back by the ground, a 180 degree

phase shift is formed. By the time the wave travels back to the radiation element the

E-filed is superposed constructively, resulting in higher gain and efficiency.


124

Another factor that affects the efficiency involved is the surface wave. Metal has a

property of supporting surface waves, which stays on the interface between the metal

and free space [87, 88]. For millimetre-wave communication those surface waves

equivalent to AC current exist along the metal surface plane. Once the surface is not

flat or smooth, the currents will radiate out and couple into the external plane wave. In

other words, if the conductive sheet is infinite, the current remains on the metal

surface. But in reality, the metal sheet has an edge and corner, so the current

propagating on it will be radiated out. As they are radiated to different directions than

the main lobe, in the far field radiation pattern, unwanted back lobes or other side

ripples are formed.

5.3.1 HIS Mechanism

In fact, the backed cavity cannot act as a reflector and improve the radiation if the

height is reduced significantly to less than a quarter of the wavelength. To reduce the

size of the antenna package, a new type of metallic electromagnetic structure was

developed early in 1999, which is called a high impedance electromagnetic structure

[44]. The structure is made of periodic metalised patterns. Unlike normal conductors,

it does not conduct ac current so it does not support surface wave propagation. The

cross section of a typical two-dimension HIS structure is shown in Fig.5-8. The

structure consists of a lattice of metal plates connecting with a metal sheet via a

vertical column conductor. Different from smooth conducting sheet that has low

impedance, this textured surface can have very high impedance.


125

Fig.5-8: Cross section of a high impedance electromagnetic surface [44]

As some AMC structures have a property of HIS, the HIS structure is also called

AMC. HIS structure is formed with finite metal plates and has different geometrics

such as square, triangular, hexagonal, and circular etc. The top view of a typical

geometric hexagonal metal plate HIS structure is shown in Fig.5-9. The size of each

period is much smaller than the wavelength, as are the gaps between each of the

lattice elements.

Fig.5-9: Top view of a hexagonal HIS structure [44]

Since the hexagonal unit structure is much smaller compared to the wavelength, the

operational principle of the structure can be explained by using an equivalent lumped

LC circuit model. Fig.5-10 shows the side views of two adjacent units. As it can be

seen, the gap between the two hexagonal plates forms a gap capacitor and the other

connecting conductive elements act as inductors. That is, the gaps between

neighbouring metal patches produce capacitance and metal paths connections induce

inductance. The structure forms a parallel LC resonant tank, as shown in Fig.5-11. At

the resonant frequency, the LC resonant tank will resonate and result in very high


126

surface impedance. In the other words, this surface is able to block the surface current

from flowing at the resonant frequency.

Fig.5-10: Single LC equivalent circuit

Fig.5-11: LC equivalent circuit for HIS structure

At around this resonant frequency, HIS can also act as a “magnetic conductor”. A

magnetic conductor is a virtual element. When the surface impedance is very high, the

tangential magnetic field is small whereas the electric field is large [89]. For example,

the Fig.5-12 shows the electric and tangential magnetic field plot on the magnetic

conductor surface. Over a specific frequency range, the magnetic conductor surface

has high impedance. Assuming an incident RF signal propagating on the surface with

E-field directing as Ei, the incident tangential magnetic field will be in Hi direction.

Due to the high impedance of the surface, the signal reflected back with the same

phase as Ei. Therefore the reflected E-field Er determined that the reflective tangential

magnetic field Hr is opposite to Hi. If perfect reflection occurs, Er = Ei, the electric

field is constructively added up whereas the magnetic field cancels each other out.


127

Fig.5-12: E and H fields on magnetic conductor surface

The magnetic conductor has properties of high impedance and low loss, which can be

used as a new type of ground plane for low profile antennas. The new structure can

reflect waves. However, different from a flat conducting metal plane, the image

current produced on the plane and the wave in free space are in phase. Therefore the

requirement of distance between the antenna and the reflector can be greatly relaxed.

Theoretically, the antenna can be placed adjacent to HIS without causing any

destructive influence to the radiation pattern as shown in Fig.5-13.

Fig.5-13: An antenna placed closely to a HIS sheet


128

In addition, the periodic texture structure supports are tightly bound, which leads the

Transverse Magnetic mode (TM mode) frequency to propagate much more slowly

along the surface. It also supports Transverse Electric mode (TE mode) that is bound

to the surface at some special frequency and radiates at some other frequency. Fig.5-

14 indicates the electric and magnetic field arrangement in TE mode propagation [90].

It can be seen from Fig.5-14, the magnetic field is extended out on the surface in

loops. The electric field is divided tangentially to both the radiation direction and the

surface. This kind of electromagnetic field arrangement does not support propagating

surface waves in a certain forbidden frequency band.

Fig.5-14: TE mode surface wave propagating on a HIS [44]

Thus the surface can be designed to have a special forbidden frequency band that

stops the surface wave generating on the high impedance conducting sheet. Without

the supporting of the surface wave in a forbidden bandwidth, a smooth radiation

pattern can be obtained and is independent to the effects of multipath interference

along the ground plane.

5.3.2 HIS design

According to different frequency and bandwidth requirements, HIS can be built to

meet the needs by properly designing the periodic structures. HIS structures work for

a large frequency range from several megahertz to tens of gigahertz. Referring to the


129

prototype of HIS in Fig.5-8, one can find that there are several parameters that affect

the surface performance. The most primary is the thickness of the HIS structure

because it plays an important role in affecting the bandwidth performance. Take the

hexagonal shape that shows in Fig.5-14 as an example, the high impedance

electromagnetic structure consists of lattice of element. Each element has the shape of

a hexagonal metal plate and is connected together to the bottom solid metal sheet via

a conducting metal column.

Fig.5-15: Cross section and top view of a hexagonal HIS structure [91]

As discussed in Section 5.3.1, the period of these metal plate elements and the height

should be much less than the wavelength so that an efficient equivalent LC resonant

circuit can be obtained. The period of the structure that shows in Fig.5-15 is marked

as D, the hexagonal metal pattern has a scale of D. The gap, g, is formed between

nearby metal plates. The whole surface has a height, h, away from the bottom sheet.

Thus high surface impedance Zs is formed. On the top view, the hexagonal side has a

length of x, which has a relationship with pattern width, w.

According to the trigonometric function,

. (21)

xxw 32

32


130

In the two layers structure, HIS can be described completely in terms of sheet

capacitance and sheet inductance because it was stated before that the HIS has an

equivalent circuit of parallel capacitors. In this section, the method of obtaining the

values of these parameters to make the most constructive improvement will be

discussed. Resonate frequency and radiation bandwidth can be expressed as follows:

Resonate frequency 𝜔0 =1

𝐿𝐶 (22)

The bandwidth equation relating to sheet capacitance and inductance was presented in

[92]. Radiation bandwidth for a narrow band antenna is percentage bandwidth and for

wide band antenna is called fractional bandwidth. The AMC structure has its own

resonant frequency as well and it can be considered a “natural frequency”, which is

𝜔0. Since in our design, 𝜔0= 60 GHz, radiation bandwidth of this structure can be

expressed as:

𝐵𝑎𝑛𝑑𝑤𝑖𝑑𝑡𝑕 =𝑓𝐻−𝑓𝐿

𝑓𝑐=

∆𝜔

𝜔0 (23)

In paper [92], it states that the electromagnetic radiation is due to the superposition of

antenna current and image current, while the radiated magnetic field is linked to the

electric field by the impedance of free space. Therefore the antenna current I can be

expressed as:

𝐼 = 𝐼𝑖𝑚𝑎𝑔𝑒 + 𝐻𝑟𝑎𝑑𝑖𝑎𝑡𝑖𝑜𝑛 =𝐸

𝑍𝑠+

𝐸

𝜂 (24)

where E is Electric field, 𝜂 is the resistance of free space, 𝑍𝑠 is the sheet impedance of

HIS and 𝑍𝑠 = 𝐿 𝐶


131

From equation 24, we can see when the sheet impedance 𝑍𝑠 = 𝜂, the radiation drops in

half. The difference of these frequencies is ∆𝜔 and it can be estimated by substituting

𝜔0 = 1 𝐿𝐶 and 𝑍0 = 𝐿 𝐶 .

Finally, radiation bandwidth can be obtained from the characteristic impedance of the

surface divided by the impedance of free space, which is shown as:

𝐵𝑎𝑛𝑑𝑤𝑖𝑑𝑡𝑕 =𝑍0

𝜂=

𝐿

𝐶

𝜇 0𝜀0

= 𝜀0𝐿

𝜇0𝐶 (25)

where L and C are sheet inductance and sheet capacitance for the observation HIS

structure, respectively. , is the resonant frequency at which the reflection phase is

zero and the surface becomes a magnetic conductor. Adjusting the parameter values

to vary the value of L and C, any RF can be obtained. In reality the actual antenna

bandwidth may be smaller than the one calculated in equation 23. The bandwidth

obtained in the above equation can be seen as equivalent to the width of surface wave

band gap. However the enhancement function of a reflection wave depends on

different surface patch shapes and sizes as well.

To calculate the sheet capacitance and inductance C and L, a part of two nearby metal

patches are firstly taken into consideration.

Fig.5-16: A section of a hexagonal HIS structure design

0


132

Fig.5-16 shows the surface sheet conductance geometry taking from HIS structure

prototype. In the structure, g is the gap width between two neighbouring hexagonal

plates and D is the distance between two centre units. Once the centre distance D >>

g, the electronic flux, , which generates around the gap can be obtained from

equation 30[44] as:

𝜍 = lim 2𝜀𝑉

𝜋cos−1

𝐷

𝑔 =

2𝜀𝑉

𝜋cosh−1

𝐷

𝑔

(26)

In this equation, is the permittivity on surface, V indicates potential voltage on

surface and electronic flux is a limitation value.

According to HIS structure shows in Fig.5-16, each hexagonal unit plate has a width

of w. Therefore, it assumes the surface has two values of different permittivity and the

individual capacitor width is w. Permittivity is in the radiation air space, while

another permittivity, , is produced by the HIS substrate. Since the sheet capacitance

has equal flux electrons on both sides, the equation of each conductor can be derived

from equation 26. Thus,

𝐶 =𝜍

𝑉=

𝑤 𝜀1+𝜀2

𝜋𝑐𝑜𝑠𝑕−1

𝐷

𝑔

)

(27)

The capacitance generated obeys the Guass‟ Law, which states that the amount of

charges on plates is proportional to the area of those plates. As the shape of each

patch is important in resonate frequency and bandwidth determination, a geometric

correction factor N is introduced to the individual capacitor calculation. The square

patch is taken as a reference, it has an area of , and , where

is the permittivity of the gap and g is the distance between two patches. According to

s

e

s

e1

e2

squareAg

AC

square

square

e


133

the reference [44], each shape has its own patch surface area, , which can be

expressed as . The geometric correction factor N is the ratio of the

designed capacitor and the regular squared capacitor. Thus,

i.e.

(28)

Therefore the sheet capacitance can be obtained from equation 28. The factor N has

different values for different shapes as shown in Table 5-1.

Table 5-1: Sheet capacitance correction factor in various geometries

Different geometry Factor N

Square 1

Triangle

Round circle

Hexagonal

Sheet inductance relates to the thickness of the structure,

(29)

where, is the thickness of the HIS structure, is the permeability for free space

and is the relative permeability of the circuit board material. Derived from three

equations above, an equation of resonate frequency is obtained by substituting

sheet capacitance and inductance substituted into equation 22,

𝜔 =1

𝐿𝐶=

1

𝐿𝑠𝑕𝑒𝑒𝑡 ∙ 𝑁𝐶𝑠𝑞𝑢𝑎𝑟𝑒

=1

𝑕 ∙ 𝜇0𝜇𝑟 ∙ 𝑁 ∙𝑤 𝜀1+𝜀2

𝜋cosh−1

𝐷

𝑔

(30)

designedA

g

AC

designed

designed

squareddesigned CNC

square

designed

A

AN

3

4

3

1

sheetL

rsheet hL 0

h0

r

cf


134

𝜔 is the centre frequency in radians, and 𝜔 = 2 ∙ 𝜋 ∙ 𝑓𝑐 ,

The centre frequency 𝑓𝑐 has an expression of

𝑓𝑐 =1

2𝜋 ∙ 𝐿𝑠𝑕𝑒𝑒𝑡 ∙ 𝑁𝐶𝑠𝑞𝑢𝑎𝑟𝑒

=1

2𝜋 ∙ 𝑕 ∙ 𝜇0𝜇𝑟 ∙ 𝑁 ∙𝑤(𝜀1+𝜀2)

𝜋cosh−1

𝐷

𝑔

(31)

Also the surface impedance can be calculated as:

𝑍𝑠 =𝑗𝜔𝐿

1 − 𝜔2 ∙ 𝐿 ∙ 𝑁𝐶𝑠𝑞𝑢𝑎𝑟𝑒 (32)

This is a new method of evaluating performance of HIS structure. It is based on

enhanced effective medium method, which was only used to evaluate the performance

of the square patch HIS structure [93]. According to the new method, other shapes of

patches can be designed more conveniently. Using this method some performance of

low profile antennas can be improved [94].

5.3.3 HIS Fabrication by PCB Technology

The basic AMC, a mushroom-shaped structure, forms high impedance on its surface

and can be fabricated by using PCB technology. The mushroom-shaped structure is a

two-dimensional structure. The two-layers HIS structure fabrication starts with

electrodepositing a copper layer on both sides of a plastic board to form a special

plastic board with metal layers on both sides. The process flow of making mushroom-

shaped structures on PCB technology can be summarized as shown in Fig.5-17. First,

a plastic board is covered with two copper layers and ready (Fig.5-17(a)). Holes are

then drilled vertically from the upper copper layer to the bottom copper layer, which


135

are then prepared for building conducting vias. A thin plated metal is applied by using

the electroless process and forms a coating around the „via holes‟ and a photoresist

pattern is then used to etch the top metal layer (Fig.5-17 (b)). Finally the metal layer

is covered on the top layer (Fig.5-17 (c)).

Fig.5-17: Process flow of making mushroom-shaped structures on PCB technology

Most HIS structures need conducting vias to the ground to form the inductance.

Therefore, most of them have following common properties

a) A sheet of dielectric materials,

b) A series of parallel conductive strips on one side,

c) A conducting material layer on another side.


136

Recently, HIS structures have been widely used to improve the performance of

antennas [95, 96]. For instance, HIS structure has been used on low observable

aircrafts because of its compact, relatively light and low cost properties.

HIS structures can be used as good electromagnetic isolators. In other words, HIS

structure can be applied to antenna arrays for anti-jamming applications. This is

because surface waves can be stopped by using HIS structures. Recently, multi-

layered HIS structures have been reported to achieve better performance, even with a

compact size [97, 98]. Each layer of patch units has its own sheet capacitance and

inductance. Certainly, a three-dimensional HIS structure requires a more complex

approach compared to a two-dimensional structure. In this work HIS structures were

designed and fabricated on Si CMOS technology.

5.3.4 Folded Dipole Antenna with HIS Cavity Backed

Simulation in HFSS

It has been discussed that theoretically the new metamaterial HIS structure has the

property of improving antenna performance without distance limitation. By applying

HIS structure as a reflecting ground in the cavity, a low profile antenna working at 60

GHz can have higher radiation gain. The antenna has the same structure as the one in

Section 5.2 while only the backed cavity has been changed. Fig.5-18 gives the

simulation structure in HFSS. Radiation antenna used is a 1.5 mm folded dipole

antenna fed by a 100 Ω CPS. The antenna is attached below a dielectric substrate with

a thickness of 250 μm. Antenna is suspending on the top centre of the cavity and a U-

shaped slot is built to have a better wave propagating from the feeder.


137

Fig.5-18: Folded dipole antenna with a HIS cavity backed in HFSS structure

The cavity is enclosed with two kinds of HIS structures, hexagonal surfaces with

conducting via and corrugated metal slab. Hexagonal shaped HIS structure has

already been discussed in Section 5.3.2. The corrugated metal slab is another type of

structure that can provide high impedance at the top surface as well [99].

The details of this corrugated structure are shown in Fig.5-19. A series of slots are cut

vertically, and each slot is narrow and has a length of one quarter-wavelength deep.

When a wave propagates to the surface, it travels down to the bottom along the slab.

Equally, the structure can be regarded as a parallel plate transmission line, which is

shorted at one end and open at the other end. Thus impedance in the top is quite high.

In this project, corrugated metal slab is built around four vertical walls in the cavity.

This corrugated structure stopped surface wave propagation through improving the

radiation directivity.


138

Fig.5-19: Corrugated metal slab structure [44]

Fig.5-20: Hexagonal HIS structure acting as reflector of the antenna

Lattices of hexagonal metal plates connect to the ground solid plane via conducting

cylinders. In order to make the HIS and the folded dipole antenna to resonate at 60

GHz, the HIS structure is designed as shown in Fig.5-20. The period width and height

are much smaller than the wavelength. As stated earlier in equation 25, the bandwidth,

, which is proportional to sheet inductance L but inverse proportional

to sheet capacitance C. Therefore method to increase the bandwidth is either to

designed

sheet

C

LBW


139

increase sheet inductance or reduce the sheet capacitance. According to equation 29,

inductance L depends on the dielectric permeability and the structure thickness. To

minimize the size of the cavity, the thickness cannot be too large. Therefore efforts

are made mainly to focus on modifying the sheet capacitance. If it is necessary, size

of the cavity can be fixed to a required size by using capacitive loading.

Fig.5-21: Top view of the HIS cavity

Fig.5-21 gives the top view of the whole cavity structure. The cavity has a size of 4.2

mm × 2.9 mm and a height of 400 μm. Metal plate has a period of 0.5 mm and a 5 μm

gap between their neighbouring elements. Conducting via has a height of 311 μm.

A full wave simulation is taken in radiation boundary conditions around the antenna.

Wave port is fixed to a regular size and an impedance of 100 Ω lumped wave is

generated in order to match the CPS and folded dipole antenna. Discrete calculating is


140

setup to sweep from 54 GHz to 72 GHz and the centre frequency is set to 58 GHz.

The smallest meshing size in the HFSS for this proposed structure is 3 μm. The

meshing size is automatically set by the solver according to electric field distribution

of the structure. The meshing size can also be re-sized as the computation progresses,

i.e., re-meshing. Consequently the meshing size varies in different part of the

structure. The reflection coefficient, S11 has a plot as shown in Fig.5-22.

Fig.5-22: HFSS simulation result for the reflection coefficient, S11

Simulation result of S11 gives a resonant frequency at 58 GHz. A bandwidth of 7 GHz

is obtained from 55 GHz to 62 GHz. Bandwidth can be a challenge, especially in

patch antenna type [100]. Since most of patch antennas has a property of narrow

bandwidth and the performance is closely related to the thickness of the substrate,

efforts were made to overcome the knottiness such as suspending the antenna on air,

using other bandwidth antenna that resonate in more than one frequency [101-103].

In this work, the antenna was designed to have a wide bandwidth because folded

dipole acting as loop antenna, can be seen as many dipoles parallel and radiate


141

together. The antenna is also suspending in the cavity with a distance to the reflector

ground, which determines the bandwidth performance. Hence contradiction turns up

about the cavity height, it has to earn a distance to increase the bandwidth but needs to

consider the cavity size at the same time. Therefore a small size cavity limits the

antenna bandwidth performance. However it still has a wide bandwidth of 7 GHz

within the commercial licensed frequency band.

A far field radiation patterns were used to study the antenna radiation properties such

as gain and directivity. Results are shown in Fig.5-23 and Fig.5-24 in both E and H

plane.

Fig.5-23: Radiation pattern of the antenna gain in dB


142

Fig.5-24: Radiation pattern of the antenna directivity in dB

Radiation patterns in Fig.5-23 and Fig.5-24 show the gain and directivity at antenna

resonant frequency respectively. It can be seen that the gain at its maximum radiation

efficiency achieved 6.93 dB, which has a 1.5 dB enhancement compared to pure

metal cavity, comparing with diagram Fig.5-5(a). The same to the directivity, it has a

1.5 dB improvement than that with a simple metal cavity, shown in Fig.5-5(b).

Furthermore the radiation pattern becomes smoother and directional because it

drastically removed the back lobe. Fig.5-25 gives the 3D pattern plot of the gain,

which is more directivity than the one in Fig.5-4.


143

Fig.5-25: Far field 3D plot of the antenna directivity

To sum up, the folded dipole antenna with high impedance electromagnetic surface

cavity backed obtains a smoother and more convergent far field radiation pattern. The

HIS cavity height is of 100 μm and a 7 GHz bandwidth is obtained. Comparison was

taken between two different cavities, structures in Fig.5-1 and Fig.5-18, respectively.

Simulation results showed that antenna operates in HIS cavity has a 1.5 dB gain

enhancement than the flat metal cavity. However, the height of conducting via (see

Fig.5-20), h = 311 μm, which makes the HIS cavity wall even thicker then the

resonant cavity height. Simulation results of this AMC cavity proved that HIS

structure can improve the antenna radiation gain with much decreased distance

between HIS reflector and antenna.

However, as discussed in section 5.3.2, height of conducting via, h determined the

sheet inductance, 𝐿𝑠𝑕𝑒𝑒𝑡 . To work at 60 GHz, 311 μm is the minimum height for h,

whereas the regular silicon bulk thickness in standard Si CMOS technology is 250 μm.


144

That is to say, it will require extra efforts to build this structure on standard Si CMOS

technology. In Section 5.4 and 5.5, two other AMC structures with HIS property will

be proposed, they are planar and easy to fabricate.

5.4 Low Profile Patch Antenna with Micro-patterned

Artificial Lattice Plane

As discussed in Section 5.3, compares to the conventional metal cavity, folded dipole

antenna structure obtained a 1.5 dB gain enhancement from „hexagonal mushroom‟

and „conjugated slab‟ shaped HIS combining structure. Two on-chip antennas with

different AMC structures were designed, fabricated and characterised using two

different semiconductor foundries and they are presented the following two sections.

To have easier feeding structure and implementation by Si CMOS technology, the

patch antenna was used instead of the folded dipole

There normally are two methods to increase the bandwidth: apply an antenna on a low

dielectric constant material and increase the thickness of the dielectric layer. However,

neither of these two methods can be applied in Si CMOS technology, therefore,

additional elements which can improve antenna radiation performance must be

considered. When patch antenna operates at its resonant frequency, electric field

concentrates along the edge of patch. To enlarge the electric field at the edges, two

parallel strips are added to form a slot. This slot makes E-field coupled and boosted

up, in order to improve the bandwidth. The low-profile patch antenna designed with


145

two adjusted strips is shown in Fig.5-26, for the two following AMC structure

application.

Fig.5-26: Improved patch antenna with two strips

5.4.1 Non-conducting Via AMC Structure Design

Our work used an AMC structure placing on the first metal layer of the wafer

fabrication process. As briefly introduced in Chapter 1, Si CMOS technology is a

frequently used to fabricate active and passive components. With this process, ICs

and devices are built upon oxidation material layer to avoid being affected by the

lossy silicon substrate. There are five main process steps applied in a standard Si

CMOS technology. They are oxidation (oxide grows on the silicon surface substrate

to form a layer of SiO2), diffusion (the impurity atoms on the surface moving into the

bulk material), ion implantation (ions are accelerated to a high velocity by the electric

field), deposition (mask is used to depose materials on silicon wafer) and etching

(remove exposed material and obtain one metal layer). Nevertheless, these five steps

are repeated on multi-layer Si CMOS technology.


146

For example, the fabrication process we used was eight metal layers Si CMOS

technology from Chartered foundry. All these eight metal layers had been built by

sequence of five-steps on the silicon material substrate. As shown in Fig.5-27, metal

layers are labelled as M1 to M8 according to their layer positions, respectively.

Thickness of each dielectric layer, SiO2, is only 1 μm.

Fig.5-27: Eight metal layer Si CMOS process layout

Fig.5-28: Top view of square AMC unit structure

Fig.5-28 shows the unit metal plate which forms the periodic AMC structures. High

impedance could be generated on the surface within a frequency band. In the previous


147

discussion of mushroom-shaped structure, a LC equivalent circuit was shown in

Fig.5-11, which derived impedance matches at LC resonant frequency. With a

conducting via connecting to the ground plane, plenty of works were successfully

done to improve the performance of passive components such as transmission lines,

antennas and filters [104-107].

Aimed to mass production and compatibility, Si CMOS process is the first choice as

the technology of component fabrication. However, structures with ground connecting

via cannot be implemented by Si CMOS technology according to the five-step

procedure. The AMC structure in this work only has one metal layer which consists of

periodic unit element showed in Fig.5-28. Capacitance and inductance of its

equivalent parallel LC circuit can be adjusted through parameters D, G. At 60 GHz,

the AMC metal period size, D, is smaller than the wavelength. The narrow gap

arranging at the edge of the square in the unit pattern requires G << D. As the antenna

designed in this work is a patch antenna that is fed by a 50 Ω microstrip line, eddy

current loss inside the shield becomes prominent at high frequency. Inserted between

the antenna plane and the silicon substrate, this structure of AMC plane blocks the

electric field from entering the substrate due to its high in-plane dielectric constant

[108]. The current density appears mainly around the edges of metal conductor,

cutting narrow slots close to the edges therefore to reduce the eddy current.

In Fig.5-29, a distinct microwave propagating path affected by the AMC structure is

shown. The AMC acting as a reflector reflected back the incident wave without phase

reversal. At the same time, the reflected wave takes part in the antenna radiation so as

the gain performance is enhanced.


148

Fig.5-29: Distinct microwave propagating path

To observe the phase shift and reflection coefficient, AMC structure was simulated in

HFSS Ver.12. A master-slave boundary was used to simulate the infinite periodic

element plane. The incident wave starts from half-wavelength away off the AMC unit

surface.

A phase plot against frequency of reflecting wave S11 was made and shown in Fig.5-

30. It can be clearly seen that the phase equals to zero at 60 GHz frequency. Plot in

Fig.5-31 sketches the phase change in simulation model. According to Fig.5-31, we

assume the incident wave has a phase Phase1. After half wavelength propagating, the

phase of wave on AMC structure surface became Phase2, which is 180 degree shift

out of Phase1. As the phase of S11 that shows in Fig.5-30 equals zero, which means

the reflected wave Phase3 is in phase with incident. Propagating back half wavelength

to the AMC surface, Phase4 has a -180 degree phase shift out of Phase3. When taking

AMC surface as reference, the incident phase, Phase2, and the reflect phase, Phase4,

has null phase shift. Therefore, an artificial magnetic conducting plane was formed on

the surface.


149

Fig.5-30: Simulated phase plot of S11

Fig.5-31: Simulated phase sketch of wave propagating

5.4.2 On-chip AMC Structure Antenna Design and Fabrication

Si CMOS substrate is lossy. To reduce the loss, AMC structure that shows in Fig.5-32

was investigated. The 1.212 mm × 1.111 mm sized AMC structure plane on the Si

substrate plays a role of a reflector. On the other hand, a patch antenna sized 0.4 mm


150

× 0.4 mm with two additional parasitic elements was designed for broadening the

bandwidth of the antenna. As Fig.5-32(b) shows, the ground plane is locating at the

bottom of silicon bulk, which indicates the low resistivity silicon is used to be the

substrate of AMC.

(a)

(b)

Fig.5-32: (a) Simulate structure of patch antenna with AMC plane. (b) Side view of

structure layout


151

Fig.5-32 shows the detail of the proposed antenna. It can be seen that the patch

antenna is located on the AMC plane. Two parallel strips adding on both sides can

wider the bandwidth. The G-S-G probe pads are added for measurement. The antenna

is fabricated through the 8 layers 0.13 μm Si CMOS process. As shown in Fig.5-27,

patch antenna is built on the top layer and the AMC structure stays on the first metal

layer. The thickness of 8 dielectric material layers is only 7 μm.

Results are shown in three parameters: the return loss S11, the operating frequency

gain and the directivity. Fig.5-33 shows simulated return loss of the patch antenna. It

can be noticed that the substrate loss is very high, especially at higher frequency. For

instance the loss above 80 GHz is more than 10 dB. For such high loss substrate, one

might not use VSWR < 2 to define the antenna bandwidth. The VSWR can be less

than -10 dB, but the RF power is dissipated by the substrate rather than radiate out to

the space. In Fig.5-33 the return loss from 72.5 GHz to 95 GHz clearly indicated that

the RF power has been radiated out to the space in addition to those dissipating to the

substrate. If we choose the bandwidth to be 10 dB lower than the power dissipated at

higher end of the frequency band, i.e., -20 dB in this study, the simulated bandwidth

shows in Fig.5-33 is approximately 5 GHz. Fig.5-34 and Fig.5-35 show the radiation

patterns of gain and directivity, respectively. In order to present the radiation patterns

in details, the patterns were cut along x-y plane, with angle ϕ in every 10 degrees. As

a result, plenty of traces are listing in the plot. Each colour shows one radiation

pattern at one cutting angle. For instance, the inner dark blue circle in Fig.5-34 shows

radiation pattern when ϕ = 90°, and while the outer red circle shows radiation pattern

when ϕ = 0°.


152

Fig.5-33: Return loss S11 of the patch antenna

Fig.5-34: Simulated antenna gain at 85 GHz


153

Fig.5-35: Simulated antenna directivity at 85 GHz

(a)


154

(b)

Fig.5-36: (a) On-chip antenna measured by probe station system (b) Fabricated on-chip

antenna with AMC structure by 8 layer 0.13 μm Si CMOS process

The full structure of this AMC patch antenna was fabricated by 8 layers 0.13 μm Si

CMOS RF technology. On-chip antenna was fabricated by Chartered foundry. After

fabrication, antenna was measured by using GSG-100 probe station system shows in

Fig.5-36. Fig.5-36(a) shows the apparatus used to measure the on chip antenna.

Antenna was placed in the centre of metal chamber. A photo of on-chip AMC antenna

is shown in Fig.5-36(b) and the centre distance between two ground pads is 100 μm.

2-port network Cascade Microtech Model 11000 probe station provides GSG-100

probe to measure the antenna, HP 8510XF millimetre-wave controller and HP 8510c

network analyser were used to observe antenna reflection coefficient. As the foundry

requires that certain level of metal density for each metal must be satisfied, additional

strips were added around the sides of the antenna after design rule check (DRC). Due


155

to these strips, extra electric field was added to the radiation and the resonant

frequency of antenna has been shifted. The extra metal strips also have effects on the

antenna bandwidth.

Fig.5-37: Measured reflection coefficient of on-chip antenna

Fig.5-37 shows the measured reflection coefficient, S11 of the fabricated on-chip

antenna. As it can be seen, the antenna works at a clearly resonant frequency of 50

GHz. The loss, especially at higher frequency end is very high, therefore, the 20 dB

bandwidth is approximately 10 GHz (20% fractional bandwidth). However, the

efficiency of this antenna is low due to the substrate loss.


156

5.5 Low Profile On-chip Antenna with Dog-bone and

UC-PBG Structure Plane

To reduce the substrate loss and increase the antenna efficiency, a novel AMC

structure is proposed in this section. The shape of the proposed unit element is similar

to a bone and is therefore called a dog-bone structure as shown in Fig.5-38. This dog-

bone shaped model was firstly introduced in 2010 [109]. The equivalent circuit model

of the structure is shown in Fig.5-39.

Fig.5-38: Unit element of dog-bone structure

Fig.5-39: Equivalent circuit of dog-bone structure plane

The resonant frequency can be given as 𝑓 =1

2𝜋 𝐿𝐶, where C and L are the capacitance

and inductance of the structure, respectively.

5.5.1 Dog-bone shaped AMC structure applied to on-chip

antenna

In our work, the dog-bone metal structure was built on a dioxide silicon layer of 0.18

μm Si CMOS technology from the TSMC foundry. The layout of this Si CMOS


157

technology is shown in Fig.5-40. The first metal layer, M1, is etched on Interlayer

Dielectric (ILD) material, and two different dioxide silicon materials with different

dielectric constants are raised over M1. After the top metal layer, M6 finished etching,

conformal coating is covered for protection. Each group of dielectric material layers

has a thickness of 0.85 μm. Because atoms in dioxide silicon material can get excited

at high temperature, it may be unstable and even crack. Therefore the height of the

dioxide silicon material must be limited. The property of the Si CMOS process is

predestined to build any conducting vias with several hundred micrometers height. To

build in the Si CMOS process, only structures without vias meet the requirement.

Fig.5-40: 0.18 μm Si CMOS process from TSMC foundry model cross section [110]

Fig.5-41 shows the structure of this on-chip antenna. It can be seen that the ground

plane is located on M1, and the dog-bone structure lies on M2. Each of them has a

metal thickness of 0.53 μm and is separated by a 0.85 μm SiO2 layer. Upon the 4.99


158

μm SiO2 substrate, patch antenna and UC-PBG structures are arranged. To operate at

a high frequency of 65 GHz, the dimensions of the dog-bone shape were designed to

have the values as shown in Fig.5-42.

Fig.5-41: Dog-bone structure layout on the Si CMOS technology process

Fig.5-42: Dimensions of the dog-bone shape working at 65 GHz


159

Fig.5-43: Simulated S11 phase plot of dog-bone unit using HFSS Ver.12

Full wave simulations of this dog-bone structure were carried out using HFSS Ver.12.

A master-slave boundary was used to simulate the infinite periodic element plane.

Distance from the wave input port to the surface of the unit plane was set to half-

wavelength. Therefore the incident wave would travel along a distance to make a

phase shift of 180 degree. The phase against frequency of reflecting RF signal S11 plot

is shown in Fig.5-43. It can be clearly seen that the phase shift is zero at 60 GHz.

5.5.2 UC-PBG Structure Applying to On-chip Antenna

To further increase the gain by suppressing the surface wave, a PBG structure is

incorporated with the proposed on-chip antenna. PBG structure was first proposed by

E.Yablonovitch in 1993 [37]. Recently it has been well developed in millimetre-wave

field. The first UC-PBG structure was proposed for microwave circuit application in


160

1999 [111]. Also in that paper [111], a periodic grid consisting of PBG cells created a

forbidden frequency range so that the surface wave was stopped and reduced.

Fig.5-44: UC-PBG unit cell structure [112]

Fig.5-44 shows the unit cell of the UC-PBG structure. Each unit cell is patched above

ground with a dielectric material substrate on the secluded substrate. The structure

mainly contains a large square, and four narrow strips connect each unit together. The

connecting arm between two units acts as the inductor on the surface and the spaces

between units provides capacitance. As shown in Fig.5-44, the parameters of the UC-

PBG structure unit cell are marked with different variables. D is the period of the unit

cell, d is the distance between two squares, g is the gap between two neighbouring

units, m is the width of the small square in the corner, a is the width of the narrow

strip and t is the space between the small square and the narrow strip. To have the

UC-PBG structure working at the desired frequency, the parameters indicated in

Fig.5-44 are derived.

The relatively simple formulas used are based on the equations presented in [113,

114]. If the periodic grid consisting of a large number of UC-PBG cells can be

considered as a HIS with a grid impedance of equation 33 [114]

𝑍𝑔𝑟𝑖𝑑 =𝜌2

4

𝑗𝜔𝐶

1 − 𝜔2𝐿𝐶


161

(33)

where ρ is the effective wave impedance of the substrate medium and ρ = ρ0 εe and

εe refers to the effective permittivity, which can be given as

εe = (εr + 1) 2 .

Where the relative constant is εr . If the unit period 𝐷 ≪ 𝑤𝑎𝑣𝑒𝑙𝑒𝑛𝑔𝑡𝑕 so that the

thickness of UC-PBG material is negligent, Grid capacitance, C, and inductance, L,

can be determined as [113, 114]:

C =2dεeε0

πln

2D

πa

L =μ0D

2πln

2D

πt

(34)

It can be seen from equation 33, if it is at resonant frequency 𝑓 =1

2𝜋 𝐿𝐶 , the factor

1 − 𝜔2𝐿𝐶 = 0, which means the grid impedance, 𝑍𝑔𝑟𝑖𝑑 equals to infinity. The surface

impedance of the UC-PBG structure can be calculated as a parallel connection of the

grid impedance and the wave impedance inside the dielectric substrate [115].

𝑍𝑠 =𝑍𝑔𝑟𝑖𝑑 𝑍𝑑

𝑍𝑔𝑟𝑖𝑑 +𝑍𝑑 ,

(35)

Where 𝑍𝑑 is the dielectric material impedance and 𝑍𝑑 = 𝜌0 𝜀𝑟 .

Therefore at resonant frequency, this periodic grid has the property of a HIS. Based

on the above equations, the UC-PBG unit cell structure was designed to have a = 24.7

μm, m = 108 μm , t = 37.5 μm , g = 35 μm , d = 250 μm and D = 315 μm in order to

operate at 65 GHz.


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5.5.3 AMC and UC-PBG On-chip Antenna Design and

Fabrication

In this work, the dog-bone shaped AMC structure plane and UC-PBG structures were

employed to combat the substrate loss and prevent surface wave propagation. The

employment of these two structures can improve antenna performance. The AMC

structure makes low profile antenna possible. Meanwhile, a UC-PBG structure was

built on the same layer surrounding the patch to stop the surface wave and further

increase the radiation gain at the operating frequency. The antenna is processed using

0.18 μm Si CMOS technology from the TSMC foundry. A wide bandwidth of 3.4

GHz from 64 to 67.8 GHz was obtained. The results showed a 1.6 dB gain

enhancement and an increase of more than 1 GHz bandwidth compared to the antenna

without AMC structures. The antenna radiation efficiency also exceeds 10%.

Fig.5-45: On-chip patch antenna with dog-bone AMC and UC-PBG model

The dog-bone structure and UC-PBG are printed on metal layers M2 and M6,

respectively, as shown in Fig.5-45. Each metal layer was separated by a 0.85 μm

dielectric layer. However, no metal is placed in layers M3 to M5 in this design, which

means the dielectric layer thickness in this work became 4 × 0.85 μm + 3 × 0.53 μm =

4.99 μm. Traditional patch antenna located a 4.99 μm distance above ground would


163

have poor bandwidth and efficiency, due to the imaging current effect. With the help

of the AMC structure, performance is improved significantly. The structure was

firstly simulated in HFSS Ver.12. The main parameters, reflective coefficient,

directivity and gain, were observed.

The S11 simulation result is shown in Fig.5-46 and indicates the bandwidth of 3.2

GHz from 64.6 GHz to 67.8 GHz. The patch antenna designed in this work was fed

by a 50 Ω microstrip line with a patch size of 1.13 mm × 1.01 mm. Two strips are

placed 5 μm away from each sides. Fig.5-47 presents the E-field existing on patch

antenna and UC-PBG patterns. Most of the E-field is located at the edge of the patch

antenna instead of leaking to the side as a surface wave.

Fig.5-46: S11 simulated result


164

Fig.5-47: E-field existing on patch antenna and UC-PBG patterns

Fig.5-48 plots the antenna gain as a radiation pattern including E-H plane. The

simulated gain is of -10.07 dB at an operating frequency of 65.5 GHz. However, the

pattern has clear directivity and small back lobe. This is due to the AMC plane

located under the patch antenna reflecting the signal to radiate without flowing

sideways. The simulation of a traditional patch antenna has been done for comparison

purposes. The gain of on-chip patch antenna at operating frequency is shown in Fig.5-

49. According to this result, the gain of the antenna with AMC and UC-PBG has

improved by 1.6 dB.


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Fig.5-48: Radiation pattern of on-chip AMC and UC-PBG antenna gain at 65.5 GHz

Fig.5-49: Radiation pattern of on-chip traditional patch antenna gain at 65.5 GHz


166

Fig.5-50: Fabricated on-chip AMC and UC-PBG structured antenna layout

In addition to the simulation, this antenna was also fabricated by the TSMC foundry

using 0.18 μm Si CMOS technology. Fig.5-50 is a photo of the fabricated antenna. A

Cascade Microtech Model 11000 probe station, an HP 8510XF millimetre-wave

controller and an HP 8510c network analyser were used as the measurement

apparatus. Both measured and simulated S11 are shown in Fig.5-51. In this plot, the

purple dashed curve represents S11 of the traditional on-chip patch antenna. The red

and blue curves represent the simulated and measured S11 results of the AMC and

UC-PBG structured on-chip patch antennas, respectively. They clearly shows that the

bandwidth increased by more than 1 GHz.


167

Fig.5-51: Reflection coefficient results

5.6 Conclusion

In this chapter, we focused on designing low profile on-chip antennas for millimetre-

wave application. As the silicon material in Si CMOS technology has low resistivity,

Si substrate is so lossy that antenna efficiency would be very low. To avoid the lossy

Si substrate, a dielectrical layer on standard Si CMOS technology is used instead.

However, the thin thickness of the SiO2 layer reduces the radiation efficiency. As has

been discussed in Chapter 4, on-chip antenna could be successfully designed if the

effect from the conductive ground plane could be eliminated. The AMC structure was

used to solve this issue.

In the beginning, two folded dipole antennas with different cavities were proposed,

simulated and discussed. Two cavities were made of different materials, one with

normal conductive metal and the other with the AMC structure. Full wave simulation

results in HFSS showed that the AMC structure provided an enhancement on a gain


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of 1.5 dB, compared to the radiation gain obtained from a normal metal cavity. The

AMC structure cavity consists of a mushroom structure and corrugated slabs. Both of

these structures are in three dimensions, which increase the total antenna size and

become hard to fabricate.

Two other AMC structures with HIS properties were designed, simulated and

fabricated from two Si CMOS foundries. Both antennas are based on low profile

patch antennas with two parasitic strips to increase their radiation bandwidth.

According to the Si CMOS technology, the on-chip antenna proposed in Section 5.4

was built with a micro-patterned artificial lattice plane and this AMC structure was

placed on the first metal layer of the process. After being fabricated by Si CMOS

process from the Chartered foundry, the proposed antenna was measured.

Measurement results showed the on-chip antenna resonant at 50 GHz with a 20 dB

bandwidth of 10 GHz. In reality, the antenna integrity has been changed by the

foundry during the fabrication to meet the metal density requirement. The Chartered

foundry requires that metal located on each layer must meet the standard density.

However the patch antenna on the top metal layer is too small to achieve the

requirement. Therefore, additional metal strips around the antenna and AMC

structures were added in order to pass the DRC required from the foundry. As a result,

the effective antenna size has been enlarged and the resonant frequency decreased.

The substrate loss due to the silicon material in this antenna is high, especially at

higher frequencies. For this high loss structure, we define the bandwidth as -10 dB

lower than the power dissipated at the higher end of the frequency band, which is –20

dB. That is to say, the bandwidth that shows in the measurement result is

approximately 10 GHz, which is 20% of the fractional bandwidth.


169

Although the bandwidth of the antenna structure in Section 5.4 is large, the substrate

loss dissipated most of the RF wave power. In Section 5.5, the AMC structure and

metal layer arrangement was improved. A novel on-chip antenna with a dog-bone and

UC-PBG structure plane was designed, fabricated and proposed. A ground plane was

put on the first metal layer of the Si CMOS process to isolate the antenna from the

lossy silicon substrate. A regularly arranged dog-bone shaped AMC structure was

placed on the second metal layer (M2), which formed an RF wave reflection without

phase reversal. The patch antenna with parasitic strips was located in the centre of the

top metal layer and it was surrounded by a UC-PBG structure to prevent surface wave

loss. Fabrication was done by the TSMC foundry and the measurement results show

that this on-chip antenna operates at 65.5 GHz with a bandwidth of 3.2 GHz.

Compared to the simulation bandwidth of the patch antenna without an AMC

structure, the AMC and UG-PBG structured antenna has gained a bandwidth rise of

more than 1 GHz and 1.6 dB higher gain.

According to the specification, the on-chip antennas proposed resonate at millimetre-

wave frequency and have a bandwidth of 3.2 GHz, which is enough for the HD data

transfer requirement of 3 Gbps. However the gain of this proposed antenna is -10.07

dB, which is 1.6 dB higher than traditional on-chip patch antenna without AMC.

Chapter 6 On-chip Antenna Fabrication and Measurement

170

Chapter 6

On-chip Antenna Fabrication and Measurement

6.1 Measurement Setup

On chip antenna is an attractive but challenging topic. The most often concerned is

the silicon bulk loss. Low resistivity of silicon substrate kills energy heavily at high

frequency. As a one-port network, reflection coefficient is obtained to observe

antenna radiation. The main measurement apparatus used in our work are shown in

Fig.6-1.

Fig.6-1: Measurement apparatus of on-chip antenna


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As on-chip antenna was designed to compact into millimetre-wave communication

devices, it needs to be stable performance even if surrounding temperature changes.

In this chapter, the relationship between these two proposed on-chip antennas and

temperature will be investigated and discussed. As the size of antenna is very small,

we use GSG-100 probe station to measure. However, it is not easy to put probe station

into a chamber room to measure the gain. Followed the temperature observation, we

chose another method to detect antenna gain. The method and results will be

presented.

6.2 Reflection Coefficient Changes with Temperature

Based on two structures presented in Section 5.4 and Section 5.5, relationship

between reflection coefficient and temperature was observed. Each antenna was

placed in the probe station, which is well located by a chamber. The temperature of

the chamber can be change within a various range from -75 Celsius to 200 Celsius by

temperature controller, as shown in Fig.6-2.

Fig.6-2: Temperature controller


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Normal steady flow of air was infused into the probe station chamber from the gas

dryer is shown in Fig.6-1. Humidity needs to be decreased to a suitable standard.

Otherwise, the surface of measuring device will raise mist or even ice at low

temperature.

6.2.1 Temperature Measurement of One Layer AMC Structured

Antenna

Firstly the on-chip antenna with one layer AMC structure was measured at five

different temperatures. This antenna was fabricated by Chartered foundry. Fig.6-3

gave the testing antenna layout.

Fig.6-3: Single layer AMC structure antenna measuring in probe station

The Reflection coefficient plot against frequency is measured and shown from Fig.6-4

to Fig.6-8. Plot was taken at five different temperatures: -57 Celsius, 0 Celsius, 20

Celsius (room temperature), 75 Celsius and 145 Celsius. The bandwidth variations


173

with temperature were investigated. Due to the Si substrate loss, 20 dB (instead of 10

dB) bandwidth was examined.

Fig.6-4: Reflection coefficient at room temperature (20 Celsius)

Fig.6-5: Reflection coefficient at 0 Celsius


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Fig.6-6: Reflection coefficient result at -57 Celsius

It can be seen from these three results in Fig.6-4, Fig.6-5 and Fig.6-6, bandwidth

keeps increasing along with temperature decreasing. However the average in-band

reflection coefficient decreased with temperature. Further two more measurements

were made to prove this trend as shown in Fig.6-7 and Fig.6-8.

Fig.6-7: Reflection coefficient result at 75 Celsius


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Fig.6-8: Reflection coefficient result at 145 Celsius

The relationship between temperature and bandwidth is summarized in Table 6-1.

Table 6-1: Average in-band S11 and bandwidth relationship with temperature

Temperature (Celsius) S11 Bandwidth

-57 -19 dB N/A

0 -21 dB 14 GHz

20 -22.5 dB 11 GHz

75 -23 dB 9 GHz

145 -24 dB 8 GHz

As the temperature increased, magnitude of reflection coefficient decreased, which

indicates less RF power is reflected, resulting better efficiency. The reason is that the

resistivity ρ ∝ temperature (T). High temperature increased resistivity of silicon

bulk in order to reduce the substrate loss. Therefore, when temperature falls to -57

Celsius, the resistivity of silicon bulk decreased, return loss increases to greater than -

20 dB. Therefore the 20 dB bandwidth at the low temperature, -57 Celsius, cannot be

detected.


176

6.2.2 Temperature Measurement of Dog-bone AMC and UC-

PBG Structured Antenna

This on-chip antenna, which shows in Fig.6-9 was measured by the same equipment

and procedure as Section 6.2.1. Fig.6-10 shows S11 plot at the temperature of 15

Celsius. Ten groups of data were collected at different temperatures. Reflection

coefficient values of antenna operating at those temperatures are measured and

recorded in Table 6-2.

Fig.6-9: Dog-bone AMC and UC-PBG antenna in temperature measurement

Fig.6-10: S11 plot at 15 Celsius temperature


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Table 6-2: Average in-band reflection coefficient S11 changing with temperature

Temperature in Celsius S11

-25 -10.881 dB

-20 -10.992 dB

0 -11.438 dB

15 -12.340 dB

25 -12.470 dB

50 -12.993 dB

70 -13.513 dB

105 -13.721 dB

120 -15.133 dB

150 -16.099 dB

Fig.6-11: Reflection coefficient against temperature plot

Plot in Fig.6-11 further proved, as already seen in Section 6.2.2 that antenna

efficiency increased along with temperature. In a fact, increasing of the resistivity

brings higher efficiency of the antenna.


178

6.3 Gain Measurement of On-chip Antenna

On-chip antennas at millimetre-wave band are very small. The probe station setup

makes it very difficult to measure the antenna radiation patterns. This is because the

surroundings of the probe station and even the probe itself (which is hundreds time

bigger than the antenna itself), as it can be seen in Fig.6-12, severely disturb the

radiation path for any possible accurate measurement. Furthermore it is impossible to

make neither transmitter antenna nor receiver antenna to rotate, which is required for

radiation pattern measurement. To overcome these measurement shortages, only

antenna gains were measured.

Fig.6-12: On-chip antenna and probe station


179

Two same on-chip antennas were used to measure the antenna radiation transmission

performance. Two on-chip antennas were placed apart and acting as transmitter and

receiver. A 2-port network is formed with these two antennas. Measurements were

taken from S-parameter values of this 2-port network by VNA. In order to have more

accurate measurement results, plenty of positions were chosen. They were indicated

from P1 to PN according to their position to the antenna from transmitter. Due to the

limitation of the probe station, the furthest measuring distance is 8 mm, any points

that is more than 8 mm away from no signal can be detected by the VNA.

Twenty positions in far-field range were measured. According to Far-Filed range

definition, area out of sphere with radius R =2L2

λ is called far-field range. As Fig.6-13

shows, two antennas operating at 65.5 GHz should be placed at least 1.11 mm apart to

measure the far-field radiation gain.

Fig.6-13: Far field determination


180

Fig.6-14: Antenna measurement position plot

Fig.6-14 plots out each measurement position, where the transmitter antenna is fixed

at the centre and the receiver antenna changes its positions to measure the

transmission performance. The rectangular in the centre represents the on-chip

antenna acting as transmitter and P1, P2 …PN representing the positions of receiver

antenna. For instance, the purple rectangular represents transmit antenna and at

position P20, the pink rectangular indicates receive antenna facing to the transmit one.

The arrow shows the direction of measurement probe and dash circle indicates the far-

field range of the transmitter antenna. Totally twenty groups of 2-port S-parameters

data are recorded and displayed in Table 6-3.


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Table 6-3: Data collection from twenty different positions

Position Distance S11 (dB) S22 (dB) S12 (dB) S21 (dB)

P1 2 mm -14.11 -13.06 -44.25 -44.64

P2 2 mm -13.21 -12.37 -56.27 -54.68

P3 2 mm -12.44 -12.52 -53.52 -53.50

P4 2.83 mm -12.90 -13.13 -56.63 -56.23

P5 2.83 mm -18.74 -25.37 -44.31 -44.38

P6 2.83 mm -13.91 -12.37 -56.36 -55.47

P7 2.83 mm -15.80 -12.96 -54.30 -54.99

P8 4 mm -12.72 -12.24 -46.13 -45.48

P9 4 mm -12.80 -13.08 -58.05 -58.50

P10 4 mm -14.11 -13.05 -53.25 -53.14

P11 5.66 mm -11.90 -11.80 -58.86 -57.82

P12 5.66 mm -12.70 -12.90 -57.90 -55.00

P13 5.66 mm -13.09 -12.45 -58.64 -56.24

P14 5.66 mm -12.30 -12.04 -57.82 -55.61

P15 6 mm -19.48 -14.37 -52.06 -50.08

P16 6 mm -11.76 -12.61 -63.11 -63.05

P17 6 mm -15.96 -13.37 -53.51 -52.01

P18 8.49 mm -11.40 -14.53 -57.74 -57.08

P19 8.49 mm -11.90 -14.53 -57.23 -55.80

P20 8 mm -13.06 -12.86 -54.84 -55.23


182

Fig.6-15: S21 measurement result at different position

Fig.6-16: S11 measurement result at different position


183

Measurement was taken under room temperature with a source power of -15 dBm. In

order to clearly show the relationship between s-parameter value and position, data in

Table 6-3 were illustrated in Fig.6-15 and Fig.6-16 to show the transmission

coefficient S21 and reflection coefficient S11, respectively. The dots plots display a

clear decibel value according to different positions around the receiver antenna. Data

of S21 in Fig.6-15 shows that the transmitter antenna radiates power towards infinite

sphere. For example, along the positive direction of x-axis, where P1, P8, P15 and P20

are measured, transmission power is in a trend of reducing. This trend also can be

applied to other directions that start from transmit antenna. However, P14 and P5 have

lower and higher transmit power than they should be, respectively. This may be

caused by the probe station metal cavity multiple reflections.

To further prove the measurement result accuracy, full wave simulation of two

antennas with the transmitter antenna in the centre and receiver antenna at position P18,

which are 8.49 mm distance apart from the transmit antenna. The simulation layout in

HFSS Ver.12 is shown in Fig.6-17.

Fig.6-17: Simulation model of gain observing in HFSS Ver.12


184

The receiver antenna, Ant.2 has a distance 8.49 mm apart from the transmitter and its

distance to x and y axis is 6 mm. Fig.6-18 is the plot of two S-parameters, S11 and S21.

M1 is a marker of S11 and M2 is the marker of S21. Simulation result shows two

antennas resonant at 65 GHz and the value of S21 is -57.35 dB.

Fig.6-18: S11 and S21 plot in simulation

Fig.6-19: Simulated radiation pattern at the plane with P18 of Ant 2 at 65 GHz


185

As it is impossible, to turn the GSG probe station in an angle to measure the radiation

pattern, two antennas were placed on the same plane for the transmission coefficient

measurement, which is shown in Fig.6-12. Therefore at position P18, power

transmitted into Ant 2 is not all power that radiates out. In Fig.6-19, radiation pattern

of gain was plot when ϕ = 135°, where P18 locates. Marker M1 indicates the exactly

gain obtained at P18. The gain at P18 equals to -15.92 dB. It should be pointed out that

this is not the maximum gain. It is the gain at ϕ = 135°. The maximum radiation

power gain is where marker M2 indicates, -9.12 dB. Simulation result in Fig.6-19

shows that the transmitter antenna has a maximum gain of -9.12 dB, as marker M2

shows. Compared to the radiation gain that shows in Fig.5-48, radiation gain of

antenna increased from -10.07 dB to -9.12 dB. The 0.9 dB gain enhancement is due to

the effect of receiver antenna.

As the antenna received power, 𝑃𝑟 , can be expressed as [116]:

𝑃𝑟 = 𝑃𝑡

𝐺𝑡𝐺𝑟 ∙ 𝜆2

(4𝜋𝑅)2

(36)

where 𝑃𝑡 is the transmit power from Ant 1, 𝐺𝑡 and 𝐺𝑟 are gain of transmit and receive

antennas, in this case, 𝐺𝑟 and 𝐺𝑡 are the same and equals to the antenna received gain.

𝜆 is the wavelength in free space and 𝑅 is the distance of two antennas. Also the

transmit coefficient S21 is the ratio of power received to power transmitted,

𝑆21 = 10log(𝑃𝑟

𝑃𝑡)

(37)

Substituted equation 37 into equation 36, the received antenna gain can be estimated.

In Table 6-3, measured S21 at P18 is -57.08 dB and the received gain can be calculated

to be -14.89 dB. Comparing to the simulation result that is shown in Fig.6-19, the


186

measured gain (calculated based on measured S21) is approximately 1 dB lower than

that simulated, which is quite acceptable given that the simulation setup is not quite

the same as the measurement one. For instance, there is no probes in the simulation

and the boundary in the simulation hardly mimics the probe station cavity and its

surroundings.

6.4 Conclusion

In this chapter, two low profile AMC Si CMOS on-chip antennas proposed in Section

5.4 and Section 5.5 were fabricated and characterised. Effects of temperature

variation on the AMC antennas were also examined experimentally.

Since the AMC on-chip antenna needs to be measured using probe station used in our

work is hundreds times larger than the antenna, it is impossible to move the probe

station into an anechoic chamber. It is also impossible to rotate the probe to measure

the radiation pattern using our existing facility. An alternative way was adopted to

measure the gain performance of the on-chip antennas by measuring transmission

coefficient, S21, between two exactly the same antennas, one as transmitter and the

other as receiver in different positions around the transmit antenna. The measurement

results seem agreeing with simulated ones reasonably well.

Chapter 7 Conclusion and Future work

187

Chapter 7

Conclusion and Future Work

7.1 Summary of the Work

Since unlicensed band around 60 GHz was announced, millimetre-wave technology

had rapidly and substantially developed. In the beginning of this thesis, considerations

of millimetre-wave devices were explained.

U-shaped slot patch antenna can be designed to operate at millimetre-wave HD

wireless communication system. In this work, U-shaped slot antenna was designed on

40 μm SiO2 substrate. Simulation results of U-shaped slot patch antenna showed that

the proposed design can achieve a gain of 2.44 dB and a bandwidth of 1 GHz at two

millimetre-wave resonant frequencies. Simulation result also proved that if the

thickness of SiO2 substrate increased to 100 μm, bandwidth of the antenna will be

wider. If two U-shaped slots were built in antenna structure, a dual band performance

can be obtained.


188

With the discovery of metamaterial technique due to its unique electromagnetic

properties, possible applications were studied. It has been proved that a fully

integrated metamaterial CPW transmission line can be applied on high resistivity Si

MMIC system using a multilayer fabrication technology. To provide more compact

on-chip antenna, metamaterial technique was implemented.

The work in Chapter 5 is aimed to design a compact antenna that operating on 60

GHz based on SiO2 substrate. There are generally three efforts that could be made to

satisfy the HD system antenna requirements, such as:

1) Design a folded dipole antenna with wide bandwidth

2) Arrange the antenna position and suspend it in a resonant cavity to enhance the

radiation performance

3) Applying a HIS structure in the cavity to further improve the performance

The folded dipole antenna is chosen instead of U-shaped slot antenna, because it has

less ground distance affect than the U-shaped slot antenna. The antenna is built

directly on the top of the substrate without a ground plane backed. Folded dipole

antenna radiates energy without the presence of ground plane, which means substrate

thickness can be less than quarter wavelength. Simulation result in Section 4.3 shows

this folded dipole antenna achieved a bandwidth of 3 GHz at the centre frequency of

60 GHz.

In order to improve the gain, a cavity is added to the back of the antenna in order to

act as a reflector and increase the radiation efficiency. This cavity is built with flat

metal planes and at the lowest cavity height of quarter-wavelength. This metal cavity


189

provides a 3 GHz bandwidth enhancement and 2 dB gain improvement comparing to

the result in Section 4.3. Therefore, the antenna has an improved bandwidth of 6 GHz

from 59 to 65 GHz with radiation gain of 5.58 dB and directivity of 5.77 dB. It has

been proved that the cavity does increase antenna radiation performances.

Back lobe in radiation pattern plot indicated that part of RF power is propagating

along the substrate as a surface wave. The technique that used in the thesis by

applying a metamaterial structure placing around antenna can stop the surface wave.

Therefore, another cavity with HIS structure was presented. It can be put adjacency to

the radiator so that antenna can be arranged on a substrate of less than quarter

wavelength thickness. Compared to the metal cavity, improvement of HIS cavity is

obtained in bandwidth, gain and directivity based on the HFSS simulation result. It is

shown that this approach also eliminates the radiation back lobes as expected and the

radiation pattern becomes smooth. However, the antenna is relatively large to

fabricate on Si CMOS substrate. Hence other metamaterials techniques on thinner and

easier structures are required to be investigated further.

In Sections 5.4 and 5.5, two planar AMC on-chip antennas were proposed and

fabricated. The planar structure is realized by printed the designin Si CMOS process.

Multilayer Si CMOS 0.13 μm and 0.18 μm technologies were used in fabrication. The

measurements were taken using VNA and GSG-100 prober. Measurement results

show that the antenna with single AMC layer can operates at 50 GHz. As for their

high loss, 20 dB bandwidth standards were used instead of 10 dB. Therefore antenna

can achieves a 20 dB bandwidth of 10 GHz. Operating frequency of the fabricated

antenna move to 50 GHz. The main reason is due to the additional strip lines that


190

placed besides the patch antenna. To satisfy the metal check requirement in Chartered

foundry fabrication, strips were needed. However, those metal strips act as series of

reactors and they increase the antenna size. The other on-chip antenna was processed

with two metamaterial structure layers. Dog-bone shaped is placed under the antenna

to avoid silicon loss and UC-PBG is placed next to patch antenna to prevent surface

wave loss. Measurement result shows a small improvement on both bandwidth and

gain.

The relationship between reflection coefficient and temperature of two fabricated on-

chip antennas were further observed in Chapeter 6. It was found that the magnitude of

the reflection coefficient S11 decreased when the temperature increases. Under the

source power scale of -15 dBm, radiation efficiency at different positions were

measured and analysed. The antenna design proposed in Section 5.5 is chosen for its

gain measurement. The measurement results proved that the antenna maximum

radiation gain is around -10 dB, which means only 10 % of the transmit power

successfully transferred out into others.

To sum up the conclusion of this work, much improved bandwidth of 3.2 GHz can

obtained, which is enough to support 3 Gbps HD data wireless transmission speed.

The Si CMOS AMC antenna presented in this work is easy to fabricate, simple to

design and appears to be attractive for its low cost mass production.


191

7.2 Suggestions for Future Research

While the research outcome is promising in the AMC on-chip antenna design, further

research is required mainly to improve the on-chip antenna performance from both

antenna for both antenna and metamaterial structure parts. Besides the structures

presented in this thesis, some other combination of metamaterial structures can be

investigated further.

Fig.7-1: Jerusalem Crosses AMC structure under patch antenna

Fig.7-2: Snowflake shape AMC structure with folded dipole antenna


192

Fig.7-3: Snowflake shape AMC structure with patch antenna

For example, Fig.7-1 shows a patch antenna with Jerusalem crossed AMC structure,

Fig.7-2 shows a folded dipole antenna with snowflakes shape and Fig.7-3 shows a

patch antenna with snowflakes shape AMC structure. They all can be fabricated with

Si CMOS technology. However, they are more complicated structure than the ones

that proposed in this thesis.

The work in this thesis has finally proposed an on-chip antenna that partly meets the

specification but it is easy to fabricate by using Si CMOS technology. In order to

improve the radiation gain and directional performance, the single on-chip antenna

can be arranged into an array. The antenna array concept is leading in the market

since single element cannot be sufficient in practical use. For example, a single


193

antenna element radiates a fix power, which is not enough as required in the long

distance communication system. In order to increase the transmission performance,

designers propose in increasing the electrical size of antenna by regularly arrange

antenna elements to form an array system.

Array antenna can be seen as many individual directional antennas that add together

and radiating in their own directions. Types of array with different arrangement of

different individual elements are mainly based on four controls below:

1. The geometrical configuration of the overall array. For example linear, circular

rectangular.

2. The relative displacement between the elements

3. The excitation amplitude of the individual elements

4. The relative pattern of the individual elements

Once the single element performance meets the requested value, adaptive array can be

built by applying the beam forming technology. Antenna array can be built in many

forms such as one-dimensional linear array and two-dimensional planar array. This

technique combines different amplitude and phase in far field, hence can increase the

total gain and direction performance. To introduce the adaptive array concept into the

system mainly because it provides an optimal gain at a certain direction, while not

wastes power in the undesired directions. For example, in the wireless system, the

distance between a TV set box and the displayer is much further than that between TV

set and the computer. An adaptive array will estimate a two different power providing

on them so that not to waste useless power.


194

Beam forming technique mainly used with adaptive antenna array, which forms a

rotatable radiation beam in far field. Nowadays, many methods of beam scanning

technologies have been developed. With this technique, transmitter can lock the

destination after searching the receive device in advance. Once the destination is

found and connected the system allows a high directivity antenna array to starts

transmitting the signal. In this way, power can be saved.

There are several mechanisms to make the beam steerable. The main objective is to

change the phase against time. This aim can be achieved by directly scanning the

phase and scanning frequency or using the digital computer system to form a digital

rotatable beam as well as mechanically changing the direction of each element in the

array. In the future work of this project, scanning phase by using a phase shifter can

be considered to be applied with the beam forming technology.

Appendix

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Appendix

List of Publications

[1] Curtis, P.D.; Y. Peng; W. Tong; Z. Hu; "Multilayered metamaterials for Silicon

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[5] Y. Peng; Z. Hu, H. Yang, T. Gao, “A Wideband On-Chip Antenna With a Single

AMC Structure Plane”, 2011 IEEE International Symposium on Antennas and

Propagation and USNC/URSI National Radio Science Meeting, Spokane, Washington,

USA. July 4-8, 2011.

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