Conference-201303-F Musavi-Control Strategies for a LLC Multi-resonant DC-DC Converter in Battery Charging Applications

8/9/2019 Conference-201303-F Musavi-Control Strategies for a LLC Multi-resonant DC-DC Converter in Battery Charging Applications

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Control Strategies for a LLC Multi-Resonant DC-DC

Converter in Battery Charging Applications

Fariborz Musavi, Marian Craciun, Deepak Gautam,

Murray EdingtonDepartment of Research, Engineering

Delta-Q Technologies Corp., Burnaby, BC, Canada

[email protected], [email protected],[email protected], [email protected]

1 Wilson Eberle and 2 William G. Dunford1 School of Eng. | 2 Dept. of Electrical and Computer Eng.University of British Columbia | 1 Okanagan | 2 Vancouver

1 Kelowna, BC, Canada | 2 Vancouver, BC, Canada1 [email protected] | 2 [email protected]

AbstractIn this paper, a control strategy is presented for a

high performance LLC multi-resonant dc-dc converter in a two

stage smart charger for neighborhood electric vehicle

applications. It addresses several aspects and limitations of LLC

resonant dc-dc converters in battery charging applications, such

as very wide output voltage range while keeping the efficiency

maximized, implementation of the current mode control at the

secondary side and optimization of burst mode operation for

current regulation at very low output voltage. The proposedcontrol scheme minimize both low and high frequency current

ripple on the battery while maintaining stability of the dc-dc

converter, thus maximizing battery life without penalizing the

volume of the charger. Experimental results are presented for a

prototype unit converting 390 V from the input dc link to an

output voltage range of 48 V to 72 V dc at 650 W. The prototype

achieves a peak efficiency of 96 %.

I. INTRODUCTION

Neighborhood Electric Vehicles (NEVs) are propelled byan electric motor that is supplied with power from arechargeable battery [1, 2]. Presently, the performancecharacteristics required for many electric vehicle (EV)

applications far exceed the storage capabilities of conventionalbattery systems. However, battery technology is improvingand as this transition occurs, the charging of these batteriesbecomes very complicated due to the high voltages andcurrents involved in the system and the sophisticated chargingalgorithms [3]. Quick charging of high capacity battery packscauses increased disturbances in the ac utility power system,thereby increasing the need for efficient, low-distortion smartchargers. The accepted charger power architecture includes anac-dc converter with power factor correction (PFC) [4],followed by an isolated dc-dc converter, as shown in Fig. 1[5]. This architecture virtually eliminates both the low- andhigh-frequency current ripple on the battery, thus maximizingbattery life without penalizing the volume of the charger

circuit. The front end ac-dc PFC converter is a conventionalCCM boost topology [6, 7]. The following dc-dc section is ahalf-bridge multi-resonant LLC converter. The half-bridgeresonant LLC converter is widely used in telecom industriesfor its high efficiency at the resonant frequency and its ability

to regulate the output voltage during the hold-up time, wherethe output voltage is constant and the input voltage might dropsignificantly [8-11].

Figure 1. Typical battery charging power architecture.

However, its application for battery charging impacts thedesign criteria significantly, as to address the following:

A. Un-controlled area operation:

The output voltage requirement for a battery charger isdrastically different and challenging compared to telecomapplications. Fig. 2 illustrates a simplified battery chargingprofile for a 48 V system. As it indicates, the battery voltage,at the dc-dc converter output, can vary from as low as 36 Vand as high as 72 V. In addition, in the case of severelydischarged batteries it is required to control current down to

almost 0A when the voltage is below about 50% of maximumoutput voltage in the Un-controlled Area of Fig. 3, wherethe LLC output V-I plane is illustrated [12].

Figure 2. Simplified adaptive 4 step lead acid battery charging profile.

This work has been sponsored and supported by Delta-Q Technologies

Corporation.

978-1-4673-4355-8/13/$31.00 2013 IEEE 1804


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Figure 3. LLC output V-I plane with un-controllered area.

B.Beat frequency quadratic pole phenomena:

This is a special characteristic for resonant converters [13-15]. Frequency-to-Output transfer function of the LLC

resonant converter contains a quadratic pole, as illustrated in[13] and [16]. Both, damping factor Q and of the quadraticpole vary with the converter operating condition. This termcould introduce either a pair of complex poles or two realpoles affecting the power stage dynamics. When it results incomplex poles, the frequency is approximately given by thedifference between the switching and the resonant tankfrequencies, therefore it is called beat frequency doublepole. It is of particular importance for a battery charger as theoperating conditions and load models vary widely, requiringcurrent or voltage regulation in any point of the highlightedarea of Fig. 3 with constant voltage or/and constant resistanceload. In order to compensate for the additional phase lag, it isrequired to reduce the bandwidth of the control loop. As a

consequence, a voltage mode converter will have a slowtransient and poor rejection of the line frequency ripple thatneeds to be addressed.

0.3 0.75 1.2 1.65 2.090.5

0.6

0.7

0.8

0.9

1

1.1

1.2

1.3

1.4

1.5

1.6

1.7

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1.9

2fnmins fnmax

Figure 4. Typical DC transfer ratio of an LLC DC to DC converter obtainedusing FHA.

C. Secondary Side Current Mode Control:

In a battery charger, it is desired to control the charge rate,which is in fact the charger current. In addition, rejecting thelow frequency ripple on the dc link bus is required. Thismeans reducing the transconductance of the dc-dc converter.And in order to satisfy these conditions, a current modecontrol with high current loop gain at twice the line frequencyis desired. Current mode control can be implemented either in

the primary side or secondary side [13, 15, 17]. Primary sidecontrol requires isolation of feedback control signal, which isusually accomplished by using an optocoupler. The maindisadvantages of using an optocoupler would be significantvariation of the control loop gain due to optocouplers poorCTR initial tolerance, reduced bandwidth and degradationwith the temperature and ageing. In order to compensate forthese variations, a larger gain margin, and in some cases phasemargin, in control loop design is mandatory. Secondary sidecontrol removes optocouplers limitations enabling morerepeatable performance. One implementation is as shown inFig.5, where the gating signals are transferred to the primaryside. However, the disadvantage is now sensing the input busvoltage across the isolation barrier for brown out and under-

voltage protection of the DC-to-DC stage.

Figure 5. Simplified seconday side current mode control.

II. BURST MODE OPERATION (NL,SC)

Burst mode operation [18] can be used for depletedbatteries that require operation of the LLC converter in the un-controlled area of the V-I plane, shown in Fig. 3. This methodis used solely for reviving neglected batteries. In this region,the charger voltage is below 1.5 V/Cell (36 V) and theswitching frequency has reached its maximum value (500kHz). At this point of operation, the converter is switched toON/OFF mode while operating at fixed frequency fsw-max. Inorder to reduce components stresses during repetitive ON-

OFF operation several precautions have to be considered:

1) Selecting half bridge topology with split resonantcapacitor as shown in Fig. 5 will ensure the capacitors arealready charged at the DC steady state level prior to startswitching, reducing the startup inrush currents.

2) Shorter duration of the first gate drive pulse at startupensures soft switching condition of the MOSFET switches atpower ON and allows fast transition to steady state values of

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the resonant inductor current. As shown in Fig. 6, the resonantcurrent reaches steady state in few switching cycles avoidinghigh peak current transitions.

3) Energy stored in resonant tank creates minor batterycurrent tail after gate pulses are stopped as shown in Fig.7,limiting choice of maximum burst frequency and/or maximumburst duty cycle.

Figure 6. Start up soft switching consideration.Ch1 = MOSFET Gate drive 5 V/div. Ch2 = Battery Current 2 A/div. Ch3=Half Bridge Node voltage 50 V/div. Ch4 = ILr2A/div.

Figure 7. Shut down battery current consideartion.Ch1 = Half Bridge Node voltage 100 V/div. Ch2 = Battery Current 2 A/div.

Ch4 = ILr2A/div.

Battery manufacturers recommend less than C/20 (5 ARMS for a 100 Ah battery) low frequency ripple current (linefrequency or double-line frequency) to minimize heatgeneration while charging. Tests performed on VRLAbatteries for UPSs with three times the recommended ripplecurrent have demonstrated that the heating effect is minimal(


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thereby reverting the converter to normal (low ripple)operation.

B. Variable frequency fixed on-time (VFFOT)

In addition to battery ripple current tolerance, batterymanufacturers provide the minimum duty cycle for pulsedcurrent charging. Accordingly, with the FFVOT controlstrategy enables operation at low O/P current ripple and highON/OFF frequency with a minimum ON duration. If inFFVOT, once the charger reaches the minimum ON durationlimit, the frequency must begin to reduce and the converterenters VFFOT. The purpose of switching the control strategyfrom FFVOT to VFFOT is to maintain the charge current atvery low value. Fig. 10 illustrates the VFFOT operationconcept and the transition from VFFOT to FFVOT modes.

Figure 10. Top: VFFOT operation concept, Bottom: Transition from VFFOT

to FFVOT.

C. Control principle and implementation

An example method of battery charging control isprovided in Fig. 11.

At the beginning, the battery charger detects if the batteryvoltage is less than 1.5 V/Cell. If the battery voltage is equal

to or more than 1.5 V/Cell, the DC-to-DC can achieve chargerate regulation in the continuous operating area, thereforecontinuous operation mode will be enabled.

If the battery voltage is less than 1.5V/cell the VFFOTmode of operation is enabled. In this mode of operation thebattery is charged with a current pulse of duration tMIN andamplitude less than ISC, charge regulation being achieved bymeans of changing the repetition rate of the current pulses fON.Then the battery current is measured and the average value iscompared to the reference current, IREF from the chargingalgorithm. If the averaged battery current is less thanIREF, thepulse repetition frequencyfON is increased by a f incrementand the resulting new repetition frequency is compared to thecurrent pulse duration. The result of these comparisons

decides if the process is repeated or if the operation mode ischanged to FFVOT mode. If a new valueIREFis received fromthe charging algorithm, it is compared to the old value. If thenewIREFvalue is less than the old one the process is restarted.If the new IREF value is more or equal to the old value themeasured battery voltage is compared to 1.5V/Cell. If thebattery voltage is less than 1.5V/Cell the process is repeated, ifthe battery voltage is more or equal to 1.5V/Cell the operationmode is changed to continuous operation mode.

Figure 11. Flowchart of battery charging control.

While operating in VFFOT mode, the battery current pulseduration tON is compared to the pulse repetition period. (Theperiod is the inverse function of the pulse repetition frequency,1/fON). If the pulse duration is more than half of the repetitionperiod, the operation mode is changed to FFVOT.

In FFVOT mode of operation the battery is charged with acurrent pulse of an amplitude less thanISCat a fixed repetition

frequencyfPWMwith a variable duration tON, charge regulationbeing achieved by means of changing the current pulseduration tON. Then the battery current is measured and theaverage value is compared with the reference currentIREFfromthe charging algorithm. If the averaged battery current is lessthanIREF, the pulse duration tONis increased by a tincrementand the resulting new pulse duration is compared to 98% ofthe pulse repetition period. The result of these comparisonsdecides if the process is repeated or if the operation mode ischanged to continuous operation mode. If a new value IREFis

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received from the charging algorithm it is compared to the oldvalue. If the new IREF value is less than the old one theprocess is restarted. If the new IREFvalue is more or equal tothe old value the measured battery voltage is compared to1.5V/Cell. If the battery voltage is less than 1.5V/Cell theprocess is repeated, if the battery voltage is more or equal to1.5V/Cell the operation mode is changed to continuousoperation mode.

Fig. 12 illustrates the area of implementation of FFVOTand VFFOT modes in un-controlled lead acid battery V-Iplane.

Figure 12. Implementation of FFVOT and VFFOT modes in battery V-I

plane.

III. CONTROL STABILITY CONSIDERATION

In order to address beat frequency and verify the stabilityof the system, both current and voltage plant stability must beverified in the extreme operating conditions, using the

previous plant modeling. Fig. 13 illustrates the block diagramrepresentation of the system with an inner current loop andouter voltage loop. Fig. 14 illustrates the uncompensated plantphase and gain frequency responses, Pi(s)at full load, 48 Vand 72 V output. The beat frequencies could be observed at 10kHz and 20 kHz, for 72 V and 48 V respectively. The closedloop cross over frequency must be placed at least one octavebelow the beat frequencies, due to excessive phase shift.

Figure 13. Block diagram representation of the system with an inner currentloop and outer voltage loop.

An overall compensated current loop phase and gain at Vo= 72 V and FL, Pi(s)Ci(s) for resistive and battery load isshown in Fig. 15. It can be observed that with battery the gainis increased to 25 dB, which will provide line frequencycurrent ripple rejection. The closed looped compensatedcurrent plant is the uncompensated plant (power stage) for thevoltage loop, as shown in Fig. 16. The compensated voltageloop transfer function is given in Fig. 17 at full load, 48 V and72 V output.

However, the battery will reduce the gain of the voltageloop, as shown in Fig. 17. Also it is observed that the cut-offfrequency drops by two decades (from 1.5 kHz to 16 Hz).

Figure 14. Plant transfer function phase and magnitude at: Vo = 48 V and Vo

= 72 V.

Figure 15. Compensated current plant transfer function phase and magnitudeat: Vo = 72 V and FL.

Figure 16. Closed current loop (Voltage plant transfer function) phase and

magnitude at: Vo = 72 V and FL.

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Figure 17. Compensated voltage plant transfer function phase and magnitude

at: Vo = 72 V and FL.

IV. SIMULATION AND EXPERIMENTAL RESULTS

A prototype of the half-bridge LLC multi-resonant

converter was built to provide a proof-of-concept and verify

the analytical work presented in this paper. Fig. 18 shows a

picture of the LLC dc-dc multi-resonant converter prototype.Table I provides the design criteria for the prototype LLC

converter. In Table II, the key components used the prototypeconverter is given.

Figure 18. Prototype of LLC dc-dc converter.

TABLE I. DESIGN SPECIFICATIONS

StageDesign Specification for LLC Resonant Converter

Parameter Designator Value

InitialDesignParametrs Input Votage Range Vin_min ~ Vin_max 370 - 410 [V]

Input Votage Nominal Vin_nom 390 [V]

Output Votage Range Vo_min ~ Vo_max 36 - 72 [V]

Output Votage Nominal Vo_nom 48 [V]

Switching Frequency fs_min ~ fs_max 150 - 450 [kHz]

Resonant Frequency fo 200 [kHz]

ResonantTank

Componnets

Transformer Ratio Nn 4:1:1

Resonant Inductor Lr 35 [H]

Resonant Capacitor Cr 28.2 [nF]

Magnetizing Inductor Lm 105 [H]

The measured efficiencies of the converter as a function ofload are given in Fig. 19, at output voltages of 48, 60 and 72V. This clearly shows that the efficiency is kept almostconstant and independent of output voltage, at full load. Thesemeasurements were taken with the output relay; commonmode EMI inductor and output fuse in the circuit.

TABLE II. COMPONENTS USED IN THE PROTOTYPE CONVERTER

Stage Components Used in LLC Resonant Converter PrototypeComponent Manufacturer Part #

PowerTrainComponents

MOSFET STMicroelectronics STB23NM60ND

Diode Rectifiers STMicroelectronics STTH2002C

Resonant Film Capacitor EPCOS MKP 28.2 [nF]

Resonant Inductor EPCOS RM12 - N97

Magnetizing Inductor EPCOS RM14 - N97

Output Film Capacitors EPCOS MKT 33.3 [F]

Controller IC On Semiconductor NCP1395

Simulation and experimental waveforms of the resonanttank current, resonant capacitor voltage and voltage acrossbottom MOSFET- Q2 are provided in Fig. 20 and Fig. 21 atVin= 390 V, and Po= 650 W. The waveforms in Fig. 20 aregiven at close the unity gain resonant frequency, fsw = 211kHz, and output voltage, Vo= 48 V. The waveforms in Fig. 21are given at fsw= 152 kHz, and an output voltage of Vo= 72V.

Figure 19. Measured effeciency vs output power for: Vo = 48 V, Vo = 60 V

and Vo = 72 V.

Fig. 22 provides example waveforms of transition fromfixed frequency variable on-time (FFVOT) control toContinuous operation mode.

Fig. 23 illustrates example waveforms of the fixedfrequency variable on-time (FFVOT) control strategy.

Fig. 24 illustrates example waveforms of the variablefrequency fixed on-time (VFFOT) control strategy.

84

86

88

90

92

94

96

98

0

100

200

300

400

500

600

700

Efficiency(%

)

Output Power (W)

Vo=48 V

Vo=60 V

Vo=72 V

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a)

b)

Figure 20. ILr, VCr, and VQ2for Vo= 48V, Po= 650 W;

a) Simulation, b) ExperimentalCh1 = VQ2100 V/div. Ch2 = VCr100 V/div. Ch4 = ILr2A/div.

a)

b)

Figure 21. ILr, VCr, and VQ2for Vo= 72V, Po= 650 W;a) Simulation, b) Experimental

Ch1 = VQ2100 V/div. Ch2 = VCr100 V/div. Ch4 = ILr2A/div.

Figure 22. Transition from Fixed frequency variable on-time (FFVOT)control to Continuous operation mode Don = 98% and Io = 7 A, Vo = 20 V.

Figure 23. Fixed frequency variable on-time (FFVOT) control strategy:

fPWM= 1 kHz, Don = 60%, Io= 5 A, Vo= 5V.

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Battery Current

(IBATT)

Resonant

Current (ILr)

Figure 24. Variable frequency fixed on-time (VFFOT) control strategy:fBurst= 31 kHz, VBATT= 3 V, On Duration = 1 Resonant Cycle, IBATT= 1.3 A.

Note that the overshoot seen in the current waveforms isdue to the small impedance of the battery simulator. Real lifedepleted batteries will have higher internal resistance so alossy damper was not deemed necessary for this mode of

operation and the waveforms looked more like those of Fig. 8.

V. CONCLUSIONS

A control strategy is presented for a high performanceLLC multi-resonant dc-dc converter in a two stage smartcharger for neighborhood electric vehicle applications. Itaddresses several aspects and limitations of LLC resonant dc-dc converters in battery charging applications, such as verywide output voltage range while keeping the efficiencymaximized, the beat frequency double pole at frequenciesclose to resonant frequency, and implementation of the currentmode control at the secondary side. The proposed control

scheme minimize both low and high frequency current rippleon the battery while maintaining stability of the dc-dcconverter, thus maximizing battery life without penalizing thevolume of the charger. Experimental results are presented fora prototype unit converting 390 V from the input dc link to anoutput voltage range of 48 V to 72 V dc at 650 W. Theprototype achieves a peak efficiency of 96 %.

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Conference-201303-F Musavi-Control Strategies for a LLC Multi-resonant DC-DC Converter in Battery Charging Applications