C DOCUME~1 ADMINI~1 LOCALS~1 Temp Plugtmp-10 Plugin-mgk Thesis Final

8/8/2019 C DOCUME~1 ADMINI~1 LOCALS~1 Temp Plugtmp-10 Plugin-mgk Thesis Final

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Blekinge Institute o TechnologyLicentiate Dissertation Series No. 2009:02

School o Engineering

on coherent and non-coherent

receiver structures for impulseradio uwb systems

Muhammad Gu ran Khan


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On Coherent and Non-coherentReceiver Structures for Impulse

Radio UWB Systems

Muhammad Gufran Khan


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On Coherent and Non-coherentReceiver Structures for Impulse

Radio UWB Systems

Muhammad Gufran Khan

Blekinge Institute of Technology Licentiate Dissertation SeriesNo 2009:02

Department of Signal ProcessingSchool of Engineering

Blekinge Institute of TechnologySWEDEN


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2009 Muhammad Gufran KhanDepartment of Signal ProcessingSchool of EngineeringPublisher: Blekinge Institute of TechnologyPrinted by Printfabriken, Karlskrona, Sweden 2009ISBN 978-91-7295-159-4Blekinge Institute of Technology Licentiate Dissertation SeriesISSN 1650-2140

urn:nbn:se:bth-00425


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v

AbstractThe demand and growth of short-range wireless communications has beentremendous in the past few years for a diverse range of applications. There issignicant interest in ultra wideband (UWB) as a physical layer technologyfor both high-data-rate (HDR), and low power, low-data-rate (LDR) short-range communications. The unique qualities of UWB such as large bandwidth,very low power spectral density (PSD) and ne time-resolution provide higherchannel capacity, reduced fading effects and position location capability in aUWB system. The extremely wide bandwidth of UWB systems also posesmany system design challenges to achieve low cost and low-complexity UWBdevices. The UWB systems have difficulties in using digital signal processing(DSP) technology, require high sampling frequencies and also face frequencydependent signal distortion. The propagation characteristics of UWB signalsalso require consideration as it differs signicantly from traditional narrow-band systems.

In this thesis, the issue of design and performance evaluation of coher-ent and non-coherent receivers for detection of impulse radio (IR) UWB sig-nals is addressed. The coherent RAKE and non-coherent transmitted refer-ence (TR) receiver structures are investigated for low power and low-data-

rate wireless sensor network applications. First, the performance of coher-ent RAKE receivers for a single-user system operating in non-line-of-sight(NLOS) scenarios in an industrial environment is evaluated. The results andperformance comparison is presented for partial RAKE (PRake) and selec-tive RAKE (SRake) using maximal ratio combining (MRC) and equal gaincombining (EGC). Secondly, the recursive structures of conventional TR andaveraged TR schemes are presented to decrease the performance loss asso-ciated with the low-complexity TR scheme for IR-UWB systems. Further,a doublet-shift TR (DSTR) signaling and detection scheme is presented forIR-UWB systems. The simulation results validate that the doublet-shift TRsignaling scheme improves the performance over conventional TR signalingscheme. Finally, dual-doublet TR (DDTR) signaling and detection schemesare proposed. The proposed dual-doublet TR schemes recover 25 to 50%energy/rate loss of conventional TR scheme. The performance of receiverstructures is evaluated in terms of uncoded bit-error-rate (BER) over thechannels measured in a medium-sized industrial environment and standardIEEE 802.15.4a multipath channels.


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Contents

Abstract . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . v

Contents . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . v i i

Preface . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . i x

Acknowledgements . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . x i

Publications list . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . xiii

Introduction . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1

Part IPerformance Evaluation of RAKE Receiver for IR-UWB Systems usingMultipath Channels for Industrial Environments . . . . . . . . . . . . . . . . . . 25

Part IIRecursive Transmitted Reference Receivers for Impulse Radio UWBS y s t e m s . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 5 7

Part IIIA Doublet-Shift Transmitted Reference Scheme for Ultra-Wideband Com-munication Systems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 81

Part IVTransmitted Reference Schemes based on Dual-Doublets for ImpulseRadio UWB Systems . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 103

Conclusions and Future Work . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 133


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PrefaceThis licentiate thesis summarizes my work within the eld of signal processingand wireless communications. The focus of the work is on sub-optimal coher-ent and non-coherent receiver design and performance evaluation for impulseradio (IR) ultra-wideband (UWB) systems. The work has been performedat the Department of Signal Processing, School of Engineering, at BlekingeInstitute of Technology. The thesis consists of four stand-alone parts:

Part IPerformance Evaluation of RAKE Receiver for IR-UWB Systems usingMultipath Channels for Industrial Environments.

Part IIRecursive Transmitted Reference Receivers for Impulse Radio UWBSystems.

Part IIIA Doublet-Shift Transmitted Reference Scheme for Ultra-Wideband Com-munication Systems

Part IVTransmitted Reference Schemes based on Dual-Doublets for ImpulseRadio UWB Systems.


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AcknowledgementsFirstly, I would like to express my sincere gratitude to my supervisor andexaminer Professor Ingvar Claesson for giving me the opportunity to be aPh. D. student in his department. I am thankful to him for providing meguidance, encouragement and constructive feedback in my research. He hasnot only helped me in my studies but also guided me in my everyday life inSweden.

I am indebted to my co-supervisor and mentor Dr. J orgen Nordberg forsupporting and guiding me in my research. His knowledge, experience andencouragement has helped me a lot during these years. This thesis has notbeen possible without his guidance and many useful discussions with him.

I would like to thank all my colleagues at the department for making anice and friendly atmosphere. I want to specially thank some of my teachersand colleagues, Professor Abbas Mohammed, Dr. Nedelko Grbic, Dr. BennyLovsrom and Dr. Benny Sallberg for their support. I would also like to thankDr. Fredrik Tufvesson and Professor Andreas F. Molisch at Lund Universityfor introducing me to the area of UWB wireless communications during myM.Sc. thesis.

I am grateful to my parents for providing me good education, teaching

me beautiful values of life and for their love and care. I am also thankful tomy brothers and sisters for their love and affection. Finally, I would like toexpress my love to my dear wife Dr. Kanwal Shehzadi for making my life sobeautiful and for being the person she is.

Muhammad Gufran Khan Ronneby, January 2009


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Publication listPart I: Parts of this work are published as:

Muhammad Gufran Khan, J orgen Nordberg, and Ingvar Claesson, Perfor-mance Evaluation of RAKE Receiver for Low Data Rate UWB Systems usingMultipath Channels for Industrial Environments, Research report, BlekingeInstitute of Technology, 2008, Issue: 4, ISSN: 1103-1581.

Muhammad Gufran Khan, J orgen Nordberg, Abbas Mohammed, and IngvarClaesson, Performance evaluation of RAKE receiver for UWB systems usingmeasured channels in industrial environments, AusWireless06 , March 2006,Sydney, Australia.

Part II: Parts of this work are published as:

Muhammad Gufran Khan, J orgen Nordberg, and Ingvar Claesson, RecursiveTransmitted Reference Receivers for Impulse Radio UWB Systems, Researchreport, Blekinge Institute of Technology, 2008 Issue: 5, ISSN: 1103-1581.

Muhammad Gufran Khan, J orgen Nordberg, and Ingvar Claesson, Detectionof Impulse Radio Ultra-Wideband Signals using Recursive Transmitted Ref-erence Receivers, Proceedings of ICUWB07, IEEE International Conferenceon Ultra-Wideband , September 24-26, 2007, Singapore.

Part III: Parts of this work are published as:

Muhammad Gufran Khan, J orgen Nordberg, and Ingvar Claesson, A Doublet-

Shift Transmitted Reference Scheme for Ultra-Wideband Communication Sys-tems, Proceedings of ICUWB07, IEEE International Conference on Ultra-Wideband , September 24-26, 2007, Singapore.

Part IV: Parts of this work are published as:

Muhammad Gufran Khan, J orgen Nordberg, and Ingvar Claesson, Trans-mitted reference scheme based on dual-doublets for UWB impulse radio,


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submitted to IET Electronics Letters , March 03, 2009.

Muhammad Gufran Khan, J orgen Nordberg, and Ingvar Claesson, An Energy-efficient Signaling and Detection Scheme for Transmitted Reference UWB Sys-tems, INCC08 International Networking and Communications Conference ,May 1-4, 2008, LUMS, Lahore, Pakistan.

Muhammad Gufran Khan, J orgen Nordberg, and Ingvar Claesson, Signalingand Detection of UWB Signals based on a Dual-Doublet Transmitted Ref-

erence Scheme, RVK08 The twentieth Nordic Conference on Radio Scienceand Communications , June 9-11, 2008, V axj o, Sweden.

Other publications in conjunction with the thesis:

Muhammad Gufran Khan, Asim A. Ashraf, Johan Karedal, Fredrik Tufves-son, and Andreas F. Molisch, Measurements and Analysis of UWB Channelsin Industrial Environments, in Proc. Wireless Personal Multimedia Com-munications (WPMC) , Sept. 2005, Aalborg, Denmark.


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Introduction

The demand and deployment of wireless mobile communication networkshas increased signicantly in the past few years. The mobile communica-tion networks are moving from third generation (3G) to Beyond 3G (B3G)networks to provide higher data-rates, spectral efficiency, scalability and im-proved quality of service [1]. The physical layer (PHY) of evolving futuregeneration mobile networks is based on orthogonal frequency division mul-tiplexing (OFDM) and multiple-input multiple-output (MIMO) techniques[1], [2]. OFDM is a modulation scheme that exploits frequency diversity in-herent in a wideband channel to achieve robust performance even in severemultipath fading channel [2]. MIMO technique permits severalfold increasein achieved data rates and spectral efficiency through spatial processing andthe use of multiple transmit and receive antennas [3].

Wireless local area networks (WLANs) based on IEEE 802.11b standardwith data-rate about 11 Mbps have also been deployed widely in recent yearsfor in-home networking. IEEE 802.11a/g WLANs have increased the data-rate up to ve times (about 54 Mbps) in just a few years [3]. Currently,IEEE 802.11n is developing a standard for next-generation WLANs with data-rates over 100 Mbps. The design of physical layer of future high-performanceWLANs is also based on MIMO and OFDM techniques [3].

The short range wireless communication technologies for connectivity inoffice, home and other indoor environments have also shown similar trendsin growth. It is envisioned that wireless personal area networks (WPANs)based on IEEE 802.15.3 standard will provide high-data-rate communicationsover a distance of 10 m to support multimedia applications. Bluetooth (IEEE802.15.1 standard) is another successful WPAN technology which provideslow-power short-range communications. The exible connectivity and an en-vironment without wires is achieved with Bluetooth, however, it can sup-port only medium data-rate applications (about 1 Mbps) over a distance of 10 m. Zigbee technology and IEEE 802.15.4 is another standard intended to

1


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2 Introduction

provide very low-data-rate (20 250 Kbps), low power and low cost wirelessnetworking over longer distances (about 30 m) [4].Recently, ultra-wideband (UWB) radio has emerged as a particularly ap-

pealing transmission technique for applications requiring either high bit ratesover short ranges or low bit rates over medium-to-long ranges [5]. The ap-proach employed by UWB radio devices is based on sharing already occupiedspectrum resources by means of the overlay principle [4]. The UWB technol-ogy offers qualities such as wide unlicensed bandwidth, low power spectraldensity, high multipath resolution and multiple access capability. These qual-ities have made UWB an attractive physical layer (PHY) technology for highdata-rate WPANs and low data-rate wireless sensor networks.

In UWB systems conventionally called impulse radio, the information istransmitted using a train of short subnanosecond pulses using e.g., pulse po-sition modulation (PPM) or pulse amplitude modulation (PAM). The char-acteristics of these wide bandwidth signals pose difficulties to design low-complexity and low-cost UWB systems. The receiver structures require highsampling frequency and wide input bandwidth analog to digital converters(ADCs) to process the signal digitally [6]. There are also other issues such asfrequency dependent signal distortion, accurate synchronization and complexchannel estimation [6], [7]. The propagation characteristics of UWB signals

are also different from traditional narrowband systems [8].The sub-optimal receiver structures provide a trade-off between perfor-mance and complexity to achieve the goal of low power and low-cost UWBsystem [9], [10]. This thesis addresses the issue of design and performanceevaluation of sub-optimal coherent and non-coherent receivers for low powerand low data-rate wireless sensor network applications. The performance of coherent RAKE receivers is evaluated using UWB channel measurements per-formed in an industrial environment. The receiver structures are proposed toreduce noise, increase energy-efficiency and improve data-rate of non-coherentTR scheme. The low-complexity non-coherent recursive TR, doublet-shift TRand dual-doublet TR receivers are proposed based on transmitted referencesignaling and detection scheme.

The Introduction aims to briey discuss the ultra-wideband (UWB) sys-tems and provide an overview of the scope of this thesis. In Section 1, ageneral overview of wireless communication system is given. Section 2 pro-vides background and characteristics of UWB system. In section 3, typesand overview of UWB systems is given briey. The characteristics of UWBchannels and introduction of IR-UWB receivers is given briey in Section 4and 5, respectively. Finally, thesis contributions are summarized in Section 6.


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Introduction 3

DigitalSource Coding Modulation

Bit to SymbolMapping

PulseShaping

Destination DecodingSymbol to bit

MappingDetection Equalization Demodulation

Tx

Rx

Figure 1: A block diagram representation of a wireless communication system.

1 Wireless Communication System OverviewA simplied overview and block diagram of a typical wireless communicationsystem is given in Fig. 1. In the transmitter, the digital source provides binaryinformation bits to be transmitted. The binary information bits are encoded

to remove the redundancy in the transmitted signal for efficient transmission,this is also called source coding . Moreover, the reliable transmission overa noisy wireless channel is achieved by introducing controlled redundancyin the signal, called channel coding . Further, the bit to symbol mappingis performed, where each symbol can be regarded as a member of a nitealphabet set containing M members. For M > 2, each symbol is made up of a sequence of two or more bits and is called M ary symbol. The bit stream isconverted to baseband waveforms using pulse shaping lter in the next stage.Finally, the bandpass modulation is used to transmit pulse-like signal on aparticular carrier frequency.

The received signal is demodulated at the receiver by frequency down-conversion to extract the baseband signal. After demodulation, an equalizer isused to mitigate the distortion effects introduced by the channel. Equalization can be described as a ltering operation to recover the transmitted signaldegraded by the channel. The detection of the recovered signal is performedin the next stage. The detected symbols are mapped back to bits and decodedto achieve binary source signal.

The relationship between the received signal and the transmitted signal of a communication system is typically called the channel [6]. The transmission


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4 Introduction

Tx

Rx

MultipathPropagation

Figure 2: An illustration of multipath propagation over a wireless channel,where Tx and Rx stand for the transmitter and receiver, respectively.

of a signal over wireless channel results in multipath components (MPCs) thatarrive at the receiver with different attenuations and delays after reectionsfrom different objects, as also illustrated in Fig. 2. This phenomena is called

multipath propagation and leads to signal distortion and fading. The ampli-tude variations in the received signal occur due to time-varying nature of thechannel impulse response. The envelope of the channel impulse response isRayleigh distributed if the channel impulse response can be modeled as azero-mean complex-valued Gaussian process [11]. This happens when there isno line-of-sight (LOS) component in the received signal; this kind of channelis called Rayleigh fading channel. However, if a LOS component exists in thereceived signal, the channel is called Rician fading channel and its impulse re-sponse can be modeled as a non-zero mean complex valued Gaussian process[11]. The Rician fading is often described in terms of a Rician factor which isdened as the ratio between the deterministic signal power and the varianceof the multipath [12]. As the dominant path decreases in amplitude, the Ri-cian distribution degenerates to a Rayleigh distribution [12]. The probabilitydensity function (pdf) of Rayleigh and Rician distribution is shown in Fig. 3.In addition to multipath propagation, there are also other limiting featuresin wireless communications such as intersymbol interference (ISI), co-channelinterference and noise. The receiver structures use channel equalization, in-terference mitigation and noise reduction techniques to recover the originaltransmitted signal.


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Introduction 5

0 2 4 6 8 100

0.01

0.02

0.03

0.04

0.05

0.06

0.07

Rayleigh

Rician

Figure 3: Probability density function (pdf) of Rayleigh and Rician distribu-tion, where Rician factor = 1 dB.

2 Background and Characteristics of UWB

The early applications of UWB technology were primarily radar related, drivenby the promise of ne-range resolution that comes with large bandwidth [13].Beginning in the late 1980s, small companies specializing in UWB started ba-sic research and development on communications and positioning systems [13].In 1993, Scholtz demonstrated the potential of impulse radio (IR) UWB toacademic community [14] and later, Win and Scholtz published their pio-neering work on time-hopping (TH) IR-UWB for multi-user communications[15], [16], [17]. In April 2002, after extensive commentary from industry, theFederal Communications Commission (FCC) issued its rst report and orderon UWB technology, thereby providing regulations to support deployment of UWB radio systems [13], [18]. The FCC regulations classify UWB applica-tions into several categories with different emission regulations in each case[13].

According to FCC denition , UWB characterizes transmission systemswith instantaneous spectral occupancy of 500 MHz or a fractional bandwidthof more than 20% [19]. The fractional bandwidth is dened as

f BW =f H f L

f c, (1)


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6 Introduction

Narrowband

P o w e r

S p e c t r a l

D e n s i t y ( d B )

Frequency (Hz)

UWB

-10 dB

f L f C f H

Figure 4: Comparison of the fractional bandwidth of a narrowband and ultrawideband communication system [6].

where f c = ( f H + f L )/ 2 with f H being the upper frequency of the 10 dBemission point, and f L the lower frequency of the 10 dB emission point [19],as shown in Fig. 4. At the physical layer level, UWB communication systemsoperate by spreading rather small amounts of average effective isotropic ra-diated power (EIRP) always less than 0.56 mW across a very wide band of frequencies relative to its center frequency [4]. The spectral mask assignedfor indoor UWB communication systems is shown in Fig. 5. The power spec-trum density limit of UWB systems is 41.3 dBm/MHz within 3 .1 GHz to10.6 GHz range.

The following are some of the characteristics of UWB systems:

Improved channel capacity. The channel capacity can be improved asUWB systems occupy a very wide frequency spectrum. The channelcapacity C (in bits per second (bps)) of the band-limited additive whiteGaussian noise (AWGN) channel increases with radio frequency (RF)bandwidth [13],

C = BRF log2(1 +P rec

BRF N o), (2)

where BRF is the RF bandwidth of the channel and P rec is the received


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Introduction 7

2 4 6 8 10 12 1480

75

70

65

60

55

50

45

40

35

Frequency (GHz)

U W B E I R P E m m

i s i o n

L e v e

l ( d B m

/ M H z )

-41.3

-51.3

-53.3

-75.3

-51.3

Part 15 Limits

Indoor Limits

Figure 5: FCC allocated spectral mask for indoor ultra wideband communi-cation systems [18].

signal power and N o is the noise power spectral density (PSD) in theRF bandwidth of the radio.

Reduced fading effects. The use of signals with gigahertz bandwidths alsomeans that multipath is resolvable down to path differential delays onthe order of a nanosecond or less, i.e., down to path length differentialson the order of a foot or less [17]. This signicantly reduces fadingeffects even in indoor environments [16], [17].

Rich multipath diversity. A pulse of width T p at the subnanosecond

scale, occupies UWB with bandwidth B 1/T p [19]. Such an ultra-short pulse gives rise to multiple resolvable copies, and thus enables richmultipath diversity [19].

High-resolution position location and tracking or radar sensing. Thelarge signal bandwidth yields a distance resolution between communi-cating devices or a radar sensing accuracy within a few centimeters [20].

Extremely low power spectral density (PSD). Average power levels in the


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8 Introduction

order of millionths of a Watt (mW) and excessive signal bandwidth yieldpower spectral densities in the order of several tens of nW/MHz [20].

The capability to highly resolve multipath combined with the ability topenetrate through materials makes impulse radio technology viable for high-quality, fully mobile short-range indoor radio systems [17]. The applications of UWB can be broadly divided into two categories depending on the data rates.The high data-rate (HDR) applications of UWB, according to IEEE 802.15.3astandardization group, are short-range (about 10 m) wireless personal areanetworks (WPANs). Alternatively, UWB radios can trade a reduced informa-tion rate for increased link range, potentially combined with accurate locationtracking capabilities, offering an operational mode dened here as low datarate and location tracking (LDR/LT) [4]. The applications for medium to longdistances (about 100 m) with relatively low data-rates and location tracking(LDR/LT) capability are identied by IEEE 802.15.4a standardization group.These applications include e.g., sensor, positioning and identication network(SPIN), wireless body area network (WBAN) etc. A SPIN is a system char-acterized by a high density (e.g., hundreds per oor) of devices (intelligentsensors or tags) in industrial factories or warehouses transmitting low-ratedata combined with position information (e.g., data rate greater than several

tens of kilobits per second and position accuracy well within 1 m) [4]. Thereare also other applications of UWB such as through-wall imaging, medicalimaging, vehicular radar and ground-penetrating radar [18].

3 Types of UWB SystemThere are two common forms of UWB called multicarrier UWB (MC-UWB)and impulse radio UWB (IR-UWB). In this thesis, IR-UWB systems are con-sidered, however, an overview of both systems is given briey in the followingsubsections.

3.1 MC-UWB SystemThe approach using multiple simultaneous carriers for transmission of UWBsignals is called MC-UWB or frequency domain UWB [6]. Orthogonal Fre-quency Domain Multiplexing (OFDM) is a special case of multicarrier trans-mission that permits subcarriers to overlap in frequency without mutual inter-ference and hence spectral efficiency is increased [6]. The multiband-OFDM


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Introduction 9

UWB is one of the proposed physical layer standard for IEEE 802.15.3aWPANs [6]. The idea of multiband-OFDM UWB is to divide the UWB spec-trum into subbands with a minimum subband bandwidth of 500 MHz, anduse multibanding in conjunction with the OFDM modulation. Each OFDMsymbol is transmitted using orthogonal sub-carriers time-interleaved acrosssubbands using time-frequency codes [21]. This approach not only providesfrequency diversity but also multiple access [21].

3.2 IR-UWB System

Impulse radio UWB communicates with baseband pulses of very short dura-tion, typically on the order of a nanosecond, thereby spreading the energy of the radio signal very thinly from near dc to a few gigahertz (GHz) [22]. Typ-ically, each symbol consists of multiple frames carrying one pulse per frame.Multiple pulses are associated with a single symbol to obtain sufficient energyper symbol while maintaining sufficiently low PSD [6].

Due to baseband (carrier-less) nature of IR-UWB signals, frequency up-conversion and down-conversion is not required in the transceiver. This re-duces the complexity and power consumption of transceiver and thus makes

IR-UWB suitable for low-complexity and low power wireless sensor networkapplications. The pulse shapes, modulation and multiple access schemes of IR-UWB system are discussed in the following subsections.

3.2.1 UWB Pulse Shape

Generally adopted pulse shapes for UWB communications include the Gaus-sian pulse, the rst derivative of Gaussian pulse (Gaussian monocycle) and thesecond derivative of the Gaussian pulse (Gaussian doublet) [19]. The secondderivative of a Gaussian function is described by [17],

p(t) = 1 4(t

m)2 e 2 (

t m

) 2 , (3)

where m is the shape factor, reducing the value of m shortens the pulse,and thus enlarges the bandwidth of the transmitted signal [6]. There are alsoother pulse shapes proposed in the literature such as the Laplacian pulse [23]and Hermite pulses [24].


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10 Introduction

Time [s]

A m p l

i t u d e

-2 -1 0 1 2

x 10-9

-1

-0.5

0

0.5

1Gaussian pulse

Gaussian monocycle

Gaussian doublet

Figure 6: Waveforms of the Gaussian pulse, Gaussian monocycle and Gaussiandoublet.

3.2.2 Multiple Access and ModulationIn multiuser IR-UWB systems, multiple access capability is usually enabledwith time-hopping (TH) and direct-sequence (DS) UWB multiple access. TheTH and DS spreading not only provide multiple access but also smooth thetransmit PSD [19]. The commonly used modulations with time-hopping (TH)scheme are antipodal pulse amplitude modulation (A-PAM), pulse positionmodulation (PPM) and on-off keying (OOK).

For binary antipodal PAM, also called binary phase shift keying (BPSK),the two possible signals generated by the transmitter are,

s(t)A P AM = E p p(t) if b = 0

E p p(t) if b = 1 (4)where p(t) is the energy-normalized pulse and E p is the energy per pulse.Similarly, the two possible signals generated by the transmitter for binaryorthogonal PPM are,

s(t)P P M = E p p(t) if b = 0 E p p(t ) if b = 1 (5)


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Introduction 11

where is the time shift introduced by PPM. The transmitted signal for OOKmodulation can be written in a similar manner as,

s(t)OOK =0 if b = 0

E p p(t) if b = 1 (6)The transmitted reference (TR) is another modulation for IR-UWB which

has been used recently with time-hopping (TH) multiple access scheme [25],[26]. The signal models for time-hopping (TH), direct-sequence (DS) andtransmitted reference (TR) IR-UWB systems are given as follows.

Time-hopping (TH) IR-UWB Signals:

In TH IR-UWB, each pulse is positioned within each frame duration ac-cording to a user-specic TH sequence [19]. The transmitted TH-A-PAMsignal of kth user can be written as [27],

s (k)T H A P AM (t) = E p

j =

(1 2b(k)

j/N f ) p(t jT f c(k )j T c). (7)

In a similar manner, the transmitted TH-PPM signal can be written as [27],

s (k )T H P P M (t) = E p

j =

p(t jT f c(k )j T c b

(k )j/N f ), (8)

where, in (7) and (8), T f is the frame duration, T c is hopping width or chipduration, N f is the number of pulses representing one binary informationsymbol b(k )j {0, 1}, is the PPM shift, and c

(k )j is the pseudorandom time-

hopping (TH) sequence in the range 0 c(k )j N h , where N h is the numberof hops. Fig. 7(a) show an example of PPM TH IR-UWB system.

Direct-Sequence (DS) IR-UWB Signals:

In DS UWB impulse radio, each information symbol is direct-sequencemodulated using a spread spectrum pseudo-noise (PN) code specic to eachuser. As these PN codes are orthogonal and known at the transmitter andthe receiver, the multiple-access interference is negligible. The transmittedDS-UWB modulated signal can be written as [28],


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12 Introduction

s (k)DS UWB (t) = E p

j =

(1 2b(k )j )

N c

i =0

c(k )i p(t jT f iT c), (9)

where T c is the chip duration, N c is the number of chips per information bitb(k)j {0, 1}, T b = T f = N cT c , c

(k )i {1, +1 }is the spread spectrum codeof user k. Fig. 7(b) show an example of DS IR-UWB system.

Transmitted Reference (TR) IR-UWB Signals:

In transmitted reference (TR) scheme, a reference and a data-modulatedsignal separated by a xed delay is called a TR doublet . Each bit is trans-mitted over N f frames with each frame carrying a TR doublet [10], [29].The transmitted signal for TR TH-UWB system using binary antipodal PAMmodulation can be written as,

s (k )T H T R (t) =

E p

j =

p(t jT f c(k)j T c)

+ (1 2b(k )

j/N f ) p(t jT f c(k)j T c T d ) , (10)

where p(t) is the transmitted Gaussian monocycle, E p is the pulse energy, T cis the chip duration, T f is the frame duration. Fig. 7(c) shows an exampleof TR TH IR-UWB system. The gure explains that each frame contains aTR doublet consisting of two pulses separated by a xed delay of durationT d . Thus, one binary information symbol b j/N f {0, 1}is transmitted by astream of 2 N f pulses. The cj {0, 1,...,N c 1}is a periodic pseudo-randomtime-hopping sequence, which is used to time-hop the pairs of reference anddata-modulated pulses in each frame.

4 Characteristics of UWB ChannelsPreviously, most of the work has been performed in the area of narrowbandchannel modeling and characterization, whereas channel modeling for UWBsystems is relatively a new area [8]. In [31], [32] and [33], UWB channelmeasurement results are presented and channel models are proposed based on


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Introduction 13

0

(a)

0

(b)

0

(c)

b1 = 0 b2 = 1

T f = N h T c

T c

T c

T c

T p

T b = N f T f

T b = N f T f

T b = N cT c

2 T b

2 T b

2 T b

T dT dT dT dT f

Figure 7: (a) PPM TH IR-UWB system for two bits b1 = 0 and b2 = 1, whereN h = 3, N f = 4 and is the PPM shift between two pulse positions. (b) DSIR-UWB system for two bits b1 = 0 and b2 = 1, where T c is chip durationand c = [1, 1, 1, 1, 1, 1]. (c) TR TH IR-UWB system for two bits b1 = 0and b2 = 1, where N h = 3, N f = 2 and T d is the delay between reference andmodulated (shown lled) pulses.


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14 Introduction

the results. The narrowband channel models can not be generalized to UWBchannels due to following important differences described in [8], [34]:

Each multipath component of UWB signal can lead to delay dispersionby itself, due to frequency-selective nature of reection and diffractioncoefficients. This effect is especially important for systems with largerelative bandwidth.

The UWB signals are received with excellent delay resolution. There-fore, it often happens that only a few multipath components make upone resolvable MPC. That implies that the central limit theorem is notfullled anymore, and the amplitude statistics of such a resolvable MPCare not complex Gaussian anymore. Similarly, there is an appreciableprobability that areas of no energy can exist during which no signi-cant amount of energy is arriving at the receiver.

The statistics of arrival times of multipath components strongly varywith the bandwidth, as well as with the center frequency of the UWBsignal.

Due to wide frequency band, the propagation signals experience fre-quency dependent effects. Specically, the path loss is described as afunction of frequency as well as of distance when the relative bandwidthis large.

5 IR-UWB ReceiversThe received signal in any communications system is an attenuated, delayed,and possibly distorted version of the signal that was transmitted plus noise and(possibly) interference [6]. For IR-UWB systems, coherent and non-coherentreceivers implemented either in digital or analog domain can be used for detec-tion of distorted and noisy received signal. The implementation of all-digitalreceiver requires high-sampling frequency ADCs, and its high-data-rate so-lutions also need large memory and high processing speeds, which makes itexpensive to implement [6]. Alternatively, a fully analog implementation of IR-UWB receiver can provide a simple and low-cost receiver. The coher-ent and non-coherent IR-UWB receivers are discussed briey in the followingsubsections.


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Introduction 15

Matched filter 1

Matched filter 2

Matched filter K

Combiner

Path searcher

Received signal Recovered signal

Training Synchronization sequence /

Figure 8: A RAKE receiver with K ngers.

5.1 Coherent Receivers

The coherent receivers correlate the received signal with a locally generatedreference (LGR) template [9]. However, to correlate the received UWB signalwith LGR template, the receiver needs to achieve the pulse-level synchroniza-tion with accuracy on the order of tens of picoseconds [35].

The coherent RAKE receivers take advantage of the time-diversity pro-vided by multipath channel. This scheme was invented by Price and Green[36] in 1958. It consists of a bank of matched lters (also called ngers ) witheach nger matched to a different replica of the same transmitted signal, seeFig. 8. The outputs of the ngers are appropriately weighted and combinedto reap the benets of multipath diversity [9]. The coherent RAKE receiversrequire accurate synchronization and the knowledge of channel impulse re-sponse.


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16 Introduction

BandpassFilter

Delay

DecisionReceived signal

dt

Figure 9: A non-coherent transmitted reference (TR) receiver structure.

5.2 Non-Coherent ReceiversThe non-coherent receivers do not require channel estimation and have low-complexity implementation at the expense of performance loss. The sub-optimal non-coherent receivers are suitable for low power and low-cost sce-narios such as LDR WPANs and wireless sensor networks. Transmitted ref-erence (TR) receiver and energy detector (ED) are two popular non-coherentdetection schemes for IR-UWB signals.

5.2.1 Transmitted Reference Receivers

The non-coherent transmitted reference receiver shown in Fig. 9 is used todemodulate the signal transmitted using TR modulation. TR receiver delaysthe reference signal and correlates it with data-modulated signal in each frame.The decision statistic is acquired by adding the correlations over N f frames.TR receivers exploit multipath diversity inherent in the environment withoutthe need for stringent acquisition and channel estimation [29].

On the other hand, TR receiver suffers a 3 dB penalty because half of thepulses are unmodulated [6]. In addition, the performance of TR receiver isalso limited by the noise-on-noise term in the correlator due to noisy referencetemplate [6], [30]. The effect of noise-on-noise term can be reduced partially

by using averaging of the reference pulses before correlation [10], [30].

5.2.2 Energy Detectors (ED)

Energy detector is a non-coherent approach for UWB signal reception, seeFig. 10, where low complexity receivers can be achieved at the expense of someperformance degradation [37], [38]. On-off keying (OOK) is one of the mostpopular non-coherent modulation options that has been considered for energydetectors [39]. OOK based implementation of energy detectors is achieved by


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Introduction 17

Bandpassfilter

Square-lawdevice dt Decision

Received signal

Figure 10: A non-coherent energy detector (ED) receiver structure.

passing the signal through a square law device followed by an integrator and adecision mechanism, where the decisions are made by comparing the outputsof the integrator with a threshold [39]. Pulse position modulation (PPM) can

also be used for non-coherent energy detector (ED) receiver.

6 Contributions of The ThesisThe thesis consists of four stand-alone parts. In Part I, performance of RAKEreceivers is evaluated for low data-rate IR-UWB systems using measured andsimulated multipath channels for industrial environments. In Part II, recursiveschemes of transmitted reference receivers are presented for noise averagingand BER improvement of IR-UWB systems. Part III presents a doublet-shifttransmitted reference scheme to improve data-rate and BER of IR-UWB com-munication systems. In Part IV, dual-doublet transmitted reference schemesto improve data-rate and energy efficiency of TR IR-UWB systems are pro-posed. In the following subsections, a short summary of each part is given.

6.1 Part I: Performance Evaluation of RAKE Receiverfor IR-UWB Systems using Multipath Channels forIndustrial Environments

In this Part, the performance of RAKE receivers is evaluated in terms of uncoded BER using the measured channel responses in an industrial en-vironment. The standard channel model proposed by IEEE 802.15.4a fornon line-of-sight (NLOS) industrial environments is also used for comparison.The industrial environment is considered as it is expected that UWB deviceswill be deployed in industrial buildings, factories and warehouses for applica-tions such as process control, supervision of storage halls, asset tagging andmanagement [40]. The multipath environment is dense for such environmentsdue to large number of metallic reectors in the surroundings. The channelcharacteristics also differ from office and residential environments and need to


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18 Introduction

be considered for system design [41]. In the past, performance evaluations forcoherent and non-coherent receivers using UWB channels has been performedonly for office or residential environments [42], [43], [44], [45], [46]. Moreover,the realistic channel measurement data has not been used in the previousperformance evaluations of IR-UWB receivers.

The performance evaluation results presented in this Part provide receiverperformance degradation from shorter (28 m) to longer (816 m) Tx-Rx sep-arations. The BER performance of RAKE receiver types such as selectiveRAKE (SRake) and partial RAKE (PRake) is compared. The number of ngers required for PRake and SRake receiver to achieve a specic BER isdetermined. The comparison between RAKE combining scheme such as max-imal ratio combining (MRC) and equal gain combining (EGC) is presented.It is found that the difference in the performance of MRC and EGC combin-ing schemes is not that signicant for SRake, while PRake has a considerablybetter performance using MRC. The results show that the performance of re-ceiver depends to a large extent on the underlying channel. It is also observedthat the simulated channels for industrial environments can not be general-ized for all such environments as the propagation characteristics can differsignicantly from one environment to another.

6.2 Part II: Recursive Transmitted Reference Receiversfor Impulse Radio UWB Systems

In Part II, recursive schemes of the transmitted reference receiver structureto improve the detection performance of IR-UWB signals are proposed. Theperformance of conventional TR scheme suffers from the noisy reference tem-plate in the correlator. The noise averaging schemes described in the literaturesuch as presented in [10] have higher implementation complexity. The pro-posed recursive schemes decrease noise in the reference pulse by recursivelyestimating the reference template. The recursive TR (R-TR) structure es-timates the reference template for the correlator frame-by-frame recursivelyfrom the previous template and the newly received reference pulse. The re-cursive ATR (R-ATR) structure estimates the reference template bit-by-bitrecursively from the previous template and the average over reference pulseswithin current bit.

The proposed R-TR receiver has low-complexity as it requires relativelyshorter delays for recursive averaging. Moreover, a slightly modied TR sig-naling sequence in conjunction with ATR and R-ATR receiver scheme is sug-gested. The use of slightly modied TR signaling sequence in conjunction


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Introduction 19

with ATR and R-ATR receiver scheme requires shorter delays for averaging.In the proposed recursive scheme, the complexity of recursion process in R-ATR scheme can be reduced by repeating the time-hopping sequence after abit duration. It has been evaluated by simulations of TR IR-UWB systemthat the R-TR and R-ATR receiver schemes outperform the conventional TRand ATR receiver over slowly-varying channels.

6.3 Part III: A Doublet-Shift Transmitted ReferenceScheme for Ultra-Wideband Communication Systems

In Part III, a low-complexity doublet-shift TR signaling and detection schemeis presented for IR-UWB systems. The conventional TR scheme has low-complexity but its performance degrades due to two drawbacks. It uses anoisy reference template in the correlator, and the use of reference pulsesdecreases the data-rate.

The proposed scheme transmits two TR doublets within a frame dura-tion by using closely-spaced pulses and shifting the positions of the pulses inthe latter doublet. This scheme utilizes the same amount of energy as theconventional TR scheme. The proposed DSTR signaling scheme also has thepotential to increases the data rate or capacity by using closely-spaced pulses

within a doublet. Moreover, the receiver structure DSTR-I for the proposedscheme is the same as the conventional TR receiver. However, the improvedDSTR-II structure has higher complexity as it requires an extra delay andcorrelation operation to demodulate the pulses within a doublet. The perfor-mance evaluation results provide a comparison of the proposed scheme withconventional TR scheme. The performance improvement of doublet-shift TRsignaling scheme over conventional TR signaling scheme is validated usingsimulation results.

6.4 Part IV: Transmitted Reference Schemes based on

Dual-Doublets for Impulse Radio UWB SystemsIn this Part, low-complexity dual-doublet TR signaling and detection schemes(scheme 1 and scheme 2) are presented for IR-UWB systems. The proposedDDTR (scheme 1) transmits three bits with four pulses using a combinationof the conventional TR and differential TR schemes. In DDTR (scheme 1),1.76 dB less energy per bit is utilized and 25% rate loss of conventional TRscheme is recovered for the same BER. The receiver structure for DDTR(scheme 1) requires an extra delay and a correlator.


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20 Introduction

The proposed DDTR (scheme 2) transmits two bits per symbol by usingtwo TR doublets in each frame. For the detection of DDTR (scheme 2), eachmodulated pulse is correlated with two reference pulses, as opposed to theconventional TR scheme which correlates each modulated pulse with only onereference pulse. The proposed DDTR (scheme 2) receiver structure requiresan extra, a correlator and two adders. It has been validated by the simulationresults that the proposed DDTR scheme requires 3 dB less energy per bit andrecovers 50% rate loss of conventional TR scheme, while giving the same BERperformance. It is also observed that the BER performance improvement of about 2 dB is achieved with proposed scheme if the same energy/rate is usedas the conventional TR scheme.

The proposed DDTR schemes have an advantage over existing energy-efficient differential TR scheme as DDTR schemes do not require a decisiondirected approach for averaging of the reference pulses. The DDTR receiversare extended to ADDTR receivers to perform averaging of the reference pulsesfor noise reduction in the template waveform.

References

[1] S. Chia, T. Gill, L. Ibbetson, D. Lister, A. Pollard, R. Irmer, D. Almod-ovar, N. Holmes, S. Pike, 3G evolution IEEE Microwave Magazine,Aug. 2008 , vol. 9, no. 4, pp. 52-63

[2] A. Doufexi, S. Armour, Design considerations and physical layer perfor-mance results for a 4G OFDMA system employing dynamic subcarrierallocation, IEEE 16th International Symposium on Personal, Indoor and Mobile Radio Communications, PIMRC 2005 , vol. 1, pp. 357-361

[3] S. Nanda, R. Walton, J. Ketchum, M. Wallace, and S. Howard, A high-performance MIMO OFDM wireless LAN, IEEE Communications Mag-

azine , vol. 43, no. 2, pp. 101-109[4] D. Porcino and W. Hirt, Ultra-wideband radio technology: potential

and challenges ahead, IEEE Communications Magazine, 2003 , vol. 41,no. 7, pp. 66- 74

[5] M.-G. di Benedetto, T. Kaiser and N. Schmidt UWB State of the Art ,Journal of Applied Signal Processing Editorial, EURASIP publishing,March 2005


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22 Introduction

[18] U. S. Federal Communications Commission (FCC), Revision of part 15of the commissions rules regarding ultra-wideband transmission systemsFirst Report and Order, ET Docket 98153, FCC 0248; Adopted: Febru-ary 2002; Released: April 2002

[19] L. Yang and G. B. Giannakis,Ultra-wideband communications, IEEE Signal Process. Magazine , vol. 21, no. 6, pp. 2654, Nov. 2004

[20] W. Hirt, Ultra-wideband radio technology: overview and future re-search, Elsevier Computer Communications vol. 26, no. 1, pp. 46-52,

1 January 2003[21] A. Batra, J. Balakrishnan, G. R. Aiello, J. R. Foerster and A. Dabak,

Design of a Multiband OFDM System for Realistic UWB Channel En-vironments, IEEE transactions on Microwave Theory and Techniques ,vol. 52, no. 9, Sept. 2004

[22] M. Z. Win and R. A. Scholtz, Impulse radio: How it works, IEEE Commun. Lett. , vol. 2, no. pp. 3638, Feb. 1998

[23] J. T. Conroy, J. L. LoCicero, D. R. Ucci, Communication techniquesusing monopulse waveforms IEEE Military Communications ConferenceProceedings, MILCOM 1999 , vol. 2, pp. 1181-1185

[24] M. Ghavami, L. B. Michael, S. Haruyama, R. Kohno, A Novel UWBPulse Shape Modulation System, Wireless Personal Communications,2002 - Springer , vol. 23, no. 1, pp. 105-120

[25] R. Hoctor and H. Tomlinson, Delay-hopped transmitted-reference RFcommunications, IEEE UWBST , pp. 265269, Baltimore, MD, 2002

[26] R. T. Hoctor and H.W. Tomlinson, An overviewof delay-hopped,transmitted- reference RF communications, in Technical Information Se-ries : G.E. Research and Development Center, Jan. 2002, pp. 129.

[27] G. Durisi, S. Benedetto, Performance evaluation and comparison of dif-ferent modulation schemes for UWB multiaccess systems, IEEE Inter-national Conference on Communications, 2003. ICC 03. , vol. 3, pp.2187- 2191

[28] Bo Hu, N. C. Beaulieu, Accurate performance evaluation of time-hopping and direct-sequence UWB systems in multi-user interference,


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24 Introduction

[39] M.E. Sahin, I. Guvenc, H. Arslan, Optimization of energy detector re-ceivers for UWB systems, IEEE 61st Vehicular Technology Conference,2005. VTC 2005-Spring. , vol. 2, pp. 1386- 1390

[40] M. G. Khan, A. A. Ashraf, J. Karedal, F. Tufvesson, and A. F.Molisch, Measurements and Analysis of UWB Channels in IndustrialEnvironments, in Proc. Wireless Personal Multimedia Communications(WPMC) , Aalborg, Denmark, Sept. 2005

[41] J. Karedal, S. Wyne, P. Almers, F. Tufvesson, and A. F. Molisch, UWB

channel measurements in an industrial environment, in Proc. IEEE Globecom , 2004

[42] A. Rajeswaran, V.S. Somayazulu, J. R. Foerster, RAKE performance fora pulse based UWB system in a realistic UWB indoor channel, Proc.IEEE International Conference on Communications (ICC 03) , vol.4,pp. 2879 - 2883, 11-15 May 2003

[43] B. Mielczarek, M. Wessman, and A. Svensson, Performance of coherentUWB RAKE receivers with channel estimators, Proc. IEEE Vehicular Technology Conference , vol. 3, pp. 1880-1884, Oct. 2003

[44] G. Durisi, S. Benedetto, Performance of coherent and noncoherent re-ceivers for UWB communications , Proc. IEEE International Conferenceon Communications (ICC 2004) , vol. 6, pp. 3429-3433, June 20-24, 2004

[45] S. Gezici, H. Kobayashi, H. V. Poor and A. F. Molisch, Optimal and sub-optimal linear receivers for time-hopping impulse radio systems, Proc.IEEE Wireless Communications and Networking Conference (WCNC 2004), vol. 2, pp. 908-913, Atlanta, GA, March 2004

[46] M. A. Rahman, S. Sasaki, Z. Jie, S. Muramatsu, H. Kikuchi, Perfor-mance evaluation of RAKE reception of ultra wideband signals over mul-tipath channels from energy capture perspective, International Work-shop on Ultra Wideband Systems, 2004. Joint with Conference on Ultra-wideband Systems and Technologies, pp. 231-235, May 2004


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Part I


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Part I

Performance Evaluation of RAKE Receiver forIR-UWB Systems using Multipath Channels for

Industrial Environments


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Parts of this work are published as:

M. G. Khan, J. Nordberg and I. Claesson, Performance Evaluation of RAKEReceiver for Low Data Rate UWB Systems using Multipath Channels forIndustrial Environments, Research report, Blekinge Institute of Technology,2008, Issue: 4, ISSN: 1103-1581.

M. G. Khan, J. Nordberg and I. Claesson, Performance evaluation of RAKEreceiver for UWB systems using measured channels in industrial environ-ments, AusWireless06 , March 2006, Sydney, Australia.


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Performance Evaluation of RAKE Receiver forIR-UWB Systems using Multipath Channels for

Industrial Environments

Muhammad Gufran Khan, J orgen Nordberg and Ingvar Claesson

Abstract

For UWB communication systems, the industrial environments arean important scenario. However, due to large number of metallic scat-terers in the environment, the multipath offered by UWB channels isdense and many multipath components have signicant energy. In thisPart, the performance evaluation of RAKE receivers for a single usersystem operating in non-line-of-sight (NLOS) scenarios in industrial en-

vironments is presented. The channels used for the evaluation are mea-sured in a medium-sized industrial environment. In addition, the stan-dard IEEE 802.15.4a channel model for NLOS industrial environment isused for comparison with the results of the measured channels. The per-formance is compared for partial RAKE (PRake) and selective RAKE(SRake) in terms of uncoded bit-error-rate (BER) with the assumptionthat the channel is known. The effect of different number of ngerson BER of PRake and SRake is studied. Moreover, the performanceof maximal ratio combining (MRC) and equal gain combining (EGC)is compared for PRake and SRake receiver. The results also provideBER performance comparison as separation between transmitter andreceiver increases. Finally, based on the simulation results, conclusionsare presented considering the performance and complexity issues.

1 IntroductionThe transmission of UWB signal over a wireless channel results in multipathcomponents (MPCs). Due to wide bandwidth, the MPCs arrive with highdelay resolution and thus receiver (Rx) is able to resolve many MPCs. The

29


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30 Part I

RAKE receiver can be used to receive the MPCs as it exploits the time-diversity inherent in multipth and attempts to collect the signal energy co-herently from the received signal paths that fall within its span [1]. However,as the number of resolved MPCs is very high in case of UWB systems, thecombining of hundreds of MPCs using RAKE receivers is not realistic. Itbecomes important to use only a subset of the received MPCs using a specicselection criteria.

The IEEE 802.15.4a group, which has developed a physical layer standardfor low-data-rate (LDR) systems, has recognized the fact that a considerable

amount of UWB devices will be deployed in industrial buildings, factoriesand warehouses [2]. The application set includes e.g., sensor networks forprocess control, supervision of storage halls, asset tagging and management.For such environments like a factory hall with multiple metallic reectors, themultipath environment is dense and almost all resolvable delay bins containsignicant energy [3]. In this case, a RAKE receiver needs to capture a largenumber of (on the order of hundred) multipath components to collect a sig-nicant amount of the received energy [2], [4]. The RAKE receiver design inthis case is a challenging task.

Previously, the performance of coherent RAKE and non-coherent receiversfor UWB systems has been evaluated in the literature, however, most of theseevaluations have been performed using UWB channels for office or residentialenvironments [5], [6], [7], [8], [9]. Moreover, as most of the evaluations arebased on simulated channel models employing Rayleigh or Lognormal fadingstatistics, it is necessary to evaluate the system performance considering realis-tic channel characteristics of the environment. Thus, the objective of this Partof thesis is to investigate and analyze the performance of RAKE receivers interms of uncoded BER for industrial environments using the measured chan-nel responses. The standard channel model proposed by IEEE 802.15.4a forNLOS industrial environments is also used for comparison. The dependence of achievable BER on the types of RAKE, the number of ngers and the RAKEcombining schemes is evaluated and compared. Further, based on the simu-lation results, conclusions are drawn considering performance and complexityissues of RAKE receiver design for these scenarios.

The rest of the Part I is organized as follows: the system model of a typicalUWB system is described in Section 2. Section 3 describes the UWB channelsused for the performance evaluation of the system and Section 4 describesRAKE receivers. The simulation parameters and results are discussed inSection 5 and nally conclusions are presented in Section 6.


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Performance Evaluation of RAKE Receiver for Low Data Rate UWB Systems ... 31

ModulationMultipathChannel RAKE

Receiver

Data

Pseudo-random Codes

Detector

s(t)

h(t)

g(t) r (t)

n(t)

Figure 1: Block diagram of the system model.

2 System ModelConsidering a binary antipodal modulation and time-hopping impulse radio(TH-IR) UWB system as multiple-access scheme, the transmitted signal froma user in the system can be represented as

s(t) = E p

j =

dj b j/N f p(t jT f cj T c), (1)where p(t) is the energy-normalized transmitted UWB pulse, E p is the energyper pulse, T f is the frame duration, T c is the chip duration, N f is the num-ber of pulses representing one binary information symbol b j/N f {1, 1}transmitted by the user. Fig. 1 and 2 show the system model and an exampleof the transmitted sequence for a single bit, respectively. The pseudorandomtime-hopping (TH) sequences {cj }are assigned to each user that share theUWB media to avoid catastrophic collisions among the pulses of differentusers. If the number of chips in a frame is denoted as N c , then the chip inter-val is chosen to satisfy T c T f /N c . This avoids pulses of different users fromoverlapping. The pseudorandom polarity codes dj {1, 1}having equalprobability provide robustness against multiple access interference [10]. Thepolarity randomization codes also help to get a zero-mean output and shape

the transmit spectrum according to FCC rules [11].The signal for a user is transmitted through a multipath channel with theimpulse response given by

h(t) =K 1

k=0

k (t k ), (2)where k are the channel tap weights, K is the number of multipath com-ponents and k is the delay associated with kth multipath component. The


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34 Part I

Delay

Amplitude

Figure 3: Multipath components with different attenuations and delays.

where is the frequency dependence factor. The distance dependence of thepath loss in dB is described by [13],

P L (d) = P L o + 10 n log10d

do(9)

where the reference distance do is set to 1 m, P L o is the path loss at thereference distance, and n is the path loss exponent. The path loss exponentdepends on the environment and on whether a line-of-sight (LOS) connectionexists between the transmitter and receiver or not [13].

The variation in the received signal power about its mean value is typicallytermed shadowing [12]. The path loss (averaged over small-scale fading) indB can be written as [13],

P L (d) = P L o + 10 n log10d

do+ S (10)

where S accounts for shadowing and is a Gaussian distributed random variablewith zero mean and the standard deviation s .

3.1.2 Power delay prole

Power delay prole (PDP) relates the power of the received signal with thedelay experienced by the multipath component and is dened as the square


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magnitudes of the impulse response of the signal averaged over a local areaas [16],

P DP ( ) = |h(t; )|2 , (11)

where |h(t; )| is the absolute value of impulse response of the signal, as alsoshown in Fig. 3. The impulse response (in complex baseband) is based on theSV (Saleh-Valenzuela) model which is given in general as [17],

hdiscr (t) =L 1

l=0

K 1

k=0

k,l exp( jk,l )(t T l k,l ), (12)

where k,l is the tap weight of the kth component (path) in the lth cluster,T l is the arrival time of the lth cluster and k,l is the delay of the kth MPCrelative to the lth cluster arrival time T l , see Fig. 4. The phases k,l areuniformly distributed, i.e. for a bandpass system, the phase is taken as auniformly random variable in the range from 0 to 2 [13].

The number of clusters L is an important parameter of the model and isassumed to be Poisson-distributed [13],

pdf L (L) =(L)L exp(L)

L!(13)

so that the mean L completely characterizes the distribution.The distribution of cluster arrival times are given by a Poisson process [13],

p(T L |T L 1) = l exp[l (T L T L 1)], l > 0 (14)where l is the cluster arrival rate.

The ray arrival times are modeled with mixtures of two Poisson processesas follows [13],

p( k,l | (k 1) ,l ) = 1exp[1( k,l (k 1) ,l )]+( 1)2exp[2( k,l (k 1) ,l )], k > 0 (15)where is the mixture probability, while 1 and 2 are the ray arrival rates.

The power delay prole (the mean power of different paths) is exponentialwithin each cluster [13],

E {|ak,l |2}= l1

l [(1 )1 + 2 + 1]exp( k,l / l ) (16)


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36 Part I

Delay

Amplitude

Clusters

1/

1/

Figure 4: Principle of Saleh-Valenzuela model.

where l is the integrated energy of the lth cluster and l is the intra-clusterdecay time constant.

Further, the mean (over cluster-shadowing) mean (over small-scale fading)energy (normalized to l ) of the lth cluster follows in general exponentialdecay [13],

10log(l ) = 10log(exp( T l / )) + M cluster (17)where M cluster is normally distributed random variable with standard devia-tion cluster around it.

Finally, the cluster decay rates are dened to depend linearly on the arrivaltime of the cluster [13],

l = k T l + o (18)

where k describes the increase of the decay constant with delay.In this model, particularly for some NLOS scenarios (office and industrial),the shape of the power delay prole (PDP) is dened differently, i.e. (on alog-linear scale) [13],

E {|a2k,l |}= (1 exp( k,l / rise )) exp( k,l / 1)

1 + rise 1

1 1 + rise (1 )

, (19)


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where the parameter describes the attenuation of the rst component, theparameter rise determines how fast the PDP increases to its local maximum,and 1 determines the decay at late times.

3.1.3 Delay dispersion

Delay dispersion is dened to occur when the channel impulse response lastsfor a nite amount of time or channel is frequency selective [18]. Delay dis-persion in multipath channels is characterized by two important parameters,mean excess delay and root mean square (rms) delay spread [18].

Excess delay is the relative delay of kth received multipath component ascompared to the rst arriving path, and is denoted as k [16]. Mean excessdelay is dened as the rst moment of the PDP given by [16],

m = P DP ( )d P DP ( )d (20)The rms delay spread is dened as the second central moment of the PDPgiven by [13],

rms =

P DP ( ) 2d

P DP ( )d

P DP ( )d

P DP ( )d

2

(21)

The delay spread depends on the distance, however, this effect is neglected inthe channel model for simplicity [13].

3.1.4 Small scale fading

The rapid uctuations of the received signal strength over very short traveldistances (of few wavelengths) or short time durations (on the order of sec-onds) is called small-scale fading [16]. In this model, the distribution of smallscale amplitudes is Nakagami [13],

pdf (x) =2

(m)(m

)m x2m 1 exp(m

x2), (22)

where m 1/ 2 is the Nakagami mfactor, ( m) is the gamma function,and is the mean-square value of the amplitude i.e., mean power. Themparameter is modeled as a lognormally distributed random variable, log-arithm of which has a mean m and a standard deviation m [13]. Both of these values can have a delay dependence [13],


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38 Part I

m ( ) = mo km (23)m = mo km (24)For the rst component of each cluster, the Nakagami factor is assumed to be

deterministic and independent of delay [13],

m = mo . (25)

The IEEE 802.15.4a channel model covers different office, residential andindustrial scenarios. The channel model which is used for performance eval-uation of RAKE receiver is for NLOS scenarios in industrial environments,referred to as CM8.

3.2 Measured UWB Channels

The UWB channel measurement campaign had been conducted together with

Lund university in MAX-Lab, Lund, Sweden, a medium-sized industrial en-vironment. The hall has a oor area of 94 70 m and a ceiling height of 10m. The hall contains many metallic objects, pipes, pumps and cylinders, seeFig. 5. The measurements were performed in the frequency domain using avector network analyzer (VNA) in conjunction with virtual antenna arrays.The frequency range measured was from 3 .1 to 8.0 GHz. 2001 frequencypoints were measured, resulting in a delay resolution of 0 .2 ns, and a max-imum delay of 408 ns corresponding to 122 m path delay. Omnidirectionalconical monopole antenna were used at transmitter and receiver, respectively.The use of virtual antenna arrays allows to create a virtual MIMO system of 7

7 antenna positions. The antenna separation was 50 mm, i.e., more than

half a wavelength at the minimum frequency in order to obtain spatially inde-pendent channels. Thus, for each measurement, 49 independent realizationsof the channel were measured over a local area resulting in a total of 49 16independent NLOS channel realizations. A total of sixteen peer-to-peer (P-P)NLOS positions were measured at four different locations with Tx-Rx sepa-rations of 2 m, 4 m, 8 m and 16 m at each location. A complete descriptionof the measurement setup and analysis of the model parameters can be foundin [2].


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Figure 5: An overview of the environment for UWB channel measurements.


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40 Part I

3.2.1 Measured Power Delay Proles

The measured transfer functions were transformed to the delay domain usinginverse Fourier transformation with a hanning window applied. The powerdelay proles (PDPs) calculated from the measurements on one antenna pairof the virtual array are called instantaneous PDPs, where the PDP from then-th transmitter to the m-th receiver is dened as [2],

P DP ( ,m,n ) = |h( ,m,n )|2 . (26)Average power delay proles (APDPs) were obtained from 49 instanta-

neous PDPs corresponding to different combinations of transmitter and re-ceiver positions on the virtual array for each of the measurement positions.Thus, APDPs were obtained as [2],

APDP ( ) =1

MN

M

m =1

N

n =1P DP ( ,m,n ). (27)

The analysis of APDPs for P-P NLOS measurements has shown the fol-lowing effects [2]:

1. For shorter distances (and a few of longer distance measurements), therst component is the strongest. A similar observation was also madein the measurement results presented in [3] for industrial environments.

2. The strong rst component is followed by a minimum in the APDPindicating that the multipath components arrive in clusters.

3. For many of the measurements at larger distances, the strongest com-ponent arrives 10

40 ns after the rst arriving component.

4. For longer distances, APDPs have soft-onset shape as also seen in [3].

5. It has been observed that some specic MPCs carry signicant powerfor larger Tx-Rx separations.

6. It is also observed that nearly all resolvable delay bins contain signicantenergy


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0 50 100 150 200 250 300 35030

25

20

15

10

5

0

Excess delay [ns]

A v e r a g e

P o w e r

[ d B ]

CM8

MG2

MG1

Figure 6: Average power delay proles of MG1, MG2 and CM8 averaged overall channel realizations.

3.2.2 Comparison of channel characteristics

The measured channel responses are divided into two groups based on theTx-Rx separation. The measurement group 1 (MG1) covers the distances inthe range of 2 to 8 m, while the measurement group 2 (MG2) consists of themeasured channel responses with Tx-Rx separation from 10 to 16 m. Thetotal number of independent realizations for MG1 and MG2 are 490 and 294,respectively.

For the performance comparison with the measured channels, 100 realiza-tions of IEEE 802.15.4a channel (CM8) are simulated using the channel modelparameters. A comparison of some characteristics averaged over different re-alizations of the channels is given in Table II. NP 10[dB] is the average numberof paths within 10 dB of the strongest path in the channel and NP {85%}denotes the average number of paths that contain 85 percent of the channelenergy. Moreover, average power delay proles (APDPs) of MG1, MG2 andCM8 averaged over all channel realizations are shown in Fig. 6.


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42 Part I

4 RAKE Receivers

4.1 Introduction

The amount of multipath energy that can be collected at the receiver and thereceiver complexity are commonly used to determine the performance androbustness of a wireless communication system [19]. The RAKE receiver isused in any kind of spread spectrum communication system to accumulatethe energy in the signicant multipath components [12]. The use of RAKEreceiver is also common in UWB systems to collect the rich multipath diversityavailable at the receiver.

A RAKE receiver consists of a bank of correlators, also called ngers, andeach nger is matched (synchronized) to a particular multipath component tocombine the received multipaths coherently [12]. In order to enable symbol-rate sampling in IR-UWB system, the received signal can be correlated witha symbol-length template signal, and the correlator output can be sampledonce per symbol [20]. The drawback of RAKE receiver in UWB systems isthat the number of multipath components that can be utilized in a typicalRAKE combiner is limited by power consumption issues, design complexity,and the channel estimation [21]. Secondly, each multipath undergoes a dif-

ferent channel in UWB systems, which causes distortion in the received pulseshape and makes the use of a single LOS path signal as a suboptimal template[22].

4.2 RAKE Types

A tapped-delay-line channel with K number of delays provides us with K replicas of the same transmitted signal at the receiver [12]. Hence, a receiverthat processes the received signal in an optimum manner will achieve theperformance of an equivalent K th order diversity system [12]. In practice,

only a subset of total resolved multipath components is used in the RAKEreceivers [21]. The RAKE types based on the number of multipath compo-nents used are given as follows [21],

All RAKE ( ARake): The RAKE receiver which combines all the K resolved multipath components is called all RAKE ( ARake ).

Selective RAKE ( SRake): The SRake receiver searches for the M best paths out of K resolved MPCs to use them as RAKE ngers.


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r (t)

dt

dt

dtvtemp (t 0)

vtemp (t 1)

vtemp (t K 1)

0

1

K 1

zi

Figure 7: A RAKE receiver structure for TH-IR UWB system.

Partial RAKE ( PRake ): The PRake receiver uses the M rst arriving paths out of K resolvable multipath components.4.3 RAKE Combining SchemesThe outputs of the correlators (ngers) are passed to the RAKE combiner.The RAKE receiver can use different combining schemes such as maximalratio combining (MRC) and equal gain combining (EGC) scheme.

4.3.1 Maximal Ratio Combining (MRC)

If maximal ratio combining (MRC) technique is used, the amplitudes of thereceived MPCs are estimated and used as weighing vector in each nger.The performance and optimality of the MRC consequently depend upon thereceivers knowledge of the channel [22]. Let = [ 0 , 1 , , K 1] be theRAKE combining weights which are different for different RAKE types.

In case of ARake, the combining weights are chosen equal to the fadingcoefficients of the channel, = [ 0 , 1 , , K 1], i.e., = (28)


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44 Part I

For SRake, if the set of indices of the M best fading coefficients withlargest amplitude is denoted by S , then the combining weights arechosen as follows [23],

= k , k S 0, k S (29)

For PRake, using the rst M multipath components, the weights of MRC combining are given by [23],

= k , k = 0 , , M 10, k = M, , K 1(30)

where M K .4.3.2 Equal Gain Combining (EGC)

In case of equal gain combining (EGC) scheme, all the tracked MPCs areweighted with their corresponding signs and combined [5]. Thus, EGC com-bining scheme only requires the phase of the fading channel [24]. In a carrier-less UWB system, determining the phase is even simpler because the phaseis either 0 or , to account for pulse inversion [25]. In a practical system,performing EGC will be simpler than MRC but there will be a performancetrade-off [24].

4.4 Analysis of RAKE ReceiverIn this subsection, it is assumed that we have perfect knowledge of the chan-nel and ARake receiver is used to receive the IR-UWB signal discussed inSection 2. Considering the ith symbol, the output of the kth nger of the

receiver can be written as

yi,k = r i (t)vtemp (t k )dt, (31)k = 0 , 1, , K 1 for each i

where vtemp (t) is assumed to be the normalized template signal matched tothe whole pulse sequence of one information symbol, i.e.,


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vtemp (t) = 1N f N f 1j =0 dj p(t jT f cj T c), (32)0 t N f T f

and r i (t) is the received signal for ith bit, which is given by

r i (t) =

E p

N f 1

j =0

dj bi g(t

jT f

cj T c) + n(t), (33)

0 t N f T f The analysis of RAKE receiver is discussed further based on the assumptionsof resolvable and nonresolvable paths.

4.4.1 Resolvable Paths

It is assumed that all paths are resolvable, that is, the minimum time between

any two paths is larger than the pulse width [12]. Let us dene the cross-correlation function between g(t) and p(t) as [22],

( ) = g(t) p(t )dt, (34)where ( ) = 0 if T p or T g . If there is a perfect match of thereceived signal with the reference signal, zero inter-frame and inter-symbolinterference, then the output of kth nger after summation over N f frames of ith symbol can be written in discrete time as

yi,k =

N f E pbi k + i,k , (35)

k = 0 , 1, , K 1 for each iwhere the last term i,k = n(t)vtemp (t k )dt is the noise at the outputof the correlator which has zero mean and variance 2n .

All ngers of the RAKE use a delayed version of the template signalvtemp (t), with a delay chosen from the vector = [ 0 . , K 1]

T , to match itto a specic multipath component. The outputs of all the ngers (correlators)for the ith symbol can be written together in vector notation as


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46 Part I

y i = N f E pbi + i , (36)where y i = [yi, 0 , , yi,K 1]

T , = [ 0 , , K 1]T , and the noise compo-

nent in vector notation is i = [i, 0 , , i,K 1]T .

Further, the diversity combining using the weight vector yields the de-cision statistic for ith bit as

zi = T y(i) (37)

= N f E pbiK 1

k=0

k k +K 1

k=0

k i,k , (38)

for ARake receiver with MRC combining, = , thus

zi = N f E pbiK 1

k=0

2k +K 1

k=0

k i,k . (39)

To determine the bit error probability (BEP) at the output of the RAKE,the approximate mean and variance of the decision statistic at the output of RAKE from (39) are evaluated as

E [zi ] =

N f E p

K 1

l=0

2k , (40)

Var( zi ) = 2nK 1

k=0

2k . (41)

In case of binary antipodal modulation, the approximate expression of BEP


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conditioned on a particular channel realization is given by [1], [22],

P e| ( , ) = Q E [zi |( , )]2Var( zi |( , ))= Q

N f E p K 1k=0 2k 22n

K 1k=0 2k

= Q

2N f E p

N o

K 1

k=0

2k

, (42)

here Q() is the standard function Q(x) = 1 / 2 x e t 2 / 2dt [22].4.4.2 Nonresolvable Paths

Assuming that the two paths may be less than a pulse width apart, the auto-correlation function of the wave is now evident [12]. The output of kth ngerfor ith symbol can be written as

yi,k = N f E pbi k + N f E pbiK 1

j =0 ,j = k

j R( j k ) + i,k dt, (43)k = 0 , 1, , K 1 for each i

where R( ) is the autocorrelation of the pulse p(t), dened as

R( ) = p(t) p(t )dt, (44)and i,k is the noise with zero mean and variance 2n

K 1j =0 j R( j k ) [12].

The outputs of all the ngers can be written together in vector form asy i = [yi, 0 , , yi,K 1]

T , then the decision statistic is formed using the diver-sity combining, i.e.,


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48 Part I

zi = T y(i)

= N f E pbiK 1

k=0

k k + N f E pbiK 1

k=0

K 1

j =0 ,j = k

k j R( j k )

+K 1

k=0

k i,k , (45)

as for ARake receiver with MRC combining, = , thus

zi = N f E pbiK 1

k=0

2k +K 1

k=0

K 1

j =0 ,j = k

k j R( j k )

+K 1

k=0

k i,k . (46)

The approximate mean and variance of the decision statistic at the output of RAKE are evaluated as [12],

E [zi ] = N f E pK 1

k=0

2k +K 1

k=0

K 1

j =0 ,j = k

k j R( j k ), (47)

Var( zi ) = 2nK 1

k=0

K 1

j =0

k j R( j k ). (48)The approximate expression of BEP conditioned on a particular channel

realization is given by [12],

P e | ( , ) = Q

2N f E p

N o

K 1k=0 2k +

K 1k=0

K 1j =0 ,j = k k j R( j

k )

2

K 1k=0

K 1j =0 k j R( j k )

,

(49)

where K 1k=0K 1j =0 ,j = k k j R( j k ) represents the additional term due topath correlations. Because the delays and amplitudes of the RAKE ngers

will be shifted when path correlations are nonzero, the development of (49) isa simplied analysis [12].


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5 Performance EvaluationIn order to compare the performance of RAKE receivers on the NLOS mul-tipath channels described in Section 4, the UWB system is simulated for anindoor industrial environment. The simulation parameters are as follows:

The binary antipodal modulation is used employing second derivativeof the Gaussian pulse. The pulse duration is kept 1 .02 ns and 0.625 ns over measured and

simulated channels, respectively.

The uncoded data rate of 4 Mbps is achieved with N f = 10 and T f = 25 ns. A frame time of T f = 25 ns results in inter-frame interference (IFI) asit is less than the delay spread introduced by the channels. The energy of channel impulse response is normalized as 2l = 1. The system is assumed to be synchronized with perfect knowledge of the channel characteristics.

For each E b/N o , performance over the channels is evaluated using 490,294 and 100 channel realizations of MG1, MG2 and CM8, respectively. The simulations are performed until 200 bit errors has been detected orat least 100 , 000 bits has been transmitted. It should be mentioned that these are uncoded bit-error-rates which canbe improved by using different coding techniques.In Fig. 8, the analytical expression of (40) and (41) are averaged for each

channel over corresponding number of available realizations. The output SNR(SN R o) is shown versus the number of MPCs captured by PRake and SRakeover MG1, MG2 and CM8. The results demonstrate an increase in SN R o withthe increase in number of ngers. It is observed that the increase in SNR withthe increase in number of captured MPCs is low for PRake as compared toSRake. Moreover, the dependence of SN R o over the channel models is alsoobserved. The gain in SN R o as the number of ngers increases is much higherin case of MG1 up to a certain number of ngers. It is noteworthy that thisis particularly depicted by PRake receiver. It is also observed that the SNRimprovement is signicant by increasing the number of ngers up to 100 for


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52 Part I

0 2 4 6 8 10 12 14 16 1810

4

103

102

101

100

Eb/No [dB]

B E R

SRake 10, MG1SRake 30, MG1SRake 50, MG1SRake 10, MG2SRake 30, MG2SRake 50, MG2SRake 10, CM8SRake 30, CM8SRake 50, CM8AWGN

Figure 10: Comparison of BER of different number of ngers of SRake withMRC combining on MG1, MG2 and CM8.

compared to the performance over MG2 and CM8. For MG1, the results of Fig. 10 demonstrate that a BER of 10 3 is achieved with only 10 ngers of the SRake with E b/N o = 16 dB. Moreover, the same BER of about 10 3 isachieved with E b/N o = 12 dB when 30 ngers of the SRake receiver are used.

When compared with PRake receiver, it is observed that SRake can achievemuch better performance. This is particularly true for channels having a shapeof the PDP similar to CM8. In case of MG2, the SRake performance usingonly 10 ngers is better than using 50 ngers of PRake. While for MG1,the results demonstrate that 10 ngers of SRake has almost the same BERas 50 ngers of the PRake. These results depict a signicant performanceimprovement associated with the SRake that uses only few ngers. However,it should be mentioned that the complexity of the SRake stems from the factthat it needs to search for the 10 ngers that carry the maximum energy.

The effect of RAKE combining scheme is analyzed in Fig. 11. The BERis compared using 30 ngers of both types of RAKE using EGC and MRCcombining schemes. The results show that the performance of MRC com-bining is better than EGC. It should be noted that, over measured channelswith PRake, there is performance improvement using MRC rather than EGC


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