An FPGA-Based Novel Digital PWM Control

3040 IEEE TRANSACTIONS ON INDUSTRIAL ELECTRONICS, VOL. 56, NO. 8, AUGUST 2009

An FPGA-Based Novel Digital PWM ControlScheme for BLDC Motor Drives

Anand Sathyan, Student Member, IEEE, Nikola Milivojevic, Student Member, IEEE,Young-Joo Lee, Student Member, IEEE, Mahesh Krishnamurthy, Member, IEEE, and

Ali Emadi, Senior Member, IEEE

Abstract—Development of advanced motor drives has yieldedincreases in efficiency and reliability. Residential and commercialappliances such as refrigerators and air conditioning systems useconventional motor drive technology. The machines found in theseapplications are characterized by low efficiency and high main-tenance. A brushless dc (BLDC) motor drive is characterized byhigher efficiency, lower maintenance, and higher cost. In a marketdriven by profit margins, the appliance industry is reluctant toreplace the conventional motor drives with the advanced motordrives (BLDC) due to their higher cost. Therefore, it is necessaryto have a low-cost but effective BLDC motor controller. This paperlays the groundwork for the development of a new low-cost ICfor control of BLDC motors. A simple novel digital pulsewidth-modulation control has been implemented for a trapezoidal BLDCmotor drive system. Due to the simplistic nature of this control,it has the potential to be implemented in a low-cost application-specific integrated circuit. The novel controller is modeled andverified using simulations. Experimental verification is carriedout using field-programmable gate arrays to validate the claimspresented.

Index Terms—Brushless dc (BLDC) motor drives, converters,field-programmable gate arrays (FPGAs), inverters, motion con-trol, pulsewidth modulation (PWM).

I. INTRODUCTION

AN ELECTRIC motor is defined as a transducer that con-verts electrical energy into mechanical energy. Electrical

motors are an integral part of industrial plants with no less than5 billion motors built worldwide every year. Residential andcommercial applications mostly use conventional motor drivetechnologies. Typically, machines found in these appliances aresingle-phase induction motors or brushed dc machines whichare characterized by low efficiency and high maintenance,respectively [1], [2]. Single-phase induction motors are lessefficient because of the ohmic loss in the rotor and due tothe phase angle displacement between the stator current andback electromotive force (EMF). In the case of dc machines,they require more maintenance due to the presence of brushes.

Manuscript received November 17, 2008; revised April 23, 2009. Firstpublished May 15, 2009; current version published July 24, 2009.

A. Sathyan was with the Department of Electrical and Computer Engineer-ing, Illinois Institute of Technology, Chicago, IL 60616 USA. He is now withChrysler Corporation, Detroit, MI 48234 USA.

N. Milivojevic, Y.-J. Lee, M. Krishnamurthy, and A. Emadi are with theDepartment of Electrical and Computer Engineering, Illinois Institute of Tech-nology, Chicago, IL 60616 USA (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TIE.2009.2022067

Replacing these inefficient motors with more efficient brushlessdc (BLDC) motors will result in substantial energy savings.

A permanent-magnet synchronous machine has several ad-vantages over other machine types. Most notably (comparedto other dc motors), they require lower maintenance due tothe elimination of the mechanical commutator. They also havehigh power density. Compared to induction machines, they havelower inertia, allowing for faster dynamic response to referencecommands. In addition, they are more efficient due to thepermanent magnets, which results in significantly lower rotorlosses. The major disadvantage with permanent-magnet motorsis their higher cost and relatively greater degree of complexityintroduced by the power electronic converter used to drive them[2], [3].

Hysteresis current control and pulsewidth modulation(PWM) control coupled with continuous control theory haveproduced the most widely used BLDC motor control techniques[4]. Hysteresis current control is essential toward achievingadequate servo performance, namely, instantaneous torque con-trol, yielding faster speed response compared to PWM control.For most applications, proportional-integral (PI) current andspeed compensators are sufficient to establish a well-regulatedspeed/torque controller. In other cases, state feedback control isneeded to achieve a more precise control of the BLDC motor.Classic control theory and linear system theory are well un-derstood but are highly complex and require extensive controlsystems knowledge to develop a well-designed controller [5].Discrete control theory allows for such controllers to be dig-itally implemented with microcontrollers, microprocessors, ordigital signal processors (DSPs). Digitizing analog controllersserves to add complexity to the overall design procedure [6],[7]. Moreover, digital implementation of a continuous controltechnique does not produce a digital controller. Instead, whatresults is a digitally implemented control technique, whichtypically results in high controller cost.

In this paper, a novel digital PWM controller has beenproposed for a BLDC motor. This controller treats the BLDCmotor as a digital system. The BLDC system is only allowed tooperate at a low duty (DL) or a high duty (DH). Speed regula-tion is achieved by alternating between low duty and high duty,which makes the concept of the controller extremely simple fordesign and implementation. In addition, this technique utilizesonly one current sensor in the dc link. This helps reduce thecost and complexity of motor control hardware and, in turn, canboost the demand of BLDC motors for commercial mass pro-duction applications. For the controller presented in this paper,

0278-0046/$26.00 © 2009 IEEE

SATHYAN et al.: FPGA-BASED NOVEL DIGITAL PWM CONTROL SCHEME FOR BLDC MOTOR DRIVES 3041

Fig. 1. Typical inverter drive system for a BLDC motor.

Fig. 2. Back EMF and phase current variation with rotor electrical angle.

characteristic equations of a BLDC machine have been used toderive the design procedure, which involves simple first-ordernonhomogenous differential equations. During steady-state op-eration, the design procedure is reduced to a few simple alge-braic equations. Computer simulations and experimental resultsare presented for proof of concept.

II. BRUSHLESS DC MOTOR DRIVE STRATEGIES

For a three-phase BLDC application, the most commontopology used is a three-phase buck-derived converter or athree-phase inverter bridge [8], [9]. The typical inverter drivesystem for a BLDC motor is shown in Fig. 1.

As shown in Fig. 1, the output stage consists of a three-phase inverter composed of switches that could be MOSFETsor insulated-gate bipolar transistors (IGBTs). If IGBTs areused, antiparallel diodes need to be connected across them forcarrying reverse currents, while MOSFETs use body diodes.MOSFETs give lower turn-off switching loss and usually lowerdiode forward drop, but that advantage may be offset by higherON-state voltage drop and turn-on switching/diode reverse re-covery loss than IGBTs. Typical waveforms for a three-phaseBLDC motor with trapezoidal flux distribution are shown inFig. 2 [10]. Approximately, the back EMF induced per phase of

Fig. 3. Sensor versus drive timing.

the motor winding is constant for 120◦, before and after whichit changes linearly with rotor angle.

In order to get constant output power and, consequently, con-stant output torque, current is driven through a motor windingduring the flat portion of the back-EMF waveform. Only twoswitches are turned on at a time, one in a high side and the otherin a low side. Thus, for a star-connected motor winding, twophases are connected in series across the dc bus, while the thirdwinding is open. The switches in Fig. 1 are operated such thateach phase carries current only during the 120◦ period whenthe back EMF is constant. Thus, there is a commutation eventbetween phases for every 60◦ electrical, as shown in Fig. 2.Effectively, it means that there is a current transition every60◦. Appropriate commutation therefore requires knowledge ofrotor position, which can be directly detected using positionsensors or estimated in sensorless manner by monitoring backEMF in the open phase [11].

Unlike a brushed dc motor, the commutation of a BLDCmotor is controlled electronically [12]. To rotate the BLDCmotor, the stator windings should be energized in a sequence.It is important to know the rotor position in order to followthe proper energizing sequence. For the present system, rotorposition is sensed using Hall effect sensors embedded into thestator. By reading the Hall effect sensors, a 3-b code can beobtained, with values ranging from one to six. Each code valuerepresents a sector in which the rotor is presently located. Eachcode value therefore gives us information on which windingsneed to be excited to turn the rotor [13], [14]. State “0” and “7”are considered invalid states for Hall effect sensors. A timingdiagram showing the relationship between the sensor outputsand the required motor drive voltages is shown in Fig. 3.

The numbers at the top of Fig. 3 correspond to the currentphases shown in Fig. 4. It is apparent from Fig. 3 that thethree sensor outputs overlap in such a way as to create sixunique 3-b codes corresponding to each of the drive phases. Thenumbers shown around the periphery of the motor diagram inFig. 4 represent the sensor position code. The north pole of the


Fig. 4. BLDC motor showing the commutation sequence.

TABLE ICLOCKWISE SENSOR AND DRIVE BITS BY PHASE ORDER

rotor points to the code that is output at that rotor position. Thenumbers are the sensor logic levels where the most significantbit is sensor C and the least significant bit is sensor A [15]. Theinput sensor state and the corresponding drive state required forcommutation can be put in the form of a state table as shown inTable I.

III. DIGITAL PWM CONTROL OF BLDC DRIVES

Speed control in a BLDC involves changing the applied volt-age across the motor phases [16], [17]. This can be done usinga sensored method based on the concept of pulse amplitudemodulation, PWM, or hysteresis control [18], [19]. Anothermethod involves control using sensorless techniques. In the lasttwo decades, many sensorless drive solutions have been offeredto eliminate the costly and fragile position sensor for BLDCmotors with trapezoidal back EMFs [20], [21].

A common control algorithm for a permanent-magnet BLDCmotor is PWM current control. It is based on the assumption oflinear relationship between the phase current and the torque,similar to that in a brushed dc motor. Therefore, by adjustingphase current, the electromagnetic torque can be controlledto meet the requirement. The general structure of a currentcontroller for a BLDC motor is shown in Fig. 5. Instantaneouscurrent in the motor is regulated in each phase by a hysteresisregulator, which maintains the current within adjustable limits.The rotor position information is sensed to enable commutationlogic, which has six outputs to control the upper and lowerphase leg power switches [22]. The current reference is de-termined by a PI regulator, which maintains the rotor averagespeed constant.

Fig. 5. Conventional PWM current control.

Fig. 6. Flowchart describing the novel digital control.

This paper presents the design, simulation, and experimen-tal verification of a novel constant-frequency digital PWMcontroller which has been designed for a BLDC motor drivesystem. In essence, the controller treats the BLDC motor asa digital system. The concept of this digital controller is verysimple. Speed regulation is achieved by using two levels ofduty cycles—a high duty (DH) and a low duty (DL). The rulesof the digital controller can be explained using the flowchartshown in Fig. 6.

Unlike a hysteresis current controller, a PWM control doesnot have an inherent current control capability. Hence, a currentlimiter has to be introduced. Current hysteresis digital con-trollers have been reported for a BLDC motor in [4]. This paperpresents a controller with no need of any state observer. Fig. 7shows the proposed digital controller. Fig. 8 shows the completeblock diagram of the motor drive system.

A proportional controller provides the reference for thecurrent limit. The current is made to stay within a maximumand minimum limit. The maximum value of Ilimit is 1.5 times


Fig. 7. Proposed digital control.

Fig. 8. Block diagram for digital PWM control for a BLDC motor drivesystem.

the rated motor current. This is because motors can handle1.5 times the rated current for a short duration of time. Theminimum value of Ilimit decides the steady-state error. For avalue equal to zero, a large steady-state error is observed inthe simulation. The minimum value of Ilimit is defined as theratio of a percentage (1%) of the rated torque to the torqueconstant.

The proportional constant K for a desired speed ripple canbe calculated as follows. In steady state, Δω ≤ |ωerr ∗ 2|. In theworst case, Δω = |ωerr ∗ 2|. For the desired speed ripple Δω,a constant Kset can be defined as

Kset =Δω

ωrated. (1)

Taking the maximum value of the speed ripple

Kset =ωerr ∗ 2ωrated

. (2)

Fig. 9. Gate switching waveforms.

As long as

ωactual < ω∗ − Δω, Ilimit = Ilimit,max. (3)

In addition, Ilimit α ωerror

Ilimit = K ∗ ωerror. (4)

By using (1)–(3) in (4), it can be shown that

K =2 ∗ Ilimit,max

Kset ∗ ωrated. (5)

In this control strategy, both the high- and low-side switchesare switched simultaneously. Both high- and low-side diodesconduct. The waveforms for this type of switching are shownin Fig. 9.


Fig. 10. Steady-state speed response.

IV. CONTROLLER DESIGN

The value of the duty ratio D can be obtained from theelectrical and mechanical equations. The value of D can beexpressed as a function of the motor parameters.

From the torque equation, we have

Tem = Jdω

dt+ bω + TL (6)

where Tem, ω(t), b, J , and TL denote developed electromag-netic torque, rotor angular velocity, viscous friction constant,rotor moment of inertia, and load torque, respectively

KtI = Jdω

dt+ bω + TL (7)

where Kt = torque constant and I = average current. Atsteady state, (7) can be written in terms of steady-state angularvelocity ωss as

I(ωss) =1

Kt(bωss + TL). (8)

At steady state, the time phase voltage equations can be writtenin terms of phase voltage Van, phase current I , winding resis-tance, and velocity constant Ke as

Van = IRa + Keωss. (9)

Substituting the value of the steady-state current from (10) anddefining the phase voltage in terms of dc-link voltage Vdc andduty D, we get

DVdc =(TL + ωssb)

KtR + Keωss (10)

where ωss denotes the steady-state angular velocity.Therefore, the duty ratio can be expressed in terms of the

motor parameters as

D =1

Vdc

[(TL + ωssb)

KtR + Keωss

]. (11)

A. Speed Ripple Calculation

Steady-state error is a function of the speed samples. It istherefore necessary to find out the effect of sampling time (TP )on the speed ripple. Fig. 10 shows the speed response at steadystate.

The maximum deviation from the reference speed (ω∗) dueto the application of high duty DH is denoted by ΔωH , andthe maximum deviation from the reference speed due to theapplication of a low duty DL is denoted by ΔωL. The speedresponse can be expressed as

ω(t) =Tem − TL

b+[ω∗ −

(Tem − TL

b

)]e−

bJ t. (12)

From Fig. 10, at time t = t2 + TP

ΔωH =Tem − TL

b+[ω∗−

(Tem − TL

b

)]e−

bJ t − ω∗. (13)

At time t = t1 + TP

ΔωL = −(

Tem − TL

b

)+[ω∗ −

(Tem − TL

b

)]e−

bJ t + ω∗.

(14)

The speed ripple can be calculated as

Δω = ΔωH + ΔωL (15)

Δω =ΔTem

b

(1 − e−

bJ TP

)(16)

TP = −τm ln

(1 − Δω

ΔTemb

)(17)

where τm = mechanical time constant

TP = −τm ln(

1 − Δω

ωNL,max

). (18)

From (18), for a desired speed ripple, the sampling time canbe calculated. ωNL,max is the maximum speed under no-loadconditions.

B. Steady-State Analysis

For the purpose of analysis, the proposed digital controllerwas considered equivalent to a proportional controller with highgain and saturation. This analysis was aimed at determining ifthe actual motor speed reaches the reference speed at steadystate. The transfer function for a BLDC motor is shown as

ω(s)V (s)

=Kt

JLa

s2 + (JRa+BLa)JLa

s + (BRa+KtKe)JLa

(19)

where V = DVin, with D being the duty cycle.


Fig. 11. Closed-loop control of the BLDC motor with the novel digital PWMcontroller.

The overall block diagram with the closed-loop control usingthe proposed novel digital PWM controller is shown in Fig. 11.The tolerated error decides the sampling in the speed loop.Based on the speed error, a high duty or low duty is selectedby the Gw block. The output of the Gw block is Dw, whichcan be Dh or Dl. The duty, along with the information aboutthe actual motor current, is the input to the Gi block, basedon which it selects a high duty, low duty, or a skip state. Theoutput of the Gi block is Di, which can be Dh, Dl, or D0. Theminimum current (imin) is used to compensate for the R andB terms so that the steady-state error is reduced. The outputof the Gi block is a PWM voltage signal which is given as theinput to the BLDC motor.

Simulations were carried out for a maximum variation fromthe reference speed of about 10 r/min. The necessary frequencyof sampling in the speed loop for the desired speed ripple can becalculated from (18). For example, a speed ripple of 10 r/minrequires a minimum sampling time of 916 μs. This was con-firmed. From measurements, this sampling time resulted in aspeed ripple of about 6 r/min.

V. DESCRIPTION OF EXPERIMENTAL SETUP

An experimental setup was constructed in order to implementand further validate the simulation results of the proposedtechnique. The following section gives a brief description ofthe requirements and final designs for the experimental setup.The experimental setup is shown in Fig. 12. Parameters of thetest machine are given in Table II.

The inverter was built using Pwrx IGBT modules, rated at50 A, 600 V, with a switching frequency of 6 kHz. Gate signalswere generated in the field-programmable gate array (FPGA)controller and buffered using an inverted Schmidt circuit. Forthe mechanical load, constant load torque was provided by amagnetic brake. Current protection was realized by adding adc-link current transducer. This sensor was used to sense thedc-link current and keep it below a maximum value in order tolimit inrush current. If the value of the dc-link current exceeds

Fig. 12. Final experimental setup.

TABLE IIDATA SHEET FOR BLDC MOTOR FROM POLY-SCIENTIFIC (BN42-EU-02)

the predefined value, the current regulator sends a signal thatautomatically sets the duty cycle to zero. Therefore, the currentis not allowed to rise further, until the value drops below thelimited value again.

An FPGA platform used for controlling the BLDC machineis Spartan 3 family, from Xilinx. Reference speed value was setdigitally, and a speed loop was used to compare the actual speedand the reference speed and based on error to determine the dutycycle for the next period. The actual speed was easily calculatedas a time between two Hall effect signals. The schematic of thecontroller simulated in the FPGA is shown in Fig. 13.

Inputs to the FPGA device include three Hall effect sensorsand reference speed information, while output signals are trig-gers for switching on and off IGBTs. In addition, the dc-linkcurrent was measured using LEM transducer, and analog signalis sent first to A/D converter, and then to FPGA, for currentprotection. In order to show the speed error, 8-b D/A (digital-to-analog) converter was used.

As shown in Figs. 12 and 13, the logic used to control theBLDC motor is very simple with few components. As a base forcomparison, when using conventional control schemes with aDSP, the cost may be as much as 20 dollars [23]. The novel dig-ital PWM control scheme uses 3 adders, 3 comparators, 5 flip-flops, 30 logic elements, and 3 counters. In mass production,an analog IC implemented using the aforementioned elementswould cost less than a dollar, which would make the BLDC


Fig. 13. Block diagram showing operations and functions implemented inFPGA device.

Fig. 14. Simulated duty, speed, and current response for a commanded speedof 2500 r/min for full-load operation.

motors affordable for manufacturers to use in their products.In addition, in conventional schemes, three current sensors arerequired [1], whereas this control scheme requires only onecurrent sensor at the dc link. Furthermore, this scheme can beeffectively implemented on an FPGA as opposed to DSPs thatare used for the implementation of complex control schemes.This makes the present technique significantly cost effective.

VI. SIMULATION RESULTS AND

EXPERIMENTAL VERIFICATION

For the verification of the control scheme, several operatingconditions were selected. Fig. 14 shows the simulated responsefor a reference speed of 2500 r/min under full-load conditions.This figure includes the variation of speed versus commandedreference, duty ratio assigned, and the resultant current.

Figs. 15–21 show the experimental results obtained for vari-ous operating conditions, and Table III summarizes these resultsfor various load conditions and reference speeds. It can be notedthat the average error is well below 5%. In addition, the absolute

Fig. 15. Experimental results for a reference speed of 2500 r/min under no-load condition.

Fig. 16. Experimental results for a reference speed of 2500 r/min. Load is30% of rated value.


Fig. 17. Experimental results for a reference speed of 1500 r/min under no-load condition.

Fig. 18. Experimental results for a reference speed of 1500 r/min. Load is30% of rated value.

Fig. 19. Experimental results for a reference speed of 2100 r/min underfull load.

Fig. 20. Speed response for change in load torque and for a reference speedof 2000 r/min.

value of the maximum speed ripple stays well within 30 r/min,or in other words, the maximum error stays below 5%. Hence,this control technique is well suited for applications where high


Fig. 21. Experimental results for a change in reference speed from 2200 to1300 r/min under no-load condition.

TABLE IIISUMMARY OF THE EXPERIMENTAL RESULTS

precision is not required. Some of the applications where thiscontrol technique can be implemented are washing machines,dryers, machine tools, pumps, refrigerators, etc.

In order to test the stability of the proposed technique, theresponse of the system was observed for a sudden changein operating conditions (Figs. 20 and 21). Fig. 20 shows theresponse for a step change in load torque, which was variedfrom 20% to 80%. Fig. 21 shows the response of the system fora step change in commanded speed. For each of these changes,the speed response was observed to settle within 5% of thecommanded value. Thus, this simple low-cost stable controlstrategy proves its effectiveness for use in applications whichare subject to disturbances.

VII. CONCLUSION

A new digital control concept for BLDC machines hasbeen introduced and experimentally verified. The aim of thispaper is to develop a low-cost controller for applications whereinefficient single-phase induction motors are used. Due to thesimplistic nature of this control, it has the potential to be imple-mented in a low-cost application-specific integrated circuit. Thecontroller exploits the characteristic of most electromechanicalsystems. Since electrical time constants are much faster relativeto the mechanical time constants, speed regulation with anacceptable amount of ripple is possible by rapid switching ofstates. Furthermore, this control strategy does not require astate observer. Under dynamic load conditions, the proposedcontroller was found to be capable of regulating speed withoutthe use of an observer. This results in a considerable reductionof size and the cost of the system.

REFERENCES

[1] A. Emadi, Handbook of Automotive Power Electronics and Motor Drives.Boca Raton, FL: CRC Press, 2005.

[2] A. A. Aboulnaga, P. C. Desai, F. Rodriguez, T. R. Cooke, and A. Emadi,“A novel low-cost high-performance single-phase adjustable-speed motordrive using PM brushless DC machines: IIT’s design for 2003 futureenergy challenge,” in Proc. 19th Annu. IEEE Appl. Power Electron. Conf.,Anaheim, CA, Feb. 2004, pp. 1595–1603.

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Anand Sathyan (S’04) received the B.S. degree(with distinction) in electrical engineering fromAmrita Institute of Technology, Coimbatore, India,in 1998, the M.S. degree (with distinction) in electri-cal engineering from Coimbatore Institute of Tech-nology, Coimbatore, in 2001, and the Ph.D. degreefrom the Department of Electrical and Computer En-gineering, Illinois Institute of Technology, Chicago.

From August 2001 to August 2004, he was aFaculty Member with the Department of ElectricalEngineering, Amrita Institute of Technology. From

August 2004 to December 2008, he was a Research Assistant with the GraingerPower Electronics and Motor Drives Laboratory, Illinois Institute of Technol-ogy. Since April 2009, he has been with Chrysler Corporation, Detroit, MI,working on hybrid and plug-in hybrid vehicles. His research interests includepower electronics, motor drives, and renewable energy systems.

Nikola Milivojevic (S’07) received the B.S. andM.S. degrees from the University of Belgrade,Belgrade, Serbia. He is currently working towardthe Ph.D. degree in the Department of Electrical andComputer Engineering, Illinois Institute of Technol-ogy, Chicago.

He has several years of experience in the field ofrenewable energy sources as well as energy storagesystems. He was engaged in research and develop-ment projects on control of ac/dc electric drives. Hiscurrent research interests include maximum power

point tracking techniques for renewable energy systems as well as low-costelectric drive control in generating applications.

Mr. Milivojevic was one of the student team leaders that won “InternationalFuture Energy Challenge 2005” with low-cost adjustable-speed motor drivesolution.

Young-Joo Lee (S’07) received the B.S. degreein electrical engineering from Korea University ofTechnology and Education, Cheonan, Korea, in 1996and the M.S. degree from Gwang-Woon University,Seoul, Korea, in 2003. Since 2006, he has beenworking toward the Ph.D. degree in the Departmentof Electrical and Computer Engineering, Illinois In-stitute of Technology, Chicago. His Ph.D. researchfocuses on integrated bidirectional converter forplug-in hybrid electric vehicles.

In 1995, he joined SunStar R&C, Incheon, Korea,where he specialized in industrial sewing machines and controllers, and motorsand controllers for industrial sewing machines. He then joined Genoray Com-pany, Ltd., Seongnam, Korea. He has over ten years of industrial experienceand has developed several commercial system controllers for sewing machinesand medical X-ray fluoroscopy equipment, which require control of brushlessdc motors, induction and stepper motors, high-frequency full-bridge converters,X-ray electron tubes, and other electric–pneumatic actuators.

Mahesh Krishnamurthy (S’02–M’08) received theM.S. degree from the University of Missouri, Rolla,in 2004 and the Ph.D. degree from the University ofTexas, Arlington, in 2008.

Since 2008, he has been with the Department ofElectrical and Computer Engineering, Illinois Insti-tute of Technology, Chicago, as an Assistant Profes-sor. His current research interests include numericalanalysis of energy conversion in electric machines;design, analysis, and control of power electronicsand drives in renewable energy; and hybrid and plug-

in hybrid vehicular applications.

Ali Emadi (S’98–M’00–SM’03) received the B.S.and M.S. degrees (with highest distinction) in elec-trical engineering from Sharif University of Technol-ogy, Tehran, Iran, and the Ph.D. degree in electricalengineering from Texas A&M University, CollegeStation.

He is the Harris Perlstein Endowed Chair Profes-sor of Electrical Engineering and the Director of theElectric Power and Power Electronics Center andGrainger Laboratories, Illinois Institute of Technol-ogy, Chicago, where he has established research and

teaching facilities in power electronics, motor drives, and vehicular powersystems. He has been named as a Chicago Matters Global Visionary in 2009.He is the principal author/coauthor of over 220 journals and conference papersas well as six books.

Dr. Emadi was the founding General Chair of the First IEEE Vehicle Powerand Propulsion Conference (VPPC’05) and is currently the Chair of the IEEEVPPC Steering Committee.

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An FPGA-Based Novel Digital PWM Control