Absorptive Low-Pass Cables: State of the Art and an Outlook to the Future

IEEE TRANSACTIONS ON ELECTROMAGNETIC COMPATIBILITY, VOL. EMC-28, NO. 1, FEBRUARY 1986

Absorptive Low-Pass Cables: State of the Art andan Outlook to the Future

FERDY MAYER, FELLOW, IEEE

Abstract-A new concept, "limited bandwidth of electrical intercon-nections," is presented and compared with the current "all-pass powerand communications networks." Low-pass performance, with an ad-justed cutoff frequency and cutoff slope, can be achieved in a new cost-effective way (compared to classical filters) through the introduction ofelectrical and magnetic losses in electrical lines and cables, and can befurther enhanced by "interfacial resonance" effects and by a "simulatedskin-effect coupled line." Typical design criteria, performance, andapplications are presented, along with their industrial and economicsignificance for the EMC-system design of the future.Key Words-Cables, absorptive, low-pass, distributed filter, measure-

ments.Index Code-D5f, D7f, E7k.

I. INTRODUCTIONT HE NEED for EMC protection in electronic systems and

the possibility of building low-transfer impedance incoaxial and other cables have drawn attention to the use oflow-pass cables in military and aerospace applications (U.S.Specification MIL-C-85485).The current state of the art of low-pass cables is presented.

Electrical models and implementation of such cables, basedupon absorptive dielectromagnetics, etc., are described.New techniques that allow the control of frequency per-

formance, by a "simulated skin effect," as well as theimplementation of low-pass cables in networks, with addi-tional impedance mismatch low-pass and bandpass absorption,are presented. Significant use of such cables is foreseen forhigh-volume markets, such as the automated plant andcomputerized office of the future.

II. Low-PASS CABLES WITH A SINGLE LINEA. Description and Characteristics

Electrical wires and cables, using the principle of magnetic(and dielectric) absorption showing distributed low-pass filtereffects, first appeared in 1957 [1].

Application of this basic concept developed rapidly. Solidabsorptive low-pass filters, described in scientific and techni-cal literature since the 1960's [2]-[5], [22], were developedfor various uses. For the application to flexible wires andcables, magnetic and dielectric absorption and a number ofother physical effects have been used to realize, or to simulate,frequency-sensitive loss effects [6]-[8], [10], [11]. The use of

Manuscript received September 30, 1983; revised October 21, 1985. Thispaper was originally presented at the 1983 IEEE EMC InternationalSymposium, Washington, DC, August 23-25, 1983.The author is with LEAD, 43 rue de 11 Novembre 1918, 94700 Maisons-

Alfort, France. (1) 48-93-44-44.IEEE Log Number 8405862.

Ski e ld

Cov) uctor0E I

Thu,VIL\S Iat"oL agj e r- 1,Clio rm

Fig. 1. Low-pass coaxial line.

frequency-selective losses is limited to applications where aclear separation between passband and stopband exists [7],[9]; due to absorption, such filters show unconditionalperformance (independent of external influences such astermination mismatch) over a broad frequency range.

Fig. 1 shows the basic structure of an absorptive "low-pass" cable: a layer of absorptive flexible composite sur-rounds the conductor, covered either internally or externallyby a thin layer of insulating material, which gives the wire itsdielectric strength.The wire, as described, needs the presence of a ground

electrode to achieve a four-pole structure with a (frequency-sensitive) attenuation. A predetermined frequency responseneeds a fixed capacitance to ground. A maximum of attenua-tion is achieved when the interelectrode space is filled with theabsorptive composite, e.g., in a coaxial structure with ametallic outer braid, as shown.

Since the 1960's, such cables have been on the market,particularly for military, aeronautical, and space applications.The recently issued U.S. Specification MIL-C-85485 repre-sents the first standard concerning low-pass cables.

Fig. 2 shows the performance of such cables, as per MIL-C-85485 (since the original preparation of this paper, the finalspecification MIL-C-85485A has been issued as a replace-ment, and generally shows lesser attenuation performance),using known absorptive composites [1]. It is characterized,first, by its attenuation per unit length as a function offrequency.The frequency range under 10 MHz is referred to as

passband, the range from 1 to 18 GHz (not shown) asstopband. It is interesting to note that attenuation inside thislatter range is difficult to measure, the theoretical calculation(see hereunder) showing values of, and over, several thousand

0018-9375/86/0200-0007$01.00 1986 IEEE

7


*0

0

W..

-

0.2

9-

90

00

70

0

5a

40

3^

20

'O

-.0

l:I054 3S/0-20)1

1

_

b I_ a 0 o

FREQUENCY (MElz )

a

0i

C)C-

I-._1iI!2 - C=nszant I

.I.

Spec. L istIlSt(f)

1i0. _ _

O-

50. 0 _ o:isnal *'

CI CI

l\""\ "\10.0 __

\I,

'!:22Sc a>/:-0-2cl:1

1. .

Fig. 2. Attenuation and transfer impedance-MIL-C-85485.

decibels per meter. Between passband and stopband there is a"transition band" where one can define a cutoff frequency(where attenuation equals 3 dB/m).The coaxial cable of Fig. 1 has a shield on the outside. The

efficiency of this shield may be defined by its basic transferimpedance (equivalent cable without losses), or by an "effec-tive" transfer impedance, i.e., the basic shielding effectcombined with the cable attenuation-as measured with a realabsorptive cable.

Such an "effective" transfer impedance Zt illustrates thatthe coupling through the shield decreases when frequencyincreases. This apparent shield improvement is of use todescribe the reduced susceptibility against electromagneticemissions from the outside (such as EMP), or, inversely, thesuppression of leakage of signals to the outside (such asTEMPEST). The MIL-C-85485A standard has been drawn upbased on cables produced in the United States. These cablesare currently used in advanced military applications.A series of new composites developed recently makes

possible improved attenuation performance (higher attenua-tions at low frequencies) with a typical cutoff frequency (3 dB/m) of 25 MHz. Fig. 3 shows this performance, compared to atypical MIL-C-85485 cable implementation.

These types of cables present a number of interestingcharacteristics (compared to classical low-pass filters) becausethey are based upon distributed loss effects rather than lumpedreactive effects.

a) Their attenuation performance is unconditional, i.e.,independence of the attenuation from matching with inter-

.1

INSERTION LOSS

110. 0

100. 0

1, 9,1 El

W. 7 p

S E . D

l-

,, 5 El.. [

'T- 4 ED. EsD

L-J 3 El El

II 2 t:. El

I D.

B I.10

FREQUENCY [MHz]

Date Of Test: 9/0/82

100 ieee

Written: 5/13/82

Sariple Length: .98 Feet.

Fig. 3. Attenuation of various low-pass lines.

faces, absence of problems with interface resonances, ofcomponent resonances and eigenresonances, as they areencountered in lumped-component filters.

b) Their attenuation extends up to and above 3 decades offrequency range (for example, 10 MHz to 20 GHz) [8].

c) Their attenuation slope versus frequency is relativelysteep above the cutoff frequency, the attenuation increasingapproximately proportionally to the 3/2 power of frequency inthe transition range.As the attenuation is proportional to the length, it is easy to

design a specific filter line (with a given cutoff frequency).Fig. 4 indicates experimental data, for various lengths of cableof Fig. 1.

B. ModelingPropagation modes in such a structure are in general of the

TM type, but the TEM approximation is valid at lowerfrequencies, i.e., when the transverse dimensions of thestructure are smaller than the actual wavelength in thecomposite.

In this case, the Kirchhoff Theory is valid, starting from asimple distributed-constant scheme (Fig. 5). Z represents theglobal series impedance of the line, with its internal conductor(1), the external conductor (4) and their skin effects (RFapproximation, but valid too for artificial skin-effect layers(7), for frequencies above one skin-effect layer at the surface).The external impedances 2 and 3 of the same conductorsrepresent the magnetic contribution of the volume of theabsorptive composite and of the insulation. Y represents theglobal parallel admittance of the line, with the two complexcapacitors in series, due to the absorptive (magneto-) dielectriclayer and the insulating layer.

Their ratio determines the complex characteristic impe-dance of the line, and their product, the propagation constant,with its real term defining the attenuation.The practical calculation is easily programmable, with the

introduction of measured data of the complex permittivity andpermeability of the absorptive composite and the insulator.

8

I

MAYER: ABSORPTIVE LOW-PASS CABLES

5 0

E 1001,

F-

c-50

10 30FREQUENCY MHz

100 300 1000

Fig. 4. Attenuation as function of length.

DL'ameter D1 D1a DI D.,

ondlJ)itv;ty (f elPern'Ltllvlty EaI 3pernea6b'Ity 1 1

zc=vpg Q

Y'=o+jF

L,1(,_

Z_rA Z ;tc Z_t_nA Zernl 1

Z;nt"^^l)- +1 V/F } Appro>x. RF W

Elco .aD

L

IEEE TRANSACTIONS ON ELLECTROMAGNETIC COMPATIBILITY, VOL. EMC-28, NO. 1, FEBRUARY 1986

---

1.`.Il

--

--

-

- .

F-

Il Hz l0 (00 1H-L 10

0o3

TABLE IIATTENUATION VERSUS INSULATOR THICKNESS

CABLE ATTENUATION IN dB/m

frequency

insulatiothickness

1 MHz

eD-0,2 0,070 0,87 25,9 594=

DI = 2,S e = 0,35D2 = 4,l D3 = 4,8

lnm

e = 0,7D3 = 5,5

0,040

0,020

insulation polyethylene .= 2,25 (1-jO)

TABLE IIIATTENUATION VERSUS COMPOSITE

Fig. 6. Permittivity and permeability of composite material.

TABLE IATTENUATION VERSUS COMPOSITE LAYER THICKNESS

CABLE ATTENUATION IN dB/m

frequency

thickness 1 MHz 10 Hiz 100 MHIz 1000 MHlze = 0 13 4 0,007 0,023 0,07 0,23 without

eD2 3,3 D3-4 0,02 0,37 13,8 320D= D 4 i5 4 B 0,04 0,60 20,3 4702,S mm.

e - 1,2 mmD2=4,9 D3-5,6 0,065 0,84 25,2 581e=0,4 1=1,81m 0,036 0,67 25,0 S81

and with a high permittivity, so as to maintain good attenua-tion.

Table II shows the influence of the insulation thickness. Athigher frequencies where the permittivity of the compositedecreases (Fig. 6), the sensitivity to the insulation thicknessdecreases too, as can be expected.

For a given thickness and quality of insulating material(design value), and a given thickness of composite, there is anoptimum value for the composite permittivity, to maximizeattenuation.

Table III shows comparative attenuations with MUSORB Iand MUSORB II, the latter showing a smaller dielectricconstant.The use of dielectrics with low permittivity and with high

dielectric losses is advantageous when higher losses at lowerfrequencies are to be achieved.

F. Characteristic Impedance; Impedance MatchingThe correct choice of the cable parameters, composite, and

insulation allows the design of specific characteristic impe-dances (for example, 50 Q) over practically the entirefrequency range. This achieves impedance matching in thepassband and a matched termination for frequencies above,where the cable behaves like a matched absorptive load. To

D, 2,s

D :LtIrnmr

achieve this result, the two layers are chosen with the propervariations of ,u* and e* and with the fact that the composite canbe designed to show a variation of ft* and e* of the same trend.Fig. 4 attenuation curves are related to such an impedance-matched cable.

G. Magnetic Saturation; Radiation EffectsThe fine subdivision of magnetic particles in the composite

introduces many nonmagnetic gaps, so the relative magneticpermeability of the basic ferrite bulk material (a few hundredto a few thousand) decreases to 15-20 (Fig. 6).

At the same time, saturating magnetic fields are multipliedon the order of a hundred times. The composite is difficult tosaturate, a huge advantage where the absorptive and/ormagnetic performance is to be maintained in strong fields(EMP).

Applications of the low-pass cables for EMP hardening posethe problem of the response of such a cable to ionizingradiation. The scattering of electronic charge in the dielectricand composite layer, and across their gap with the conductors,produces parasitic signals (Photo Compton Currents). Aprotection against this can be built into the low-pass cable (fordetails, see [15]).

III. Low-PASS CABLES WITH COUPLED LINESA. Description and Characteristics

The simple low-pass cables described use dielectric andmagnetic losses in a two-layer structure. The absorptiveeffects allow the design of low-pass cables with usable

I

10o

100

)K6H 10

10 MHz 100 MHz Ghz

0,60

0,40

20 ,3

14 ,9

470

347

Su . '- 'IVI J

I _ _

I

L

10

It

I

ri


absorption in the low tens-of-megahertz range with attenuationincreasing approximately with the 3/2 power of frequency.

Fig. 7 illustrates the concept of using coupled linesdeveloped after the work of Schlicke in the field of EMCcapacitors [16]. The cutoff frequency is decreased and theslope of attenuation versus frequency is increased to the powerof 2.A constant-distributed resistor R is connected in parallel

with the series inductor / of the line; or by duality principle, aresistor r' is connected in series with the parallel capacitance cof the cable.The attenuation a of such a line can easily be calculated as

(12c02 )1/2IVR +1ae=8.686 XcI in dB/m

where X is 2 ir times the frequency.Fig. 7 also shows the general shape of the attenuation curve,

with an initial slope proportional to the square of frequency;reaching a point of slope inversion (a maximum in the ca/fscale) for a frequency

fm= 0.28 1 (or fm=0.28 ri)where the attenuation rises to

aem=8.686 * 2.22 ffm * 4 .In the same way, the combination of both, for a value of fmdecreased by 44 percent, shows a value of am 41 percenthigher.

In practice, coupling between / and R is unavoidable in theimplemenitation of this "simulated skin effect, " as the value ofR is complex, varying with frequency (skin effect).

Fig. 8 shows the attenuation on a practical line (using thesimple line described earlier), with a simulated skin-effectlayer R, for different values of R. The curve for infinite R (R= O) corresponds to values shown above.The slope of 2 at the beginning, as well as the limited

increase of attenuation at higher frequencies, is clearlyillustrated in Fig. 8. At low frequencies, the line is theequivalent of a central-copper-conductor line with an externalcopper braid, both determining the attenuation of 4*10-3 dB/m. At very high frequencies, the line is equivalent to a centralconductor of the resistive layer R and the external copperbraid, both separated by an insulating layer. Ignoring loss inthis layer, attenuation will reach the asymptotic value of

Rt =8.686 in dB/m26R(Zc)

where 61 (Zc) is the real part of characteristic impedance of theouter line.The results are shown for three values of R (R = 150, 15,

1.5 QIm). The characteristic impedance accordingly goestoward VR/co. Calculated values for Rm and fm by thissimple model (shown in Fig. 7) are close to the value

Capev- Cood uc tor M3netLC Cocsa .Sk eld

/~-

Resistive Skeacatk I kT61 rnsu .tLv)3.of R ,ohm La4er

z= 1 4

2= *~~~=jI= I1 x 1I'

jL"L^ R

a~ ~ rI

.cwsr'

I:RIQyll]:Nt()'

Fig. 7. Low-pass coaxial coupled lines.

E ivv

- (0coC-

0s1

001

4 -

Fig. 8. Attenuation of typical coupled-line cable.

calculated by the exact model, representing reality, and so areconvenient for practical use [15].

Such cables allow a decrease by a factor of approximately10 in the frequencies of low-pass cables, for a givenattenuation. For example, a cutoff frequency (3 dB/m) of 15MHz for the simple line (R = oo) will become 1.5 MHz withthe same line having simulated skin effect, optimized with R- 20 Q/m. The disadvantage of this is a limitation of theabsorption at higher frequencies.

It is interesting to note that higher frequency field compo-nents do not penetrate the magnetic composite; in other words,magnetic saturation is reduced even further for high-strengthfields.

Prototypes of the above model have been realized, and haveproven the above model predictions. This type of line is underdevelopment for military applications [17].

IV. Low-PASS CABLES WITH HELICOIDAL LINESThe modeling of both types of lines described clearly shows

the importance of the series inductance of such lines, and therelated losses.

I 1

0

i- >.:5=:;-. -Z)"

.1,


For a magnetic simulated-skin-effect line, an increase of100 times of inductance / allows a decrease of fm by 100;attenuation a,m will be reduced only 10 times, or, for the sameattenuation, frequency would be divided by 10.

Helical lines, with conductors wound around a magneticcomposite core, covered by a sheath of the same composite,are possible with existing manufacturing processes.One can achieve low-pass cables with cutoff frequencies per

meter down to 300 kHz. The resistive layer R is appliedaround the helix or helices (one conductor or several conduc-tors in parallel). This type of line is under development formilitary applications [17].

V. Low-PASS CABLE NETWOFKSA. Characteristic Impedance MismatchWe considered earlier the case of low-pass lines matched to

the rest of the interconnecting network. An interestingpossibility exists for deliberately "mismatching" where multi-ple reflections will occur and also where related resonancesare more or less suppressed by the low-pass effects of theabsorptive lines. Such networks have been mentioned earlier[7], [12], [18], [19].

Let us consider a composite line, formed by cascading twoalternating sections of length 11 and 12, with different character-istic impedances Z01 and Z02 and with corresponding attenua-tions a1 and a2, variable with frequency.

Set K = Z02/Z01, and -= ea+jO, where a + jf is thepropagation constant of line 1 or 2. If one neglects endeffects-i.e., considering an infinite repetition of the two Z0land Z02 lines-the calculation of the voltage ratio betweeninput and output for the two sections yields [12]

function simplifies to

ein, K__2_eut 2(2QK Ir/) r2)

and K > 50, a1 > 2 dB, ax2 > 2 dB.We can therefore make the following interesting observa-

tions.a) At low frequencies, the attenuation of such a line is

relatively low, and approximately equal to the sum of thelosses of the two elements.

b) For increasing frequency, the attenuation rises faster thanthe losses in the elements, since phase rotations in the sectionsintroduce mismatch losses.

c) For a frequency close to the frequency for which oneelement represents a quarter wavelength, the attenuationshows a maximum. Higher in frequency, the attenuationdecreases, reaching a minimum for a length equal to a half-wavelength. (The value of attenuation is then a function of thelosses in the two sections, and is, in general, greater than theirsum.)

d) The attenuation now represents a periodic fluctuationbetween maxima (at frequencies corresponding to odd multi-ples of a quarter-wavelength), and minima (at frequenciescorresponding to even multiples).

e) The amplitude of such fluctuations decreases progres-sively when at the same time the attenuation of the elementsincreases with frequency.

f) The attenuation of the composite line oscillates around acurve whose value is equal to the sum of the attenuation of thetwo elements, plus twice the mismatch losses.

ein K+j 2 1 (I')-K+l I(r 72)eout 2VK 2-2q 2

{ F(1++7'2_)-(2K-1)2(2+72)1 2 K212 2t 1/21K+j12( 2T 'qi?72 2VK

The terms m1 and 772 represent simply the attenuation (withcorresponding phase rotation) of the two elements, withoutany reflective losses. The term 20 log (K + 1)/2VK representsthe "mismatching losses" in decibels, which, for a K of highvalue (K . 10), is approximately equal to

oa = 10 logl0 KI - 6 in dB.

The importance of multiple reflections on the behavior ofthe line depends on the attenuation of one element. Ifattenuation is small, the second reflection returns with theamplitude comparable in magnitude to the first. On the otherhand, for high attenuation-and this is the particular case inwhich we are interested-second and subsequent reflectionsare negligible. If, in addition, K is significant, the transfer

The minimum attenuation will occur only if both linesrepresent an integral number of half-wavelengths at the samefrequency. The maximum attenuation will be obtained whenthe individual elements are a quarter-wavelength at the samefrequency.

g) It can be further shown that if one element shows noattenuation, the total attenuation is due to the attenuation of theother element. But even for a very small attenuation in that oneelement, total attenuation will be much greater than the sum ofindividual losses.

h) For a given total attenuation of the two elements, thegreatest loss in the composite line will be achieved if the loss isdivided equally between the two elements.

i) For a limited number of sections, and for sections ofdifferent lengths and K, the situation becomes more compli-

12


cated, but, basically, attenuation-versus-frequency curves canbe designed cancelling out the fluctuations mentioned, orincreasing the attenuation in a given frequency range.The principle of simple addition of partial-response curves

is valid, as long as losses are high enough to cancel out highermode reflections [20].

j) Attenuation for two elements before the first maximum(element length X/4) is proportional to frequency, i.e., smallerthan with previous low-pass cables for one resonating element.But for a higher number of sections, the slope of attenuation isproportional to the number of sections.

k) High values of attenuation can be achieved, even in the"transition-frequency" range of low-pass cables. The "mis-matched line" can thus decrease the cutoff frequency, with theadditional possibility of shaping the attenuation-versus-fre-quency curve, and more particularly of increasing its slope atthe low-frequency end. A few examples will demonstrate thestatement.

Tests were made with two types of low-pass lines-the firsta straight coaxial line as described in Section II (Z -50 Q),the second using a helical cable with a small capacitance toground (Z-_ 1900 Q) (Fig. 9). In all these tests, the low-frequency attenuation is essentially due to the dc resistance ofthe cable implementations.

Fig. 10 shows attenuation versus frequency of the helicalline, placed in a 50-Q (MIL-STD-220A) test setup. Thequarter-wavelength (for the first resonance) corresponds to 3.6MHz, with an attenuation peak of 15 dB (whereas the basicattenuation of that cable, at 3.6 MHz, is approximately 1.4dB), an attenuation value which is consistent with the equationmentioned earlier.The multiple of half-wavelengths gives attenuation minima

whose values are close to the basic value of attenuation of thehelical line (dashed curve).The amplitude of fluctuation declines with increasing

frequency and, above approximately 60 MHz, no reflectedwave reaches the output.

Fig. 11 represents the same type of helical line witk adouble length; resonant frequencies are halved and theattenuation is approximately doubled, as expected.

Fig. 12 represents two identical helical lines, separated witha length of straight low-pass cable. Overall attenuationcorresponds to the sum of two elements of the helical line plusa length of the straight coaxial line. Obviously, attenuations at3.8 MHz and odd multiples have been emphasized (addition oftwo quarter-wavelength attenuations), but even multiples showlow attenuation minima (7, 14 MHz).

In Fig. 13, we have halved the length of one of the helicalelements to achieve a first quarter-wavelength for both 3.6 and7.2 MHz, thus achieving a broader absorption spectrum, withattenuated minimum peaks.

Obviously, by synthesis, maximum continuous attenuation,or attenuation curves of a given shape, can thus be achieved.For example, four helical lines (1.04, 0.52, 0.26, and 0.13 m)separated by simple coaxial low-pass elements (1.65, 1.65, and1.65 m) will achieve a 3-dB cutoff frequency of approximately70 kHz, an attenuation of 20 dB at 1.0 MHz, 30 dB at 1.3MHz, and an attenuation above 60 dB from 2.0 MHz to 20

(CPLe' CondLctor

N6 s ret,ve Covo>rsolttL COrec-f 3 0 nM

Fig. 9. Hi}

0

,SkIeLcd5h-4L- SlOrH,C - VG pF/Z - 9oo-fa

Fo.v-o Dielectric-S SSrn

.gh-2, low-pass coaxial line.

* * T I I *1 . , , 1 . .- .2 4 , SI oMH020Z ltOoI1GHz

FrequencyFig. 10. Attenuation of mismatched line I = 0.52 m. -

Fig. 11. Attenuation of mismatched line I 1.05 m.

1 2. 4t (6 lonHz20 tO 40 100 2OO lootoO 4GHzFrequency

Fig. 12. Attenuation of two mismatched lines I, = 12 = 0.52 m.

t

13


I MHz

:Od._i

'-.. -Y

101Hz 100 M Hz

I~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~~c

__1211LJI 223 Lh' - jj~- I-jI-I ]-f-jjfIiAll fI

.OI -- . --10 tlH220 tO I40 1o zoo70 oo 1 GHzFreque cy

Fig. 13. Attenuation of two mismatched lines 11 = 0.52 m, 12 = 0.26 m.

GHz. (All these tests used normal low-pass lines, withoutsimulated skin effect).We are currently studying the modeling of arbitrary lengths

of mismatched lines (Project LOSSYNET). Complex net-works of power/communication lines with low-pass perform-ance can be achieved at the least cost. Indeed, it can be shownthat, in such networks, mismatched interfacing with the partialuse of straight low-pass coaxial lines can give results equal tothose of a network with overall (more expensive and heavier)helical lines.An overall straight coaxial-line version for MIL-C-85485A

cables is in development, taking advantage of an increasedslope (above 2) of attenuation in the transition band.B. Lumped Impedance Mismatch

Obviously, lumped impedances, such as series chokes andshunt capacitors, can also achieve characteristic impedancedisruptions within a line. Here, too, it can be demonstratedthat multiple reflections occur and that a lossy cable canimprove the performance of otherwise imperfect components(self-resonance of an inductor, or capacitor!).We shall give an example of a typical application in a cord-

set power line filter using lumped capacitors.

VI. APPLICATIONS

A. General AspectsMore and more industries use electronics-from a simple

control box or instrument to complex electronic systems. Bydefinition, all these devices are susceptible to electromagneticdisturbances where the disturbance may arise through thepower supply (conducted) or directly through the environment(induced or radiated).With the use of low-pass wires (a hookup wire, inside the

device) and low-pass cables (interconnecting cables, net-works), disturbance frequencies (if separated in the spectrum)are absorbed.

This characteristic is built in, i.e., the wire or cable fulfillsthis mission, without any special connections, grounding, orshielding. It appears and is used like any ordinary wire orcable. Thus the following become evident.

a) The absorbed electromagnetic disturbance cannot reach

the susceptible device, i.e., one "insulates" noisy sourcesfrom sensitive equipment.

b) Because of this absorption, parasitic (inductive orcapacitive) couplings between lines or between line and "hotspots" are also decreased at the same time.

c) Because of this absorption, radiation from noisy lines andcables is suppressed, with a special reference to open-linestructures.

d) With the implementation of given cutoff frequencies, onecomes to the new concept of "limited-bandwidth design ofinterconnections" (in place of all-pass interconnections), aninrteresting new EMC concept, where each interconnectionconducts a useful frequency range, but not more.

Figs. 14 and 15 show several industrial cable implementa-tions. I The first represents a three-phase power cable, with aprotected ground conductor (acting as "ground choke" incomputers and industrial control systems). The second showsa multiconductor signaling cable, where each conductor isprotected.

In these examples, each individual conductor is coveredwith an absorptive layer. In conjunction with other conductorsand the ground conductor (and sheath), these lines areprotected from both differential and common-mode distur-bances.

Other variants, where an absorptive layer encompassesseveral wires or where it covers the outer braid of a coaxialline, absorb selectively common-mode disturbances; con-versely, selective absorption of the differential mode aloneis possible-the coaxial cable of Section II is an example.

Intermediary solutions are easy to implement, where one oranother of the modes of propagation is attenuated by a givenamount [21].B. Examples of ApplicationsThe results below are illustrated in the time domain using

the cable of Fig. 14 (copper cross section 4 x 5 mm2;attenuation as mentioned in Section II; length 50 m).The test setup used a pulse generator, with a rise time of a

few nanoseconds. The output of the line is shown in Fig. 16with a delay of 1.44 Its corresponding to a propagation speedof 3.5-107 m/s approximately-i.e., a reduction by a factor of8.7 relative to free space-and a rise time (and fall time) (10-90 percent of the amplitude) slowed down to about 350 ns, dueto the high-frequency absorption (fC = 1 MHz for a 3-dBattenuation by the 50 m length). This lengthened rise time, ofcourse, decreases the amplitude of shorter pulses.

The second sketch shows a decrease of the amplitude to one-half for a pulse length of l10 ns; the third sketch shows adecrease of 32 dB for a pulse of 4.3 ns length.The dispersion, due to propagation parameters which

change with frequency, is visible: indeed, a 5-ns pulse, is"lengthened" to over 100 ns.

It is clearly apparent that such a low-pass cable acts through

1 Manufacturing: Cableries de Saint Etienne et Phoe6enne, St-Etienne,France; Radio Tresses Cables, Genay, Lyon, France; Filotex, Draveil,France; Societe Nationale des Poudres et Explosifs (S.N.P.E.), Paris, France.For further information please contact Dr. F. Mayer, LEAD, 43 rue de 11Novembre 1918, 94700 Maisons-Alfort, France. Telephone: (1) 48-93-44-44.

:11 I, I:

A'tO52." t IDdi,

hIL-STD-220DP

ii1,ll} I /// nDi'st

t 1- ',,I II,

| 5 I ' X ' T - I e I : . ! I . | , . i X . | | . BMuan- 11g

1 i t-11 1 1. 1 1 I t 2 -44

14

w

-0-Ns...

V I

a 4 6o

I:

;I

...i

IIlJA


Fig. 14. Low-pPass power cables.

Fig. 15. Multiconductor low-pass signaling cable.

delayl k rseI

I

IO

3-Phas-. tInustriVl Ca6le 4xSAwA'Ie. S rfd-

..4'*"c

t,r I0 I iZ

SDCJ 2SD" -IO;2' iI 10 f SC 5 rl SE

Fig. 16. Time response of low-pass power cable.

dispersion (lengthening pulses and increasing rise time), aswell as through absorption, which destroys a major part of theenergy content of the pulse.

For design purposes, the classical formula (for first-ordernetworks) can be used

0.35rise time r-=2 in seconds

where f, is the cutoff frequency, as defined earlier.The power network on board an automobile is an example

of an environment where EMC problems may be generateddue to the increasing use of electronic control systems. Fig. 17shows the very broad emission spectrum caused just by thefront-window washer in a typical car (engine stopped, levelmeasured at the battery). Curve a relates to the emission levelof a normal car (with classical copper-wire power and controldistribution network); curve b relates to the same networkimplemented with nonshielded low-pass hookup wire. Actu-ally, level b is below the noise level of the instrumentationabove 30 MHz.

15


Whcss ICeL4- wassher (vio E1Ii c?) c(! Q c: nz%ultP, -30

Fig. 17. RF Spectrum of an automotive appliance.

A third example is the use of high-performance helicalcables (Section III), with a cutoff frequency off, = 300 kHzfor 1 m length, typically for military applications.An approximate simulation, with a signal input of high

amplitude, gives the results shown in Fig. 18. The pulse risetime of 2 ns is lengthened to 2 ys, and the amplitude at theoutput of the 1--m line is reduced to 11 percent of the outputsignal. Such a cable [ 17] is used for the EMP hardening ofmilitary systems and missiles.We further mention the use of a length of the coaxial line

(Section II) linked to classical filter components and classicalfilters (Fig. 19). The dashed curves show the filter improve-ment at higher frequencies, achieved by the series connection(at the input or output of the device) of a small length of thelow-pass line. This improves the frequency behavior of suchequipment up to 20 GHz and higher.

In a fifth example, we show the result of the same coaxiallow-pass cable when connected between two lumped filtercomponents-i.e., two small capacitors with very short leads(as used in cord-set filters), of a capacitance value 47 nF (Fig.20). In Fig. 20 curve a shows the attenuation with a low-losscoaxial cable. A first attenuation maximum occurs at thefrequency where this line represents a quarter-wavelength,i.e., 29 MHz, where the capacitors are active; at higherfrequencies, above the self-resonant frequency of the capaci-tors (20-25 MHz), the basic attenuation decreases (thecapacitors appear like inductors) and regularly spaced reso-nances occur when the signal reflects at the cable-componentinterface. Curve b shows the attenuation when the low-losscoaxial line is replaced by the low-pass coaxial line of SectionII. A first-resonance maximum attenuation occurs at 8.6 MHz,improving filter performance in the 3-15-MHz range; abovethe component resonance, attenuation remains above 60dB because of the low-pass cable attenuation, giving a definiteimprovement in performance of the filter, equivalent to the useof two special (and expensive) feedthrough capacitors.We expect the rapid development of such improvements in

all lumped filters and cord-set filters in the near future.Obviously, similar improvements can be made with lumped-series-inductor filters or inductor/capacitor combinations.

L. g. c2Sg1stc

< t rr~~~~l,sr" rP rf~~~~3,2tF

^s

F..en 1hicbtEwf0ilo

Fig. 18. Response of 1-in helical cable to EMP waveform (simulation).

04I

J:

(0SD10-30

to

Line FiLter,, O,f / /, f/ //

_ ' uf'to 206HC..

l tniN,W d -f A

g ': r"

N N\

L8n(' CD Fe v- - ---- Pf

MHz s IO lo I&1 I0

Fig. 19. Attenuation of typical black-box powerline filters.

16


lk T i7 " 5e| II6 a 10MH,20\ 'tO 40 100 ZOO -1 GH-

FrequencyFig. 20. Attenuation of an absorptive 7r filter.

We envisage further applications in the following fields:a) EMC in antinuclear shelters;b) EMC between power control equipment and electronic

controls on particle accelerators;c) EMC in huge computer centers, for electronic data

handling, and banking operations;d) EMC in microwave-oven manufacturing (and testing);e) EMC in the new electronic computerized offices;f) EMC in manufacturing plants, with electronic robots

and microprocessor control;g) EMC in electronic controllers for heavy industry; andh) EMC in all hardened aeronautical and space applica-

tions.REFERENCES

[1] F. Mayer, "Fil allumage antiparasite," France Patent 1 205 158, Sept.30, 1957, and additions 74 223, Sept. 30, 1958, and 80 097, July 13,1961; also F. Mayer, "Antiparasite electric cable," U.S. Patents3 191 132 and 3 309 633 serial Mar. 24, 1959, CIP Jan. 10, 1963.

[2] P. Schiffres, "A dissipative coaxial RFI-filter," IEEE Trans. Electro-magn. Compat., vol. EMC-6, no. 1, pp. 55-61, Jan. 1964.

[3] W. B. Warren, Jr. and W. F. Woodward, "Development of UHF-

filters with low spurious response levels," in Proc. 10th TriserviceConf. EMC (Chicago, IL), Nov. 1964, pp. 668-673.

[4] H. W. Denny and W. B. Warren, "Lossy transmission line filters,"IEEE Trans. Electromagn. Compat., vol. EMC-10, no. 4, pp. 363-370, Dec. 1968.

[5] J. H. Bogar and E. M. Reyner, "Miniature low-pass EMI filters,"Proc. IEEE, vol. 67, Jan. 1, 1979.

[6] F. Mayer, "Electromagnetic compatibility antiinterference wires,cables, and filters," IEEE Trans. Electromagn. Compat., vol. EMC-8, pp. 153-160, Sept. 1966.

[7] F. Mayer, "Absorptive lines as RFI-filters," IEEE Trans. Electro-magn. Compat., vol. EMC-10, p. 224, June 1968.

[8] F. Mayer, "RFI-suppression components: State of the art, newdevelopments," IEEE Trans. Electromagn. Compat., vol. EMC-18,pp. 59-70, May 1976.

[9] H. Schlicke, "Survey," IEEE Trans. Electromagn. Compat., vol.EMC-10, pp. 181 -186, June 1968.

[10] F. Mayer, "Improvements in or relating to devices for the transmissionof electrical energy," U.S. Patent 3 573 676, Nov. 26, 1965.

[11] H. Schlicke, "Compatible EMI-filters," IEEE Spectrum, vol. 4, pp.59-68, Oct. 1967.

[12] H. G. Tobin, "Two conductor low-pass transmission lines," IIT-Res.Inst. Technol. Center, Chicago, IL, Rep. 5167-F, Aug. 30, 1963.

[13] U.S. Patent Application 855 593, Mar. 12, 1979, CIP 202 654, Oct.31, 1980.

[14] ,"Cables Passe-bas a Absorption: Etat de l'Art et leur importancepour les systemes de l'avenir," presented at the French Nat. EMCConf. Tregastel, June 1-3, 1983.

[15] U.S. Patent Application 429 032, Sept. 30, 1982.[16] H. Schlicke, "Simulated skin effect filters," IEEE Trans. Electro-

magn. Compat., vol. EMC-6, pp. 47-54, Jan. 1964.[17] R and D Contract LEAD-Societ6 Nationale Industrielle Aerospatiale,

Rep. 43/211 584 JOD, Aug. 25, 1982, and supplement, Mar. 29, 1983.[18] LEAD, Maisons-Alfort, France, Res. Rep., "Pseudoresonant tech-

niques," Jan. 23, 1966; also Trend Rep., "Electrical wires andcables," sec. 6.1, July 15, 1978.

[19] J. J. Max, "Distributed low-pass filters for EMI filtering," in Proc.Colloq. Electromagn. Compat. (Zurich, Switzerland), Mar. 8-10,1983 pp. 223-228.

[20] LEAD-DIELI, Maisons-Alfort, France, Predevelopment study,"Brute force filters: BRUTO," Mar. 21, 1980.

[21] For details, see MUSORB product line specifications and U.S. PatentApplication 351 493, Feb. 23, 1982.

[22] H. M. Hoffart, "Electromagnetic interference reduction filters,"IEEE Trans. Electromagn. Compat., vol. EMC-10, no. 2, pp. 225-232, June 1968.

[23] A. R. Martin, "A new concept for EMI protection of cables andharnesses," EMC Tech., pp. 60-65, Apr.-June 1983.

17

Documents

Absorptive Low-Pass Cables: State of the Art and an Outlook to the Future