A Low Phase-Noise VCO Using an Electronically Tunable Substrate Integrated Waveguide Resonator

3452 IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES, VOL. 58, NO. 12, DECEMBER 2010

A Low Phase-Noise VCO Using an ElectronicallyTunable Substrate Integrated Waveguide Resonator

Fan Fan He, Ke Wu, Fellow, IEEE, Wei Hong, Senior Member, IEEE,Liang Han, Student Member, IEEE, and Xiaoping Chen

Abstract—In this paper, an -band low phase-noise voltage-controlled oscillator (VCO) using a novel electronically tunablesubstrate integrated waveguide (SIW) resonator is proposed anddeveloped for RF/microwave applications on the basis of thesubstrate integrated circuits concept. In this case, the resonantfrequency of the SIW cavity resonator is tuned by differentdc-biasing voltages applied over a varactor coupled to the cavity.Measured results show that the tuning range of the resonator isabout 630 MHz with an unloaded of 138. Subsequently, anovel reflection-type low phase noise VCO is developed by takingadvantage of the proposed tunable resonator. Measured resultsdemonstrate a frequency tuning range of 460 MHz and a phasenoise of 88 dBc/Hz at a 100-kHz offset over all oscillation frequen-cies. The VCO is also able to deliver an output power from 6.5 to10 dBm. This type of VCO is very suitable for low-cost microwaveand millimeter-wave applications.

Index Terms—Phase noise, substrate integrated waveguide(SIW), tunable resonator, varactor, voltage-controlled oscillator(VCO).

I. INTRODUCTION

R ecently, substrate integrated waveguide (SIW) structureshave attracted a lot of attention. The SIW can be synthe-

sized in a substrate by metallic via arrays utilizing a standardprinted circuit board (PCB) or low-temperature co-fired ceramic(LTCC) process. The microwave and millimeter-wave compo-nents based on the SIW, which can be easily integrated withother planar circuits, have the advantages of high- factor, lowinsertion loss, and high power capability. Therefore, a numberof applications based on the SIW technique have been reported

Manuscript received February 15, 2010; revised July 27, 2010; accepted Au-gust 02, 2010. Date of publication October 21, 2010; date of current versionDecember 10, 2010. This work was supported in part by the Natural Sciencesand Engineering Research Council of Canada (NSERC), by the National 973Project of China under Grant 2010CB327400, and by the National Natural Sci-ence Foundation of China (NSFC) under Grant 60921063. This paper is anexpanded paper from the Asia–Pacific Microwave Conference, Singapore, De-cember 7–10, 2009.

F. F. He is with the Poly-Grames Research Center, Department of ElectricalEngineering, École Polytechnique de Montréal, Montréal, QC, Canada H3C3A7, and also with the State Key Laboratory of Millimeter Waves, College ofInformation Science and Engineering, Southeast University, Nanjing 210096,China (e-mail: [email protected]).

K. Wu, L. Han, and X. Chen are with the Poly-Grames Research Center, De-partment of Electrical Engineering, École Polytechnique de Montréal, Montréal,QC, Canada H3C 3A7 (e-mail: [email protected]).

W. Hong is with the State Key Laboratory of Millimeter Waves, College ofInformation Science and Engineering, Southeast University, Nanjing 210096,China (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TMTT.2010.2081550

in [1]–[10], especially a high- resonator that was designedusing the SIW technique [11], [12].

Based on the SIW resonator, several microwave and mil-limeter-wave oscillators with fixed oscillation frequencies havebeen developed. In [13], a feedback-type oscillator using aSIW cavity resonator as the frequency selector, which has aphase noise of 73 dBc/Hz at a 100-kHz offset, was developed.A -band Gunn diode oscillator with a phase noise lowerthan 91.23 dBc/Hz at a 100-kHz offset was also reported in[14]. Moreover, SIW cavity resonator oscillators have beendeveloped on the analysis of a relationship between the factorand phase noise [15]. However, most radar and communicationapplications need a voltage-controlled oscillator (VCO) asthe local oscillator (LO) source. Thus far, VCOs based on theSIW resonator have not been reported because it is difficultto design tunable SIW resonators. Although one tunable SIWresonator was developed in [16], it cannot be tuned continu-ously. Recently, we have proposed a continuously electricallytunable SIW reflective cavity resonator that can be used todesign tunable devices such as VCOs and tunable filters [17].This resonator makes use of a typical SIW cavity resonatorthat is combined with a surface mounted varactor to realizethe desired tuning function. Compared to the coaxial packagecomponents combined with the SIW [16], the fabricationcomplexity decreases greatly in this case.

In this paper, an -band low phase-noise VCO is designed,fabricated, and measured with an electronically tunable SIWresonator that is optimized to achieve a wider tuning range than[17]. Described in Section II are the design and analysis of theproposed tunable SIW resonator with its simulated and mea-sured results. In Section III, a reflective VCO based on the pro-posed resonator is designed and measured. The oscillation fre-quency is tuned by applying the reverse dc voltage applied overthe varactor. All the structures in this paper are simulated bymeans of the simulation tools CST 2010 and ADS 2009. Cir-cuits designed are fabricated on a Duroid 6002 substrate with adielectric constant of 2.94 and a thickness of 0.508 mm.

II. DESIGN AND ANALYSIS OF THE SIW TUNABLE RESONATOR

Fig. 1 illustrates the top view of the physical configurationof the electrically tunable SIW reflective cavity resonator. Thewhite and yellow (in online version) areas stand for the substrateand metal covers of the substrate, respectively. The separate cir-cular metal cover is used to provide a dc bias for the varactor.In Section II-A, we will explain how to mount the varactor andset the dc-bias line in detail. As for circuit design, the cavityand its external coupling to the cavity using microstrip line are

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HE et al.: LOW PHASE-NOISE VCO USING ELECTRONICALLY TUNABLE SIW RESONATOR 3453

Fig. 1. Top view of physical configuration of the electrically tunable SIW re-flective cavity resonator. � � �� mm, � � �� mm, � � �� mm,� � ��mm,� � ��mm,� � ��mm,� � ��mm,� � ��mm,� � �� mm, � � �� mm, � � �� mm, and � � � mm.

firstly developed, and then the cavity coupling to the varactor isdesigned.

A. Design of the SIW Cavity and Its External Coupling to theMicrostrip Line

According to [13], the propagation properties of the-like mode of the SIW are very similar to the

mode of a rectangular waveguide. As a result, an SIW cavitycan be designed by using the following equation:

(m/W) (1)

As shown in Fig. 1, the metallic holes are replaced by metallicslots for brevity, where W is the width of these slots. Resonantfrequency of the mode is 9.01 GHz for mm and

mm. The unloaded quality factor of the cavity can beapproximated by the following equation:

(2)

where quantities and are given by Pozar [19] for a rectan-gular waveguide cavity. An estimate of is obtained by usingeffective dimensions of the SIW cavity in the above formulas.In this case, we can use (2) to calculate the unloaded factorof the mode SIW cavity to be 374.

The energy is coupled to the cavity by means of effective cur-rent probes with the microstrip line. The current probes are builtby moving (or removing) metallic slot on one side of the cavityto make a place for an insert, as illustrated in Fig. 1. The probeis merely a prolongation of the microstrip line in the cavity thatis then short circuited. As the probes are comparable in size tothe cavity, they can change the frequency of resonance. A 3-Delectromagnetic simulator is necessary to accurately design thecavity. The strength of the coupling of a current probe mainlydepends on depth and width and of the probe. Itis also important to note that for a microstrip line coupling withthe cavity, the probe is merely a prolongation of the microstripline in the cavity that is then short circuited. Fig. 2 shows the

Fig. 2. Electric field distribution of the electrically tunable SIW reflectivecavity resonator.

electric field distribution of the tunable SIW reflective cavityresonators. Fig. 3 shows the simulated and measured . Wecan see that the simulated and measured resonant frequenciesare slightly different because of the deviation in dielectric con-stant and the fabrication error. The measured resonant frequencyis 8.955 GHz. In [19], a one-port reflection technique was pro-posed to extract the unloaded and loaded from the mea-sured return loss. At the resonant frequency, the coupling coef-ficient can be obtained from the return loss as

(3)

where is the return loss at the resonant frequency. Fromthe equation above, we can get that

(4.1)

or

(4.2)

The coupling coefficient in (4.1) and (4.2) correspond to theunder-coupling case and the over-coupling case, respectively.Through the response circle in the Smith chart (Fig. 3), these twosolutions can easily be distinguished. Usually, a small responsecircle excluding the origin of the Smith chart signifies an under-coupling case; for an over-coupling case, the response circle islarge and encloses the origin. In this design, the over-couplingis indicated as shown in Fig. 3. According to dB,

has been calculated from (4.2). Fig. 3 also helps us tocalculate the loaded . Therefore, the unloaded

can be calculated by

(5)


Fig. 3. Simulated and measured return loss of the SIW cavity resonator in theSmith chart.

B. Coupling Varactor to SIW Resonant Cavity

In the SIW structure, the metallic slots connect the topmetallic cover to the bottom metallic cover, and thus the topmetallic cover cannot be used for the dc-bias line or connectedwith the dc-bias line of active devices. Therefore, it is necessaryto use a separate metallic cover to connect dc bias for activedevices. In our circuit, the circular metallic cover with a diam-eter of is used for dc bias, where is the width of the gapand is the distance from the center of the separated circleto the center of the cavity. The bias line outside of the cavitycan be connected to the circle metallic cover through a bondingwire. The circle metallic cover provides dc bias for the varactor,its cathode is connected on the circle metallic cover, and itsanode is connected on the other top metallic cover (ground).Fig. 2 shows that the electric field in the cavity can be coupledto the separated metallic cover through the gap so the varactorwill mainly be excited by a magnetic coupling. If distancechanges from 0.6 to 3 mm, the simulated resonant frequencywill shift downward from 9.17 to 9.03 GHz, but the almostkeeps constant. When is equal to 1.8 mm, the measuredoscillation frequency is 9.113 GHz and is equal to 125.Measured and simulated results indicate that the ring slot willincrease the resonant frequency and of the SIW cavity.

In the following, we will investigate how the physical lengthand the tuning capacitance of varactor affect the tuning

range of the resonant frequency. Fig. 4 shows simulated oftwo cases when is swept from 1.1 to 0.3 pF. When isequal to 3 mm, the resonant frequency sweeps from 8.441 to8.861 GHz, which results in a tuning range 4.85%; when isequal to 0.6 mm, the resonant frequency increases from 8.510to 9.005 GHz and the tuning is about 5.65%. Simulated resultsindicate that the increased tuning range will increase as the dis-tance decreases. The increment of the tuning range also in-dicates the coupling strength increases between the cavity andcapacitor .

In the design of electrically tunable SIW cavity resonators fordemonstrative purposes, an Aeroflex/Metelics silicon varactordiode MSV34060-0805-2 is used. Parameters of the varactor

Fig. 4. Simulated �� versus � when �� mm and �� mm.

TABLE IPARAMETERS OF THE VARACTOR DIODE

diode are listed in Table I. The tuning capacitance of varactorcan be obtained from 0.3 to 1 pF when the reverse voltage

varies from 30 to 0 V. The distance mm is chosen inthis case.

Fig. 5 displays measured versus dc-bias voltage for thevaractor. The resonant frequency of the tunable reflective res-onator changes from 9.32 to 9.95 GHz, while the dc-bias voltageis swept from 0 to 30 V. The tuning range is about 630 MHzor 6.54% and changes from 55 to 53. Thus, if the varactoris mounted, the unloaded therefore varies from 132 to 138.Compared to characteristics of the SIW resonant cavity, the var-actor has some side effects on the unloaded of this resonator.In the design, measured results do not agree well with simu-lated results due to the parasitic effects introduced to circuit aftermounting of varactor and connecting the dc-biasing line. Theseparasitic effects are very hard to model since they are not con-sistent. Thus, we do not propose the equivalent circuit of thetunable resonator in this paper. Finally, this proposed resonatorprovides a way to resolve the problem of tuning of the SIWcavity. The measured results will be used to simulate the pro-posed VCO in Section III.

III. DESIGN AND MEASUREMENT OF THE SIW VCO

Fig. 6 describes the physical configuration of the reflectiveSIW resonator VCO. This configuration consists of a reflec-


Fig. 5. Measured �� of the reflective tunable resonator.

Fig. 6. Physical configuration of the reflective SIW VCO.

tive tunable SIW cavity resonator, a transistor with an outputmatching network, and a quarter-wavelength interdigital capac-itor. On the other hand, the VCO design is based on the neg-ative resistance concept using a common-source series feed-back element to generate the negative resistance. The activedevice used is an Agilent ATF36077 ultra-low-noise pseudo-morphic high electron-mobility transistor (pHEMT). The feed-back element generating instability in the VCO is a short stubwith a length . A 65- microstrip line with length usedat the gate side is to establish the required negative conduc-tance and meet the oscillation conditions and

. The frequency tuning is realized witha variable capacitance mounting on the SIW tunable cavity.Since the SIW is inherently grounded, this oscillator has the gatevotage . The dimensions of the VCO are mm,

mm, mm, and mm. Fig. 7 shows aphotograph of the fabricated VCO.

The proposed VCO is measured using a test fixture (Wiltron3680) and a spectrum analyzer (Agilent E4440A) in terms ofperformance parameters including the oscillation frequency,output power, second harmonic suppression, and phase noise.Fig. 8 shows the measured oscillation frequency and the outputpower versus the reverse bias voltage applied on the varactor,

Fig. 7. Photograph of the fabricated VCO.

Fig. 8. Measured and simulated oscillation frequency and measured outputpower versus the reverse bias voltage � .

Fig. 9. Measured phase noise at 100-kHz offset and second harmonic suppres-sion.

while the applied bias voltages of the pHEMT is Vand mA. The power consumption is about 37 mW.The tuning range of the oscillation frequency is varying from9.356 GHz (0 V) with an output power of 6.4 dBm to 9.816 GHz(13.3 V) with an output power of 9.3 dBm. The center oscilla-tion frequency is 9.586 GHz (6 V) and the tuning range is 4.8%.When voltage is more than 13.3 V, the oscillation frequency


Fig. 10. Measured phase noise when � � �� V.

Fig. 11. Measured spectrum at 9.81 GHz.

suddenly changes to and keeps at 9.98 GHz. This phenomenaindicates that the oscillation conditions are just met at 9.98 GHzif we apply more than 13.3 V. Measured phase noise of theVCO is better than 88 dBc/Hz at an offset frequency of 100 kHzover the entire tuning frequency range, as shown in Fig. 9. Thesecond harmonic is suppressed more than 33 dB comparing tothe fundamental oscillation frequency. The best suppression of50 dBc occurs at 9.816 GHz (13.3 V). Therefore, the secondharmonic has less of an effect on the fundamental oscillation.Fig. 10 plots measured smoothed phase noise when the os-cillation frequency is 9.4425 GHz (2 V). Fig. 11 shows themeasured spectrum when the oscillation frequency is 9.81 GHz.

For the comparisons among other VCOs, the figure of merit(FOM) is used as

(6)

TABLE IIPERFORMANCE OF REPORTED VCOs

where is the oscillation frequency, is the offset, isthe phase noise at offset , and (mW) is the dc power con-sumption of the VCO. The measured -band VCO has a FOMof 184 dBc/Hz. Table II lists the performance of state-of-artVCOs based on integrated circuit (IC) technology.

IV. CONCLUSION

In this paper, an -band VCO based on a novel tunable SIWcavity resonator has been developed. First, the tunable SIWcavity resonator has been proposed and analyzed. The proposedtunable resonator not only realizes a tuning function by ad-justing the dc-biasing voltage of the varactor, but also retainsthe inherent high- characteristics of the SIW cavity reonnator.A novel planar VCO based on the proposed resonator is then de-signed and fabricated. Measured results show that our proposedVCO has many advantages such as low cost, easy planar inte-gration, and low phase noise. The VCO will be very useful incost-effective wireless systems.

ACKNOWLEDGMENT

The authors would like thanks the Rogers Corporation,Rogers, CT, to provide free samples of the RT/Duroid 6002substrate, The authors are also grateful to S. Dubé and A. Traian,both with the Poly-Grames Research Center, Montréal, QC,Canada, for fabricating experimental prototypes. The authorswish to thank N. Yang and N. Van Hoang, Poly-Grames Re-search Center, for their help during this work.

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Fan Fan He was born in Nanjing, China. He receivedthe M.S. degree in electrical engineering from XidianUniversity, Xi’an, China, in 2005, and is currentlyworking toward the Ph.D. degree in electrical engi-neering at both Southeast University, Nanjing, China,and the École Polytechnique de Montréal, Montréal,QC, Canada.

He is currently an exchange student with the ÉcolePolytechnique de Montréal. His current research in-terests include advanced microwave and millimeter-wave components and systems.

Ke Wu (M’87–SM’92–F’01) is currently a Professorof electrical engineering and Tier-I Canada ResearchChair in RF and millimeter-wave engineering withthe École Polytechnique de Montréal, Montréal,QC, Canada. He also holds the first Cheung Kongendowed chair professorship (visiting) with South-east University, the first Sir Yue-Kong Pao chairprofessorship (visiting) with Ningbo University,and an honorary professorship with the NanjingUniversity of Science and Technology and the CityUniversity of Hong Kong. He has been the Director

of the Poly-Grames Research Center and the Director of the Center forRadiofrequency Electronics Research of Quebec (Regroupement stratégiqueof FRQNT). He has authored or coauthored over 630 referred papers and anumber of books/book chapters. He holds numerous patents. He has servedon the Editorial/Review Boards of many technical journals, transactions, andletters, as well as scientific encyclopedia as both an editor and guest editor. Hiscurrent research interests involve substrate integrated circuits (SICs), antennaarrays, advanced computer-aided design (CAD) and modeling techniques, anddevelopment of low-cost RF and millimeter-wave transceivers and sensorsfor wireless systems and biomedical applications. He is also interested in themodeling and design of microwave photonic circuits and systems.

Dr. Wu is a member of the Electromagnetics Academy, Sigma Xi, and theURSI. He is a Fellow of the Canadian Academy of Engineering (CAE) and theRoyal Society of Canada (The Canadian Academy of the Sciences and Humani-ties). He has held key positions in and has served on various panels and interna-tional committees including the chair of Technical Program Committees, Inter-national Steering Committees, and international conferences/symposia. He willbe the general chair of the 2012 IEEE Microwave Theory and Techniques So-ciety (IEEE MTT-S) International Microwave Symposium (IMS). He is the cur-rent chair of the joint IEEE Chapters of the IEEE MTT-S/Antennas and Propaga-tion Society (AP-S)/Lasers and Electro-Optics Society (LEOS), Montréal, QC,Canada. He was an elected IEEE MTT-S Administrative Committee (AdCom)member (2006–2009). He is the chair of the IEEE MTT-S Transnational Com-mittee. He is an IEEE MTT-S Distinguished Microwave Lecturer (2009–2011).He was the recipient of many awards and prizes including the first IEEE MTT-SOutstanding Young Engineer Award and the 2004 Fessenden Medal of IEEECanada.

Wei Hong (M’92–SM’07) was born in HebeiProvince, China, on October 24, 1962. He receivedthe B.S. degree from the Zhenzhou Institute ofTechnology, Zhenzhou, China, in 1982, and theM.S. and Ph.D. degrees from Southeast University,Nanjing, China, in 1985 and 1988, respectively, allin radio engineering.

Since 1988, he has been with the State Key Lab-oratory of Millimeter Waves, Southeast University,where he is currently a Professor and the AssociateDean of the Department of Radio Engineering.

In 1993 and from 1995 to 1998, he was a short-term Visiting Scholar withthe University of California at Berkeley and the University of California atSanta Cruz, respectively. He has authored or coauthored over 200 technicalpublications. He authored Principle and Application of the Method of Lines (inChinese) (Southeast Univ. Press, 1993) and Domain Decomposition Methodfor EM Boundary Value Problems (in Chinese) (Sci. Press, 2005). He has beenengaged in numerical methods for electromagnetic problems, millimeter-wavetheory and technology, antennas, electromagnetic scattering, RF technologyfor mobile communications, etc.

Prof. Hong is a Senior Member of the China Institute of Electronics (CIE).He is vice-president of the Microwave Society and Antenna Society, CIE.He has served as a reviewer for many technical journals, including the IEEETRANSACTIONS ON ANTENNAS AND PROPAGATION. He is currently an associateeditor for the IEEE TRANSACTIONS ON MICROWAVE THEORY AND TECHNIQUES.He was a two-time recipient of the First-Class Science and TechnologyProgress Prize issued by the State Education Commission (1992 and 1994),the Fourth-Class National Natural Science Prize (1991), and the First- andThird-Class Science and Technology Progress Prize of Jiangsu Province. Inaddition, he was also the recipient of the Foundations for China DistinguishedYoung Investigators and the Innovation Group awards of the National ScienceFoundation of China.


Liang Han (S’07) was born in Nanjing, China. Hereceived the B.E. (with distinction) and M.S. degreesfrom Southeast University, Nanjing, China, in 2004and 2007, respectively, both in electrical engineering,and is currently working toward the Ph.D. degree inelectrical engineering at the École Polytechnique deMontréal, Montréal, QC, Canada.

His current research interests include advancedcomputer-aided design (CAD) and modeling tech-niques and the development of multifunctional RFtransceivers.

Xiao-Ping Chen was born in Hubei Province, China.He received the Ph.D. degree in electrical engineeringfrom the Huazhong University of Science and Tech-nology, Wuhan, China, in 2003.

From 2003 to 2006, he was a Post-DoctoralResearcher with the State Key Laboratory of Mil-limeter-waves, Radio Engineering Department,Southeast University, Nanjing, China, where he wasinvolved with the design of advanced microwaveand millimeter-wave components and circuits forcommunication systems. In May 2006, he was

a Post-Doctoral Research Fellow with the Poly-Grames Research Center,Department of Electrical Engineering, École Polytechnique de Montréal,Montréal, QC, Canada, where he is currently a Research Associate. He hasauthored or coauthored over 30 referred journals and conference papers andsome proprietary research reports. He has been a member of the Editorial Boardof the IET Journal. He holds several patents. His current research interests arefocused on millimeter-wave components, antennas, and subsystems for radarsensors.

Dr. Chen has been a reviewer for several IEEE publications. He was the re-cipient of a 2004 China Postdoctoral Fellowship. He was also the recipient ofthe 2005 Open Foundation of the State Key Laboratory of Millimeter-waves,Southeast University.

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A Low Phase-Noise VCO Using an Electronically Tunable Substrate Integrated Waveguide Resonator