A Highly Linear Fully Integrated Powerline Filter for Biopotential Acquisition Systems

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IEEE TRANSACTIONS ON BIOMEDICAL CIRCUITS AND SYSTEMS 1

A Highly Linear Fully Integrated Powerline Filter forBiopotential Acquisition Systems

Hussain A. AlzaherAlzaher, Noman TasadduqTasadduq, and Yaqub MahnashiMahnashi

Abstract—Powerline interference is one of the most dominantproblems in detection and processing of biopotential signals. Thiswork presents a new fully integrated notch filter exhibiting highlinearity and low power consumption. High filter linearity ispreserved utilizing active-RC approach while IC implementationis achieved through replacing passive resistors by R-2R laddersachieving area saving of approximately 120 times. The filter designis optimized for low power operation using an efficient circuittopology and an ultra-low power operational amplifier. Fullydifferential implementation of the proposed filter shows notchdepth of 43 dB (78 dB for 4th-order) with THD of better than

70 dB while consuming about 150 nW from 1.5 V supply.

Index Terms—Active-RC filters, biopotential acquisition sys-tems, powerline interference, ultra-low frequency filters.

I. INTRODUCTION

I N recent years, there has been significant developmentin the various fields of biomedical circuits and systems

[1]–[17]. Ultra-low frequency filters has wide range of ap-plications in such systems. Examples include hearing aid[6], photoplethysmogram [7], electrocardiogram (ECG) [8],wearable breathing detector [9], wireless reflectance pulseoximeter for unfiltered photoplethysmograms [10], [11], elec-tronic patch for wearable health monitoring by reflectancepulse oximetry [12], acquisition of various neurophysiologicalsignals [13]–[16], and neural spike detection [17].

In biopotential acquisition systems, common-mode rejection(CMR) is one of the most important performance parameter. Itis of particular interest at the frequency of the primary source ofinterference that is the 60 Hz or 50 Hz of the ac main. Althoughmodern differential amplifiers customary exhibits a common-mode rejection ratio of 80 to 90 dB, common-mode to differ-ential mode conversion known as “the potential divider effect”due to mismatch in both external and internal components inthe signal path degrades CMR performance. Traditional tech-niques to improve CMR performance are: shielding, isolationand driven right leg configuration. The most effective shieldingtechnique is guarding with the average of the input signals. It

Manuscript received April 07, 2012; revised October 15, 2012; accepted Jan-uary 11, 2013. This work was supported by the King Fahd University of Petro-leum & Minerals and King Abdulaziz City for Science and Technology – ProjectARP-29-99. This paper was recommended by Associate Editor A. Bermak.

The authors are with the Electrical Engineering Department, King Fahd Uni-versity of Petroleum & Minerals, Dhahran 31261, Saudi Arabia (e-mail: [email protected]).

Color versions of one or more of the figures in this paper are available onlineat http://ieeexplore.ieee.org.

Digital Object Identifier 10.1109/TBCAS.2013.2245506

results in good interference suppression with just one extra am-plifier, however, it may lead to signal loss and distortion and itis problematic when long measuring cables are used. [18]

On the other hand, improving the isolation between the de-vice ground and the patient ground helps to improve the systemCMR. Isolated measurements can be problematic, even if thecommon-mode voltage is kept small, because the interferencevoltage across the isolation is not rejected sufficiently [18]. Adriven right leg configuration is the most practical way to reducethe common-mode voltage if reduction of interference currentis not sufficient [18] and [19]. The main drawback of a drivenright leg circuit is it being potentially unstable [20]. In practicaldesigns, the electrode impedance and loop stability become lim-iting factors in common-mode feedback (CMFB) efficiency. Aproper driven right leg circuit may accomplish CMR improve-ment of about 33 dB [19]. The second drawback with this ap-proach is that beside the CMFB amplifier, a voltage buffer withrelatively high biasing current is needed to drive the relativelysmall resistance connected to the patient’s body.

Alternatively, a technique that is not only capable of im-proving CMR but pay special attention to the primary sourceof interference (power-line noise) would be more attractive.This can be achieved through adopting a fully differential notchfilter. While the fully differential operation would positivelycontribute to the overall CMR, the selective characteristics ofthe notch response would remove the strong interference at50 Hz or 60 Hz even in presences of the potential divider effect.In addition, this approach circumvents the stability problemof the right leg configuration leading to much higher powerline attenuation while maintaining low power consumption.Therefore, the CMR requirement of other parts would be con-siderably relaxed as they would be responsible for removingthe other less significant sources of common-mode noise. How-ever, one must be careful with such approach so that real-worldsignals are not compromised with this type of filtering. Theline frequency interference concurrently occurs within thesame band where biopotential and other physiological signalshave most of their energy [21], [22]. Thus, it has considerableeffects and plays crucial part on the quality of these signals.Examples include ECG, electroencephalogram (EEG), andelectromyogram (EMG) recordings with frequency rangesof 100 Hz, 250 Hz, and 1000 Hz, respectively [23]. Weakbiomedical signals in order of to mV can be monitored atthe body surface and pre-amplification and filtering are manda-tory before further digital signal processing (DSP). A typicalsystem for processing biopotential signals consists of sensorsor transducers converting the physiological signals into electricsignals, a low noise preamplifier, a powerline notch filter, and a

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2 IEEE TRANSACTIONS ON BIOMEDICAL CIRCUITS AND SYSTEMS

Fig. 1. A typical system for processing biopotential signals.

lowpass filter followed by an analog to digital converter (ADC),as shown in Fig. 1.

The amplifier with low input-referred noise typically providesgain in order of 10–100. Clearly, a notch filter is required as thepowerline interference is often larger than the desired biopoten-tial signals. EEG superimposes the most stringent requirementsuch that the notch filter should provide attenuation of 40 dBto achieve a signal-to-noise ratio (SNR) of 0 dB. Obviously,more attenuation would relax the design of subsequent stages.The lowpass filter is needed to reduce the out-of-band noise andinterferers. The notch filtering can be achieved using either adigital filter or an analog filter. But the digital notch filters areuseful when the powerline interference is smaller than the ac-tual signal [24], which is normally not the case. Also, the use ofdigital notch filters in some biomedical systems [25] and [26]was found to degrade the overall resolution of the system due tothe reduction in the dynamic range of the analog front end [27].

The general form of a second-order notch filter transfer func-tion is given by

(1)

where is pole frequency as well as the notch frequency (ze-roes at ), is the pole quality factor, and is the gain.The selectivity performance of a notch filter is characterized byits notch depth, ideally infinity, and . In order to utilize verylarge-scale integration (VLSI) techniques in biomedical instru-mentation, implementation of this system in a single integratedchip is required. This has been a challenging design problemdue to the difficulty in developing efficient methods to achievelarge time constant to implement the notch filter. Such notchfilter would require resistances in the order of and capaci-tances in the range of nF, and therefore is impractical for imple-mentation on an IC chip [28]. Components with relatively largervalues would suffer from nonlinearities and parasitics that mayprevent proper circuit operation [29]. Therefore, special low fre-quency design techniques must be adopted in order to realize theextra-large time constants of the filter.

A new notch filter design avoiding drawbacks of availablesolutions and providing improved characteristics is presented inthis work. The main desired requirements in addition to integra-tion in a single chip are as follows: (i) Low power consumptionin order to reduce amount of heat, decrease battery size and in-crease battery life, (ii) Low input referred noises to process theweak physiological signals and (iii) High linearity to avoid gen-erating harmonics that could be more dangerous than the pow-erline interferences. In fact, the second requirement would berelieved in presence of pre-amplification and would converge to

requirement (iii). Programmability is also needed to adjust thefilter zero frequency to its nominal value compensating for in-accurate component values, process variations, and temperaturechanges.

A preliminary version of this work providing initial simula-tion results was presented in [30]. This paper further expandsthe assessment of the available solutions and elaborates more onthe systematic development of the proposed solution, and hencehighlights its inherent advantages more. This work presentsthe results of a more optimized design and provides additionaltheoretical verifications. Detailed experimental results andmeasurement validations using real biomedical signals arealso provided. The following section describes the availablesolutions and summarizes their performance characteristics.Consequently, Section III presents the systematic methodologyused to develop the proposed solution. The proposed topologyand non-ideal analysis are discussed in Sections IV and V,respectively. Designs of the basic building blocks used toimplement the filter are presented in Section VI. Experimentalresults verifying the presented theory are given in Section VII.

II. AVAILABLE SOLUTIONS

Operational transconduactance amplifier (OTA) or gm-Cbased filters such as [27], [31]–[33] are widely used in biopo-tential acquisition systems. For EEG application a 5th orderelliptic lowpass notch filter using LC ladder approach basedon OTAs was reported in [27] and [31]. The designs cover thefour bands of brain waves that constitute the basic EEG signal:

wave (0.5 Hz 4 Hz), wave (4 Hz 8 Hz), wave (8 Hz13 Hz), and wave (13 Hz 40 Hz). It employs both currentdivision and current cancellation technique for the design ofOTA. Working in weak inversion region, it manages nA/Vtransconductance enabling it to select small capacitor values.The single ended filter of [27] is fabricated in 0.35 CMOSprocess, and achieves 66 dB notch attenuation at 50 Hz with astopband attenuation of 36 dB above 50 Hz, while consumingtotal power of 11.1 . The filter is associated with THD of

50 dB for an input voltage of 25 mV and frequency of 8 Hz.Whereas, the filter of [31] is simulated using 0.6 CMOSprocess, and achieves 58.5 dB attenuation at 50 Hz with astopband attenuation of 32 dB above 50 Hz. The power andlinearity characteristics of the filter presented in [32] are notgiven.

Capacitance scaling technique may be utilized for realizinglarge time constant [33]. Capacitance multiplier circuits, de-pending often on transconductance ratios or transistor sizing,can provide 10–200 times of the basic capacitance value. An-other method to realize low frequency notch filters uses ac-tive-RC filters with Twin-T resistor configurations. But theycannot practically achieve frequencies as low as 60 Hz since2 resistance and 10.6 nF capacitance could be required [29].Alternatively, time constant multiplier (TCM) circuit attaininglarge time constants using resistors and capacitors that can be in-tegrated was suggested in [29] but it uses 10 opamps. The filterexhibits Q of 1/2, and achieves a notch depth of 45 dB whennormal opamps are used.

Also, chopper stabilization technique, which has been usedfor flicker-noise reduction has been adopted to realize the


ALZAHER et al.: A HIGHLY LINEAR FULLY INTEGRATED POWERLINE FILTER FOR BIOPOTENTIAL ACQUISITION SYSTEMS 3

notch filter [23] and [34]. In this approach the interferencealong with the desired signal are shifted up in frequency andthen the powerline signals are filtered at relatively higherfrequencies. The output chopper demodulates them back tothe baseband after filtering. This approach reduces the capac-itor values by modulation to the powerline frequencies (e.g.

) and upto 40 dB notch attenuation isreported. Simulation results in 90 nm CMOS process shows apower consumption of 75 mW from supply voltage of 3 V.

Switched capacitor (SC) based integrators for realizing largetime constants are presented in [35]–[37]. In [35] high time con-stant SC integrator circuit using T-cells is presented with a ca-pacitance spread of 25. Experimental results using clock fre-quency of 100 kHz shows S/N ratio of 60 dB with minimal ca-pacitance of 0.4 pF. The integrator in [36] uses capacitor ratiosto realize large time constant and unlike the T-network approachis insensitive to parasitics. The integrator is utilized to design a60 Hz notch filter which is operated using a clock frequency of128 kHz. Experimental results show standard deviation of 0.5%in the notch frequency. The design in [37] presents a gain andoffset compensated SC based integrator. A 60 Hz notch filter ispresented and simulation and analytical results show improve-ments in the gain error.

III. PROPOSED APPROACH

Pre-amplification of the weak biomedical signals, as demon-strated in Fig. 1, relaxes the noise performance of the filter.But it demands more linearity in order to process large inputsignals. Filter designs using gm-C or OTA-C are in conflictwith use of pre-amplifications. Transconductors, in general, ex-hibits low linear ranges particularly for low supply voltages[38], [39]. Actually, there are tradeoffs between small transcon-ductance, low-noise performance, and dynamic range. This isbecause small transconductance often requires reducing the bi-asing current which automatically results in smaller input linearrange and more noise.

Linearity of capacitor multiplier circuits (CMC) depends onactive elements (mainly gm circuits or transistors). Also, thereis a clear conflict between linear range and multiplication factor.Even if the linearity issue is circumvented by using active-RCapproach [29], this solution would be power inefficient sinceCMC or TCM would be sub-circuit of the overall filter design.For example, adopting TCM requires 10 opamps to realize thenotch filter in [29]. Chopper stabilization is a complicated so-lution and hence power inefficient. It requires modulation anddemodulation, clock generation, and two notch filters instead ofthe original one (because of modulation). In addition, this ap-proach suffers from generating unwanted spikes, and hence re-quires auxiliary lowpass spike filter [23].

In fact, opamp based circuits is considered to be the optimumsolution because of its inherent superior linearity. However, SCis expected to be unsuitable for applications requiring large timeconstants (of the order of millisecond or more). This is becauseadvanced processes suffer from gate leakage problems. In ad-dition, SC circuits are associated with switching noise limitingtheir applications. On the other hand, MOSFET-C technique ex-hibits low linearity (similar to gm circuits). Therefore, it can bededuced that the optimum solution would be based on active-RC

Fig. 2. Configuration of the R-2R ladder in this work.

(opamp based filters). However, while expanding this approachto demonstrate area-efficient solution the following criteria hasto be fulfilled: (i) avoid dynamic switching, (ii) preserve highlinearity, and (iii) maintain low power by minimizing numberof opamps. These requirements would exclude SC or chopperstabilization methods, MOSFET-C approach, and opamp basedTCM methods. R-2R ladders, which is conventionally used indata converters, was previously adopted to promote bandwidthtuning of a lowpass filter over wide frequency range [40]. Thiswork proposes utilization of R-2R ladders to allow realizationof large time constants. It is observed that R-2R ladders arecapable of achieving large time constant because their equiva-lent resistance is much greater than their total actual resistance.They preserve the high linearity (poly resistor) and maintainzero DC power consumption. The optimum use of R-2R laddersin this application is achieved when their configurations resultin the largest possible equivalent resistance (i.e. when only theleast significant branch current is connected to the virtualground) as shown in Fig. 2.

Therefore, the maximum resistance can beincreased by increasing the size of the ladder and/or thevalue of the basic resistance . In this case there is no needto use any switch, saving a huge silicon area. The total areaneeded to make an n-bit R-2R would be that of

. The relative saving in area achieved through the use of aR-2R ladder is proportional to .Therefore, it can be seen that this saving is independent ofvalues and it improves considerably as n increases. Saving inarea of approximately 11, 35, 117 and 400 times is achievedfor , 10, 12, and 14, respectively. In conventional useof R-2R in data conversion, the main error sources are due tomismatches of the switch-on-resistances which are avoided inour proposed solution. Also, the actual value of the equivalentresistance is not important in the presence of tuning feature. TheR-2R ladder can be incorporated into the circuit by replacing thepassive resistors controlling the filter parameters of the originalfilter with R-2R ladders. This can be applied as long as theseresistors are connected to virtual ground, which simulates theproper operating condition of the R-2R ladder [41].

IV. PROPOSED FILTER TOPOLOGY

Biquads based on a single-opamp are more power efficientthan their multi-opamp based counterparts. However, such min-imal opamp realizations require passive components matchingconditions to realize a notch response. Without these constrainsthe filter would exhibit different transfer functions. In particular,the depth of the notch will be significantly reduced for inaccu-rate component values. Examples include the general biquadsbased on the Sallen-Key [42] and Delyiannis-Friend filters [43]



Fig. 3. The proposed filter based on Tow-Thomas Biquad with R-2R ladders.

and [44]. The second important factor that prevents the use ofsuch topologies for this application is the absence of virtualgrounds required to include R-2R ladders in the filter design.

In fact, the needed virtual grounds are available inmulti-opamp biquads developed from the two-integratorloop topologies namely Kerwin–Huelsman–Newcomb (KHN)[45] and Tow–Thomas [46]. There are two methods to obtainthe notch filter from the basic circuit of KHN and Tow-Thomas.The first approach is to sum the HPF and LPF in KHN, or theinput signal and BPF in Tow-Thomas. In this case an additionalopamp and three resistors would be required by the adder cir-cuit. It can be shown that, unlike KHN filter, the Tow-Thomascan be designed for pre-required gain and by independentlyselecting two resistors. Then, the pole frequency can be inde-pendently adjusted by tuning all resistors simultaneously. Inaddition, the three opamps in the Tow-Thomas topology havetheir non-inverting input grounded and therefore their invertinginputs will be held ideally at virtual ground. This feature per-mits the use of opamps with small common-mode input range.The second technique, more efficient, requiring no additionalopamp, injects the input signal into proper internal nodes toproduce the required notch function. The notch filter basedon Tow-Thomas biquad and feedforward technique shown inFig. 3 is selected for this application. It requires an additionalcapacitor to realize the notch function compared with basicTow-Thomas biquad.

Assuming the opamps to be ideal, routine analysis shows thatthe filter exhibits the following transfer function:

(2)

For equal gain of unity in LF and HF sides of the notch thefollowing conditions and arerequired. Selecting and as in the basictopology, the pole (the notch) frequency, and pole factor aregiven by

(3)

(4)

Therefore, the notch frequency can be adjusted without dis-turbing by adjusting either all capacitors or all resistors simul-taneously. Note that is not selected to be equal to whenit is observed that its value is important for optimizing the re-sponse of the filter as will be seen in the next section.

V. NON IDEAL ANALYSIS

Equation (2) is obtained assuming the opamp gain to be in-finity. Since the frequency of operation is low, the proposedfilter will not suffer from the opamp’s gain bandwidth productproblem. However, adopting practical opamps with finite gainswill result in deviations from the ideal response. This section in-vestigates the effect of finite opamp gain on the proposed filterperformance. In fact, active-RC filters are designed based onthe larger open-loop gain of opamps and closed-loop configura-tions. Therefore, as the open-loop gain of opamps increases, theerror between the ideal and practical responses of the filter be-comes less and its linearity and noise performance will improve.Assuming finite opamp gains of , non-ideal analysis of the cir-cuit of Fig. 2 yields the transfer function of (5). (See equationat bottom of page.) It can be seen that as (5) reduces to(2).

It is clear that finite generates an s-term in numerator andtherefore the notch will not exhibit absolute zero. Neglecting theless important error in the and terms of denominator, (5)is used to estimate the notch depth as given by (6).

(6)

Clearly, can be improved by increasing ratio, , and/ordecreasing . Hence, it is advantageous to select as largeas possible. The effect of the mismatch of the components ontothe notch depth is also studied. The mismatch in the passivecomponents is described by and

(5)



Fig. 4. Fully differential opamp (modified from [48]).

where and are the averages of the two ideallymatched passive components. These relations can be used toobtain (7). (See equation at bottom of page.) Subsequently, (7)can be approximated as follows:

(8)

Note that and can be positive or negative. Assumingpassive components mismatch of 1%, which are typically ob-served for small-area diffused devices [46], would lead to anotch depth of 46 dB whereas mismatches of 0.5% would re-sult in notch depth of 52 dB. As matching accuracy is improvedusing large devices and careful layout, the notch attenuation in-creases. For example, a 0.1% matching accuracy would improvethe notch depth to 86 dB.

VI. DESIGN OF FILTER COMPONENTS

Designs of the basic building blocks used to implement thefilter are presented in the following subsections.

A. Opamp

The power consumption of the opamp must be optimized forlow power operation. Also, fully integrated biomedical systemsincorporate fully differential architectures to enhance the perfor-mance in terms of supply noise rejection, signal swing, and har-monic distortion A fully differential version of a two stage classAB opamp is shown in Fig. 4 [48]. Since both input and outputstages are class-AB, it can work with very low biasing currents,hence providing very low power solution. The opamp was re-designed in 0.35 technology and simulated using supplyvoltages of 0.75 V. The opamp was optimized to achieve gainof more than 100 dB. This is achieved when is set to 1 nAleading to a total current of approximately 50 nA. The opamp iscompensated to have a phase margin of better than 70 resultingin a unity gain frequency of 100 kHz.

B. Passive Components

The value of CR required to achieve 60 Hz notch frequencycan be determined for a specific ladder size and . Assumingmaximum capacitance of 50 pF for and , Table Igives the required value of for several ladder sizes. Also, thevalue of is given to maintain for different cases.

(7)



TABLE IPASSIVE COMPONENT VALUES AS FUNCTION OF VARIOUS LADDER SIZES

TABLE IISEVERAL POSSIBLE VALUES OF � FOR � � ��, � � ��

Table I shows that ladder sizes must be more than 10 bitsto have reasonable values of . For , the ratio

is relatively small and hence will limit the notchdepth. This is confirmed by the simulation results where a notchdepth of only 42.5 dB is achieved. Considering the passive com-ponents for the cases of 13 and 14 bits, it can be seen that theformer choice is more area efficient. This is clear since this caserequires additional capacitance of just 2 pF for while it re-quires half of the area for the ladder forming and uses onebit less for ladders making through . Comparing the casesof ladder size of 12 bits with that of 13 bits, it is expected that thelater case will provide better notch depth. However, former caseis more area efficient as the additional 10 pF in value is lessthan the area of adding 1 bit for four ladders and double the areafor the ladder forming . In addition, the frequency responseof the 12 bit ladder design shows a notch depth of 49.67 dBwhich is less than the case of 13 bits just by 0.07 dB.

Table II shows the several different values of (assumingmaximum capacitance of 50 pF) and for achieving notch fre-quency of 60 Hz when using 12 bit ladders of .Also, it gives values required to set . Effect of in-creasing to improve the notch depth for is verifiedthrough simulations. It has been found that as is increased,more depth is attained. In fact, selecting shows a10 dB improvement in the notch depth compared with the caseof equal capacitors. Thus, the design case adopting 12 bits lad-ders, , , and

is considered to be optimum.

C. Tuning

It can be seen from (3) and (4) that can be tuned withoutdisturbing via adjusting either all resistors ’s and/or all ca-pacitors simultaneously. The R-2R ladders can be programmedto tune the notch frequency over wide range given the opportu-nity to accommodate different possible applications [49]. Forinstance, frequency tuning range of more than three decadescan be achieved with 10-bit ladders. However, this techniquewould be unsuitable for fine tuning since every bit change woulddouble the notch frequency.

Fig. 5. Die photograph of the proposed filter.

Fine tuning can be achieved using resistors and/or capacitormatrices with small areas. Note that the conventional use ofsuch matrices exhibit large areas as they are intended for coarsetuning in addition to fine tuning, wherein the coarse tuning re-quires large component spread. Simultaneous tuning of all re-sistors would require four resistor arrays while tuning capac-itors require three capacitor arrays. Therefore, three capacitormatrices are adopted to tune the filter notch frequency. Tuningrange from 40 Hz to 80 Hz for notch frequency is selected. Thisallows for compensating of 33.3% variation in nominal fre-quency. To achieve resolution accuracy of approximately 1%(0.6 Hz), capacitor matrices of 6-bits are incorporated.

Unlike the high-frequency (noise generating) switches of SCtechnique, the switches of the capacitor arrays are quasi-staticwhich are suitable for sensitive analog signal processing. Theembedded digital tuning feature of the proposed approach al-lows direct programmability by digital tuning. Digital automaticfrequency tuning scheme such as that described in [50] can beemployed to compensate for both components and temperaturevariations.

VII. EXPERIMENTAL RESULTS

The proposed filter was fabricated using 12 bit ladders withand , , and

in 0.35 CMOS process. Die photograph of the pro-posed design occupying an area of about 1 is shown inFig. 5. In fully differential structure, there is no need for the in-verter since signals can be inverted by means of proper crosscoupling between the positive and negative paths. The lad-ders were made up of 14-bit, the additional 2 bits are employedto allow programming the quality factor of the filter from 1/2 to2.

A. AC Signal Testing

Experimental results for the frequency response of the pro-posed filter with tuning are shown in Fig. 6. It shows that thefilter achieves notch attenuation of 43 dB. Capacitor matricesof 6 bits are used to provide notch frequency tuning from 40 Hzto 80 Hz with resolution accuracy of about 0.6 Hz. Also, exper-imental results show THD of about 70 dB due to input signalsof 100 mV. Also, it was found that the gain of the high passband



Fig. 6. Experimental results showing notch frequency tuning.

Fig. 7. Experimental results of cascaded 4th-order filter for Q of 1/2.

filter is flat for frequencies up to approximately 50 kHz whichis much more than the desired biopotential signal bandwidths.Two sections of the filter were connected in cascade to realize4th-order design in order to enhance the notch depth. Measure-ment results are shown in Fig. 7 where 78 dB notch depth isrecorded.

In addition, the response of second-order filter for isshown in Fig. 8 wherein a notch depth of 36 dB is achieved.It is found that by cascading two of these sections the low sidepassband frequency extends up to 40 Hz and the notch depthbecomes about 68 dB.

B. Physiological Testing

Experimental tests validating the function of the notch filteron the quality of biomedical signal recording was also carriedout following a similar procedure as that of [51]. The filterwas used for processing real data of an arrhythmic ECG signaltracing [52]. Adopting a suitable interfacing setup, a distortedECG signal with 60 Hz interference could be generated as

Fig. 8. Experimental results showing Q tuned to 2.

Fig. 9. Time domain output showing distorted ECG signal (CH1) and resultsafter the notch filtering (CH2).

shown in Channel 1 of Fig. 9. The corresponding time domainoutput signal of the notch filter is shown in Channel 2. Obvi-ously, clear baseline signal is obtained.

The results shown in Fig. 9 demonstrate the role of the pro-posed notch filter in eliminating the line noise. It can be seenthat although the line noise has been substantially eliminated,the ECG waveforms show baseline drift. This is expected be-cause eliminating all components at the line frequency will notremove other sources of common-mode noise. However, thenotch filter is supposed to be used as a part of the processingsystem as depicted in Fig. 1. Consequently, the low noise ampli-fier is expected to further reject the other less significant sourcesof common-mode noise. Also, the lowpass filter shall suppressout-of-band noise and interferers leading to a better overall noiseperformance. A notch filter with a relatively low Q may resultin losing other signal components of interest that happen to fallin the notch band. Obviously, the higher the Q of the filter, theless is the loss. The measuring results show that Q of the pro-posed filter has been sufficient to allow observing the importantfeatures of the ECG that are the P, Q, R, S, and T waves.



Fig. 10. Power spectrum of a real Arrhythmic signal showing significant pow-erline interference (CH 1) as well as attenuation due to the proposed notch filter(CH 2).

TABLE IIICOMPARISON OF THE PROPOSED FILTER

The power spectrums of the input and output signals are alsorevealed in Channel 1 and Channel 2 of Fig. 10. It can be seenfrom the results that with the help of the notch filter the 60 Hzinterference has been eliminated (61 dB attenuation). In addi-tion, the time delay between input and output signals was alsomeasured and found to be less than 5 .

Table III summaries the characteristics of the proposed filter.The main characteristics of notch and lowpass notch suggestedin [8], [23], [27], [31] and [34]. It is clear that the proposed filtermanages to show 10 dB and 20 dB improvement over its coun-terpart of [8] and [27] in terms of THD. Also, its power con-sumption is considerably much lower than the previous solu-tions. The notch attenuation is increased by cascading two sec-tions of the proposed biquad. For the case of , the

4th-order filter provides 18 dB and 12 dB more notch depththan [8] and [27]. For the case of where the proposedfilter and that of [27] exhibit same lower side passband fre-quency (40 Hz), the proposed filter provides slightly better per-formances in terms of notch depth while consuming signifi-cantly much lower power consumption.

VIII. CONCLUSION

This work presents a new design for implementing the power-line notch filter for biopotential acquisition systems. The paperstudies and assesses the available solutions. This has lead toa novel approach avoiding disadvantages of other works pro-viding improved characteristics. A new fully integrated notchfilter is proposed. R-2R ladders are adopted to allow the realiza-tion of large time constant in small area and they are employedin a proper filter topology. The proposed filter design is system-atically identified to be the optimum. Main claims are supportedwith analytical proofs. Also, the operation and results are ver-ified through IC fabrication and experimental results. Experi-mental results show significant improvement in terms of powerconsumption and linearity.

ACKNOWLEDGMENT

The authors would like to thank the Editor, Associate Editor,and the reviewers for their valuable comments that have en-hanced this work.

REFERENCES

[1] H. Okuno and T. Yagi, “Image sensor system with bio-inspiredefficient coding and adaptation,” IEEE Trans. Biomed. CircuitsSyst., vol. 6, no. 4, pp. 375–384, Aug. 2012.

[2] N. Ollivier-Henry, W. Gao, X. Fang, N. A. Mbow, D. Brasse, B. Hum-bert, C. Hu-Guo, C. Colledani, and Y. Hu, “Design and characteris-tics of a multichannel front-end ASIC using current-mode CSA forsmall-animal PET imaging,” IEEE Trans. Biomed. Circuits Syst., vol. 5,no. 1, pp. 90–99, Feb. 2011.

[3] S. Ramakrishnan, P. E. Hasler, and C. Gordon, “Floating gate synapseswith spike-time-dependent plasticity,” IEEE Trans. Biomed. CircuitsSyst., vol. 5, no. 3, pp. 244–252, Jun. 2011.

[4] S. Narasimhan, H. J. Chiel, and S. Bhunia, “Ultra-low-power and ro-bust digital-signal-processing hardware for implantable neural inter-face microsystems,” IEEE Trans. Biomed. Circuits Syst., vol. 5, no. 2,pp. 169–178, Apr. 2011.

[5] M. T. Salam, M. Sawan, and D. K. Nguyen, “A novel low-power-im-plantable epileptic seizure-onset detector,” IEEE Trans. Biomed. Cir-cuits Syst., vol. 5, no. 6, pp. 568–578, Dec. 2011.

[6] I. Deligoz, S. R. Naqvi, T. Copani, S. Kiaei, B. Bakkaloglu, S. Je, and J.Chae, “A MEMS-based power-scalable hearing aid analog front end,”IEEE Trans. Biomed. Circuits Syst., vol. 5, no. 3, pp. 201–213, Jun.2011.

[7] A. Wong, K. Pun, Y. Zhang, and K. Nang Leung, “A low-power CMOSfront-end for photoplethysmographic signal acquisition with robust DCphotocurrent rejection,” IEEE Trans. Biomed. Circuits Syst., vol. 2, no.4, pp. 280–288, Dec. 2008.

[8] S. Lee and C. Cheng, “Systematic design and modeling of a OTA-Cfilter for portable ECG detection,” IEEE Trans. Biomed. Circuits Syst.,vol. 3, no. 1, pp. 53–64, Feb. 2009.

[9] P. Corbishley and E. Rodriguez-Villegas, “A nanopower bandpass filterfor detection of an acoustic signal in a wearable breathing detector,”IEEE Trans. Biomed. Circuits Syst., vol. 1, no. 3, pp. 163–171, Sep.2007.

[10] K. Li and S. Warren, “A wireless reflectance pulse oximeter with dig-ital baseline control for unfiltered photoplethysmograms,” IEEE Trans.Biomed. Circuits Syst., vol. 6, no. 3, pp. 269–278, Jun. 2012.

[11] K. Li, S. Warren, and B. Natarajan, “Onboard tagging for real-timequality assessment of photoplethysmograms acquired by a wireless re-flectance pulse oximeter,” IEEE Trans. Biomed. Circuits Syst., vol. 6,no. 1, pp. 54–63, Feb. 2012.



[12] R. G. Haahr, S. B. Duun, M. H. Toft, B. Belhage, J. Larsen, K.Birkelund, and E. V. Thomsen, “An electronic patch for wearablehealth monitoring by reflectance pulse oximetry,” IEEE Trans. Biomed.Circuits Syst., vol. 6, no. 1, pp. 45–53, Feb. 2012.

[13] M. Mollazadeh, K. Murari, G. Cauwenberghs, and N. Thakor, “Mi-cropower CMOS integrated low-noise amplification, filtering, and dig-itization of multimodal neuropotentials,” IEEE Trans. Biomed. CircuitsSyst., vol. 3, no. 1, pp. 1–10, Feb. 2009.

[14] M. Mollazadeh, K. Murari, G. Cauwenberghs, and N. Thakor, “Wire-less micropower instrumentation for multimodal acquisition of elec-trical and chemical neural activity,” IEEE Trans. Biomed. Circuits Syst.,vol. 3, no. 6, pp. 388–397, Dec. 2009.

[15] C. M. Lopez, D. Prodanov, D. Braeken, I. Gligorijevic, W. Eberle, C.Bartic, R. Puers, and G. Gielen, “A multichannel integrated circuit forelectrical recording of neural activity, with independent channel pro-grammability,” IEEE Trans. Biomed. Circuits Syst., vol. 6, no. 2, pp.101–110, Apr. 2012.

[16] E. Greenwald, M. Mollazadeh, C. Hu, W. Tang, E. Culurciello, andN. V. Thakor, “A VLSI neural monitoring system with ultra-widebandtelemetry for awake behaving subjects,” IEEE Trans. Biomed. CircuitsSyst., vol. 5, no. 2, pp. 112–119, Apr. 2011.

[17] A. Rodríguez-Pérez, J. Ruiz-Amaya, M. Delgado-Restituto, and Á.Rodríguez-Vázquez, “A low-power programmable neural spike detec-tion channel with embedded calibration and data compression,” IEEETrans. Biomed. Circuits Syst., vol. 6, no. 2, pp. 87–100, Apr. 2012.

[18] A. C. M. Rijn, A. Peper, and A. Grimbergen, “High quality recording ofbioelectric events. II: A low-noise, low-power multichannel amplifierdesign,” Med. Biomed. Eng. Comput., vol. 28, pp. 389–397, 1990.

[19] L. Fay, V. Misra, and R. Sarpeshkar, “A micropower electrocardio-gram amplifier,” IEEE Trans. Biomed. Circuits Syst., vol. 3, no. 5, pp.312–320, Oct. 2009.

[20] B. Winter and J. Webster, “Driven-right-leg circuit design,” IEEETrans. Biomed. Eng., vol. BME-30, no. 1, pp. 62–66, Jan. 1983.

[21] J. C. Huhta and J. G. Webster, “60-Hz interference in electrocardiog-raphy,” IEEE Trans. Biomed. Eng., vol. BME-20, no. 2, pp. 91–101,Mar. 1973.

[22] J. L. Bohorquez, M. Yip, A. P. Chandrakasan, and J. L. Dawson, “Abiomedical sensor interface with a sinc filter and interference cancel-lation,” IEEE J. Solid-State Circuits, vol. 46, no. 4, pp. 746–756, Apr.2011.

[23] C. Ma, P. Mak, M. Vai, P. Mak, S. Pun, W. Feng, and R. P. Martins,“A 90 nm CMOS bio-potential signal readout front-end with improvedpowerline interference rejection,” in Proc. IEEE Int. Symp. CircuitsSyst., May 2009, pp. 665–668.

[24] J. Piskorowski, “Digital notch filter with time-varying quality factorfor the reduction of powerline interference,” in Proc. IEEE Int. Symp.Circuits Syst., May 2010, pp. 2706–2709.

[25] Y. Bai, W. Chu, C. Chen, Y. Lee, Y. Tsai, and C. Tsai, “Adjustable60 Hz noise reduction by a notch filter for ECG signals,” in Proc. 21stInstrumentation and Measurement Technology Conf., May 2004, pp.1706–1711.

[26] P. S. Hamilton, “A comparison of adaptive and nonadaptive filters forreduction of power line interference in the ECG,” IEEE Trans. Biomed.Eng., vol. 43, no. 1, pp. 105–109, Jan. 1996.

[27] X. Qian, Y. P. Xu, and X. Li, “A CMOS continuous-time low-passnotch filter for EEG systems,” Analog Integr. Circuits Signal Process.,vol. 44, no. 3, pp. 231–238, Sep. 2005.

[28] R. Schaumann, M. S. Ghausi, and K. R. Laker, Design of Analog Fil-ters: Passive, Active RC, and Switched Capacitor. Englewood Cliffs,NJ, USA: Prentice-Hall, 1990, ch. 5/7.

[29] B. K. Casper, D. J. Comer, and D. T. Comer, “An integrable 60-HzNotch Filter,” IEEE Trans. Circuits Syst. II, Analog Digit. SignalProcess., vol. 46, no. 1, pp. 74–77, Jan. 1999.

[30] H. Alzaher, N. Tasadduq, and Y. Mahnashi, “An Integrated Active-RCPowerline Notch Filter for Biopotential Acquisition Devices,” in Proc.Int. Conf. Biomed. Elect. and Devices, Feb. 2012, pp. 65–70.

[31] C. Ling and M. Luo, “Research of the ASIC used for EEG signal de-tecting,” in Proc. Int. Conf. Anti-Counterfeiting, Security and Identifi-cation, 2008, pp. 316–319.

[32] D. J. Comer, D. T. Comer, B. K. Casper, and D. S. Korth, “A LowFrequency, Continuous-Time Notch Filter Using Gm-C Circuits,” Int.J. Electron., vol. 86, no. 11, pp. 1349–1357, Nov. 1999.

[33] C. L. Hsu, M. H. Ho, Y. K. Wu, and T. H. Chen, “Design of lowfre-quency low-pass filters for biomedical applications,” in Proc. Asia Pa-cific Conf. Circuits and Systems, 2006, pp. 690–695.

[34] C. Ma, P. Mak, M. Vai, P. Mak, S. Pun, W. Feng, and R. P. Martins,“Frequency-bandwidth-tunable powerline notch filter for biopotentialacquisition systems,” Electron. Lett., vol. 45, no. 4, pp. 197–199, Feb.2009.

[35] W. M. C. Sansen and P. Van Peteghem, “An area-efficient approach tothe design of very-large time constants in switched-capacitor integra-tors,” IEEE J. Solid-State Circuits, vol. SC-19, no. 5, pp. 772–780, Oct.1984.

[36] K. Nagaraj, “A parasitic-insensitive area-efficient approach to realizingvery large constants in switched-capacitor circuits,” IEEE Trans. Cir-cuits Syst., vol. 36, no. 9, pp. 1210–1216, Sep. 1989.

[37] N. A. Radev and K. P. Ivanov, “Comparative analysis of two gain- andoffset-compensated very large time constant switched-capacitor inte-grators,” in Proc. Int. Conf. Microelectronics, Oct. 2000, pp. 51–54.

[38] A. Durham, W. Redman-White, and J. Hughes, “High-linearity contin-uous time filter in 5-V VLSI CMOS,” IEEE J. Solid-State Circuits, vol.27, no. 9, pp. 1270–1276, Sept. 1992.

[39] G. Roberts and A. Sedra, “All current-mode frequency selective cir-cuits,” Electron., Lett., vol. 25, no. 12, pp. 759–761, June 1989.

[40] H. A. Alzaher and M. Ismail, “Digitally tuned analogue integrated fil-ters using R-2R ladder,” Electron., Lett., vol. 36, no. 15, pp. 1278–1280,Jul. 2000.

[41] H. A. Alzaher, H. O. Elwan, and M. Ismail, “A CMOS highly linearchannel-select filter for 3G multistandard integrated wireless re-ceivers,” IEEE J. Solid-State Circuits, vol. 37, no. 1, pp. 27–37, Jan.2002.

[42] R. Sallen and E. Key, “A practical method of designing RC active fil-ters,” IRE Trans. Circuit Theory, vol. CT-2, pp. 74–85, 1955.

[43] T. Delyiannis, “High-Q factor circuit with reduced sensitivities,” Elec-tron. Lett., vol. 4, no. 26, p. 577, Dec. 1968.

[44] J. Friend, “A single operational amplifier biquadratic filter section,”IEEE ISCT Digest Tech. Papers p. 189, 1970.

[45] J. Kerwin W, P. Huelsman, and R. Newcomb, “State-variable synthesisfor insensitive integrated circuit transfer functions,” IEEE J. Solid-StateCircuits, vol. 2, no. 3, pp. 87–92, Sep. 1967.

[46] J. Tow, “Active RC filters—A state-space realization,” Proc. IEEE, vol.56, no. 6, pp. 1137–1139, Jun. 1968.

[47] P. Gray and R. Meyer, Analysis and Design of Analog Integrated Cir-cuits, 5th ed. Hoboken, NJ, USA: Wiley, 2009.

[48] E. López-Morillo, R. G. Carvajal, H. ElGimili, A. Lopez-Martin, J.Ramirez-Angulo, and E. Rodríguez-Villegas, “A very low-power classAB/AB op-amp based sigma-delta modulator for biomedical applica-tions,” in Proc. Midwest Symp. Circuits and Systems, 2006, vol. 2, pp.458–462.

[49] H. Alzaher, A. Hussein, and N. Tasadduq, “Dual-mode bluetooth/wire-less local area network channel-select filter for direct conversion re-ceivers,” IET Circuits, Devices Syst., vol. 5, no. 3, pp. 189–195, May2011.

[50] H. Khorramabadi, M. Tarsia, and N. Woo, “Baseband filters for IS-95CDMA receiver applications featuring digital automatic frequencytuning,” in Proc. Int. Solid State Circuits Conf., San Francisco, CA,USA, 1996, pp. 172–173.

[51] S. Lee and C. Cheng, “Systematic design and modeling of a OTA-Cfilter for portable ECG detection,” IEEE Trans. Biomed. Circuits Syst.,vol. 3, no. 1, pp. 53–64, Feb. 2009.

[52] Chart-O-Matic: View Signals and Annotations, PhysioNet, created byMIT Room E25-505A [Online]. Available: http://www.physionet.org/cgi-bin/chart

Hussain A. Alzaher received the B.S. and M.S.degrees (Hons.) from the King Fahd University ofPetroleum & Minerals (KFUPM), Dhahran, SaudiArabia, in 1994 and 1997, respectively, and thePh.D. degree (Hons.) from The Ohio State Univer-sity, Columbus, OH, USA, in 2001.

He is Full Professor of electrical engineeringat KFUPM. His research interests include appli-cations of electronic circuit techniques for longand short range wireless communications such asmultistandards mobile phones, Bluetooth, WLAN,

and WiMAX. His recent interests include designs of very-low frequencyfilters for biomedical instrumentation systems and baseband chain for DigitalVideo Broadcasting-Handheld (DVB-H). He is the author or coauthor ofapproximately 60 journal papers.



Dr. Alzaher was the recipient of the 2003 Showman Award for the best YoungArab Researchers.

Noman Tasadduq received the B.S. and M.S. de-grees in electrical engineering from NED University,Karachi, Pakistan, and the King Fahd University ofPetroleum & Minerals (KFUPM), Dhahran, SaudiArabia, respectively.

Currently, he is a faculty member in the ElectricalEngineering Department at KFUPM. His main fieldsof interest are analog signal processing circuits, cur-rent mode circuits, and analog filter design.

Yaqub Mahnashi received the B.S. and M.S. degreesin electrical engineering from the King Fahd Univer-sity of Petroleum & Minerals (KFUPM), Dhahran,Saudi Arabia, in 2008 and 2012, respectively.

Currently, he is a faculty member in the ElectricalEngineering Department at KFUPM. His main fieldsof interest are analog filter design and analog circuittechniques for biomedical applications.

Documents

A Highly Linear Fully Integrated Powerline Filter for Biopotential Acquisition Systems