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Electric Power Systems Research 81 (2011) 1132–1143
Contents lists available at ScienceDirect
Electric Power Systems Research
journa l homepage: www.e lsev ier .com/ locate /epsr
four-leg unified series–parallel active filter system for periodicnd non-periodic disturbance compensation
ehmet Ucar, Sule Ozdemir, Engin Ozdemir ∗
epartment of Electrical Education, Technical Education Faculty, Kocaeli University, Umuttepe Kampus, Izmit 41380, Kocaeli, Turkey
r t i c l e i n f o
rticle history:eceived 10 August 2010eceived in revised form 3 December 2010ccepted 4 January 2011vailable online 22 January 2011
a b s t r a c t
This paper presents a three-phase four-leg (3P4L) unified series–parallel active filter (USPAF) system,compensating for both periodic and non-periodic disturbances using a generalized non-active powertheory (GNAP) based control strategy. The 3P4L USPAF system is realized by the integration of series andparallel active filters (AFs), composed of the two 3P4L voltage source inverters (VSIs) sharing a commondc-link capacitor. The GNAP theory was implemented previously in the parallel AF. In this study, the
eywords:ctive filterour-leg voltage source inverterarmonicson-periodic
theory is proposed for the 3P4L USPAF system to compensate non-sinusoidal periodic and non-periodiccurrents and voltages. Distorted source voltages and unbalanced non-linear load currents compensationwere verified simultaneously through the 3P4L USPAF system experimental prototype. Sub-harmonicand stochastic non-periodic current/voltage compensations were analyzed through simulations withMatlab/Simulink software. The simulation and experimental results showed that the theory proposed
m wariodi
ub-harmonicnbalance
for the 3P4L USPAF syste(3P4W) systems under pe
. Introduction
In recent years, the increasing use of power electronic devicesnd unbalanced/non-linear loads has led to the generation of non-inusoidal periodic and non-periodic current/voltage disturbancesn electrical power systems. Generally, power electronic convertersenerate harmonic components with frequencies that are integerultiplies of the line frequency. However, in some cases, such as
ine commutated three-phase thyristor based rectifiers, arc fur-aces and welding machines are typical loads, the line currentsay contain both frequency lower than the line frequency (sub-
armonic) and frequency higher than the line frequency (stochasticon-periodic, the wave-shape and amplitude are constantly chang-
ng) components but not integer multiple of the line frequency1–5]. These waveforms are considered as non-periodic, although
athematically the currents may still have a periodic waveform,ut in any event, the period of the currents is not equal to the periodf the line voltage [1,2]. The non-periodic components can occur asell in the source voltage. The effects of non-periodic components
f current and voltage are similar to that caused by harmonics.hey may contribute power loss, disturbances, measurement errorsnd control malfunctions, thus degradation of the power quality inistribution systems [2].
∗ Corresponding author. Tel.: +90 262 3032248; fax: +90 262 3032203.E-mail addresses: [email protected], [email protected]
E. Ozdemir).
378-7796/$ – see front matter © 2011 Elsevier B.V. All rights reserved.oi:10.1016/j.epsr.2011.01.001
s applicable to non-active power compensation in three-phase four-wirec and non-periodic disturbances.
© 2011 Elsevier B.V. All rights reserved.
Unified series–parallel active filter (USPAF) systems have beenwidely studied to compensate the disturbances of source voltageand load current simultaneously. The USPAF system consists of aseries active filter (AF) and a parallel AF combined with a com-mon dc-link. The parallel AF compensates the current disturbancesand regulates the dc-link voltage, while the series AF compensatesthe voltage disturbances [6–9]. Fig. 1 shows the general power cir-cuit configuration of the USPAF system. The USPAF structures areused in the literature such as two three-phase three-leg (3P3L)voltage source inverter (VSI) with split capacitor [7,10–14], a 3P3LVSI for the series AF and a three-phase four-leg (3P4L) VSI forthe parallel AF [15,16] and two 3P4L VSI topology [17] for solv-ing power quality disturbances at the point of common coupling(PCC).
The conventional 3P3L VSI with split capacitor topology has aneutral line directly connected to the midpoint of the dc-link. Thistopology has been widely used in the USPAF systems due to itssimplicity to compensate the zero-sequence components in thethree-phase four-wire (3P4W) power system. However, this topol-ogy needs two capacitors and an extra control loop to maintain azero voltage error difference between both the capacitor voltages,resulting in a more complex control loop to maintain the dc-linkvoltage at constant level [16]. In addition, it has poor dc-link volt-
age utilization performance and requires the dc-link voltage to bemaintained to be at least twice as large as the peak value of theinverter nominal output voltage [17]. Furthermore, due to the rip-ple voltage of the dc-link caused by harmonic power resulting fromthe negative and zero-sequence currents, the capacitance of the dc-
M. Ucar et al. / Electric Power Systems
3∼
Source
Sensitive loads
iLiS iPF
+vSvSF vL–
Series AF Parallel AF
USPAF system
Cdc
LS
LSF LPF
LL
LL
Non-linear loads
Vdc
CSF RSF CPF RPF
N1/N2
Other non-linear loads
PCC
ls
Tsslctcvttataph
vlstDdwlcctbsdoUvplp
2
btUapd
does not change the compensation results as long as it is an integral
Fig. 1. General power circuit configuration of the USPAF system.
ink should be large enough to satisfy the permitted ripple currentpecification of the dc-link capacitor [18].
In this paper, two 3P4L VSI based USPAF system is proposed.he 3P4L VSI topology uses an additional leg to control the zero-equence component compared to the conventional 3P3L VSI withplit capacitor topology. The zero-sequence component is circu-ated in the system via the fourth leg, the dc-link voltage oscillationsan be made small [18]. Therefore, smaller dc-link capacitor thanhose of the conventional topology is required and the dc-linkapacitor voltage balancing algorithm is not necessary. The dc-linkoltage utilization performance of the 3P4L topology is better ando obtain a specific output voltage, a lower dc-link voltage thanhe conventional topology suffices [17,19]. This also provides andvantage for combined operation of the proposed 3P4L USPAF sys-em and distributed generation system such as photovoltaic arraysnd wind turbines, which is connected to the dc-link, due to com-ensate voltage interruption, as well as voltage sag, voltage swell,armonics and reactive power.
In the previous studies, the control algorithms for current andoltage compensators were often based on the assumption that theoad currents and the source voltages were periodic. The USPAFystems have been applied to the compensation of voltage fluctua-ions related non-periodic waveform in recent years [6,10,20–22].ifferent non-active power theories in the time domain have beeniscussed in [23]. The generalized non-active power (GNAP) theoryas applied for the compensation of the periodic and non-periodic
oad current with the parallel AF [24–26], the static synchronousompensator (STATCOM) [27] and voltage and current unbalanceompensation using an active filter [28]. The theory does not specifyhe characteristics of the voltage and current, they can theoreticallye any waveshape. The main objective of this paper is to compen-ate the non-sinusoidal periodic and non-periodic current/voltageisturbances using the 3P4L USPAF system based on the GNAP the-ry. The simulation and experimental results show that the 3P4LSPAF system using the proposed theory can regulate the loadoltage, compensating the source voltage harmonics and the non-eriodic voltage components while simultaneously eliminating the
oad current harmonics, unbalance and non-periodic current com-onents.
. Generalized non-active power theory
The instantaneous non-active power theory was first presentedy Fryze [29] for periodic (but non-sinusoidal) waveforms in theime domain. The GNAP theory [25] implemented on the 3P4L
SPAF system is based on Fryze’s idea of non-active power andn extension of the theory proposed in [30] for periodic and non-eriodic waveforms in the time domain. In this paper, all vectors areenoted by lower case bold letters. Voltage vector v(t) and currentResearch 81 (2011) 1132–1143 1133
vector i(t) in an m-phase system,
v(t) = [v1(t), v2(t), . . . , vm(t)]T , (1)
i(t) = [i1(t), i2(t), . . . , im(t)]T . (2)
The instantaneous power p(t) and the average power P(t), isdefined as the average value of the instantaneous power p(t) overthe averaging interval [t − Tc, t], that are:
p(t) = vT (t)i(t) =m∑
k=1
vk(t)ik(t), (3)
P(t) = 1Tc
∫ t
t−Tc
p(�)d�. (4)
In Eq. (4), the averaging time interval Tc can be chosen manu-ally for different cases such as a periodic system with harmonics, aperiodic system with sub-harmonics, and a non-periodic system. Aspecific value of Tc can be chosen to fit the application or to achievean optimal result for each case. The instantaneous active currentia(t) and the instantaneous non-active current in(t) are given inEqs. (5) and (6) respectively.
ia(t) = P(t)
V2p (t)
vp(t) (5)
in(t) = i(t) − ia(t) (6)
In Eq. (5), voltage vp(t) is the reference voltage which is chosenon the basis of the characteristics of the system and the desiredcompensation results. Vp(t) is the corresponding rms value of thereference voltage vp(t), that is:
Vp(t) =
√1Tc
∫ t
t−Tc
vTp(�)vp(�)d�. (7)
The instantaneous active power pa(t) and the instantaneousnon-active power pn(t) are defined by following equations,
pa(t) = vT (t)ia(t) =m∑
k=1
vk(t)iak(t), (8)
pn(t) = vT (t)in(t) =m∑
k=1
vk(t)ink(t). (9)
The average active power Pa(t) and average non-active powerPn(t) are defined by averaging the instantaneous powers over timeinterval [t − Tc, t], that are:
Pa(t) = 1Tc
∫ t
t−Tc
pa(�)d�, (10)
Pn(t) = 1Tc
∫ t
t−Tc
pn(�)d�. (11)
The rms values of the system voltage, the active current ia(t),the non-active current in(t) and the current i(t) are given in [25].The apparent power S(t), the apparent active power Pp(t), and theapparent non-active power Q(t) are also defined based on the rmsvalues of the voltage and currents. In the GNAP theory, the stan-dard definitions for an ideal three-phase sinusoidal power systemuse the fundamental period T to define the rms values, the averageactive power and the non-active power. The averaging interval Tc
multiple of T/2, where T is the fundamental period of the system,if there are only harmonics in the load current in the periodicdisturbances conditions. The non-active current is completely com-pensated and purely sinusoidal source current with unity power
1134 M. Ucar et al. / Electric Power Systems Research 81 (2011) 1132–1143
Table 1Tc selections for different compensation objectives [25].
Compensation objective vp Tc Resulting active current ia(t)
Single-phase or polyphase reactive current v T/2 or T Unity power factor and sinusoidal for sinusoidal vS
Single-phase or polyphase reactive current and harmonic current vf T/2 or T Sinusoidal regardless of vS distortionTnnn
fnnrsS
itscctadlafaTi
3
VcA
3
sda
Fwmptr
v
the desired load voltage vS1+. Then, the compensation referencevoltages (v∗
SFa, v∗SFb
,v∗SFc) of the series AF are derived by Eq. (20).
va(t) = P(t)2
ip(t) (18)
Instantaneous reactive power for polyphase system vNon-periodic disturbance current vf
Sub-harmonic current vf
Stochastic non-periodic current vf
actor is achieved. However, in other cases, if the disturbance ison-periodic such as a three-phase load with sub-harmonics, or aon-periodic load, Tc has significant influence on the compensationesults, and the power and energy storage rating of the compen-ator’s components [27]. The Tc selection procedure is explained inections 4.2 and 4.3 under these conditions.
The choice of the time averaging interval Tc is also significantn the energy storage design consideration of the 3P4L USPAF sys-em. Choosing a longer Tc results in a smoother source current withmaller amplitude; however, this requires that the compensatorurrent increase as well as the energy storage requirement of theompensator. If Tc is large enough, increasing Tc further will notypically improve the compensation results significantly. Gener-lly, there is no need to increase Tc to a larger value as the smallecrease in total harmonic distortion (THD) is often not worth the
arger capital costs (higher ratings of the compensator componentsnd therefore higher capital expenses) [25]. This depends on therequency of the non-periodic or periodic signal. Tc is identifieds an offline process based on the specific application as given inable 1. Also, vp(t) can be source voltage vS(t) itself or vf (t), whichs the fundamental component of vS(t), as shown in Table 1.
. Control of the 3P4L USPAF system
The proposed 3P4L USPAF system is realized with two four-legSIs with a dc-link capacitor and the GNAP theory based voltage andurrent control strategies. This theory, used at first for the parallelF, is now proposed for the 3P4L USPAF system.
.1. Series AF control strategy
The series AF, which uses 3P4L VSI, control block diagram ishown in Fig. 2. In this control strategy, the positive sequenceetector generates auxiliary control signals (ia1+, ib1+, ic1+) used asreference current ip(t) for the GNAP theory.
Block diagram of the positive sequence detector is shown inig. 3. The fundamental frequency ω1 (2�50) is used in a sineave generator to produce sin(ω1t) and cos(ω1t) signals at unityagnitude. The source voltages (vSa, vSb, vSc) are input of the
ositive-sequence detector. These voltages are transformed into
he synchronous dq reference frame by using Eq. (12) with theeference frame rotating at the fundamental frequency ω1.Sdq = Tdqabc
vSabc, (12)
vS
Positive sequencedetector (Fig. 3)
Sinusoidal load
voltage calculation
(18) (Fig. 4)i1+
Carrier-based PWM
voltage controller (Fig. 5)
2
Vam(19)
X
X
÷
vSF
*SFv
*LmV
∑
+vS1+
1
va QSF−
Fig. 2. The series AF control block diagram.
c → 0 Instantaneously unity power factor for polyphase system(T/2) Reduced amplitude and near sine wave with unity power factorT Pure sine wave or smoothed sine wave with unity power factorT Smoothed sine wave with near power factor
where the transformation matrix is shown in Eq. (13).
Tdqabc
= 23
[sin(ω1t) sin(ω1t − 120◦) sin(ω1t + 120◦)
cos(ω1t) cos(ω1t − 120◦) cos(ω1t + 120◦)
]⎡⎢⎣
va
vb
vc
⎤⎥⎦(13)
By this transform, fundamental positive sequence component,which is transformed into dc quantities in d and q axes, can beextracted by Eq. (14) and then transformed back into the abc refer-ence frame using Eq. (15).
v̄d,q = 1Tc
∫ t
t−Tc
vd,q dt (14)
vabc1+ = Tabcdq v̄d,q; (Tabc
dq = Tdqabc
−1) (15)
As shown in Fig. 3, the v1+(t) is divided by its amplitude Vdqmusing Eq. (16) and the output signals of the positive-sequencedetector (ia1+, ib1+, ic1+), have unity amplitude and in phase withthe fundamental positive-sequence component of the source volt-ages (vSa1+, vSb1+, vSc1+) are obtained. Also, effective value of thisreference current ip(t) is given in Eq. (17).
Vdqm =√
v̄2d + v̄2
q (16)
Ip(t) =
√1Tc
∫ t
t−Tc
iTp(�)ip(�)d� (17)
The average power is calculated given Eq. (4) by using the refer-ence currents and the source voltages. The sinusoidal load voltageva(t) is derived by using Eq. (18) [31]. Fig. 4 shows block dia-gram of the sinusoidal load voltage calculation which is appliedto the series AF control. As clearly shown in Fig. 2, the va(t) isdivided by its amplitude Vam using Eq. (19) and multiplied thedesired load voltage magnitude VLm* for converting the va(t) to
Ip (t)
vSdqabcT
(13) X
÷i1+
abcdqT
(15) (14)
dv
qvqv
v1+ (14)
Vdqm (16)
dv
Sine generator [sin(ω1t) cos(ω1t)]
Fig. 3. Block diagram of the positive sequence detector.
M. Ucar et al. / Electric Power Systems Research 81 (2011) 1132–1143 1135
X
X
÷2pI
∑ cT1/s1/
∑ cT1/s1/+
−
+−
vS
i1+
P
vaip
1
Tc delay
V
v
vtpIcrstw(
v
w
v
v
v
wew
3
s
Hysteresis current
controller
*PFi
XPI
iPF
vS1+
Sinusoidal source current
calculation (5)
iL
2
∑
+−
∑+
−
∑−+
vdc
1/Vm
ia in
ica
QPF
iPFn
*PFn
i
Tc delay
Fig. 4. Block diagram of the sinusoidal load voltage calculation.
am = 23
√v2
aa + v2ab
+ v2ac (19)
∗SF (t) = vS(t) − vS1+(t) (20)
The obtained reference voltages compared with the series AFoltages in a simple carrier-based PWM controller, shown in Fig. 5,he voltage references (v∗
an, v∗bn
, v∗cn) are generated for the three
hases, all with respect to “n” the center point of the fourth leg.n addition, voltage feed-forward (with a Kv gain) is added to theontroller for the purpose of good reference tracking. The fourth legeference voltage v∗
no is defined by Eq. (21) to achieve the optimumwitching sequence using an offset voltage concept [32]. Then, allhe inverter phase output terminal reference voltages (v∗
ao, v∗bo
, v∗co)
ith respect to the virtual dc-link midpoint of the split capacitor“o” point) are defined in Eq. (24).
∗no =
⎧⎪⎪⎪⎪⎨⎪⎪⎪⎪⎩
− v∗max2
, vmin > 0
− v∗min2
, vmax < 0
− v∗max + v∗
min2
, otherwise
(21)
here v∗min and v∗
max are defined as following,
∗min = min(v∗
an, v∗bn, v∗
cn), (22)
∗max = max (v∗
an, v∗bn, v∗
cn). (23)
∗xo = v∗
xn + v∗no, (24)
here x = a, b, c. Finally, the series AF switching signals QSF are gen-rated by comparing the modulation signals with triangular carrierave.
.2. Parallel AF control strategy
The parallel AF, which uses 3P4L VSI, control block diagram ishown in Fig. 6.
Fig. 5. Carrier-based PWM sc
dc voltagecontrol
*dcV
Fig. 6. The parallel AF control block diagram.
The average power is calculated given Eq. (4) by using loadcurrents and fundamental positive sequence source voltages(vSa1+, vSb1+, vSc1+) over the averaging interval [t − Tc, t]. Desiredsinusoidal source currents (iSa1+, iSb1+, iSc1+) are derived by using Eq.(5) and similar block diagram as shown in Fig. 4. The instantaneousnon-active current in(t) is calculated as in Eq. (6). An additionalactive current ica(t) also required to meet the losses in Eq. (25) isdrawn from the source by regulating the dc-link voltage vdc to thereference voltage V∗
dc. The PI controller is used to regulate the dc-
link voltage vdc as shown in Fig. 6. The error between the actual dcvoltage and its reference value is treated in the PI controller and theoutput is multiplied by a sinusoidal fundamental template of unityamplitude for each phase of the three phases. The compensationreference currents (i∗PFa, i∗
PFb, i∗PFc) of the parallel AF are obtained by
Eq. (26). The reference neutral current is obtained in terms of phasecurrents with Eq. (27). The reference currents are compared the par-allel AF currents and applied to hysteresis current controller. Thus,the parallel AF switching signals QPF are obtained.
ica(t) = vS1+Vm
[KP(V∗
dc − vdc) + KI
∫ t
0
(V∗dc − vdc) dt
](25)
i∗PF (t) = in(t) − ica(t) (26)
i∗PFn = −(i∗PFa + i∗PFb + i∗PFc) (27)
4. Simulation and experimental results
The proposed 3P4L USPAF system prototype is designed anddeveloped in laboratory to validate the GNAP theory. A three-phasedelta-star (�-Y) step-down transformer rated at 380/110 V and25 kVA is used to provide a 3P4W experimental system voltages. Inthis system, the line-to-neutral voltage is 110 V and the frequency
heme for the series AF.
1136 M. Ucar et al. / Electric Power Systems Research 81 (2011) 1132–1143
f the 3
iU
cptltowfa
bvnis
Fig. 7. Power circuit block diagram o
s 50 Hz. The power circuit and control block diagram of the 3P4LSPAF implementation is given in Figs. 7 and 8, respectively.
In Fig. 7, the non-linear load-1 (RL loaded three-phase halfontrolled thyristor rectifier with firing angle 30◦, RC loaded three-hase diode rectifier and RC loaded single-phase diode rectifier) ishe load group that requires ideal source voltages. Also, the non-inear load-2 (RC loaded three-phase diode rectifier) is connected tohe PCC to create source voltage distortion, and resembles the effectf other loads on a radial network. The three-phase source voltagesith distortion are synthesized by increasing system impedance
rom 59 �H to 4 mH and connecting the non-linear load-2 to PCCs clearly shown in Fig. 7.
The 3P4L USPAF system has 3P4W power circuit configuration
ased on the two four-leg VSI with a dc-link capacitor. The dc-linkoltage is adjusted at 350 V via parallel AF. The parallel AF is con-ected in parallel with the load while the series AF is connectedn series with the utility and the load through three single-phaseeries injection transformers. Both AFs are digitally controlled using
Fig. 8. Control block diagram of the 3P4
P4L USPAF system implementation.
a dSPACE DS1103 controller board. The dSPACE controller boardincludes a real-time processor and the necessary I/O interfacesto carry-out the control operation. This hardware supports thereal time interface (RTI) tool that allows programming via Mat-lab/Simulink. In this way, all the control circuit components areimplemented graphically within the Simulink environment.
In the 3P4L USPAF system, the load currents and the paral-lel AF currents are measured by utilizing TEG NA-50P hall-effectcurrent sensors, source voltages and the series AF voltages are mea-sured by utilizing the TEG NV-25P hall-effect based isolated voltagesensors and the dc-link voltage is measured by AD210 isolationamplifier for control and protection purposes. All the measuredsignals are scaled in the signal conditioning board which provides
the measured signals at the required voltage level for the dSPACEanalog digital converter (ADC) unit. The parallel VSI uses the vari-able frequency hysteresis current controller. The series VSI usesthe carrier-based PWM voltage controller. The proposed algorithmfor the USPAF system requires sampling time of 20 �s to executeL USPAF system implementation.
stems
tsI
SdgaspTaaicittlaplscU
loVVofppVppv
Dbtsm
TT
M. Ucar et al. / Electric Power Sy
he Matlab/Simulink generated C-codes in real-time. The generatedwitching signals are taken out of DS1103 with the help of digital/O channels.
The two four-leg VSI of the 3P4L USPAF system consists of eightEMIKRON SKM75GB128D dual-pack IGBT modules which areriven by 6-channel CONCEPT 6SD106EI and 2-channel 2SD106AIate drivers. The IGBT gate drivers have analog dead time gener-tion facility and also monitors the collector–emitter voltage forhort-circuit failure condition. The dc-link capacitance rating is pro-ortional to the maximum energy storage variation of the capacitor.he energy exchange is different for a given dc-link voltage vari-tion in particular application [26]. Different capacitance valuesre required to meet different compensation cases. Because thenstantaneous non-active power is zero at all times under periodiconditions, the current flowing into or out of the dc-link capacitors zero. Therefore, a small capacitor can meet the requirement ofhis case. Under non-periodic conditions, the average power P(t) isime varying. The net energy flowing in the dc-link capacitor is noonger zero over one cycle. The capacitor must have sufficient stor-ge capacity under these conditions to be able to absorb the loadower fluctuations. In the USPAF system design presented, the dc-
ink capacitor size needed is determined through Matlab/Simulinkimulations. The electrolytic capacitor 2350 �F (two 4700 �F, 450 Vapacitors connected in series) is connected dc-link of the 3P4LSPAF system.
The parallel AF currents and the series AF voltages are also uti-ized in the overcurrent and overvoltage protection board. If a faultccurs, all IGBT gate signals are set to zero level so that the twoSIs are disabled. Owing to the switching of the parallel and seriesSI’s, the compensating currents and voltages have unwanted high-rder harmonics. High-pass passive filters represented by RPF, CPF
or parallel AF and RSF, CSF for series AF in Fig. 7 are connected torevent the flow of switching harmonics into the PCC. The cou-ling inductances LPF and LSF are necessary to limit di/dt in theSI’s. Also, pre-charge resistors are utilized at the ac side of thearallel AF to limit the inrush current during the startup and by-assed after the dc-link capacitors are charged to their steady-statealue.
The experimental waveforms were recorded by TextronixPO3054 digital oscilloscope and the harmonic analyses were done
y Fluke 434 power quality analyzer. Fig. 9 illustrates the pho-ographs of the 3P4L USPAF system laboratory prototype. Table 2hows the circuit parameters used in the simulation and experi-ent.able 2he 3P4L USPAF system parameters.
Components Symbol Parameters
Power sourceVoltage, frequency VSabc , fs 110 V, 50 HzImpedance LS 59 �H
DC-linkCapacitors Cdc 2350 �F (2 × 4700 �F
connected in series)Reference voltage Vdc* 350 V
Parallel AFFilter LPF , RPF , CPF 2.5 mH, 5 �, 10 �FSwitching frequency fSWp 10 kHz
Series AFFilter LSF , RSF , CSF 0.7 mH, 5 �, 80 �FSwitching frequency fSWs 10 kHzInjection transformer N1/N2, S 2, 5.4 kVA
Non-linear load-1Three-phase thyristor rectifier LL , Ldc , Rdc 3 mH, 5.7 mH, 22 �Three-phase diode rectifier LL , Cdc , Rdc 3 mH, 100 �F, 30 �Single-phase diode rectifier LL , Cdc , Rdc 3 mH, 330 �F, 45 �
Non-linear load-2Three-phase diode rectifier Cdc , Rdc 8800 �F, 15 �
Research 81 (2011) 1132–1143 1137
The proposed 3P4L USPAF non-active power compensation sys-tem is simulated, and an experimental setup is also built, so thatdifferent cases can be studied in simulations or experiments. Thefirst case for periodic current and voltage compensation (Section4.1) is tested in the experimental setup and the last two casesfor (Sections 4.2 and 4.3) are simulated in Matlab/Simulink soft-ware since they are difficult to be carried out in an experimentalsetup. The IEEE–519 standard limits of 5% on THD in voltage or cur-rent is set on the THD of source currents and load voltages aftercompensation in this study.
4.1. Distorted source voltage and unbalanced non-linear loadcurrent compensation
The harmonic currents will produce voltage distortion thatcan affect other sensitive loads at PCC as they interact with theimpedance of an electrical distribution system. For compensationof periodic current and voltage with fundamental period T, usinga compensation period Tc that is a multiple of T/2 is enough forcomplete compensation [25]. The non-linear load-1, which requiresclean supply voltage, is connected 3P4W power system to cre-ate current unbalance and harmonics, and also the non-linearload-2 is connected to the PCC to create supply voltage distor-tion as shown in Fig. 7. The three-phase distorted load voltagesbefore compensation are demonstrated in Fig. 10(a). After com-pensation choosing the period as Tc = T/2, the three-phase sourcevoltages with distortion is compensated to the sinusoidal wave-forms shown in Fig. 10(b). The THD of the load voltages, whichaveraged 9.0% before compensation, is about 2.7% after compen-sation, which is well within the limits specified by IEEE–519. InFig. 10(c) there are shown from top to bottom, phase-a sourcevoltage, injected voltage, compensated load voltage and dc-linkvoltage. By means of the parallel AF current compensation, the THDvalues of the source voltages are decreased about from 9.0% to 7.3%.After the series AF compensation, the load voltage THD values aredecreased to about 2.7%. Also, the dc-link voltage has a desiredreference value. The three-phase unbalanced non-linear sourcecurrents and source neutral current before compensation are pre-sented in Fig. 10(d). After compensation choosing the period asTc = T/2, it is evident that the three-phase source currents are nearlysinusoidal with constant amplitude that is shown in Fig. 10(e).Moreover, the neutral line current is almost eliminated. The THDof the source currents, which was about 27.8% before compensa-tion, is about 4.2% after compensation. Fig. 10(f) illustrates fromtop to bottom phase-a load current, injected current, source cur-rent and dc-link voltage waveforms during a load change. It canbe seen that phase-a load current increases by about 25% after theload change and the parallel AF maintains the source current atsinusoidal wave and the dc-link voltage at set reference value of350 V.
The harmonic spectra of the phase-b load voltage beforeand after compensation are shown in Fig. 11(a) and (b) respec-tively. The harmonic spectra of the phase-b source current beforeand after compensation are shown in Fig. 12(a) and (b) respec-tively. The experimental compensation results are summarized inTables 3 and 4.
4.2. Sub-harmonic current and voltage compensation
Sub-harmonics (frequency less than the fundamental fre-
quency) are caused by arc furnaces, cycloconverters, welders,rectifiers feeding fluctuating or cyclic load, motors working withcyclic load and wind generators [33]. Sub-harmonic current/voltagecompensation using the GNAP theory, when the fundamental fre-quency is an odd multiple of the sub-harmonic frequency, the
1138 M. Ucar et al. / Electric Power Systems Research 81 (2011) 1132–1143
TS
Fig. 9. Photographs of the 3P4L USPAF experimental test setup. (a) Top view of the experiment desk. (b) Bottom view of the experiment desk.
able 3ummary of distorted source voltage compensation.
Load voltage (vL) Before compensation After compensation
RMS (V)Phase-a 101.2 110.3Phase-b 100.8 109.5Phase-c 101.6 109.7
THD (%)phase-a 9.6 2.8phase-b 8.4 2.6phase-c 9.2 2.7
Table 4Summary of load current compensation.
Source current (iS) Before compensation After compensation
RMS (A)Phase-a 12.4 16.4Phase-b 16.1 16.4Phase-c 12.1 16.3Neutral 5.2 1.2
THD (%)Phase-a 29.1 4.3Phase-b 26.7 4.0Phase-c 27.7 4.4
Unbalance (%)Neg. seq. 9.5 0.6Zero seq. 9.8 0.3
Power factor 0.94 0.99
M. Ucar et al. / Electric Power Systems Research 81 (2011) 1132–1143 1139
F r loada Sourcp
mpmboTm
ig. 10. Experimental results of distorted source voltage and unbalanced nonlineafter compensation. (c) Source, series AF, load and dc-link voltage waveforms. (d)arallel AF, source currents and dc-link voltage waveforms.
inimum Tc for complete compensation is 1/2 of the commoneriod of both fs and fsub. When fs is an even multiple of fsub, the
inimum Tc for complete compensation is the common period ofoth fs and fsub [26]. If Tc is chosen as an integral multiple of the peri-ds of all the frequencies in p(t), the average value P(t) is a constant.herefore, ia(t) is purely sinusoidal and in phase with the funda-ental component of v(t). If Tc is not chosen this way, there are
Fig. 11. Harmonic spectra of phase-b load voltage. (a)
current compensation. (a) Load voltages before compensation. (b) Load voltagese currents before compensation. (e) Source currents after compensation. (f) Load,
still sub-harmonic components in ia(t), the non-active componentis not completely eliminated.
In this study, source voltage and load current contain sub-harmonic of 10 Hz frequency and 20% amplitude are given in Eqs.(28) and (29) for phase-a. Fig. 13 shows the sub-harmonic cur-rent and voltage compensation simulation results. Three-phasesub-harmonic source voltage and three-phase sub-harmonic load
Before compensation. (b) After compensation.
1140 M. Ucar et al. / Electric Power Systems Research 81 (2011) 1132–1143
Fig. 12. Harmonic spectra of phase-b source current. (a) Before compensation. (b) After compensation.
Fig. 13. Simulation results of sub-harmonic voltage and current compensation. (a) Three-phase sub-harmonic source voltage waveforms. (b) Positive sequence detector outputwaveforms. (c) Three-phase load voltages after compensation. (d) Three-phase sub-harmonic load current waveforms. (e) Three-phase source currents after compensation.(f) DC-link voltage.
stems
c
v
i
tptpcdF
4
riccmpTfr
Fss
M. Ucar et al. / Electric Power Sy
urrent waveforms are shown in Fig. 13(a) and (d), respectively.
Sa = (155 + 15.5 sin(2�10t))sin(2�50t) (28)
La = (21 + 2.1 sin(2�10t))sin(2�50t) (29)
The positive sequence detector output waveforms under thisest case are shown in Fig. 13(b). The 3P4L USPAF system com-ensates the sub-harmonic component by choosing Tc = 2.5T. Thus,he three-phase voltage at the load terminals and the three-hase source current are almost sub-harmonic free and withonstant amplitude as presented in Fig. 13(c) and (e). Also, thec-link voltage which follows the reference value as shown inig. 13(f).
.3. Stochastic non-periodic current and voltage compensation
The arc furnaces may contain stochastic non-periodic cur-ents (frequency higher than fundamental frequency but not annteger multiple of it) because of their rapidly changing loadurrent characteristics [25,34]. Also, static frequency converters,ycloconverters, sub-synchronous converter cascades, induction
achines and fluctuating loads may cause the stochastic non-eriodic current/voltage waveform [34]. Theoretically, the periodof a non-periodic load is infinite [25]. The GNAP theory is valid
or voltage and current of any waveshape, and the non-active cur-ent can only be completely eliminated when Tc = t and t → ∞ (ia(t)
ig. 14. Simulation results of stochastic non-periodic voltage and current compensationequence detector output waveforms. (c) Three-phase load voltages after compensation.ource currents after compensation. (f) Load neutral current waveform. (g) Source neutra
Research 81 (2011) 1132–1143 1141
is the shape as and in phase with vp(t) so that unity power fac-tor is achieved). However, this is not practical in a power system,and Tc is chosen to have a finite value (1–10 times that of thefundamental period). Additionally, the non-active components inthese loads cannot be completely compensated by choosing Tc asT/2 or T, or even several multiples of T. Choosing that period asmay result in an acceptable both source current and load volt-age which are quite close to a sine wave. If Tc is large enough,increasing Tc further will not typically improve the compensa-tion results significantly [26]. In this work, phase-a source voltageand load current components are given in Eqs. (30) and (31)[34].
vSa = 155 sin(2�50t) + 11.6 sin(2�104t − 120◦)
+ 15.5 sin(2�117t − 120◦)
+ 7.7 sin(2�134t) + 7.7 sin(2�147t)
+ 31 sin(2�250t − 120◦) (30)
iLa = 21 sin(2�50t) + 6.3 sin(2�104t − 120◦)
+ 8.4 sin(2�117t − 120◦)
+ 4.2 sin(2�134t) + 4.2 sin(2�147t)
+ 10.5 sin(2�250t − 120◦) (31)
. (a) Three-phase stochastic non-periodic source voltage waveforms. (b) Positive(d) Three-phase stochastic non-periodic load current waveforms. (e) Three-phasel current after compensation. (h) DC-link voltage.
1142 M. Ucar et al. / Electric Power Systems Research 81 (2011) 1132–1143
(Cont
rsns
taTra3
Fig. 14.
Fig. 14 shows the stochastic non-periodic voltage and cur-ent compensation choosing the period as Tc = 5T. The three-phasetochastic non-periodic source voltage, the three-phase stochasticon-periodic load current and load neutral current waveforms arehown in Fig. 14(a), (d) and (f), respectively.
The positive sequence detector output waveforms under thisest case are shown in Fig. 14(b). After compensation, load voltages
nd source currents are balanced and almost sinusoidal with lowHD as clearly shown in Fig. 14(c) and (e). The source neutral cur-ent have been reduced considerably as presented in Fig. 14(g). Inddition, the dc-link voltage is maintained at the reference value50 V as shown in Fig. 14(h).inued ).
5. Conclusion
The increasing applications of non-linear and disturbing loadsconnected to the electrical power system are responsible for thepresence of periodic and non-periodic disturbances on the linecurrents and voltages. In this paper, the GNAP theory, whichis applicable to sinusoidal or non-sinusoidal, periodic or non-
periodic, balanced or unbalanced electrical systems, is presentedand applied to the 3P4L USPAF system. The theory is adapted fordifferent compensation objectives by changing the averaging inter-val Tc and applied to the 3P4L USPAF experimental setup systembuilt and tested in the laboratory. The dSAPCE DS1103 controller
stems
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M. Ucar et al. / Electric Power Sy
as used to implement the proposed approach in real-time. Theistorted source voltage with unbalanced non-linear load currentompensation was tested in the experiments. The sub-harmonicnd the stochastic non-periodic current and voltage compensationsere simulated in Matlab/Simulink. Simulation and experimental
esults verify the validity of the GNAP theory for the best compen-ation performance of non-sinusoidal periodic and non-periodicurrent/voltage disturbances with the 3P4L USPAF system.
cknowledgement
This work is supported by TUBITAK Research Fund (no:08E083).
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