20 the Essence of ThreePhase INTELEC2011

8/13/2019 20 the Essence of ThreePhase INTELEC2011

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2011 IEEE

Proceedings of the 33rd IEEE International Telecommunications Energy Conference (INTELEC 2011), Amsterdam, Netherlands,October 9-13, 2011.

The Essence of Three-Phase PFC Rectier Systems

J. W. Kolar

T. Friedli

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The Essence of Three-Phase P Recti er Systemsnn K n

r tr S st ms a rat r S , r9 r , S w t er a , kar @em.ee.et .

this paper three-phase PFC re ti er topologies withsi usoidal i put urre ts a d o trolled output voltage are derived fromk ow si gle-phase PFC re ti er systems a /or passive three-phasediode re ti ers. The systems are lassi ed i to hy rid a d fully a tivePWM oost-type or u k-type re ti ers a d their fu tio ality a d asio trol o epts are rie y des ri ed. This fa ilitates the u dersta di gof the operati g pri iple of three-phase PFC re ti ers starti g fromsi gle-phase systems a d orga izes a d ompletes the k owledge asewith a ew hy rid three-phase u k-type PFC re ti er topology de omiated as Re ti er. I additio a alyti al formulas for al ulati gthe urre t stresses o the power semi o du tors of sele ted topologiesare provided a d re ti er systems o eri g a high pote tial for i dustrialappli atio s are omparatively evaluated o er i g the semi o du torstresses the loadi g a d volume of the mai passive ompo e ts a dthe DM a d CM EMI oise level.

Fi ally ore topi s of future resear h o three-phase PFC re ti ersystems are dis ussed su h as the a alysis of ovel hy rid u ktype PFC re ti er topologies the dire t i put urre t o trol of u ktype systems the multi-o je tive optimizatio of PFC re ti er systemso er i g e ie y a d power de sity a d the i vestigatio of thesystem performa e se sitivity to semi o du tor a d passive ompo e tste h ology.

a d o verter PWM re ti er PFC re ti er PFCoost u k three-phase si gle-phase Re ti er Re ti eroverview review.

I . NTRODUCTION

The power electronics supply of high power electrical systems from

the three-phase ac mains is usually carried out in two stages, i.e.the mains ac voltage is rst converted into a dc voltage and thenadapted to the load voltage level with a dc dc converter with orwithout galvanic isolation (cf.Fig 1). O en only one direction ofpower ow has to be provided; furthermore, coupling to the mainsis typically implemented over only three conductors, i.e. without aneutral conductor.

In the simplest case, the recti cation can be done by unidirectionalthree-phase diode recti ers with capacitive smoothing of the outputvoltage and inductors on the ac or dc side (cf.Fig 2 . The lowcomplexity and high robustness (no control, sensors, auxiliary supplies or MI ltering) of this concept must, however, be weighedagainst the disadvantages of relatively high effects on the mains andan unregulated output voltage directly dependent on the mains voltagelevel.

The mains behavior of a power converter is characterized in generalby the power factor an or the fundamental current-to-voltagedisplacement angle< and the total harmonic distortion of the inputcurrent,T which are related by the equation

1 o J +T( )

The conduction state of the passive recti ers shown inFigs 2(a)and (b) is essentially determined by the mains line-to-line voltages,whereby only t o diodes carry current at the same time, except thecommutation intervals. This means that each diode of the positiveand negative bridge halves carries current only for one third ofthe mains period, i.e. for 120. Hence, the phase currents for

9 8 1 4 12 0 0/11 $26 00 2011 IEEE

industrially applicable values of the smoothing inductance show 60wide intervals with zero current that result in a relatively high lowfrequency harmonic content or aT 0%. In order to avoidvoltage distortions resulting from voltage drops across the inner(inductive) mains impedance or the excitation of resonances in thedistribution grid aT < % at rated power is o en required.For aircra on-board power supplies, relatively high inner mainsimpedances exist and thus even stricter limits, i.e. aT < % (cf.DO F, MI -4 ), have to be ful lled. This mains current qualitycan be achieved only by means of active Power Factor Correction(PFC) recti er systems.

It should be noted that for three-phase systems, the generally

used designationP Rect er is partly misleading, since passiverecti ers, for industrially used values of the smoothing inductance[XLl /uN0.0 . . .0.1 according toFigs 2( ) and ] , alreadyexhibit a high power factor of 0. . . .0. because of thelow phase-shi of the power-forming mains current fundamentalcomponent and the associated phase voltage (cf. ( ) for o 1andT 0%, [ ], [ ]). PFC recti ers hence achieve with regardto the mains current (at rated operation) above all a reduction of thecurrent harmonics but only a slight improvement in the power factor >0. at the rated operating point is typically targeted). further important aspect of the use of active (PFC) recti er

systems i s the possibility to control the output dc voltage to a constantvalue, independent of the actual mains voltage ( uropeUN r s00 V; S , Japan

UN r s200 V,

UN r sdenominates the

rms value of the mains line-to-line voltage). converter stage onthe output side (cf.Fig 1 can thus be dimensioned to a narrowvoltage range. The mains voltage range must be considered only forthe dimensioning of the recti er stage (the delivery of a given ratedpower, e.g. at half of the input voltage, leads to a doubling of theinput current that must be mastered by the power semiconductors,

PC )

P C C

b P C BC E

Fig. 1. Block diagrams of typical converter con gurations for supplyingelectrical loads from the three phase ac mains.a Three phase ac-dc converterwith non isolated dc-dc converter (e.g. for the coupling of dc distribution systems to the three phase mains or as a mains interface for high power isolatedloads, e.g. lighting systems).Three phase ac-dc converter with isolated dcdc converter (e.g. for telecom power supplies, welders, or induction heatingsystems) Three phase ac-dc converter and three phase dc-ac converter(inverter) without isolation (e.g. for variable speed drives).


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2P

Lb uCa) n

600 - -- --- --400 m ! m 20

0

200 0

4000 0 1 20c) (ms)

I1.0 U n

r - t -

/= 2

1m H

0.9C: 0 8

0.7

0 0

-

:f= 20 m0 0 1 0 20 0 2

x O

b

200

0

200

4000

0

L p

u C un

0

0

0 0ms

0 9

0 87A

iI , = H0 7

0 0

,,,,,,: 20mlH0 0 0 20 0J O

Fig. 2. Passive three phase diode recti cation.a S oothing inductors on the ac side. S oothing inductor on the dc side. and d Correspondinginput current wavefor s.e and Resultant global average valuep of the output voltagep and power factor at the input. (Si ulation para eters:rms line to line voltage N ,rm V, mains frequency N Hz, smooth ing capacitance on the dc side and moothing inductance H . . . H .)passive power components and the MI lter) or a relatively highand well-de ned voltage level is available for the generation of theoutput (load) voltage [cf.Fig l(c)].

The requirements placed on active PFC recti er systems may thusbe summarized as follows

sinusoidal input current according to regulations regarding themains behavior of three-phase recti er systems ( -3-if 16 ); in industry, however, typicallyindependentof the concrete application, aT < % isrequired (at the rated operating point);ohmic fundamental mains behavior ( o > . );regulated output voltage; depending on the required level ofthe output dc voltage relative to the mains voltage, a systemwith boost-, buck- or buck-boost-type characteristic has to be

provided;mastering of a mains phase failure, i.e. for interruption orvoltage collapse of one mains phase, continued operation atreduced power and unchanged sinusoidal current shape shouldbe possible;unidirectional power ow, perhaps with (limited) capability ofreactive power compensation. O en, because of the supplyof a purely passive load (e.g. telecom power supply), onlyunidirectional energy conversion has to be provided or as foraircra on-board power supplies, no feedback of energy intothe mains is permitted;compliance with speci cations regarding electromagnetic, especially conducted interference emissions by means of suitableMI ltering.

The designationthree-phase P rect er chosen in this paperimplies both sinusoidal mains current shaping and regulation of thedc output voltage. Here, it should be noted that an active harmoniclter [3] of lower rated power arranged in parallel to a passive

recti er system would also enable a sinusoidal mains current, butno regulation of the output voltage. ccordingly, because of thesystem-related advantage of a constant supply voltage of a load sideconverter, a PFC recti er system is o en preferred over active lteringdespite the larger implementation effort, i.e. the conversion of theentire output power.

Parallel to the development of single-phase PFC recti er circuits,numerous concepts for three-phase PFC recti er systems have beenproposed and analyzed over the last two decades. However, thetopological relationships between the circuits and a comprehensive

classi cation have received relatively little attention. Furthermore, thebasic function of the circuits was typically treated by space vectorcalculation, analogous to three-phase drive systems, which is notimmediately comprehensible on the basis of knowledge of dc powersupply technology or single-phase PFC recti er circuits.

The goal of the present work is hence to develop the conceptsof three-phase PFC recti ers, starting from known single-phasePFC recti er systems, and to explain as clearly as possible theirbasic function and control, without reference to analysis techniquesbeing speci c to three-phase converter concepts. Details of the PulseWidth Modulation (PWM) and a detailed mathematical analysis areomitted, i.e. only the operating range of the systems is clari ed withregard to output voltage and mains current phase angle. Furthermore,the dimensioning of the power semiconductors, the main passivecomponents, and the MI lter is brie y discussed.


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To keep matters short, the considerations remain limited to unidirectional systems and here to those circuits that come into questionwith regard to implementation effort for industrial applications orhave already found such applications. umerous, only theoreticallyinteresting circuit proposals of high complexity an or high component loading are hence not considered. In particular, no circuitsare discussed that fundamentally demand low frequency passivecomponents e.g. dimensioned for sixfold mains frequency. Passivesix- or twelve-pulse recti er systems [5] , and hybrid recti er circuitswith passive harmonic current injection networks [ ] , [ ] are thusalso not treated.

In the following, inS ction II, a comprehensive classi cationof unidirectional three-phase recti er systems is presented that forcompleteness also includes purely passive systems. For PFC recti er circuits, a division is made between circuits that are fullyactive and hybrid, i.e. partially mains-commutated, and partiallyself-commutated systems. With regard to the basic structure, phasemodular and direct three-phase systems are distinguished and subsequently treated in more detail inS ction III and S ction IV withreference to selected examples. part from systems with boost-typecharacteristic, buck-type PFC recti er systems are also discussed,

which were not considered in [8], but will be of special interest infuture in connection with the charging of electric vehicle batteries orthe supply of dc distribution grids. The ordering and complementationof the knowledge base of three-phase buck-type PFC recti er sy stemsleads to a new hybrid circuit concept(SWI Recti er) that ischaracterized by low complexity of the power circuit and control, andthus of particular interest for industrial applications . InS ction V, inthe sense of support for the dimensioning of the circuits, the currentstresses of the main circuit components are brie y summarizedin the form of simple analytical expressions, and the Differential

Mode ( M) and Common Mode (CM) MI ltering of the systemsdiscussed. Finally, inS ction VI, a comparative evaluation of selectedboost-type and buck-type PFC recti er systems is presented, which isintended as an aid for the choice of concepts in industrial developmentprojects. In conclusion, inS ction VII, with regard to future furtherincreasing requirements on the ef ciency and power density of thesystems, and the further spreading of active mains interfaces, researchsubjects in the eld of PFC recti er circuits are identi ed.

I I . LA IFICATION OF NIDIRECTIONAL HREE-PHA EECTIFIER SY TEM

In Fig 3, a classi cation of unidirectional three-phase recti ercircuits is shown that for completeness also includes purelypassivesystemswhich

contain no tu -off power semiconductors, i.e.work purely mains-commutated, andemploy low frequency, i.e. passive components for output voltage smoothing and mains current shaping and, where applicable,mains or auto-transformers for the phase-shi of several converter stages working in parallel or series (multi-pulse recti ercircuits).

Furthermore, since here only diode and not thyristor circuits areconsidered,

there is no possibility of output voltage regulation.n approximately sinusoidal mains current and/or partial elimina

tion of low frequency harmonics in the input current is thus onlyobtainable with m lti-pulse systems, i.e . for 12-, 1 - or 6-pulserecti er circuits. In industry, m lti-pulse recti ers, because of theirlow complexity and great robustness, are mainly used at high power>100 kW) as mains interfaces, where the supply is typically direct

e t h ee h se e t e ste st ss ve Systems Hy Systems ve F Syste s

ISingle D ode Bridge Rectifier Systems Multi-Pulse Rectifier Systems

DC Side Inductor (Part al) Trans o Auto ans Bas dAC Side Inductors AC o DC Side In e phase T ansfo mePassive rdHa onic Injection Passive Pu se Mu ip ication

Electron c Reactance BasedRectif er Systems

Singl Diod Bridg&DC Sid El ctron. Ind.Singl Diod Bridg&AC Sid El c ron. Ind. or Cap.Multi-Puls R i r Syst mEmployin Electron. Inter-p ase Trans . I

I Combination of Diode Rectifier Har onic Inject onPhase Modula Syste sand DC/DC Converter Systems Systems DirectT ee-P ase Syste s

Boost ype lSingle Diod Bridg&DC-DC Out u StageHa -Control d Diod

BridgMulti-Puls R cti r Sys m(Trans or Au otrans -Bas dwith DC /DC Ou u S ag mpJ.AC Sid or DC Sid Ind.

Passiv /Hybr. o Ac v 3dHa . In c N work V-R ti r or 6-R ti rBoost- o Buck-Typ or Uncontro d Ou u V-Arrang m nt wit Conv rt r

D od Bridg or Mu uls Sys m wi

3rdArti ial Star-Point Conn ction

Harmon c In . ( uls Mu 1 3 2-P as S o - rans -Bas dII pressed Input Cur ent mp essed nput Voltage

Buck ype (Boost ype) (Buck- ype)

S ng D od Bridg I I&DC-DC Ou u StagHa -Con oll d D odB dge

DCM C M I DVM I CVMSingle-Sw tc wo-L v l R t r S ngl Switc r Switc R cti er

R c er Yor6-Swi Converter Six Switc Recti erwo Switc Re ti eR r ee- eve Converte ( ENNA Recti erSix-Sw tch Converte

t

Fig. 3. Class i cation of (unidirectional) three phase recti er topologies into passive, hybrid, and active systems with boost or buck type characteristic. Foreach recti er subgroup, references to publications are given in [4], in which the corresponding circuit concept was presented rst or a detailed description isprovided. The boxes of the converter groups, discussed in this paper, are highlighted by shading. For details of recti er topologies not discussed in this papeplease see also [4].


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4from the medium voltage mains whose low inner impedance allowshigher input current harmonics to be accepted.

The coupling and/or partial integration of a passive recti er and ofan active circuit part implemented with power semiconductors thatcan be actively switched off, leads tohybrid rect er circuits.

These systems fundamentally allow a regulation of the outputvoltage and a sinusoidal control of the mains current; however, alimitation to voltage regulation is possible (e .g. in the case of a diodebridge with downstream dc dc converter) or to sinusoidal currentshaping (activelter-type3rdharmonic current injection, cf.S c IVor [9] [ 4]). Furthermore, low frequency lter components of passiverecti er systems may be replace /emulated by high- frequency PWMconverters of relatively low rated power ( lectronic Inductor [ 5],[ ]) , e. g. i n the sense of an increase o f the power density. With anac side arrangement of these electronic reactances, via a change in theinductance or capacitance value in operation, a limited possibility ofvoltage regulation exists (Magnetic nergy Recovery Switch (M RS)concept [ ], [ 8]).

3rd harmonicinjection concepts form a major group of hybridrecti er circuits. Here, current is injected by a passive or activeinjection network always into that phase which would not carry

current in case of conventional diode bridge recti cation. The currentwaveforms of the two other phases are shaped in such a way that as aresult, sinusoidal current ows in all phase s. The recti er function ofthese sy stems is implemented by a diode bridge on the input side. Theactive network for current shaping, injection, and voltage regulation,arranged on the dc side, may thus be considered essentially asa dc dc converter working on a time-varying (six-pulse) dc inputvoltage. The circuits are hence relatively simple, i.e. may be analyzedwithout speci c knowledge of three-phase converter systems andexhibit relatively low complexity, also regarding control. The essentialcharacteristics o f hybrid recti er circuits may thus be summarized asfollows

mains-commutated (diode circuits) and force-commutated, circuit sections implemented with power semiconductors that canbe actively switched off;low frequency an /or switching frequency passive components;output voltage regulation and/or sinusoidal mains current shaping by turn-off power semiconductors.

In the present work, only those hybrid recti er circuits are considered which exhibit regulated output voltageand sinusoidal mainscurrent, and exclusively switching frequency passive components.

Integration of turn-off power semiconductors into the bridge-legsof a passive system nally leads toactive P rect er systems.ssential features of these systems are

forced commutation (only for systems with impressed outputcurrent partly natural commutations occur, depending on theposition of the switching instant in the mains period);exclusively switching frequency passive components;regulated output voltag .

s described in more detail inS c IV, these systems exhibit ingeneral bridge topologies and here bridge-legs of same structure, i.e .phase symmet , and a similar con guration of the power semiconductors in the positive (connected to the positive output voltage bus)and negative bridge halves, i.e .bridge symmet .

part from these direct three-phase versions (cf.Fig 3 , however,an implementation is also possible via a combination of singlephase PFC recti er systems in star(Y)- or delta( )-connection[cf.Figs a) and ]. These phase-modular versions, however, leadto three individual dc output voltages, i.e. a single output voltage canonly be formed via isolated dc dc converters that are connected tothe recti er outputs. n advantage of the phase-modular version is

0 - - - - - - - - - -0

asBoost Type c r

00

'

:as

c y c

00 0 Fig. 4. Output voltage range of direct three phase PFC recti er systemswith boost or buck type characteristic in dependence of the rms line toline mains voltage NIIrm (considered mains voltage range:NIIrm

V . . . V) N;1 enominates the peak value of the line toi e voltage. n addition, the lwer output voltage limit of a single phase boost typePFC recti er system > / N, connected between a mains phaseand the mains neutral, is shown.

the possibility of realizing a three-phase system starting from existingalready developed single-phase systems. However, it must be takeninto account that with the star-connection a coupling of the phasesystem results. For the delta-connection, the high input voltage ofthe modules has to be considered, which is de ned by the mainsline-to-line voltage and not by the phase voltage.

part from the topological distinction, a classi cation of thesystems must also be carried out with regard to the available outputvoltage range, i.e . fundamentally into circuits with boost- or bucktype characteristic. s shown inFig 4, the lower or upper outputvoltage limits of the systems are de ned by the mains line-to-linevoltage. The voltage range, not covered by the two basic converter

forms, is usually realized in industry by means of a downstream dcdc converter with boost- or buck-type characteristic. lternatively, athree-phase extension of buck-boost [ 9] , Cuk- or S PIC-converters[ 0] could be used. Because of the high complexity of the resultingcircuit, however, this approach is only of theoretical interest and ishence not described in more detail.

With regard to phase-modular recti ers, it must be pointed out thatbuck-type converter systems enable a current shaping only in a partof the mains period [ ]. Hence, for sinusoidal mains current, a boostfunction must be provided which results in a lower limit of the outputvoltages of the modules.

ote Systems wi th galvanic isolation of the output voltage are nottreated in this paper. In many cases, isolation is achieved by a dc dcoutput stage at high frequency, or is required directly at the supply ofthe systems for voltage adaptation, e.g. for connection to the mediumvoltage level . lternatively, a transf rmer may be integrated directlyinto the recti er structure. Such high frequency isolated three-phaseac dc matrix converter concepts [ ] [ 4] , however, are characterizedby a relatively high complexity of the power circuit and modulationand hence are of limited importance for industrial applications.

III . PHA E- ODULAR ECTIFIERStarting from the basic circuits of symmetrical three-phase loads,

three-phase PFC recti er systems can be implemented by star- ordelta-connection of single-phase PFC recti ers. The phase-modularsystems thus formed will be termed in the following Y- or -recti ersaccording to the circuit structure. The phase modules here may exhibita conventional topology or could be implemented as bridgeless, i.e.


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Fig. 5. Block diagram of phase modular PFC recti er systems [25]-[27].aStar(Y) connectionY recti er and delta( ) connection recti er withoutput side isolated dc-dc converters. nstead of a common EM input lterfor all phases as used fora and , in terms of full modularity, also separateEM lters could be implemented for each recti er module.

dual-boost converters, or ac- switch converters (cf.Fig 11 . They maycontain an MI lter or advantageously, a three-phase MI ltercould be installed common to all phase systems.

A. Y-Rect erFig 6 shows the circuit topology of a Y-recti er with a bridgeless

topology of the phase modules and an equivalent circuit of the ac sidesystem part. If the MI lter is three-phase (not shown inFig 6and the star-point ' is not connected to an arti cial star-point,which could be formed e.g. by lter capacitors, a switching frequencyvoltageUN Noccurs between ' and the mains star-point

ccording toia ( )d UaN U N + UN Nib UbN (U5N+ UN N)

i ( )UN UcN + UN N

( )

the impression of the ac currents is via the differences of the mainsphase voltages and the voltagesUN i formed at theinput of the recti er stages, so that the star-point voltageUN Nwithconsideration of

results in

(. . .)a + b + 0

1UN N (U N +5N+ UcN ) .

(3)

(4)

Therefore, with a free star-point' a part of the recti er inputvoltageUN advantageously does not form a current ripple, so thatthe values of the boost inductance may be reduced compared to

5P

a la ,an

pb b

bnp

C cCcn

aN .N'

Fig. 6. a Basic structure of the Y recti er. Equivalent circuit of the acsystem part without the EM input lter.

a xed star-point for the same ripple amplitude. This advantage isgained at the expense of a CM voltageUN Nof the modules, whichrequires an appropriate CM lter.

s clearly shown byFig 6(b) and with regard to the fundamentalof the input voltage required for the impression of the input current,the same conditions are present for the phase modules as for single

phase PFC recti ers supplied from a mains phase. Therefore, despitethe high peak value of the line-to-line voltage of the uropean lowvoltage grid NIl 6 V an or N Il 00 V), the output

wt

I ac c u c

Fig. 7. a Time behavior of the instantaneous fundamental power of thephases of a Y recti er. Redundant switching states concerning the resultingvariation of the phase currents fora > , b < , < [valid withinN ], which can be used for balancing the dc output

voltages ,a ,b , of the phase modules .


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rr *o

o /3L U i r ; == - verage DFig. 8. Structure of the Y recti er control: superimposed control of the average value of the dc output voltagesU ,a , b , of the phase moduleswith subordinate control of the phase currents and 2 out of 3 balancing of the dc output voltages of the phase modules. Equal signal paths of all phasesare represented by double lines. The dependence of the formation of the voltages on the sign of the respective phase currents is considered by aninversion of the switching signals for negative phase currents.

voltage of the phase modules can be selected e.g. at, 00 Van or the power transistors may be realized with 600 V superjunction power MOSF Ts.

The control structure of the system is shown inFig 8 Thereference conductance* is de ned by the output voltage controllerK (s). Multiplication of*by the normalized mains phase voltagesleads to the phase current reference valuesi to be set by thesubordinate current controllersKr(s). With regard to the fundamentalinput current, the system thus behaves as a symmetrical ohmic loadwith resistancesR j * in star-connection. ccordingly, forasymmetrical mains, there occurs in a phase with lower voltagea lower current amplitude, or a reduced supply of power to the

respective output. This must be considered for setting the referencevalue of a downstream dc dc converter.Because of the phase-modular structure, three output voltages are

to be regulated. Therefore, the voltage control is split into two parts.On the one hand, the power drawn from the mains, i.e.* is de nedfrom the average control error of the three output voltagesU , . Onthe other hand, a balancing of the output voltages is implemented,whereby in each case only the phase with the highest positive andthe highest negative voltage value is taken into account [ 5], [ 8],[ 9]. s shown inFig 7 only for these phases, e.g.and c,a higherinstantaneous output power ow is present and hence the possibilityof changing the output voltage value. Dependent on the differencebetween ,aand , an offset of the reference phase currentvalues i is formed which, however, cannot be set by the phasecurrentcontrollers because of the free star-point' ia + ib + i 0is unalterable. The phase currents thus keep their sinusoidal shapeand the symmetry to the time axis . However, as could be shown by amore detailed analysis , fori >0, the switching state 100 is mainlyused i nstead of 011 for the formation of the voltagesU requiredfor current impression. Similarly, fori


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of asymmetry [3 ], [33]. The disadvantage of this concept, basicallyknown from the star-point formation in electrical networks, however,lies in the requirement of an additional inductive component ofrelatively large volume and weight.

In Fig 9(a), the demonstrator of a highly compact version ofa single-phase bridgeless PFC recti er system [cf.Fig 11(b)] isshown; Fig 9(b) shows a highly ef cient version of the samerecti er topology. Starting from these systems , Y-recti ers with powerdensities of up to kW/ or e ciencies of > % may berealized.

B. Rect erFor delta-connection of the PFC recti er modules [cf.Fig 5(b) ] ,

the subsystems are decoupled, in contrast to the star-connection (Yrecti er). The control can therefore be carried out, individually foreach subsystem, in the same way as for single-phase PFC recti ers.Balancing of the modules with respect to power consumption is ofadvantage in the sense of symmetrical loading of the mains, butis not absolutely necessary. However, the line-to-line voltage of themains appears at the input to the modules. Hence, a relatively highoutput voltage

, >v N, ,rm(typ.

,700 V . . .00 V

for the uropean low-voltage mains, taking into account voltagetolerances) or a h igh blocking capability of the power semiconductorshas to be provided. lternatively, the semiconductor blocking voltagestress could be halved by means of a three-level topology. lsoa buck converter could be placed in front of each boost converterstage, i.e. in each phase, a buck-boost converter with a commoninductor could be implemented. This would allow the output voltagelevel of the individual modules to be chosen similar as for the Yrecti er with 00 V [34] . Then, only the transistors of the buck stagehave to be designed for line-to-line voltages. However, an additionalpower transistor then lies in the current path, which leads to higherconduction losses.

t the input of the recti er stages of the -recti er modules, for

a two-level implementation of the boost output stages, voltages(5)

( j designates the switching state of the power transistorsjJ } are formed that, apart from the switching stateab b a 0, contain a switching frequency zero sequence

voltage component

U"b Ub+ UU U + UUca a+ U .

( )

s can be immediately seen via a delta-star transformation for the

formation of the phase currentsi only the voltagesUb aan /or the equivalent star-point phase voltages1 U"N U"b Uca1 'UN U"b1 'UcN Uca

with

are effective. The zero sequence voltage component1

U U"b + + ca

( )

(8

(9)

L +

'n+

b : on+

na

Fig. 10. a Basic structure of the recti er, with thyristor bridges at theinput of the modules to provide the nominal output power in case of phaseloss three level boost stages are employed to reduce the voltage stress of thepower semiconductors. Simpli ed ac side equivalent circuit with the zerosequence component of the input current ripples of the modules.

thus drives only a switching-frequency current within the deltacircuit. This means that for a three-phase MI lter, the modulation i sbest designed through suitable synchronization of the carrier signalsof the modules such thatU is maximized or a maximum fraction

of the switching-frequency input current ripple of the modulesis held within the delta circuit [ ], [35]. There then results aminimum ripple of the phase currentsi and the MI lter effort isminimized. However, this advantage should be weighed against fullmodularity/independence of the subsystems (also regarding switchingand modulation), that is obtainable only with the con guration of anindividual MI lter per module.

n essential advantage of the -recti er is the availability of thefull rated power even for a failure of one mains phase. For this

purpose, the modules must be connected as inFig lO(a) via threephase thyristor bridges to the mains, and the thyristor bridges, oninterruption of a mains phase, are switched over to the two remainingphases (cf. [ ], [34]). However, this concept is applicable only fora suitably high loading capacity of the remaining mains phases.

C. DiscussionPhase-modular systems allow the knowledge on single-phase PFC

recti er systems to be exploited relatively directly an /or allow forthe development of a three-phase PFC recti er system with low effort.However, this advantage can be realized only with a fully modularstructure, i.e. with the arrangement of an individual MI lter permodule, so that the modulation methods for the reduction of ripple inthe phase currents described above cannot be used. However, in any


9/28

case, a balancing of the modules is required to assure a symmetricalloading of the mains. Here, the additional effort for measurement andsignal processing required for the Y-recti er should be noted.

Basically, due to the modular structure, three individual dc outputvoltages are formed that only with the aid of downstream isolateddc dc converters can be employed for the supply of a single load.Furthermore, each module requires ltering of the power ow, whichpulsates with twice the mains frequency, i.e. electrolytic capacitorsof suitably high capacitance must be provided on the output side. Onthe other hand, the assurance of a mains-holdup to master the failureof a mains voltage half-cycle anyway requires a relatively high outputcapacitance. Furthermore, by division of the overall sy stem into threesubsystems, a compact construction is supported and the cooling ofthe power components is simpli ed.

The essential advantage of the Y-recti er is the lower voltage stressof the power semiconductors or the relatively low level of the outputdc voltages. However, a. direct coupling of the phase modules ispresent, which especially for mastering of a phase failure requires aclose co-ordination of the modules and nally a control circuit for theoverall system. Hence, the advantage of modularity cannot be usedfor the control. Industrially, the system will thus probably remain

of minor importance. In contrast, the modules of the -recti erwork in a decoupled manner, and via a relatively simple expansionof the circuit topology, the full rated power is available on phasefailure. The disadvantage of the relatively high output voltage and/orrequired blocking voltage capability of the power semiconductorsin the modules ought to be alleviated in future by the availabilityof 1200 V SiC power JF Ts or SiC power MOSF Ts. Then alsoan additional buck converter stage could be implemented for eachmodule, which enables to maintain the output voltage level given bysingle-phase PFC systems and therewith the use of already developeddc dc converter circuits. On the whole, then, an excellent potentialfor industrial application of this system can be discerned.

IV. IRECT HREE-PHA E PWM ECTIFIER O OLOGIE

In the following, the topologies of important direct three-phaseboost- and buck-type recti er systems will be derived and brie y described with regard to their basic function and control. Boost sy stemswill be developed by three-phase extension of known single-phaseboost-type PFC circuits (cf.Fig Han /or Fig. in [30] ); the circuitstructures of the buck-type systems follow by extension of passivethree-phase diode recti ers with tu -off power semiconductors.

In general, the de nition of three-phase converter topologies,should be under consideration of a high level of symmetry of theresulting circuit structure. Because of the identical nature of thephases of the supplying mains (pure ac voltages of same shape andamplitude), it is obvious to provide the same structure for the circuitbranches connected to the phase terminals (phase symmetry). On the

other hand, the symmetry of the positive and negative half-cycles ofthe ac input phase currents to be impressed by the recti er systemnaturally leads to an identical arrangement of power semiconductorsin the positive and the negative half of the phase-legs. In connectionwith the dc voltage to be formed, corresponding topologically to apositive and a negative output terminal, there thus results a threephase bridge topology with symmetrical positive and negative bridgehalves (bridge symmetry). For a dc dc converter, connected to therecti er output, this symmetry does not necessarily need to bemaintained. Here, a decision could be made e.g. by analysis andcomparison of the CM MI emissions and the conduction losses ofa symmetrical or asymmetrical topology.

It should be noted that recti er systems which violate one or bothsymmetry requirements, e .g . with the aim of reducing the implementation effort, can also enable the impress ion of mains ac currents and

1

n

n

Fig. 11. Single phase boost type PFC recti er systems the three phaseextensions of the circuits leads to direct three phase hybrid or active PFCrecti er systems with boost type characteristic. Conventional PFC recti er,

bridgeless (dual boost) PFC recti er, and PFC recti er with ac switch.

the formation of a regulated output dc voltage . However, a sinusoidalshape of the phase currents is possibly not feasible (cf.S c IV A2 forsystems showing phase symmetry but no bridge symmetry), and/or

the output voltage or the modulation range of the circuits is limitedcompared to symmetric structures. Furthermore, in general a morecomplex modulation results (cf. e.g. [3 ] as an example of a systemwith bridge symmetry but no phase symmetry). In addition, withmissing phase or bridge symmetry, differing loadings of the individualpower semiconductors occur. symmetrical circuits are hence treatedwithin the scope of this work only as an intermediate step in thederivation of symmetrical circuits .

In the following for all circuits, i.e. also for systems employingpower semiconductors with high blocking voltage stress (de nedby the mains line-to-line voltage), power MOSF Ts are shown asswitching elements. This should point out the generally existingrequirement of high switching frequency or high power density. nimplementation of the power semiconductors would be possible withSi super-junction MOSF Ts with a blocking voltage of 00 V [ 3 ]or in future with SiC JF Ts (in a cascode con guration, [38] 0])or SiC MO SF Ts [4 ] ). lte atively, 1200 V IGBTs, possibly withSiC freewheeling diodes could also be employed, however, withconsiderably lower switching frequency, due to the relatively highswitch-off losses.

A. Boost-Type Systems three-phase extension of the conventional single-phase boost

type PFC recti er [cf.Fig H(a)], i.e. the addition of a third bridgeleg to the input recti er bridge, results in a hybrid recti er structureshown inFig 12(a). The system enables a control of the outputvoltage, but the input current exhibits the characteristic block shapeof passive diode recti cation withT 0% [cf. Fig 12(b)] .


10/28

ib I c l

n

$0 :

s

pib L c

c) n

4

0:

4

d ms)

Fig. 12. Three phase extension of the single phase system shown inFig. 11(a a System structure and corresponding mains voltage and mainscurrent if the dc-dc boost converter stage operates in Continuous ConductionMode (CCM). System structure if the boost inductor is shi ed to the acside and divided over the phases.d Corresponding mains voltage and currenta refers to the local average value of the phase currenta for operation of

the system in Discontinuous Conduction Mode (DCM).

If the boost inductance is moved to the ac side and distributed overthe phases [cf.Fig 12(c)] and the mode of operation is changed toCM at constant duty cycle of the power transistors, the switching

frequency peak values of the discontinuous phase currents follow asinusoidal envelope. However, as shown by more detailed analysis,low-frequency harmonics of the phase currents continue to occur[4 ]. modulation of the transistor switch-on time with sixfoldmains frequency [43 ], [44] an or operation in Boundary ConductionMode (BCM) also cannot completely eliminate the low-frequencycurrent distortion, since the smallest phase current in each case alwaysreaches zero prior to the other two phase currents and thus exhibitsa zero current interval at switching frequency [4 ]. ccordingly, arelatively high current quality is only attainable for high voltagetransfer ratios an or a relatively short demagnetization time of the

9inductors compared to the transistor switch-on time, i .e . for outputvoltages > 1 kV when operating in the uropean low-voltagemains. Because of this limitation, and the high peak current loading ofthe power semiconductors and the large MI lter effort, this circuitconcept has not been successful in industry.

Fundamentally, it should also be noted here that for three-phaseconverter systems, because of the relatively high power, operation inCCM is clearly preferable. ccordingly, the phase-shi ed operationof several CM converter stages, which would be possible for thesystem shown inFig 12(c), [45], [4 ], and is frequently used forsingle-phase sy stems (for power levels up to typ 1 kW) is of minorimportance.

) Hybrid 3rdHarmonic urrent Injection P Rect er:n improvement in the input current quality of the circuit in

Fig 12(a) is only possible by extension of the controllability, i. e. byaddition of a further power transistor (cf.Fig 13). The currents inthe positive and negative dc buses,i and i_ can then be controlledindependently and proportional to the two phase voltages involved ineach case in the formation of the output voltage of the diode bridge.If the difference ofi and i_ is then fed back via a current injectionnetwork (three four-quadrant switches, of which in each case onlyone is switched on) into the mains phase which would not carrycurrent for simple diode recti cation, a sinusoidal current shape canbe assured for all mains phases as shown below [4 ].

Because of the symmetries of the supplying mains voltage system, the mathematical proof of the sinusoidal current shaping canbe limited to a 60 -wide interval of the mains period with e. g.UaN>UbN>U Nfor which the injection switchbYb(in general,the injection switch of the phase with the smallest absolute voltagevalue) is continuously switched on.

By suitable modulation of a current proportional to the mainsphase voltageUaNcan then be impressed in or in the conductingdiode a

( 0)

whereby for the local average, i.e. the fundamental frequency component

( )

has to apply. Correspondingly, by suitable modulation of_ acurrent

= i ( )proportional toU Ncan be impressed in_ or - with

( 3)

The fundamental frequency conductance* determining the recti er input an or power, is thereby de ned by the output voltage

controller. The switching of and _ must not be carried out ina co-ordinated manner since freewheeling ofi an or i_ is alwayspossible via the freewheeling diodes and _ or the diodesantiparallel to and _ and the injection switchbYb.

For the injection currentiy follows by irchhoffs current lawi iy = 0 or ia + i iy = 0, and on consideration ofib = iy (injection switchbYb is switched on)

ib = ( ia + i ) and Zb = (Za +Z ) . ( 4)With ( ) , ( ) andUa + UbO+ U O= 0 (symmetrical sinusoidalmains phase voltages) there then results

( 5)

ccordingly, for all phases a current shape proportional to the corresponding mains phase voltage is achieved. ( quation ( 4) could also


11/28

0L i P

a i

.b CL i_ _ n

La ib l Ii

)

,

w

- .

ig. 13. a Bas c structure of the hybr d d harmon c current nject onrect er [47]. Local average equ valent c rcu t of the act ve system partfor aN> bN> N Waveforms of the phase voltage aNthecorrespond ng phase currenta and the njected current t would beadvantageous to add a second freewheel ng d ode_ n the negat ve busto m n m ze the sw tch ng frequency voltage var at on of the output.However, th s would result n ncreased conduct on losses.

be stated directly with reference to the current sumia + ib + i 0forced to zero because of the free mains star-point , but was derivedhere starting from the dc side in order to illustrate the function ofthe current injection.)

s would be clear from an analysis of further 60 sections ofthe mains period, the injection currenti exhibits threefold mainsfrequency. Thus, and in the sense of the classi cation chosen here,the recti er system should be calledhybrid 3rd harmonic currentinjection P rect er.

The feedback and/or injection of current occurs inFig 13 alwaysinto only one phase, which is selected by proper gating of the fourquadrant injection switches. lternatively, the current injection couldalso be carried out by means of a purely passive injection network,e.g. a resonance network or a transformer circuit with low zerosequence impedance (cf.INNE OTA Recti er, [48]). However, itcan then not be chosen in which phase a current is injected, but thefeedback current can only be divided up into equal parts that are theninjected into the phases. s shown in [4], a sinusoidal phase currentwaveform proportional to the mains voltage can also be attainedtherewith. However, the passive injection network shows a relativelylarge volume and weight Furthermore, in comparison to the circuit inFig 13(a) a threefold amplitude of the injection current is required,so that the semiconductors of the converter stages that impress the

currents i and _ must be dimensioned for a signi cantly highercurrent loading. Hence, considering the high power density o endemanded in industry, an active current injection i s clearly preferable.

With regard to the operating range of the sy stem it must be statedthat the output voltage must be selected signi cantly higher thanthe peak value of the line-to-line mains voltage, i.e. as>

N, ,rm as given in [49], p. 595, Sec. II-D. s shown above,ohmic fundamental mains behavior can be attained, but no phasedisplacement of the mains current relative to the mains voltage ispossible. However, it has to be emphasized that the sy stem allows acontinuation of operation with sinusoidal current shape (at reducedpower) on failure of a mains phase. ll injection switches then haveto be blocked and a simultaneous gating of and _ provided,so that the same conditions exist as for a single-phase PFC recti ersystem operating on a mains line-to-line voltage.

Hybrid3rdHarmonic urrent njection Active- ilter Rect er:Starting fromFig 13(a) and following a circuit concept known frompassive injection networks [ ] , [50] , the two inductorsand _could be lumped together into a single inductor lying in theinjection path. The resulting circuit structure is shown inFig 14[ 3], [ 4]. The output diodes and _ can now be omitted, sincea simultaneous switching on of the transistorsor _ would leadto a short-circuiting of the mains, in contrast toFig 13 and is thusnot allowed.

ccordingly, the systems shows a relatively low implementationeffort, however, at the expense of a missing output voltage control.s can be immediately seen from the absence of diodes and_ the output voltage is now determined directly by the diode

bridge and hence exhibits a six-pulse shape. However, as will beshown below, assuming a constant power load, the possibility ofimpressing sinusoidal mains phase currents remains. Thus the systemdoes not provide the full functionality of an output voltage-regulatedPFC recti er, but the function of a passive recti er with integratedactive lter and hence should be called ahybrid3rdharmonic currentinjection active lter rect er.

If a load with constant power consumption is supplied, there resultsa load current varying in antiphase to the six-pulse recti er outputvoltage. s shown below, this leads to a sinusoidal shape of all mainsphase currents a er overlaying with the injection current.

Consider a 60 interval of the mains period withUaN>UbN>U Nas in S c IV A . For the current to be injected into phase

( )

applies. The mains frequency voltage drop across the inductorcan in a rst approximation be neglected for the formation of

0 . ( )

ccordingly, we have for the voltage to be formed at the output of

the bridge-leg( 8)

If in any case one of the transistors or _ is switched on, +_ 1 applies for the relative switch-on times or_ 1

and hence, for the voltage formation of the bridge-leg

UaN+ 1 U N a + U N. ( 9)Correspondingly, there follows the duty cyclewith ( 9) as

and thus for the current in

. . * *Ub

s b UbN UbN Ua

( 0)

( )


12/28

a ib

p

c

L

wI

_ ." ."Jy :I

Fig. 14. a Hybr d rd harmon c current nject on act ve lter rect eraccord ng to [4], [14]. Local average equ valent c rcu t of the act ve parof the system for aN> bN> N Waveform of the phase voltageaNthe phase current a the nject on current y and the load current

o/ at a constant output power o

Considering the fundamental input currents that have to be generatedat the input

a = G*UaNb = G*UbN = G*UN

( )

the low frequency component of the current consumption of theconstant power load can be expressed via

i = Po = aUa +bUb= G*UaNUa + UbNUbUa Ua Ua*( Ub)= G UaN + UbN -Ua

( 3)

( 4)

For the resultant low frequency current component drawn from phasethere then follows with ( )

( 5)

directly the desired (sinusoidal) waveform proportional to the mainsvoltage. Furthermore, there applies with ( ), ( 5),

( )

and UaN + UbN +U N= 0 for phasec( )

so that the sinusoidal shape of all phase currents has been proved.It must be emphasized that the circuit inFig 14(a) allows a

sinusoidal regulation of the mains currents only in case a converteroutput stage, e.g. in the form of a dc dc converter or a PWM inverterstage, is present that assures constant power consumption. Theimplementation e t of this concept should therefore be evaluatedonly with regard to the overall system.

Furthermore, it should be noted that the waveform ofi requiredfor the formation of a sinusoidal mains current only results if nosmoothing capacitor (of higher capacitance) is connected betweenconstant power load and recti er stage. oad variations are thuspassed on directly to the mains.

) ! -Switch Rect er:If the circuit inFig U(e) is extended with a third bridge-leg and adelta-connection of four-quadrant switches is added with respect tothe operating principle and the three-phase symmetry, there followsthe topology inFig 15(a) of the ! -switch recti er [ 5 ] [53 ] . Thefour quadrant switches enable to in uence the conduction state of

the diode recti er and thus to control the ac side voltage formationvia pulse width modulation. The ! -switch recti er is an active PFCrecti er circuit since the commutation of the diode bridge occurs, incontrast to the circuit inFig 13, at switching frequency.

Similar to single-phase PFC recti er circuits, the voltage formed atthe input of a recti er bridge-leg, e.g.UaM' M'designates a virtualmid-point of the output voltage), depends on the switching state of the(entire) converter and on the direction of the phase currentia.Thisdoes not represent a limitation since a currentia in phase with themains voltageUaNhas to be impressed. eglecting the voltage dropacross the input inductorU N UNhas to be ensured. Therewithia = G*UaNand UaNalways have the same polarity.

xcept for a simultaneous switch on of all four-quadrant switchessah = She = Sea= 1), one phase terminal, e.g. is always

connected with the positive or negative output voltage bus,orccordingly, the circuit shows a two-level characteristic with regard

to the voltage formation. s is immediately clear considering thediode recti cation on the input side, the output voltage has to beselected according to

>v N, ,rm . ( 8)In case of a failure of a mains phase, the two four-quadrant switchesassociated to the phase that failed have to be permanently disabled.Then again a single-phase PFC recti er circuit with an ac sideswitch is present, which operates however from a line-toline voltage,but still allows a regulation of the output voltage and a sinusoidalimpression of the mains currents.

It is interesting to understand that the operating range of the :switch recti er is not restricted to ohmic fundamental mains behavior(as could be expected due to the diode recti er) but allows theformation of current phase displacements in the angular interval

0 ib


13/28

12p

a i LLL

S nL

: Upnb)

u uab ,uiN

la

w

Fig. 15. a C rcu t topology of the sw tch rect er [51]. Also a br dgecon gurat on of s x t rans stors w th ant parallel d odes and short c rcu tedoutput term nals could be used nstead of a delta connect on of four quadrantsw tches. Equ valent c rcu t of the system fora> b< < .e.for + Waveforms of the ma ns phase voltages, localaverage phase currentaand sect ons of the l ne to l ne voltages.

OThe voltagesuacanduabcan be formed for the given polarities ofthe currents. However, as a result of the phase displacement betweenthe phase quantities and the line-to-line quantities of 0 (comparee.g.UaNand Uabor UaNand a Ua> 0 and Uab> 0 applies,not only for = 0 0 but also for = 60 60 .Therewith, the balance of the voltagesUa uac> 0 and Uabuab>0 can be also achieved for phase voltages with a displacementof 0 against the phase current system.

It is advantageous to use the circuit inFig 16 for the control of thesystem. There, all phase currents are continuously controlled opposedto alternative control concepts [54]. The voltage reference valuesformed at the output of the phase current controllers are convertedwith a - transformation into the effectively adjustable line-to-linevoltage reference values

* * *ab= U N UbU5c= U5 U .* * *Uc = UcN U

(30)

The polarity of the phase currents or phase voltages, i.e. the information of the actual 60 sector of the mains period has then tobe considered for the control of the individual power transistors,however, no sector dependent switch-over of the entire controlstructure is required. This results in a higher input current qualityas a switch-over of the control, potentially causes current distortionsat the switching instants.

Rect er:If the delta( )-connection of the four-quadrant control switches ofthe -switch recti er is replaced by a star-connection, and the star

point of the switch arrangement is connected to a capacitively formedmid-pointMof the output voltage in terms of highest possiblesymmetry, an active three-level PWM recti er sy stem (cf.Fig 17(a),[4 ], [55]), known as Recti er, results. Functionally equivalent alternative implementations of the bridge-leg structure with alower blocking voltage stress of the power diodes are depicted inFigs 17(b) (d) [55] [5 ].

s for the -switch recti er, the ac side voltage formation of thesystem is again dependent on the sign of the phase currents. However,the recti er phase input, e.g. now can be also connected to the midpointMof the output voltage besid s the positive and the negativeoutput voltage bus. Thus, three voltage levels are available for theformation ofUaM,which justi es the designation of the system as athree-level converter.

major advantage of the three-level characteristic is that for theselection of the blocking voltage capability of the power transistoronly half of the peak value of the line-to-line voltage and not thetotal value is relevant. Furthermore, as a result of the higher numberof levels ofU M,the differenceUaN remains limited to smallvalues. Thus a smaller mains current fundamental ripple results [cf.

Fig. 16. Control scheme of the sw tch rect er w th super mposed output voltage controllerK sand subord nate nput current controllersKr sw thfeed forward of the ma ns phase voltages The rect er nput phase voltage reference values generated by the phase current controllers are converted ntol ne to l ne voltage reference values by us ng a Y transformat on and then appl ed to the system by modulat on of always only two of the four quadrantsw tchesSaba' Sbeb' Sea [cf. Fig. 15(e ]


14/28


15/28

14

b

Fig. 19. Hardware demonstrator of a kW VIENNARecti er.Measured phase current . %) and corresponding mainsphase voltage . Operating parameters: mains rms line to line voltageNIIrms400Vmains frequency N 800Hz, output voltage80 Yswitching frequency 2 0kHz. Scales:200V/div, div,0 ms/div. state-of-the-art hardware demonstrator of the Recti er

is shown inFig 19. The switching frequency of the system is0kHz. Such a high switching frequency, however, is only

useful if an extremely lowT of the input currents has to beachieved at high mains frequencies (e.g. for More lectric ircra[53], N 360Hz . . .800Hz). o advantage is given with regard topower density compared to a system with 7 kHz for forcedair cooling (cf.Fig 35).

4) Hybrid Ha - ontrolled Active ull- ont lled S -SwitchP A D Bridge ircuit:If instead of the conventional single-phase PFC recti er structurebridgeless (or dual-boost) converter topology inFig U(b) is extended to three-phases, a half-controlled hybrid phase-symmetricalrecti er circuits shown inFig 20(a) results.

This circuit topology does not allow to impress a sinusoidal inputcurrent within the entire mains period because of the lack of bridgesymmetry. If, e.g. a60interval withU >0 and U U 0 U 0 ic vUN,ll,rms (33)is required, the same as for the systems inFig 15(a) and Fig 17.Furthermore, the system can deal with a mains phase failure, thusrepresenting a mains interface that can be used in an extremely exible manner. Hence the entire circuit structure is also commerciallyavailable as a power module (generally denominated as "six packpower module) and is widely used in industry.

It should be emphasized that the circuit has a relatively simplestructure, despite the high functionality, i.e . in particular, the bidirectionality does not result in an increase in the number of switchescompared with the unidirectional str ctures derived above. Thisbecomes especially clear on using Reverse Conducting (RC)-IGBTs,which apart from the power transistor, also include the anti parallelfreewheeling diode in one chip. The same applies (in future) e.g. forSiC JF Ts (in cascode connection). The circuit is thus of particularinterest, despite the limitation made here to unidirectional systems

as the wide phase angle range may be exploited also in the purelyrecti er mode, e.g. for reactive power compensation.) Discussion:

ccording S c IV A , the implementation of a three-phase boosttype PFC recti er is possible with a current injection conceptbased on passive diode recti cation or, according toS c IV A2-S c IV A4, by control of the conduction state of a diode bridge withtu -off power semiconductors, which allows a direct impression ofsinusoidal ac currents.

The direct ac current impression is preferable compared with theimpression of two dc currents (in combination with a current injectioninto the third phase) for an industrial system as a switching atthe sector borders, potentially causes distortions is not required. Inaddition, active recti er systems are not limited to a purely ohmicfundamental mains behavior, therewith e.g. the capacitive reactive


16/28

p

a iac ,"

np

i

b nFig. 20. a Half controlled (hybrid) boost type three phase ac-dc bridgecircuit. Full controlled active three phase ac-dc bridge circuit (bidirectionalsix switch active PFC recti er).

power of the MI input lter can be (partly) compensated, or ingeneral a higher exibility for the cur ent control is given.

Thus, for the comparative evaluation inS c VI C, only the switch recti er, the Recti er and the (bidirectional) sixswitch boost- type PFC recti er are considered. The -switch recti ercould here be also omitted with regard to the system complexity.Given by its structure, however, the system cannot generate a shortcircuit of the output voltage in case of a faulty control of the powertransistors. Therewith, an advantage is given regarding the operationalsafety compared with the six- switch converter. The evaluation of thesystem furthermore is reasonable in terms of completeness of theanalysis.

e o :Wi

= lUrn lUpnb J c

Fig. 21. a Time behavior of the phase voltages within a mains period.Active par of the system for sector 1 aN> , bN N< ) with thepossibility of controlling only one phase current.Active par of the systemfor sector 2 aN bN> , N< ) with the possibility of controllingtwo phase currents, i.e. all phase currents.

15B. Systems with Buck-Type ha cteristic

s single-phase buck-type ac dc converters do not enable toprovide sinusoidal cur ents over the entire mains period [ ], thederivation of buck-type PFC recti er structures has to refer directlyto three-phase diode recti er circuits with dc side inductor [cf.Fig 2(b)]. The diode bridge has to be extended with tu -off elementsby considering phase- and bridge-symmetry requirements, such thatthe mains phases to be connected with the dc side can be arbitrarilyselected. lternatively, also a system based on the injection principlecould be conceptualized with reference toS c IV A .

) Active Six-Switch Buck-Type P Rect er: power transistor has to be added in series to each diode in the

circuit shown inFig 2(b) to enable a full, i.e. a voltage-independentcontrollability of the power transfer. The resultant converter structure is shown inFig 22(a), which is known from cu ent dc-linkconverters used for drive systems [ 3]. The diode-transistor seriescombinations represent here unidirectional, bipola blocking switches,which can be also replaced by RC-IGBTs in terms of a reductionof the conduction losses. However, a limitation of the switchingfrequencies to relatively low values (in the range of10kHz) wouldthen be required due to the relatively high switching losses [ 4].

If a transistor of the positive bridge half, e. g. transistorand atransistor of the negative bridge half, e.g. are switched on, theoutput inductor currentI is drawn from phase and fed back intophasec,

ia +ib 0i

(34)

[cf. Fig 22(b)]. In addition, the line-to-line voltageUa is switchedto the output, i.e. is used for the formation of the output voltage

U U N UcN Uc Ua . (35)

If solely both transistors of a bridge-leg are switched on,ia ibi 0 applies andU 0 i.e. the system is in a freewheeling state.The conduction losses in the freewheeling state could be reduced byimplementation of an expl icit freewheeling diode.

By proper modulation, the output currenti can thus be distributedto the three phases in such a manner that a e low-pass lteringof the pulse-width modulated discontinuous input cur entsi ifi sinusoidal mains cur ents result (inFig 21(a) only the ltercapacitors F of the MI lter on the mains side are shown).Furthermore, the output voltage , which is formed by low-pass lteringof with the output inductor and the output capacitor can beadjusted with the relative duration of the freewheeling state startingfrom zero to values

< N, ,rm . (3 )

The limited output voltage range is explained by the fact that twoline-to-line voltages have to be used in each pulse period for theformation of the output voltage in order to supply each mains phasewith cur ent. In order to maximize the achievable output voltage,here always the two largest voltages, e.g.U and Uf are selected(valid within a60interval of the mains periodN ( 30 +30)cf. Fig l (c . Both voltages have a phase displacement of60.Therefore, only voltage valuesU f V N, ,rm areavailable for the pulse period at the intersection of both voltages.Correspondingly, the output voltage range is de ned by (3 ). Thefull controllability of the system generally allows an arbitrary phase


17/28

1L p

n

i

b) CFLFig. 22. a Circuit topology of the active six switch buck type PFC recti er.

Equivalent circuit of ac par of the system. Note: if power transistors wereonly implemented in the positive or negative bridge half a circuit analogousto Fig. 20(a would result, which also would not allow for a sinusoidal currentimpression due to the limited controllability.

displacement between the mains current and the mains voltage of

(3 )

However, with an increasing phase displacementthe line to linevoltages switched to the output have lower instantaneous values,an or

< N, ,rm o < (38)applies. Correspondingly, e.g. for< 90follows 0 asis immediately clear considering the power balance between the acand the dc side. The output voltage range (3 ) is thus only valid for< 0 an or in order to ensure a wide output voltage range, thephase displacement


18/28


19/28

18again a60interval o f the mains period withaN> bN> cNorCN(0 60) is considered. The active part of the circuit in thismains sector is depicted inFig 26(b).

The recti er system should represent a symmetric three-phase loadof (fundamental) phase conductanceG*to the mains. ccordingly,the local average values of the (discontinuous) input currents may bewritten as

1a G*aN15 G* bN1c G*cN

(4 )

iNNThe reference output currentI*, t o be impressed by thebuck converter is then given under the assumption of a symmetricalthree-phase mains system by

is relevant. er simpli cation, the output voltage may be written as

uxzO+ab Ocb (4 ) multiplication of (4 ) withI results in

(48)

the instantaneous power which under the assumption of symmet

rically loaded mains shows a constant value P. ccordingly,at a constant currentI also a constant voltageu and thus due to0 a constant output voltageupnUpnis generated.s the previous derivation shows, the operation of the system is

limited to purely ohmic fundamental mains behavior,

(49)

[cf. (4 )]. The output voltage range is limited by the minimal value(4 ) of the six-pulse diode bridge output voltage,

Ndesignates the amplitude of the phase voltages ,Upnis the outputvoltage). n ideal output current controller an or1 I *isassumed for the further considerations. The currents in the phasesand care impressed by a respective switching (PWM) of

S+and _

whereby the duty cycles result with (4 ), (4 ), and (43) as

Upn0+1- u N3 A aUNUpn0+1- u N3 A cUN

(43)

(44)

Considering the partitioning of the current in the nodeand a+b+ c 0 or 1a+ 15+ 1c 0 the injection current(45)

results. Thus, the correct current is injected into the third phase (herephase For the formation of the output voltage,

uxzO+aN+ (1 0+bN (o ucN+ (1 0_ bN)(4 )

Upn< UN,ll,rms (50)and therewith identical with the output voltage range for six-switchactive buck-type PFC recti er systems.

pos sible implementation of the control circuit of the sys tem, witha superimposed output voltage controllerK ( and a subordinateoutput current controllerK J( ) is shown inFig 27. ltimately,with the output current controller the current forming voltageuxzisde ned, where advantageously a feed-forward of the output voltageu;n Uis applied. The adjustment ofxzis obtained with anappropriate selection of the duty cycle of the pulse width modulationof the transistorsS+and _ [cf. (43)]. There, the (normalized)voltages Nand Nare used as modulation functions accordingto (44) (cf. alsoFig 27 .

The pulse width modulation ofS+andS_can be implemented inphase or antiphase. The switching frequency ripple ofyis minimizedfor carrier signals D+and Dthat are in phase. For carrier signalsthat are in opposite phase, a minimal output current ripple but a

*

p

n

Fig. 27. Control structure of the S Recti er with a superimposed output voltage controllerK sand a subordinate output current controllerKJ swith feed forward of the output voltage. The voltage required to control the output current is formed through modulation ofS+and S_such that in bothconducting branches of the diode bridge a pulse width modulated current results. The local average value of this current is proportional to the correspondingmains phase voltage. A four quadrant switch is switched on by the sector control and injecting always into the mains phase with the smallest absolute voltagvalue.


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ibcCF

p

nFig. 28. Combination of an active lter type3rdharmonic current injectionrecti er and a dc-dc buck type converter stage to an active buck type PFCrecti er system. The system is characterized by a minimal number of powersemiconductors in the main current path, and only a low frequency variationof the output CM voltage. The dc-dc buck conver er should be advantageouslyimplemented as interleaved conver er.

maximum ripple ofiy results, which needs to be considered for thedesign of the lter capacitorsF at the input.

It should be noted that a hybridrdharmonic injection PFCrecti er circuit can also be built by combination of an activelter-type rdharmonic injection recti er and a simple dc dc buckconverter stage (cf.Fig 28 . The buck stage to be controlled e.g. fora constant output current or a constant output voltage then ensures,independent of the pulsation of the voltage with sixfold mainsfrequency, a constant output power. The advantage of this circuittopology is that only a single power transistor is lying in the maincurrent path, i .e . in particular at high output voltages with a relativelyshort freewheeling interval, low conduction losses occur. In addition,the negative output voltage terminal is always connected to the mainsvia a diode of the lower bridge half of the diode recti er. Therefore,no output CM voltage with switching frequency is generated. Theimplementation effort of the CM MI lter can thus be reduced.Only the parasitic capacitors of the power semiconductors lead tohigh-frequency CM noise currents (cf. related consideration of boosttype PFC recti er systems in[53] .

4) Discussion:The impression of the mains current of the considered buck-typePFC recti er systems is obtained with so far known current controlschemes always only in an indirect manner. ccordingly, contrary tothe boost-type PFC recti ers (cf. IV A , concepts based on thecurrent injection principle and direct active systems can be consideredas equivalent regarding the achievable input current quality.

Therefore, for the comparative evaluation in VI C, bothconcepts, i.e. the six-switch buck-type PFC recti er and theSWIRecti er, are considered. The three-switch buck-type PFC recti eris omitted due to the higher conduction losses and the less uniformdistribution of the semiconductor losses compared to the six-switchbuck-type PFC recti er.

C. Systems with Boost- Type and Buck- Type Cha cteristics shown in Fig 4 the output voltage range of boost-type PFC

recti ers is not immediately adjoining the output voltage range ofbuck-type systems. Voltages in the range

UN, ,rm


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2represents the ratio of the amplitude of the three-phase voltage orcurrent system on the ac side and the dc output voltage, an or thedc output current

voltages of interest can be determined [ 4] by averaging over themains period. The resultant equations for the individual topologiesare compiled inFig 30.

M= iuf (5 ) B. DM and M EM ilter

Uu UNrepresents the amplitude of the fundamental of thediscontinuous phase voltage at the recti er input of a boost-typesystem,iu is the amplitude of the fundamental of the discontinuousphase currents at the recti er input of a buck-type system).

With the relative switch-on time (duty cycle) and the input current(for boost-type recti ers), an or the output current (for buck-typerecti ers), the instantaneous conduction states of the power semiconductors are de ned, and the local average current values canbe calculated by averaging over a pulse period. Based on that, theglobal average and root mean square (rms) values of the currents and

; - +'I )

14

ab L

L

I 6+ r lD l " 2 ".+ (3 4Ai

7p

n

10,.& I .2 . I .rm 5+ 2

\ p

The input inductors of the boost-type PFC recti er systems,discussed inS c IV A, are to be considered as the rst stage ofa multi -stage MI lter placed on the ac side similar to the inputlter capacitors of the systems with buck-type characteristic. The

conducted MI noise is suppressed with this lter such that thestandards of conducted noise are ful lled in the frequency range of1 0kHz . . .30MHz (e.g. CISPR11 .(Depending on the application,another MI lter might be required on the dc side [ 5], [ ], whichhowever is not discussed here for the sake of brevity.)

Three-phase recti er circuits inherently generate a CM voltagebetween the mid-point o f the output voltage (the output voltage buses)

L pIS/ D.

iab

dn

1Mb[ s l A

V

+ " I I

' piaL ia

b LL

)[0 i IS,3\'g

f lID /

\ .{

I l V ia L

b LL

c n

n

e

Llog =

To ."

; +{ I,.

+4: In m / 0 ." i i + 4: m - D\ -i

b

f-g /\ ;

e

- T, / o " i,j ( - -- . < -

n

Fig. 30. Circuit topologies and current stresses of the main components of the power circuit of selected active boost and buck type PFC recti er systems .a Six switch boost type PFC recti er [cf.Fig. 20( ], switch recti er (cf. Fig. 16 , VIENNARecti er (cf. Fig. six switch buck type PFCrecti er (cf. Fig. 21 , e S ISSRecti er (cf. Fig. 25 , and active lter type 3rdharmonic current injection recti er (cf.Fig. 13


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and ground. The CM voltage waveform for a passive diode recti ercircuit with inductors on the dc side is depicted inFig 31(a). Foractive recti er circuits, the CM voltage has a pulsed waveform [cf.Figs 31(b) and ] thus, CM currents result due to the parasiticcapacitances to ground.

For fully active boost-type PFC systems, e.g. for theVIENNARecti er, the CM voltage originates from the recti er ac side phasevoltages employed for the current impression that do not add to zero(except for the switching stateSaM=S5M=ScM=1.Thus, as forthe Y-recti er, shown inS c III A, a CM voltage

1MN=- ( aM+ 5M+ cM) = CM (53)is generated between the mid-point of the output voltage and the(grounded) mains star-point, which could contain a low-frequencycomponentUCM,but contains in any case a switching frequencycomponent CM

CM=UCM+ CM (54)(cf. for theVIENNA Recti er also [53], Fig. 3.4 and Fig 5. 3). ltering of CM can be achieved by connectingMvia a capacitorCM Mto an arti cial mains star-point

'(representing the ground

potential), formed by a star-connection of lter capacitorsF andinsertion of a CM inductorLCM lin series to the boost inductors[cf. Fig 3Z(a)]. low-frequency variation of the potential ofMisthus no t prevented. In addition, contrary to a placement of CM ltercapacitors to ground, ground currents are minimized. dditional CMlter stages have to be implemented on the mains side for the lteringof the noise currents that result from the parasitic capacitances of thepower semiconductors to the heat sink [53].

For fully active buck-type PFC systems, within each pulse periodtwo line-to-line voltages are switched to the recti er output for theformation of the output voltage and for the distribution of the outputcurrent to the mains phases . This again leads to a CM voltageCMat switching frequency. ( CM voltage during the freewheeling

interval can be avoid by symmetrical splitting of the output inductorto the positive and negative output bus) . The concept described abovefor boost-type converters can advantageously also be used for theltering of CM of buck-type systems [cf.Fig 3Z(b)] where the

CM inductance has to be inserted on the dc side.For determining the switching frequency component of the M

voltage, which is relevant for the design of the M ltering, forboost-type systems within each pulse period, the formation of theinput current has to be considered. .g. for phase

ia ( )LT = aN - aM+ CM = aN - aN= aN -UaN - aN

(55)

(5 )

applies. The pha se current consists of a fundamental and a switchingfrequency component

where the fundamental componentza is formed according toZaL = aN - aN

and the voltage to be suppressed by the M ltering isi aLd = aN = DM .

(5 )

(58)

(59)

The ltering of DM (each phase shows an equal spectral composition of the related M voltage) is performed with the boost inductorsand with ac side capacitorsDM lbetween the phases, wherebytypically two lter stages are required [cf.Fig 3Z(a)]. dditionally,

ms400 -

200G 0u; 200

)400

r 20 0 ---_---- _ ---

-4000 5 10 15 20

c ms

21

Fig. 31. CM voltage at the output of three phase recti er systems referencedto the grounded star point of the mains.a Passive diode recti er withsmoothing inductor on the dc side [cf.Fig. 2( ] VIENNARecti er [cf.Fig. 18(a ] Six switch buck type PFC recti er [cf.Fig. 22(a ]

damping elements for reducing the resonance peaks [ ] in respectof the control stability of the system have to be added, which alsoprevent the excitation of the lter by harmonics of the mains voltage.

For buck-type systems, the M noise i s generated by the pulsatinginput currents at switching frequency and is attenuated by the inputlter capacitors F and the ac side lter inductorsLDM land an

additional input lter stage.Regarding the volume of the MI lter, it has to be noted that, e.g .for boost converter systems, a constant voltage is decomposed intoits spectral components by the pulse width modulation, i.e. into amains-frequency fundamental component and harmonic componentsgrouped around multiples of the switching frequency with sidebands .Only the fundamental frequency is used for the impression ofthe phase current, i.e. the switching frequency harmonics must besuppressed with an appropriate MI (input) lter. The harmoniccomponents , i. e. ult imately the difference between the constant outputvoltage and the actual low-frequency voltage component to beformed, e.g.UaN, show similar rms values. Considering in additionthat the MI input lter has to conduct the input current of theconverter, a signi cant fraction of the total converter volume is expected to be determined by the MI input lter. This i s con rmed byimplemented systems, where the volume of the MI lter (includingthe boost inductors or buck input lter capacitors) typically represents0% to 0% of the total converter volume (cf.Fig ZO . Here,

it should be pointed out that the required lter attenation can becalculated analytically in a simple manner by determining the spectraldecomposition of the recti er input voltage into a fundamental anda total noise voltage [ 8] [80].

VI . OM ARATI E ALUATIONIn the foregoing sections, boost- and buck-type PFC recti er

systems suitable for industrial application have been identi ed andbrie y discussed. In the following, a comparative evaluation ofselected circuit concepts with regard to e ciency, volume, and


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24/28


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24- Six Sw tch Boos

Type Rect ferD Sw cRec fer ( )

ENNRec fer

"( )

Ah(mm')

\( - )

I O]- d ; : r 1 l LlJ - bp 10]( - )

M(dB .V) /

(%

P "(cm]/kW)

P "(cm /kW)

Fig. 33. Comparative evaluation of two alternative active boost type PFCrecti er systems, i.e. the six switch recti er according toFig. 20( and theVIENNARecti er according toFig. 17(a The characteristic of the switchrecti er is shown by a dashed line.

boost inductors compared with the two-level topologies. Only a smalldifference between the individual systems is given regarding thevolume of the output capacitor as the two- and three-level convertershave similar rms values of the capacitor currents and in any case aseries connection of two electrolytic capacitors is required because ofthe output voltage of =700 V. The center tap for the V ENNARecti er thus is inherently available.

In summary, the six-switch converter is characterized by a very

simple structure of the power circuit and the V ENNA Recti er by arelatively small overall volume or a high power density. In addition,for the V ENNA Recti er (as well as for the -switch recti er) a shortcircuit of the dc-bus through a faulty control of a power transistor isno t possible and for both topologies, power transistors with relativelyslow parasitic anti-parallel body diodes can be used.

The use of the -switch recti er, which is relatively complex withregard to the circuit structure and modulation, can be only justi edwhen a three-level topology does not provide signi cant advantagesdue to a low mains voltage or if an unidirectional topology is requiredthat prevents energy feedback into the mains by its hardware structureand not only by control. Power supplies in aircra could serve as anexample here.

. omparison of the Active S -Switch Buck- Type P Rect er andthe S Rect er

In Fig 34 a conventional six-switch buck-type PFC recti eraccording toFig 22(a) and a S Recti er according to Fig 26are compared. Both systems show, with respect to the total chip arearequirements, the volume of passive components, the e ciency, andthe conducted MI noise, only very little differences. n increasein e ciency of the six-switch structure would be easily possible byusing an explicit freewheeling diode across the dc link. For the SRecti er, a reduction of the number of power semiconductors can beachieved through modi cation of the circuit topology according toFig 28. In addition, the mains commutated injection switches couldbe implemented with RC-IGBTs with a low forward voltage drop asan alternative to the SiC JF Ts.

S x Sw cBuc TypR c f rS I R c f r ( )

T( - )

A( )

( - )

TO]- i : . + = + P - bp Ij(-) ( - )

[)

/(%

[

P"(c J/kW)

Fig. 34. Comparative evaluation of two alternative buck type PFC recti ersystems, i.e. of the active six switch recti er according toFig. 22(e and thehybrid ISSRecti er according toFig. 26

In summary, the main advantage of the S Rec ti er is not seenin a higher performance but in a dc dc converter like circuit structure.ccordingly, basic knowledge of the function of a passive dioderecti er of the input stage of the system is su cient to implementa three-phase PFC recti er with sinusoidal input current and a controlled output voltage. In particular, no space vector based modulationscheme, which is frequently applied to three-phase converters andtypically leads to dif culties when dealing the rst time with three

phase systems, has to be implemented.

VII. C N LU N

s shown in this paper, a three-phase PFC recti er functionalitycan be implemented besides a phase-modular approach with

1 direct control of the conduction state of a three-phase recti erthrough integrated power transistors or parallel control brancheswith active power semiconductors, i.e. as an active recti er or

2 by shaping the output currents of a three-phase diode recti eron the dc side and feedback/injection of the current differencealways in that phase which would not conduct current forconventional passive diode recti cation, i. e. as a hybrid recti erwith 3rdharmonic current injection.

Following these basic concepts, direct three-phase recti er circuitswith boost- or buck-type characteristic are realizable. hese circuitsadvantageously have a bridge topology (at the input) with bridge-legsof identical structure and thus feature an overall bridge symmetry.

For both circuit categories, over the last two decades, a variety ofcircuit topologies have been proposed. However, in the opinion of theauthors, from the category of the boost-type recti er systems, onlythe conventional (bidirectional) s ix-switch converter and the V ENNARecti er and from the systems with buck-type characteristic onlyagain the six- switch structure and the S Recti er, proposed inthis paper, are of interest for industrial application. Compared to thesefour topologies, other circuit concepts show a (signi cantly) highercomplexity of the power and/or the control circuit, or have highcomponent stresses at a lower complexity and a limited operating


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range with regard to output voltage range and/or current-to-voltagephase displacement angle at the input. This is of particular importancewhen operating at unbalanced mains systems or in case of failure ofa mains phase.

The selected circuit topologies enable very high e ciencies as aresult of the excellent conduction and switching characteristics ofmodern Si and SiC power semiconductors. So -switching conceptsare thus not necessary and would also not be accepted by industrydue to the increase in complexity resulting from the auxiliary circuitbranches with additional losses and due to the typically complexstate sequence within a switching period. In general, in industry onlycircuit topologies are practicable that are well understood not onlyby the inventors but also by a su ciently large number of engineers.

In terms of system complexity, it should be noted that the restriction to unidirectional power ow does not allow a reduction e.g. ahalving of the number of active semiconductors or a simpler controlscheme. The reason is that ultimately also unidirectional structureshave to conduct phase currents of both directions and to generatevoltages with both polarities. Only for three-level converters, a clearadvantage of unidirectional converters (V ENNA Recti er) is givencompared with bidirectional converters; for the unidirectional systemsix transistors (with anti-parallel diodes) and six diodes are required,whereas the implementation of a topologically similar bidirectionalT-type three-level converter system [84] requires twelve transistors(with anti-parallel diodes).

The main three-phase PWM recti er circuit topologies, exceptthe S Recti er, have been already theoretically investigatedand experimentally veri ed in the literature. Therefore, for furtheracademic research, mainly the following relatively narrow topicsremain

Direct mains (input) current control of buck-type PFC recti ercircuits. [For these systems typically only the output voltage andthe output current is directly controlled and/or the mains currentis not explicitly included in a feedback loop; thus, particularlyfor high mains frequencies ( 00 Hz), current distortions canoccur at the intersections of the line-to-line voltages.]Parallel operation of a higher number (more than two) convertersystems. (High output power levels are o en implementedby parallel connectio

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